EP3073695A1 - Verfahren zum verarbeiten eines analogsignals aus einem übertragungskanal, insbesondere eines fahrzeugsignals, über online-übertragung durch trägerstrom - Google Patents

Verfahren zum verarbeiten eines analogsignals aus einem übertragungskanal, insbesondere eines fahrzeugsignals, über online-übertragung durch trägerstrom Download PDF

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Publication number
EP3073695A1
EP3073695A1 EP15190536.1A EP15190536A EP3073695A1 EP 3073695 A1 EP3073695 A1 EP 3073695A1 EP 15190536 A EP15190536 A EP 15190536A EP 3073695 A1 EP3073695 A1 EP 3073695A1
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European Patent Office
Prior art keywords
signal
digital
coefficients
frame
processing
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EP15190536.1A
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English (en)
French (fr)
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EP3073695B1 (de
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Mark Wallis
Yoann Bouvet
Pierre Demaj
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STMicroelectronics Rousset SAS
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STMicroelectronics Rousset SAS
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B3/00—Line transmission systems
    • H04B3/54—Systems for transmission via power distribution lines
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03828—Arrangements for spectral shaping; Arrangements for providing signals with specified spectral properties
    • H04L25/03834—Arrangements for spectral shaping; Arrangements for providing signals with specified spectral properties using pulse shaping
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2602—Signal structure
    • H04L27/2605—Symbol extensions, e.g. Zero Tail, Unique Word [UW]
    • H04L27/2607—Cyclic extensions
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2656—Frame synchronisation, e.g. packet synchronisation, time division duplex [TDD] switching point detection or subframe synchronisation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2662—Symbol synchronisation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2668—Details of algorithms
    • H04L27/2669—Details of algorithms characterised by the domain of operation
    • H04L27/2671—Time domain
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2689—Link with other circuits, i.e. special connections between synchronisation arrangements and other circuits for achieving synchronisation
    • H04L27/2691—Link with other circuits, i.e. special connections between synchronisation arrangements and other circuits for achieving synchronisation involving interference determination or cancellation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00—Arrangements affording multiple use of the transmission path
    • H04L5/0001—Arrangements for dividing the transmission path
    • H04L5/0003—Two-dimensional division
    • H04L5/0005—Time-frequency
    • H04L5/0007—Time-frequency the frequencies being orthogonal, e.g. OFDM(A), DMT
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03993—Noise whitening
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/06—Dc level restoring means; Bias distortion correction ; Decision circuits providing symbol by symbol detection
    • H04L25/067—Dc level restoring means; Bias distortion correction ; Decision circuits providing symbol by symbol detection providing soft decisions, i.e. decisions together with an estimate of reliability
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2668—Details of algorithms
    • H04L27/2673—Details of algorithms characterised by synchronisation parameters
    • H04L27/2675—Pilot or known symbols

Definitions

  • Embodiments and embodiments of the invention relate to the transmission of information over a communication channel, and in particular when this channel is an electrical line, the transmission of information by line power line (PLC: Power Line Communications), and more particularly the improvement of the processing of such a signal in reception when it is noisy by a narrow band noise signal (Narrow Band Interferer).
  • PLC Power Line Communications
  • Embodiments and embodiments of the invention are compatible with the various standards governing online powerline communication, in particular but not exclusively the PLC-G3, PRIME (PoweRline Intelligent Metering Evolution) standards or the IEEE standard. 1901-2.
  • Powerline carrier technology aims to transmit digital data by exploiting the existing infrastructure of the power grid. It allows remote reading of electricity meters, exchanges between electric vehicles and charging stations, and the management and control of smart grids.
  • In-line carrier technology incorporates in particular narrow-band power line communication (N-PLC) which is generally defined as communication over a power line operating at transmission at 500 KHz.
  • N-PLC narrow-band power line communication
  • N-PLC communication thus generally uses the frequency bands defined in particular by the European Electrotechnical Standardization (CENELEC) or the Federal Communications Commission (FCC).
  • CENELEC European Electrotechnical Standardization
  • FCC Federal Communications Commission
  • the transmission frequencies are between 42 and 89 KHz in the PRIME standard whereas they are between 35 and 91 KHz for the PLC standard. G3.
  • the electrical cables carrying the powerline signals in line are in a very harsh environment. They are particularly subject to disturbances of the type white noise, colored noise or noise pulse. In addition, they are not protected against interference. As a result, any FM / AM radio signal or any wireless communication can lead to the presence of harmonics of these signals in the useful frequency band used by narrow band PLC communications.
  • noise signals which are generally Narrow Band Interferer signals, that is to say having a lower frequency band than the frequency band of the wanted signal, then disturb the synchronization phase of the signal.
  • receiver connected to the power line during which the receiver must be able to synchronize to find in particular the beginning of the useful data of the frame of symbols carried by the useful signal.
  • the signals mentioned in this document have characteristics very different from the signals used in line carrier communications. Indeed, the UWB signals, (and in particular those with direct sequence spread spectrum) have a spread of the power of the signal transmitted over a wide frequency band in order to drown this power in the ambient noise or in other communications.
  • the Power Spectral Density (PSD) of a UWB signal is generally defined as less than -41 dBm / MHz.
  • the signals used in the PLC communications are signals modulated according to a multicarrier modulation, for example a quadrature modulation on orthogonal carriers (OFDM modulation: Orthogonal Frequency Division Multiplexing, according to an Anglo-Saxon name commonly used by those skilled in the art) , but using only a subset of carriers among a larger set of available carriers.
  • OFDM modulation Orthogonal Frequency Division Multiplexing
  • the size of the inverse Fourier transform and the direct Fourier transform is equal to 512, while only 97 subcarriers (the subcarriers 86 to 182 ) are used for transmission in the PRIME standard.
  • the size of the inverse Fourier transform and the direct Fourier transform is equal to 256 whereas only 36 subcarriers (the subcarriers 23 to 58) are used in the PLC-G3 standard.
  • a processing of an analog signal coming from the channel of transmission that can improve the performance of the synchronization phase that the useful analog signal is noisy or not by at least one narrow-band noise signal.
  • coefficients of a filter are determined on the fly in the time domain. predictive of an autoregressive model of the signal and the signal is filtered on the fly in the time domain by a finite impulse response (FIR) digital filter whose coefficients are those of the predictive filter.
  • FIR finite impulse response
  • the overall signal coming from the channel is filtered, whether or not this signal contains the noise signal, which makes it possible to performing a non-coherent processing that is to say not requiring a perfect synchronization in time and in phase between the moment when the noise signal is estimated and the moment when this noise signal is subtracted from the overall signal .
  • the receiver will thus be able to synchronize from the filtered signal and a reference signal, for example a known symbol.
  • a method of processing an analog channel signal from a transmission channel for example a power line.
  • the channel analog signal is likely to comprise a useful signal modulated on a subset of a set of available carriers, such as the useful signals complying with PRIME or G3-PLC standards.
  • This useful signal carries at least one frame of symbols according to a frame structure and this useful signal is optionally noisy by at least one narrow-band noise signal.
  • the noise signal is a peak of noise at a single frequency contained in the frequency band of the wanted signal, but more generally, a narrow-band noise signal is a noise signal whose frequency band is less than the frequency band of the wanted signal.
  • the channel analog signal can quite at any given time have no useful signal or include only at least one noise signal, or a useful non-noisy signal or a noisy useful signal.
  • the method according to this aspect then comprises a digital analog conversion of the analog channel signal and a synchronization processing including a filtering process.
  • the analog channel signal that will undergo the analog / digital conversion may for example be the analog signal directly from the channel or, as is generally the case, the analog signal delivered by an analog input stage (notably comprising filters bandpass, low-pass filters and an amplifier) connected to the transmission channel.
  • an analog input stage notably comprising filters bandpass, low-pass filters and an amplifier
  • the filtering processing includes time-domain on-the-fly determination of a limited number of coefficients of a predictive filter of an autoregressive model of a digital channel signal from said analog-to-digital conversion and filtering at the same time. stealing of the digital channel signal in the time domain by a finite impulse response digital filter whose coefficients are those of the predictive filter.
  • the digital channel signal on which the filtering processing is performed is not necessarily the digital signal directly from the analog / digital conversion but can be for example the digital signal from the analog / digital conversion and possibly under-sampled.
  • the signals modulated on a subset of carriers from a set of available carriers have characteristics totally different from the UWB or spread spectrum signals. They have in fact a much higher power level than a UWB signal or spread spectrum and it is then preferable to take precautions in the filtering so as to avoid completely filter the useful signal in the absence a narrow band noise.
  • the number of coefficients of the filter is advantageously less than or equal to a limit number which is chosen so as to form a finite impulse response filter whose frequency response comprises, in the presence of the noise signal, a notch at frequency band level of the noise signal, and whose frequency response has, in the absence of a noise signal, a relatively flat profile in the frequency band of the wanted signal so as to allow attenuation of the useful signal less than one chosen value, for example 6 dB, which depends of course on the intended application.
  • the number of coefficients will also preferably be equal to this limit number so as to more easily take into account several narrow-band noise signals at different tones.
  • the inventors have observed that when each symbol comprises a cyclic prefix, the acceptable limit number of filter coefficients is of the order of three quarters, and preferably of half, the length of the cyclic prefix expressed in number of samples.
  • the cyclic prefix mainly makes it possible to eliminate intersymbol interference and is a technique that consists of copying part of a symbol to place it upstream of this symbol.
  • the poles of the predictive filter resulting from the autoregressive model of the signal become the zeros of the finite impulse response filter.
  • the FIR filter can only attenuate the frequencies corresponding to these zeros.
  • the coefficients of the predictive filter are calculated at a calculation frequency of between 2 and 5 times, and preferably between 2 and 3 times, the frequency maximum of the digital channel signal.
  • the determination of the coefficients of the filter and the actual filtering are performed in the time domain and on the fly, that is to say, as the analog channel signal arrives. This makes it possible to not miss a symbol whether the signal is noisy or not.
  • the method according to this aspect also includes detecting at least one indication for identifying at least one location of said frame structure from the filtered channel digital signal and a reference signal.
  • Said indication may be for example the recognition of a known symbol of the preamble of a frame and the reference signal may be this known symbol, the detection then taking place for example by sliding correlation operations.
  • said on-the-fly determination of said coefficients and said on-the-fly filtering comprise a grouping of the samples of said digital signal into successive groups of samples, a determination of a current block of coefficients using the current group. of samples, and an application on said current group of the finite impulse response filter having said current block of coefficients so as to obtain a filtered current group of samples.
  • each frame comprises a preamble comprising known symbols and preceding the remaining part of the frame, and the filtering treatment is applied at least to detect said at least one indication at the level of the preamble of at least a frame.
  • the filtering processes are applied at least to detect said indication at the level of the preamble of each frame.
  • the filtered digital signal At the level of the filtered digital signal, it is not possible to know whether this filtered signal results from a noisy useful signal or a non-noisy useful signal. Also, it is particularly advantageous to perform after the detection of said indication, a check of the presence or absence of the noise signal, for example from at least one known symbol of the unfiltered useful signal. Indeed, this will make it possible to improve the processing of the subsequent symbols of the frame.
  • This verification may include direct Fourier transform processing on said known unfiltered symbol and analysis of the power of each carrier.
  • This verification is for example carried out on at least one symbol of the preamble.
  • the processing of the remaining part of the frame is advantageously performed on the digital signal of the unfiltered channel, which makes it possible to decode the symbols of the remaining part of the frame on an unfiltered signal that is to say not attenuated.
  • the processing of the remaining part of the frame will be performed on the filtered channel digital signal.
  • This processing of the remaining part comprises a direct Fourier transform processing, a demapping process providing for each carrier a value of the modulation coefficient (each symbol comprises modulation coefficients, or "bins", respectively associated with the carriers) and a determining for each modulation coefficient a soft decision of said value. It is then particularly advantageous to force the confidence indications of the modulation coefficients associated with the carriers whose frequencies correspond to those of the noise signal to zero.
  • a receiver comprising an input stage intended to be connected on a transmission channel and configured to deliver an analog channel signal from the transmission channel, the channel analog signal being capable of comprising a useful signal modulated on a subset of a set of available carriers, carrying at least one frame of symbols according to a frame structure and optionally noisy by at least one narrow-band noise signal, an analog / digital conversion stage for performing analog / digital conversion of the analog channel signal, and a processing stage comprising filtering means including a calculation module configured to determine on the fly a limited number of coefficients of a predictive filter of a self regressive model of a digital channel signal from the conversion stage analog / digital, a finite impulse response digital filter whose coefficients are those of the predictive filter, for on-the-fly filtering of the digital channel signal in the time domain and detection means configured to detect at least one indication allowing identifying at least one location of said frame structure from the filtered digital signal and a reference signal.
  • each symbol comprises a cyclic prefix and the number of coefficients of the filter is less than or equal to a limit number which of the order of three quarters, preferably of the order of one half, of the length of the prefix cyclic expressed in number of samples.
  • the calculation module is configured to calculate the coefficients of the predictive filter at a calculation frequency of between 2 and 5 times, preferably between 2 and 3 times, the maximum frequency of said digital channel signal.
  • the processing stage comprises grouping means configured to group samples of said digital channel signal into successive groups of samples, the calculation module is configured to determine a current block of coefficients using the group. sample stream, and the digital filter is configured to input said current group of samples so as to output a filtered current group of samples.
  • each frame comprises a preamble comprising known symbols and preceding the remaining part of the frame
  • the processing stage comprises control means configured to deliver said digital channel signal.
  • at least the filter means for the detection means detect said at least one indication at the preamble of at least one frame.
  • control means are configured to deliver said digital channel signal to the filtering means at least so that the detection means detect said at least one indication at the level of the preamble of each frame.
  • the receiver further comprises verification means configured to perform after the detection of said at least one indication, a verification of the presence or absence of said noise signal from at least one symbol useful unfiltered signal.
  • the verification means comprise a direct Fourier transform stage configured to perform a Fourier transform processing directly on the said at least one symbol and analysis means configured to perform an analysis of the power of each carrier.
  • the verification means are configured to perform said verification on at least one symbol of the preamble.
  • the processing stage further comprises additional processing means configured to perform a processing of said remaining portion of each frame and in the absence of the noise signal, the control means are configured to deliver said remaining portion of the frame directly to the additional processing means without passing through the filtering means.
  • each symbol comprising modulation coefficients respectively associated with the carriers
  • the additional processing means comprise a direct Fourier transform stage, a demapping means providing for each carrier a value of said modulation coefficient and a suitable module. determining for each modulation coefficient a confidence indication of said value, and forcing means configured to, in case of presence of said noise signal, force to zero the confidence indications of the modulation coefficients associated with the carriers whose frequencies correspond to those of the noise signal.
  • control means are configured to deactivate the module for calculating the coefficients of the filter during the processing of said remaining portion of the frame (gel of the filter coefficients).
  • the useful signal is a signal modulated according to an OFDM modulation.
  • the transmission channel may be an electrical line, the useful signal then being an in-line carrier current signal.
  • a method for filtering an analog channel signal coming from a transmission channel comprising analog-to-digital conversion of the analog channel signal and a filtering process including an on-the-fly time-domain determination of coefficients of a predictive filter of an autoregressive model of a digital channel signal from said analog / digital conversion and on-the-fly filtering of the digital channel signal in the domain time by a finite impulse response digital filter whose coefficients are those of the predictive filter.
  • the number of coefficients is advantageously limited as indicated above and / or the coefficients of the predictive filter are calculated at a calculation frequency of between 2 and 5 times, preferably between 2 and 3 times, the maximum frequency of the digital channel signal.
  • a receiver intended to be connected on the transmission channel and comprising means, such as those defined above, configured to implement such a filtering method.
  • CPL line carrier
  • FIG 1 illustrate schematically an example of a transmitter capable of transmitting a useful signal SU on an electric line LE by line carrier current.
  • the transmission chain comprises for example an ENC encoder, for example a convolutional encoder, receiving the data to be transmitted source encoding means.
  • INTL interleaving means are connected to the output of the encoder and are followed by means of "mapping" (mapping means) which transform the bits into symbols according to a transformation scheme depending on the type of modulation used, for example a BPSK type modulation or more generally QAM modulation.
  • Each symbol contains modulation coefficients associated with carriers that will be modulated accordingly.
  • the symbols are delivered at the input of processing means MTFI intended to perform a fast inverse Fourier transform (IFFT) operation.
  • IFFT fast inverse Fourier transform
  • the modulated carriers form an SNS subset of carriers among an available set of carriers (set which corresponds to the size of the inverse Fourier transform).
  • the size of the inverse Fourier transform is equal to 256 while the modulated carriers of the subset SNS lie between the ranks 23 and 58, which corresponds to a frequency band F1-F2. between 35 and 91 KHz.
  • the sampling frequency here is equal to 400 KHz leading to a carrier spacing equal to 1.5625 KHz, which thus makes the orthogonal frequencies (OFDM modulation).
  • the size of the inverse Fourier transform is equal to 512 while the number of carriers of the subset SNS is equal to 97, which provides for the useful signal a frequency band ranging between 42 and 89. KHz.
  • the modulation coefficients associated with the unused carriers are equal to 0.
  • the OFDM signal in the time domain is generated at the output of the processing means MTFI, and means MCP add to each OFDM symbol in the time domain, a cyclic prefix which is a copy at the head of the OFDM symbol of a number of samples at the end of this symbol.
  • the length of the cyclic prefix is 30 samples for a sampling frequency of 400 KHz while it is 48 samples for a sampling frequency of 250 KHz in the PRIME standard.
  • the signal is then converted into a digital-to-analog converter CNA then processed in an ETA stage, commonly designated by those skilled in the art under the term “Analog Front End”, where it undergoes in particular a power amplification, before being transmitted on the electric line LE.
  • CNA digital-to-analog converter
  • the receiver RCP here comprises an analog input stage ET1 whose input terminal BE is connected to the electric line LE.
  • This analog input stage ET1 conventionally comprises a bandpass filter BPF, a low pass filter LPF, and amplification means AMP.
  • the output of the stage ET1 is connected to an analog / digital conversion stage CAN whose output is connected to the input of a processing stage ET2.
  • the processing stage ET2 here comprises automatic gain control means AGC making it possible to control the value of the gain of the amplification means AMP of the stage ET1.
  • the signal SAC delivered at the output of the analog stage ET1 and at the input of the analog / digital conversion stage CAN designates an analog channel signal coming from the transmission channel (electrical line) LE.
  • FIG 4 schematically the frequency spectrum of such an analog SAC channel signal.
  • this signal SAC comprises the useful signal SU conveying the data transmitted from the transmitter and whose frequency band is situated between the frequencies F1 and F2 corresponding to the numbers of the modulated carriers.
  • the signal SAC also optionally includes a noise signal SB narrow band, which possibly noises the useful signal SU.
  • the noise signal SB comprises a single tone located at the frequency F3. It can, however, in practice be distributed on the frequency carrier F3 as well as on some adjacent carriers.
  • the signal SU is a dome-shaped signal whose level is much higher than the noise level AWGN of the channel in the absence of a signal.
  • the level of the noise signal SB is higher than the level of the useful signal SU.
  • the ET2 processing stage also comprises a LPF2 low-pass filter followed, although this is not necessary, by means of MSCH subsampling means.
  • the sampling frequency of the signal in upstream of the MSCH means is noted Fs while the sampling frequency of the output signal of the MSCH means is noted Fss.
  • the SNC signal at the output of the MSCH means here a digital channel signal which is derived from the analog / digital conversion of the analog signal SAC channel and on which will be applied in particular a filtering treatment in MSL filtering means as one will see it in more detail below.
  • the frequency Fc designates in the following the calculation frequency at which will be calculated in particular the filter coefficients of the MFL filtering means.
  • the specified sampling frequency Fs is 400 KHz for an FFT size of 256.
  • the subsample of the signal at a frequency Fss lower than Fs and d performing all the filtering operations at the calculation frequency Fc equal to Fss makes it possible to reduce the implementation complexity of the processing stage, and in particular the filtering means, and also makes it possible to perform Fourier transform processing Direct Fast (FFT) having a reduced size compared to the specified size of 256.
  • FFT Fourier transform processing Direct Fast
  • the TRM frame comprises a preamble PRM here comprising eight known symbols SYNCP followed by an opposite phase symbol SYNCM itself followed by a half symbol SYNCM.
  • the TRM frame then comprises a header (header) HD followed by a PLD field containing useful data symbols to be decoded and more known by those skilled in the art under the Anglo-Saxon name of "payload".
  • the symbols of the header HD contain in particular control information for the decoding of the data of the PLD field as well as the number of bytes to be decoded in the PLD field.
  • the PRM preamble of the TRM frame allows the receiver to synchronize, that is to say to obtain an indication to find the structure of the frame in order to locate the beginning of the HD header.
  • the MFL filtering means will be applied, at least during the synchronization phase of the receiver and possibly as will be seen in more detail hereinafter during the decoding phase of the remaining part of the TRM frame (PLD header and field ) in the case where the presence of a noise signal is proven.
  • the MFL filtering means will determine, on the fly, the coefficients of a predictive filter of an autoregressive model of the digital signal of the SNC channel, then will filter on the fly the digital channel signal in the time domain by a digital impulse response filter. finite whose coefficients are those of the predictive filter.
  • a signal can be modeled using convoluted white noise with an autoregressive filter.
  • the sign * denotes the conjugate complex.
  • N must be large enough to include any periodic content of the signal and to randomize any non-periodic content.
  • N can be equal to the size of the symbol, possibly downsampled, which also corresponds to the size of the Fourier transform.
  • FIR filter finite impulse response filter
  • the coefficients A n of the FIR filter are the coefficients A n of the predictive filter of the autoregressive model mentioned above.
  • the set of coefficients of the filter is advantageously limited, that is to say less than or equal to a limit number and preferably equal to this limit number.
  • the limit number of coefficients is chosen by those skilled in the art, taking into account the application and the envisaged specifications, so that, as illustrated schematically on the figure 6 , the frequency response H1 of the filter in the presence of a narrow-band noise signal has a notch around the frequency F3 of the noise signal and that the frequency response H2 of this filter has, in the absence of the noise signal , a relatively flat profile in the frequency band F1, F2 of the useful signal of to obtain a signal attenuation lower than an acceptable limit attenuation.
  • This acceptable limit attenuation depends on the implementation and the dynamics supported by the different processing means. Those skilled in the art will be able to choose this acceptable limit attenuation according to these conditions.
  • the acceptable limit attenuation can be of the order of 6 dB.
  • the length of the cyclic prefix for a sampling frequency Fs of 400 KHz is 30 samples.
  • the on-the-fly filtering process in the time domain provides a grouping of samples (step 70) so as to form a current group GR of N samples.
  • step 71 the coefficients of the predictive filter are calculated by executing the Levinson algorithm according to the above-mentioned sequence for m varying from 0 to the limit value of the number of coefficients.
  • step 72 the current group GR of N samples in the time domain is filtered with the finite impulse response filter whose coefficients are those which have just been calculated for the predictive filter.
  • a GRF group of N filtered samples is then obtained.
  • the calculation frequency Fc of the filter coefficients (equal to the frequency Fs or possibly to the frequency Fss in the case of sub-sampling). not too large compared to the maximum frequency of the digital channel signal (possibly downsampled.
  • the MFL filtering means functionally comprise MGR means configured to group the samples into groups of samples, an MCL module for calculating the coefficients of the predictive filter and an FIR module implementing the finite impulse response filter.
  • these various means and modules may for example be made in software in a microprocessor.
  • the filtered digital signal SNF delivered by the filtering means MFL is in particular used by synchronization means MSYNC, of conventional structure and known per se, to allow the receiver RCP to synchronize, that is to say by for example, to find the structure of the frame and its temporal timing so that the HD header and the PLD field can be properly decoded.
  • the synchronization means MSYNC perform sliding correlation processing between the filtered digital signal SNF and a reference signal SREF which is in this case a known symbol of the frame, for example a known symbol of the preamble such as the symbol SYNCP.
  • the indication IND representative of the frame structure and of a synchronization performed will for example be the occurrence of the transition between the last SYNCP symbol of the preamble and the symbol SYNCM.
  • This indication IND will be transmitted to the additional processing means MTRS of the processing stage ET2 so as to allow the decoding of the symbols of the header HD and the field PLD of the frame.
  • the processing stage ET2 comprises MVRF verification means configured to check the presence or absence of the noise signal in the useful signal, once the synchronization is performed.
  • this verification will be performed on the preamble of the unfiltered SNC channel digital signal, and more particularly on one of the symbols of the preamble, for example the unfiltered SYNCP symbol.
  • a direct fast Fourier transform FFT is carried out in a step 86 so as to carry out a transformation of the time domain towards the frequency domain, and then (step 91) a power analysis is carried out on the lines of the frequency spectrum obtained at the output of the Fourier transform.
  • step 92 it is examined in step 92 whether certain frequency lines have a power or a level greater than a fixed threshold TH.
  • the verification means MVRF comprise MTFD means configured to perform the direct Fourier transform processing as well as MAL analysis means.
  • these means can again be made for example in a software manner within a microprocessor.
  • the MTFD means are advantageously those already present in the additional processing means MTRS.
  • control means embodied herein by way of illustration, by a multiplexer MUX controlled by a signal SC delivered by a control module MC connected at the output of the verification means MVRF and representative of the presence or absence of the noise signal, will enable or disable the MFL filter means for further processing of the remaining portion of the frame.
  • the control module can be made for example by a logic circuit or in a software way.
  • the digital signal of the SNC channel is delivered directly to the additional processing means MTRS while in the presence of the noise signal SB, it is the signal filtered digital SNF which is delivered to additional processing means MTRS.
  • these complementary processing means MTRS comprise means MCPR configured to remove from each symbol the cyclic prefix, followed by means MTFD configured to perform the FFT fast Fourier transform.
  • the MTFD means are followed by demapping means DMP (demapping means) providing for each carrier a value of the corresponding modulation coefficient (bin).
  • DMP demapping means
  • MCSM MCSM module configured to determine for each modulation coefficient an indication of confidence (soft decision: "soft decision") of said value.
  • This module is conventional and known per se and uses for example an algorithm of the LogMAP type.
  • the additional MTRS processing means also comprise DINTL deinterleaving means followed by a DCD decoder, for example a Viterbi type decoder, followed by CRC means able to carry out a parity check.
  • the output of the CRC means is connected to the output terminal BS of the MTRS means which is connected to the means forming the MAC layer of the receiver.
  • the additional processing means MTRS receive at the input the filtered digital signal, that is to say in the presence of a noise signal on the digital channel channel signal, it is particularly advantageous that the indications of confidence (soft decision ) associated with the bins on which the noise signal is present as well as those possibly associated with neighboring bins, are set to zero. Indeed, such null soft decisions are seen as being neutral decisions for the error correction algorithm implemented in the Viterbi decoder.
  • the MTRF processing means comprise MFC forcing means configured to perform this forcing to zero.
  • all the means and modules of the additional processing means MTRS can be realized by software modules within a microprocessor.
  • the decoding of this remaining part of the frame it is preferable, for the decoding of this remaining part of the frame, to freeze the coefficients of the FIR filter c that is to say, not to update them as and decoding the remaining bet of the frame.
  • control module MC can deliver a gel signal SC1 to the calculation module MCL of the filter coefficients.

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  • Engineering & Computer Science (AREA)
  • Signal Processing (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Power Engineering (AREA)
  • Physics & Mathematics (AREA)
  • Spectroscopy & Molecular Physics (AREA)
  • Noise Elimination (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
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EP3073695B1 (de) 2018-01-24
US20170288918A1 (en) 2017-10-05
CN106027439A (zh) 2016-10-12
CN110224721A (zh) 2019-09-10
FR3034274A1 (fr) 2016-09-30
US9729199B2 (en) 2017-08-08
CN106027439B (zh) 2019-07-23
CN110224721B (zh) 2021-06-04
CN205249269U (zh) 2016-05-18
US20160285509A1 (en) 2016-09-29
US10050672B2 (en) 2018-08-14

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