EP2892137B1 - Circuit de démarrage d'alimentation à découpage présentant normalement sur la source de courant commandée - Google Patents

Circuit de démarrage d'alimentation à découpage présentant normalement sur la source de courant commandée Download PDF

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Publication number
EP2892137B1
EP2892137B1 EP15150212.7A EP15150212A EP2892137B1 EP 2892137 B1 EP2892137 B1 EP 2892137B1 EP 15150212 A EP15150212 A EP 15150212A EP 2892137 B1 EP2892137 B1 EP 2892137B1
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Prior art keywords
voltage
startup
power supply
current
switching power
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German (de)
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EP2892137A1 (fr
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Mark Jutras
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Bel Fuse Macao Commercial Offshore Ltd
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Bel Fuse Macao Commercial Offshore Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/36—Means for starting or stopping converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0006—Arrangements for supplying an adequate voltage to the control circuit of converters

Definitions

  • startup circuit When input power is initially applied to a switching power supply, it is necessary to deliver energy to control circuits for the purpose of starting operation.
  • the circuitry used for this purpose is commonly referred to as the "startup circuit". Because the power supply is not yet generating output power during startup, the startup circuit obtains power from the input in some manner.
  • One of the challenges in a switching power supply is providing the initial energy to power the control circuitry when the operating voltage limits for the control devices are far below the input voltage range of the power supply. This problem exists for DC/DC converters that receive input power from 24V or 48V nominal DC sources, for example. This is also a consideration in AC/DC power supplies that operate from input voltages that can be as high as 305 VAC and in typical applications range from 90 VAC to 264 VAC.
  • a capacitor In order to start a switching power supply when the input voltage is large relative to the normal operating voltage of primary-side-referenced control circuits, a capacitor can be charged from the input source through either a resistor or a current source. A circuit monitors the capacitor voltage and turns on the control circuitry when the capacitor voltage reaches approximately the upper operating voltage of that circuitry. The control circuitry once enabled draws energy from the charged capacitor. When startup completes and normal operation commences, the normal operating circuitry of the power converter can supply the energy required to maintain the voltage across this startup capacitor to a value within the operating limits of the control circuitry.
  • the circuitry that monitors the startup capacitor voltage and turns on the control circuitry may require only a small current (e.g., 500uA to 1mA) during startup, and the resistor or current source that feeds the startup capacitor is sized to provide this current at a minimum input voltage that is present during startup.
  • This sizing can result in much higher power dissipation by the circuitry at the larger maximum input voltage occurring during normal operation.
  • this higher power dissipation occurs irrespective of the output loading of the power converter, i.e., whether the converter is online and providing power to a load or is offline or "standby" and not providing power to a load.
  • US 2012/163047 describes a startup circuit which includes a MOSFET that is connected between a startup power source and an auxiliary power source, and a JFET that has a drain terminal connected to a drain terminal of the MOSFET.
  • a pinch-off voltage controller controls a pinch-off voltage of the JFET.
  • the startup circuit is capable of keeping the pinch-off state of the JFET with a minimum power loss.
  • US 2013/127431 describes a startup circuit which delivers regulated startup current to a control integrated circuit in a switch mode power supply system.
  • the startup circuit automatically disconnects the startup current when the switched mode power supply system Control IC starts switching the transformer or inductor used as the energy storage element in the SMPS system.
  • Offline power supplies are subject to market or regulatory requirements that place limits on no-load power loss. Thus, it is desirable to reduce no-load power losses as much as possible, including losses attributable to startup circuitry such as described above.
  • a switching power supply comprising: a storage capacitor (Caux) coupled to a power magnetic element (T1) to maintain a steady state value of an auxiliary voltage (Vaux) during steady state operation of the switching power supply, the steady state operation being preceded by a startup period in which an input-side DC voltage (Vin) of the switching power supply rises from zero toward a steady state operating value; switching and control circuitry configured to couple energy from the input-side DC voltage to the power magnetic element during the steady state operation, the switching and control circuitry including control circuitry powered by the auxiliary voltage; and a startup circuit configured and operative during the startup period to generate a startup value of the auxiliary voltage to enable the control circuitry to establish the steady state operation, the startup circuit including (i) a startup current source coupled between the input-side DC voltage and the storage capacitor to provide charging current (Ic(Q1)) thereto based on the absence of an inhibitory control signal (Vg(Q3)), the startup current source including an emitter-switched current
  • a disclosed power converter includes a startup circuit that achieves reduced no-load power loss while providing the desired function of establishing an operating voltage for control circuitry during a startup period.
  • the startup circuit has a normally on characteristic such that it automatically provides startup charging current for a startup capacitor once the input voltage has risen sufficiently high, without operation of any separately powered control circuitry.
  • the control circuitry begins operating as the startup capacitor voltage reaches an operating value, and it generates an inhibitory signal that disables the startup circuit, stopping the flow of the startup charging current and reducing the power dissipation of the startup circuit to a desired low value.
  • the normally on characteristic is achieved through use of an emitter switched current source employing a normally on switching device, such as a depletion-mode junction field-effect transistor (J-FET).
  • J-FET depletion-mode junction field-effect transistor
  • Such a device has a source-drain channel that conducts current in the absence of a control voltage on the gate of the device, and this feature is exploited for use during startup when control circuitry is not yet operating. Additionally, the source-drain channel can be cutoff by application of a sufficiently high control or bias voltage to the gate of the device, and this feature is exploited for use after startup to prevent current flow and reduce power dissipation by the startup circuit during subsequent normal operation.
  • a J-FET is a good candidate as a switch device used to enable and disable a startup current source.
  • a J-FET device can be used to establish the flow of startup capacitor charging current.
  • a low voltage from the control circuitry can be applied to the gate of the J-FET to turn it off and disable the charging current.
  • One issue with readily available J-FETs, however, is relatively low voltage rating (e.g., less than 50V) which is far below the typical voltages that need to be switched for many startup current source applications.
  • a J-FET is used as a switch in of an emitter-switched current source employing a bipolar transistor as the main current-control element. This configuration reduces the voltages experienced by the J-FET even in higher-voltage applications.
  • Another aspect of a disclosed circuit is a convenient dual use of a resistive divider network both to bias the emitter-switched current source and to provide a sufficiently low sensing voltage proportional to the input voltage for use by the control circuitry during normal operation. Since the input voltage is typically in excess of the voltage rating of the control circuit elements a voltage divider is used to generate a signal that is proportional to the input voltage but scaled to a range that does not exceed the control circuit voltage limits. The resistors that are used to derive this circuit element are sized to have an insignificant impact on no load power loss. Consequentially the divided down input voltage used as a monitoring signal is normally far below the voltage ratings of available J-FETs.
  • emitter switching provides the advantage of using a depletion mode device as a switch in the startup current source without exceeding the voltage limitations of readily available, low cost, discrete J-FET devices.
  • a disclosed startup circuit combines the resistor divider that generates a monitored signal proportional to the input voltage with an emitter-switched startup current source using a depletion mode J-FET in series with the emitter of a higher voltage NPN transistor.
  • the circuit provides an input voltage monitoring signal during steady state operation, while also disengaging the startup current source without additional circuit losses.
  • the emitter-switched bipolar transistor could alternatively be controlled with a series MOSFET or a series bipolar transistor in other embodiments.
  • FIG. 1 is a schematic block diagram of a portion of a switching power supply. It includes switching and control (SW/CNTL) circuitry 10, a power transformer T1, a startup circuit 12, and optionally an input voltage source 14.
  • the transformer T1 has main primary and secondary windings Wpri and Wsec, as well as a third or "auxiliary" winding Waux connected to a capacitor Caux.
  • the switching and control circuitry 10 receives a DC voltage Vin as well as a voltage Vaux developed on the capacitor Caux.
  • the voltage Vin is provided by the input voltage source 14 when present, and otherwise it may be an input from a separate voltage source.
  • Pertinent operation of the power supply is divided into two periods, an initial startup period in which Vin is rising from zero to a normal operating value, and a subsequent steady-state operating period in which Vin is at its normal operating value and the power supply is providing a steady DC output voltage to separate powered circuitry (not shown).
  • the switching and control circuitry 10 includes circuitry (not shown in Figure 1 ) that receives its operating power from the Vaux input; examples are described below.
  • the combination of the winding Waux and capacitor Caux function as a simple power source for this circuitry.
  • no or little current is provided to the main primary winding Wpri and therefore no or little power is available via the winding Waux.
  • the startup circuit 12 operates during this period along with Caux as the power source, until operation has proceeded to the point that the normal steady-state mechanism employing winding Waux is available and becomes operative.
  • FIG 2 shows the startup circuit 12 according to one embodiment. Its main purpose is to generate an unregulated supply voltage Vaux usable by the switching and control circuitry 10 ( Figure 1 ) during an initial startup period of operation before all normal operating voltages have been established. Vaux is generated by supplying a charging current Ic(Q1) to the capacitor Caux, which occurs in response to another current Ic(Q2) that flows during an initial part of the startup period. Detailed operation is described below.
  • a normally on transistor Q3 which may be implemented as a depletion-mode junction FET (J-FET) for example.
  • J-FET depletion-mode junction FET
  • Q3 conducts during startup to allow generation of Ic(Q2), and at the end of startup it is rendered non-conducting by application of an inhibitory control signal in the form of a positive gate voltage Vg(Q3) from a Vg generator (Vg GEN) 22.
  • Vg GEN Vg generator
  • the transistors Q1 - Q3 and related circuitry forms a startup current source that pulls power from the input source to generate the charging current Ic(Q1) for the storage capacitor Caux.
  • the startup current source includes two sub-level current sources - an emitter-switched current source formed by Q2, Q3 and related circuitry that generates Ic(Q2), and a second current source (referred to as an output current source) that responds to Ic(Q2) to generate the charging current Ic(Q1).
  • the current Ic(Q2) may be seen as an enabling current that enables Q1 to conduct the charging current Ic(Q1).
  • Another feature of the startup circuit 12 is the ability to measure the input voltage Vin by sensing the voltage at the junction of a resistive divider circuit formed by the resistors R2 and R3. This voltage is shown as K*Vin, where K is equal to R2/(R2+R3). It will be appreciated that this relationship does not actually hold during startup when Q2 is conducting base current through R3. However, at the end of startup when Q3 is shut off, Q2 also stops conducting and its base current drops to a very small parasitic value. Assuming sufficiently small values for R2 and R3, this base current is swamped by the current through R2 and R3, and the above relationship is valid.
  • Q3 is a P-Channel depletion mode J-FET.
  • a depletion mode FET is on (conducting) when zero volts is applied to its gate, and is turned off when a voltage in excess of a cutoff voltage is applied to its gate.
  • Vin is equal to zero
  • Vg(Q3) has zero volts applied and Q3 behaves as if it were a resistor connected from the emitter of Q2 to the return potential.
  • Vg(Q3) Once the voltage on the base of Q2 becomes high enough to establish current flow through Q2's base-emitter junction, begins conducting. This will establish current flow through a voltage-creating (V-C) element 20 connected between Vin and the base of Q1.
  • V-C voltage-creating
  • collector current flows in Q1.
  • This collector current is proportional to the voltage across Re(Q1), which is equal to the voltage across the V-C element 20 minus the base-emitter voltage drop (V BE ) for the conducting Q1.
  • the Q1 collector current Ic(Q1) flows in a path that allows it to charge Caux.
  • the voltage divider consisting of R2 and R3 puts a voltage on the base of Q2 that is proportional to the input voltage Vin when Q3 is off.
  • Q3 When Q3 is conducting this same voltage is clamped to a maximum value of the V BE voltage of conducting Q2 plus the voltage drop across conducting Q3.
  • the voltage on the base of Q2 For Q2 to turn on when Q3 is on, the voltage on the base of Q2 must be higher than the VBE voltage for Q2.
  • Q2's base voltage must also be sufficient to provide the voltage across Q3 required to maintain the desired Ic(Q2) current at the maximum Q3 on resistance.
  • a minimum divided down voltage of approximately 1V will satisfy the ability to properly turn on Q2 and provide desired Ic(Q2) current under most practical applications of this circuit.
  • K*Vin is only proportional to Vin when Q3 is turned off and no Q2 base-emitter current is flowing. This will be the condition after power supply startup is established since Q3 will be turned off during steady state operation. Thus, K*Vin can be used as a proxy for Vin during steady state operation.
  • a bipolar transistor or an enhancement mode MOSFET could alternatively be used as the switching transistor Q3 to obtain the benefits of the emitter-switched configuration.
  • the J-FET implementation is a practical choice because of its normally on characteristic (conducting in the absence of gate voltage). With the addition of a voltage divider from Vin to the base of a BJT or to the gate of a MOSFET the emitter-switched implementation could also be configured around those devices. Such implementations might be useful in applications having lower input voltages.
  • Figure 3 shows an alternative embodiment in which a voltage divider is established with R3 and R1//R2 that puts a voltage on the base of Q2 that is much lower than Vin and is biased up by a diode drop established by diode D3.
  • D3 is used in the case that Vin is divided to a value less than a Q2 VBE voltage drop at the minimum operating input voltage.
  • This arrangement is used to satisfy two competing design goals. First, to limit steady state power losses in offline AC/DC applications, it may be desirable to use a value for R3 in excess of a few mega-ohms. However, the signal K*Vin is supplied to an A/D converter channel whose input impedance may be less than 10K ohms for accurate sensing.
  • D3 is intended to increase the voltage on the base of Q2 to a workable level, while D4 subtracts the error introduced by D1 from the input voltage monitoring signal K*Vin.
  • the added diode drop on the base of Q2 will be sufficient to turn Q2 on when the emitter of Q2 is essentially connected to the return potential thru Q3.
  • the voltage across R2 will be a diode drop below the voltage on the base of Q2, and the voltage K*Vin across R2 is essentially proportional to Vin when Q3 is tuned off and no Q2 base-emitter current is flowing. This is the intended condition after power supply startup is established.
  • Figure 4 shows three different types of input voltage source 14 with which the startup circuit of Figure 2 or Figure 3 can be used. Because of the use of the normally on device Q3 and the attendant lack of a need for a bias voltage, the startup circuit 12 is able to produce startup current through a significant portion of a rectified sine wave.
  • the configuration of Figure 4(a) is applicable in a power supply that employs active power factor correction (PFC), so the startup circuit can provide startup voltage for a controller used for active PFC control.
  • the auxiliary winding Waux can be a winding on a boost choke or a winding on a transformer of the switching power supply that the PFC circuit feeds.
  • the configuration of Figure 4(b) is a DC source generated by rectification of an AC voltage input and a hold up capacitor.
  • the configuration of Figure 4(c) is just a DC source.
  • Figure 5 shows different types of voltage creating (V-C) elements 20 and surrounding circuitry that may be used in conjunction with Re(Q1) to determine the value of Q1's collector current.
  • Figure 5(a) shows a simple resistor, which is convenient and inexpensive but has the disadvantage of producing a voltage drop and thus a startup current that increases with Vin. This would result in a startup time that decreases with input voltage.
  • Figure 5(b) shows a negative-temperature-coefficient (NTC) resistor whose resistance decreases in value with an increase in temperature. This would be a possible consideration if it were desired to lower the startup current at higher temperatures in order to reduce stress.
  • NTC negative-temperature-coefficient
  • an NTC resistor or device is one whose pertinent characteristic (e.g., resistance) varies significantly as a function of temperature, and in particular as an inverse function of temperature.
  • a typical NTC coefficient might be on the order of -1% to - 10%, for example, meaning that the resistance decreases by that proportional amount per degree C (e.g., a 10K resistance is reduced to 9.0K for a 10-degree temperature rise when the NTC is -1%).
  • a PTC resistor or device has a coefficient of similar magnitude but opposite sign, so its resistance increases correspondingly with temperature. This is in contrast to regular resistors or devices that exhibit much lower degrees of temperature dependence.
  • a typical thin-film resistor might have a temperature coefficient on the order of 10 -4 that provides little or no observable effect on circuit operation over a normal operating temperature range.
  • NTC and PTC devices respond much more dramatically to temperature changes to cause more pronounced and desired changes to circuit operation.
  • Zener diode can be used as shown in Figure 5(c) .
  • one of the potential pitfalls of a Zener diode is the current required to get a predictable breakdown voltage across the device.
  • Most Zener diodes require lma to 5ma of current to establish a predictable clamp voltage. That would require Rc(Q2) to be relatively small and as a result this resistor will be subjected to excessive power loss stress when conducting current and thus would need to be sized properly. This may not be an issue with this configuration because this would be a pulsed current event since the goal of the circuit is to terminate that current flow under steady state conditions.
  • a voltage reference such as a Texas Instruments TLV431 could be used - two variants of this alternative are shown in Figure 5(d) .
  • a device equivalent to a TLV431 only requires 100uA of cathode current to establish a predictable voltage drop reducing the burden on Rc(Q2). If precise startup current is required, adding a diode in series the voltage creating element will cancel Q1's VBE drop placing a more precise representation of the voltage creating element across Re(Q1).
  • Figure 5(e) shows two variants of this configuration.
  • a circuit as simple as two or more series diodes, as shown in Figure 5(f) could also be used as a voltage creating element 20. The first diode creates a voltage drop to cancel Q1's VBE while the additional diode(s) result in a net voltage across Re(Q1) creating current flow.
  • Figure 6 shows two alternative implementations for the resistor Re(Q1), which is the resistive element that provides the emitter current path for Q1.
  • Q1's collector current is approximately equal to the voltage applied across this resistor divided by its resistance value.
  • Figure 6(a) shows a simple resistor - a good choice for some implementations.
  • An alternative shown in Figure 6(b) is a resistor with a positive temperature coefficient (PTC), to provide a startup current that decreases with temperature but is essentially constant relative to varying input voltage a PTC resistor could be used as this element. In that case the PTC would be applied in conjunction with a configuration from one of Figures 5(d), 5(e) and 5(f) for the V-C element 20.
  • PTC positive temperature coefficient
  • Rc(Q1) is used to distribute the power loss in the Ic(Q1) path when Q1 is enabled.
  • Rc(Q1) also limits current in that path in the event that Q1 fails in a short-circuiting manner, to prevent catastrophic failure of the components connected across Caux.
  • Rc(Q1), Rc(Q2), and R3 are shown as single resistive elements in Figure 2 . Under certain applications Vin may be such that the voltage drop across these elements is in excess of their voltage rating. In such situations these resistive elements can be replaced with multiple resistors in series in order to distribute the dropped voltages such that no single resistor is operated in excess of its voltage rating. Using multiple series resistors may also be a consideration relative to power dissipation or pulse power ratings.
  • Figure 7 shows some alternative ways in which the control signal Vg(Q3) of Figure 2 can be generated, specifically some practical methods for generating the signal that each turn Q3 off after Vaux reaches a desired voltage.
  • the arrangement of Figure 7(a) is a resistor voltage divider that will establish a voltage on the gate of Q3 that begins to cutoff Q3 current flow as Caux charges.
  • This is a simple circuit that will work in some implementations but has the disadvantages of no hysteresis and also creates an analog cutoff rather than a digitally switched cut off. With the analog cut off Q3 will not be abruptly turned off but rather slowly turned off as increasing voltage on the gate of Q3 pinches off the junction as the voltage across Caux increases. Both of those disadvantages can be corrected by using the comparator configuration shown in Figure 7(b) .
  • a microcontroller provides the ability to generate optimum control of the turn off signal, and can be configured for this purpose as shown in Figure 7(c) (microcontroller shown as "PIC12F617").
  • PIC12F617 microcontroller shown as "PIC12F617"
  • the microcontroller can manage the turn off for Q3.
  • a voltage K*Vin proportional to the startup voltage is monitored by either an analog to digital converter or a comparator within the microcontroller.
  • a software algorithm is then used to decide when to terminate the startup current source by applying voltage to the gate of Q3 through a digital output.
  • a traditional power supply control IC with start-up hysteresis can be used to turn off Q3.
  • a Texas Instruments controller UC3842 is shown receiving power from the energy stored in Caux. When Vaux reaches a voltage that turns on the UC3842, its reference pin will jump to 5V and this can be used as the voltage to turn off Q3.
  • FIG 9 depicts operation of the startup circuit when an overload fault occurs on one of the outputs of the switching power supply controlled by a circuit powered from Vaux.
  • An additional signal, Verror is drawn on this graph.
  • Verror is the error voltage for the switching power supply and is used to set operating duty cycle to maintain the desired output voltage.
  • the control of the power supply is digital rather than analog there is an equivalent operating parameter that exists in the control software that can be queried for similar behavior.
  • this signal is generated by comparing the controlled output voltage to a reference with an amplifier that has sufficient gain.
  • the digital control approach it is a calculated parameter.
  • this signal is coupled to the primary side through an isolation device such as an opto-coupler.
  • One advantage of using the disclosed startup circuit is the ease of turning on and off the startup current source to reduce power dissipation when it is not needed.
  • Q3 is controlled with a microcontroller the ease of turning on and off the startup current source can be used as an advantage in solving a problem that commonly occurs with regulated power supplies that employ an auxiliary winding to generate Vaux during steady state operation.
  • This auxiliary winding is not regulated but rather coupled to a winding that produces a regulated voltage. Since the auxiliary winding is on the primary side and the regulated winding is typically on the secondary side safety requirements force the transformer (or coupled inductor) construction to be such that the coupling between these windings is compromised.
  • the poor coupling can be such that the auxiliary winding is not capable of providing the energy required to keep Vaux above the minimum operating voltage of the control circuitry. This can be solved with the Figure 2 circuit when it is controlled by a microcontroller that also monitors the error voltage used to set regulation.
  • a mode of operation that can be executed by a microcontroller is shown.
  • the microcontroller can run software that distinguishes between allowing the mode of operation shown in Figure 9 or forcing the mode of operation shown in Figure 10 based on the relationship between Verr and Vaux.
  • the mode of operation in Figure 10 prevents Vaux from falling below a minimum value when the power supply should be sustaining its output. These can be determined by using Verror as a proxy as to whether the regulated output is in regulation which is the case when Vaux is not at its saturation level.
  • the microcontroller can control the Figure 2 circuit to behave as a repetitively pulsed current source to keep Vaux above a minimum threshold. This is not the primary intended use of the disclosed startup circuit but rather another possible operating mode.
  • FIG 11 is a detailed schematic diagram of an example power supply using a startup circuit in a way that provides a useful monitoring of Vin and takes advantage of a switched current source turn off to limit quiescent power loss after startup.
  • the startup circuit shown primarily in part 11B of Figure 11 , is implemented using one set of the multiple options described above.

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Claims (14)

  1. Alimentation à découpage, comprenant :
    un condensateur de stockage (Caux) couplé à un élément électromagnétique (T1) pour maintenir une valeur de régime permanent d'une tension auxiliaire (Vaux) au cour d'un fonctionnement en régime permanent de l'alimentation à découpage, le fonctionnement en régime permanent étant précédé d'une période de démarrage au cours de laquelle une tension CC côté entrée (Vin) de l'alimentation à découpage passe de zéro vers une valeur de fonctionnement en régime permanent ;
    un circuit de découpage et de contrôle (10), configuré pour coupler l'énergie de la tension CC côté entrée à l'élément électromagnétique au cours du fonctionnement en régime permanent, le circuit de découpage et de contrôle comportant un circuit de contrôle alimenté par la tension auxiliaire ; et
    un circuit de démarrage (12), configuré et opérationnel pendant la période de démarrage pour générer une valeur de démarrage de la tension auxiliaire pour permettre au circuit de contrôle d'établir le fonctionnement en régime permanent, le circuit de démarrage comportant(i) une source de courant de démarrage couplée entre la tension CC côté entrée et le condensateur de stockage pour y fournir un courant de charge (Ic(Q1)) sur la base de l'absence d'un signal de contrôle inhibiteur (Vg(Q3)), la source de courant de démarrage caractérisée par une source de courant commutée par émetteur (Q2, Q3) dotée d'un transistor de commutation normalement passant (Q3) qui, le signal de contrôle inhibiteur y étant appliqué, est tel qu'il conduit un courant de validation (Ic(Q2)) en l'absence du signal de contrôle inhibiteur et ne conduit pas le courant d'activation en présence du signal de contrôle inhibiteur, le courant de validation contrôlant la transmission du courant de charge par la source de courant de démarrage, et comportant (ii) un générateur (22) du signal de contrôle inhibiteur, le générateur étant configuré et opérationnel au cours de la période de démarrage pour générer le signal de contrôle inhibiteur en réponse à l'atteinte de la valeur de démarrage par la tension auxiliaire sous l'effet de l'action de charge de la source de courant de démarrage et pour maintenir le signal de contrôle inhibiteur lors d'un fonctionnement ultérieur en régime permanent,
    dans lequel la source de courant de démarrage inclut une source de courant de sortie (Q1) qui génère le courant de charge en réponse au courant de validation, et dans lequel la source de courant commutée par émetteur comporte un transistor bipolaire (Q2) doté d'un émetteur disposé en série avec le transistor normalement passant, le transistor bipolaire conduisant le courant de validation sous forme d'un courant collecteur en l'absence du signal de contrôle inhibiteur lorsque le transistor normalement passant est aussi en état de conduction, dans lequel le transistor bipolaire cesse de conduire en présence du signal de contrôle inhibiteur quand le transistor normalement passant n'est pas conducteur.
  2. Alimentation à découpage selon la revendication 1, dans lequel la source de courant de sortie inclut un deuxième transistor bipolaire (Q1) comportant une jonction base-émetteur polarisée dans le sens direct par le courant de validation pour permettre au deuxième transistor bipolaire de conduire le courant de charge sous forme de courant collecteur.
  3. Alimentation à découpage selon la revendication 2, dans lequel la source de courant de sortie inclut une résistance d'émetteur (Rc(Q1) sur un émetteur du deuxième transistor bipolaire et un élément créateur de tension (20) sur une base du deuxième transistor bipolaire réagissant au courant de validation pour établir la polarisation dans le sens direct de la jonction base-émetteur du deuxième transistor bipolaire, l'élément de création de tension et la résistance d'émetteur ayant un rapport prédéterminé établissant une valeur prédéterminée souhaitée du courant de charge.
  4. Alimentation à découpage selon la revendication 3, dans lequel l'élément de création de tension comprend soit (i) une résistance à coefficient de température négatif, configurée pour réduire la valeur du courant de charge en fonction d'une hausse de température, soit (ii) une résistance à coefficient de température positif pour réduire la valeur du courant de charge en fonction d'une hausse de température.
  5. Alimentation à découpage selon la revendication 1, dans lequel le circuit de démarrage inclut un réseau diviseur à résistance comportant un noeud intermédiaire couplé à une base du transistor bipolaire pour contrôler la conduction du courant de validation, le diviseur à résistance relié entre la tension d'entrée et un noeud de référence et générant un signal de détection de tension sur le noeud intermédiaire, et dans lequel le circuit de contrôle est configuré et opérationnel pour utiliser le signal de détection de tension pour qu'il représente la tension CC côté entrée dans le contrôle du fonctionnement du circuit de découpage et de contrôle.
  6. Alimentation à découpage selon la revendication 5, dans lequel le réseau diviseur à résistances inclut une thermistance dont la résistance varie avec la température, la thermistance étant configurée et agencée de sorte à contrôler la polarisation de la source de courant commutée par émetteur pour réduire un niveau du courant de charge en fonction de la hausse de température.
  7. Alimentation à découpage selon la revendication 1, dans lequel le transistor normalement passant est un transistor à effet de champ et à appauvrissement (Q3), le signal de contrôle inhibiteur étant appliqué à sa grille.
  8. Alimentation à découpage selon la revendication 1, dans lequel le générateur est soit (i) un circuit passif configuré et opérationnel pour générer un signal de tension analogique qui s'accroît au cours de la période de démarrage et est considéré être le signal de contrôle inhibiteur lorsque la tension auxiliaire atteint la valeur de démarrage, soit (ii) un circuit actif configuré et opérationnel pour générer un signal de tension binaire devenant à la fin de la période de démarrage une première tension binaire considérée être le signal de contrôle inhibiteur.
  9. Alimentation à découpage selon la revendication 8, dans lequel le circuit actif est doté d'un microcontrôleur.
  10. Alimentation à découpage selon la revendication 1, comprenant en outre un circuit de redressement (10) configuré et opérationnel pour générer la tension CC côté entrée à partir de la tension de source CA qui y est fournie.
  11. Alimentation à découpage selon la revendication 1, comprenant en outre un transformateur (T1) comportant des enroulements primaire (Wpri) et secondaire (Wsec), l'enroulement primaire étant couplé pour recevoir la tension CC côté entrée, l'enroulement secondaire étant couplé à une sortie de la tension d'alimentation pour fournir une puissance de sortie à une tension de sortie CC, et dans lequel l'élément électromagnétique inclut un enroulement auxiliaire (Waux) du transformateur.
  12. Alimentation à découpage selon la revendication 1, dans lequel le générateur du signal de contrôle inhibiteur est réalisé par un microcontrôleur qui exécute une routine de traitement responsable de la génération du signal de contrôle inhibiteur au cours de la période de démarrage.
  13. Alimentation à découpage selon la revendication 1, dans lequel le circuit de découpage et de contrôle génère une tension d'erreur utilisée pour définir un rapport cyclique de fonctionnement pour maintenir la tension de sortie souhaitée, la tension d'erreur augmentant jusqu'à un niveau de saturation en cas de condition de surcharge, au cours de laquelle la tension de sortie chute en deçà d'une valeur de référence prédéterminée, et dans lequel la source de courant commutée par émetteur est contrôlée (a) de sorte à être maintenue dans une condition non conductrice jusqu'à ce soit atteint un creux souhaité de la tension auxiliaire après que l'alimentation à découpage a été coupée en raison de la condition de surcharge, et (b), au stade du creux souhaité, de sorte à devenir conductrice pour que l'énergie soit recouvrée dans le condensateur auxiliaire et pour que l'alimentation à découpage tente un démarrage normal.
  14. Alimentation à découpage selon la revendication 13, dans lequel la coupure et la tentative de démarrage de l'alimentation ont lieu à multiples reprises dans un mode de fonctionnement « hoquet », le mode « hoquet » étant un mode sélectionnable par un microcontrôleur qui génère le signal de contrôle inhibiteur, en exécutant une routine de traitement au cours de la période de démarrage, et dans lequel le microcontrôleur peut séparément sélectionner un mode « sans hoquet » dans lequel la tension auxiliaire ne peut atteindre la valeur de creux souhaitée dans des circonstances où l'alimentation doit soutenir sa propre tension de sortie, le mode « sans hoquet » étant effectué par pulsation répétée de la source de courant de démarrage pour maintenir la tension auxiliaire au-dessus d'un seuil prédéterminé et étant sélectionné selon que la tension d'erreur est ou n'est pas au niveau de saturation.
EP15150212.7A 2014-01-06 2015-01-06 Circuit de démarrage d'alimentation à découpage présentant normalement sur la source de courant commandée Active EP2892137B1 (fr)

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US201461923870P 2014-01-06 2014-01-06
US14/193,014 US9337720B2 (en) 2014-01-06 2014-02-28 Switching power supply startup circuit having normally on emitter-switched current source

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US9337720B2 (en) 2016-05-10
US20150194875A1 (en) 2015-07-09
CN104767370B (zh) 2018-06-05
CN104767370A (zh) 2015-07-08
EP2892137A1 (fr) 2015-07-08

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