EP1938084A2 - Dispositif de detection magnetique a diverses frequences de fonctionnement interne - Google Patents

Dispositif de detection magnetique a diverses frequences de fonctionnement interne

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Publication number
EP1938084A2
EP1938084A2 EP06809445A EP06809445A EP1938084A2 EP 1938084 A2 EP1938084 A2 EP 1938084A2 EP 06809445 A EP06809445 A EP 06809445A EP 06809445 A EP06809445 A EP 06809445A EP 1938084 A2 EP1938084 A2 EP 1938084A2
Authority
EP
European Patent Office
Prior art keywords
frequency
magnetic sensor
signal
phase
magnetic
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP06809445A
Other languages
German (de)
English (en)
Inventor
Josephus Arnoldus Henricus Maria Kahlman
Jeroen Veen
Bart Michiel De Boer
Theodorus Petrus Henricus Gerardus Jansen
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Koninklijke Philips NV
Original Assignee
Koninklijke Philips Electronics NV
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Koninklijke Philips Electronics NV filed Critical Koninklijke Philips Electronics NV
Priority to EP06809445A priority Critical patent/EP1938084A2/fr
Publication of EP1938084A2 publication Critical patent/EP1938084A2/fr
Withdrawn legal-status Critical Current

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Classifications

    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R33/00—Arrangements or instruments for measuring magnetic variables
    • G01R33/02—Measuring direction or magnitude of magnetic fields or magnetic flux
    • G01R33/06—Measuring direction or magnitude of magnetic fields or magnetic flux using galvano-magnetic devices
    • G01R33/09—Magnetoresistive devices
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B82—NANOTECHNOLOGY
    • B82Y—SPECIFIC USES OR APPLICATIONS OF NANOSTRUCTURES; MEASUREMENT OR ANALYSIS OF NANOSTRUCTURES; MANUFACTURE OR TREATMENT OF NANOSTRUCTURES
    • B82Y25/00—Nanomagnetism, e.g. magnetoimpedance, anisotropic magnetoresistance, giant magnetoresistance or tunneling magnetoresistance
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01N—INVESTIGATING OR ANALYSING MATERIALS BY DETERMINING THEIR CHEMICAL OR PHYSICAL PROPERTIES
    • G01N15/00—Investigating characteristics of particles; Investigating permeability, pore-volume or surface-area of porous materials
    • G01N15/06—Investigating concentration of particle suspensions
    • G01N15/0656—Investigating concentration of particle suspensions using electric, e.g. electrostatic methods or magnetic methods
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01N—INVESTIGATING OR ANALYSING MATERIALS BY DETERMINING THEIR CHEMICAL OR PHYSICAL PROPERTIES
    • G01N27/00—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means
    • G01N27/72—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means by investigating magnetic variables
    • G01N27/74—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means by investigating magnetic variables of fluids
    • G01N27/745—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means by investigating magnetic variables of fluids for detecting magnetic beads used in biochemical assays
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R33/00—Arrangements or instruments for measuring magnetic variables
    • G01R33/02—Measuring direction or magnitude of magnetic fields or magnetic flux
    • G01R33/06—Measuring direction or magnitude of magnetic fields or magnetic flux using galvano-magnetic devices
    • G01R33/09—Magnetoresistive devices
    • G01R33/093—Magnetoresistive devices using multilayer structures, e.g. giant magnetoresistance sensors
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R33/00—Arrangements or instruments for measuring magnetic variables
    • G01R33/12—Measuring magnetic properties of articles or specimens of solids or fluids
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R33/00—Arrangements or instruments for measuring magnetic variables
    • G01R33/12—Measuring magnetic properties of articles or specimens of solids or fluids
    • G01R33/1269—Measuring magnetic properties of articles or specimens of solids or fluids of molecules labeled with magnetic beads

Definitions

  • Magnetic sensor device with different internal operating frequencies
  • the invention relates to magnetic sensor devices comprising a magnetic field generator, a magnetic sensor element, and a detector module that are operated with different frequencies. Moreover, the invention relates to the use of such a magnetic sensor device and to methods for the detection of at least one magnetic particle with such magnetic sensor devices.
  • a microsensor device which may for example be used in a microfluidic biosensor for the detection of biological molecules labeled with magnetic beads.
  • the microsensor device is provided with an array of sensors comprising wires for the generation of a magnetic field and Giant Magneto Resistances (GMRs) for the detection of stray fields generated by magnetized beads.
  • the signal of the GMRs is then indicative of the number of the beads near the sensor.
  • the magnetic field is generated with a high first frequency fi in order to improve the signal-to-noise ratio (SNR) by avoiding 1/f noise
  • the GMR sensor is operated with an alternating current of a second frequency f 2 that allows to separate parasitic crosstalk components from the desired magnetic signal in the GMR output.
  • an input signal of a third frequency f 3 is needed for a demodulator that extracts the desired magnetic signal from the (amplified) GMR output.
  • a problem of such a magnetic sensor device is that phase noise in any of the signals with the frequencies fi, f 2 , and f 3 decreases the signal-to- noise ratio.
  • At least one magnetic field generator that is operated with an input signal (e.g. a current) of a first frequency fi and that is used for generating a magnetic field in an adjacent investigation region.
  • the magnetic field generator may for example be realized by a wire on a substrate of a microsensor.
  • At least one magnetic sensor element that is operated with an input signal (e.g. a current) of a second frequency f 2 and that is associated with the aforementioned magnetic field generator in the sense that it is in the reach of effects caused by the magnetic field of the magnetic field generator.
  • the magnetic sensor element may particularly be a magneto-resistive element of the kind described in the WO 2005/010543 A 1 or WO 2005/010542 A2, especially a GMR, a TMR (Tunnel Magneto Resistance), or an AMR (Anisotropic Magneto Resistance).
  • At least one detector module for example a demodulator, operated with an input signal of a third frequency f 3 for separating the desired signal component, which is related to the magnetic field generated by the magnetic field generator, in the output of the magnetic sensor element.
  • Said output and the input signal of the detector module may for example be multiplied to generate a DC component proportional to the desired signal.
  • a reference generator for generating a reference signal (e.g. a voltage or a current) with a reference frequency f ref .
  • a supply unit for deriving signals with the first frequency fi, the second frequency f 2 , and the third frequency f 3 from the aforementioned reference signal, the supply unit being coupled to the magnetic field generator, the magnetic sensor element, and the detector module for supplying them with the corresponding input signals.
  • An advantage of the described magnetic sensor device is that its supply unit derives all three required input signals with the different frequencies fi, f 2 , and f 3 from one common reference frequency f ref . Frequency and phase drifts between the three input signals are therefore minimized, thus improving the signal-to-noise ratio and the stability of the magnetic sensor device significantly.
  • the supply unit is preferably designed such that there is a predetermined phase relation between the signals derived by it.
  • the magnetic sensor device may particularly comprise a feedback control loop for controlling the supply unit in such a way that a predetermined phase relation is kept between at least two of the input signals of the magnetic field generator, the magnetic sensor element, and the detector module.
  • the feedback control loop is designed such that it keeps the predetermined phase relation between all said three input signals.
  • the input signals are by definition signals that prevail directly at the corresponding component (i.e. the magnetic field generator, magnetic sensor element, and detector module).
  • phase noise phase drift
  • the feedback control loop preferably comprises a phase detector for comparing the phases of two input signals. A difference in these phases can thus be detected and counteracted for by an appropriated control of the supply unit.
  • the supply unit comprises at least one digital frequency divider that is fed with the reference signal.
  • Digital frequency dividers are known in the state of the art in various embodiments. Their common feature is that they transform a signal with a frequency f ref at their input into a signal with a frequency f ou t at their output, wherein the output frequency f ou t is a fraction of the input frequency f ref .
  • Digital frequency dividers have the advantage that the frequency ratio as well as the phase shift between their input and their output is very stable and can easily be selected via external control lines.
  • the supply unit preferably comprises three such digital frequency dividers for generating all three frequencies fi, f 2 , and f 3 from the reference signal with frequency f ref .
  • the supply unit may optionally comprise a driver unit that is coupled to the digital frequency divider and that transforms the output of said frequency divider into a desired waveform.
  • the driver circuit may comprise a band-pass filter that eliminates high frequency and DC components from the output of the frequency divider.
  • the driver circuit may comprise a look-up table, a combinatorial network or a high-speed microprocessor comprising a digital sample of the desired waveform and a digital-to-analog (DA) converter to convert the digital sample into an analog waveform.
  • DA digital-to-analog
  • the invention comprises a magnetic sensor device with the following components (wherein similar comments as with respect to the first aspect of the invention apply for identical entities): a) at least one magnetic field generator operated with an input signal of a first frequency fi; b) at least one associated magnetic sensor element operated with an input signal of a second frequency f 2 ; c) at least one detector module operated with a model signal of a third frequency f 3 for selectively processing a desired signal component, which is related to the operation of the magnetic field generator, in the output of the magnetic sensor element; d) a tracking module for adjusting the model phase and/or the model amplitude of the model signal with respect to the phase of the desired signal component.
  • the tracking module is adapted to adjust the model phase and/or the model amplitude of the model signal via an optimization of a cost function that is determined from the desired signal component and the model signal.
  • a cost function that is determined from the desired signal component and the model signal.
  • the "model signal” often (but not necessarily) serves as a model or image of the (unknown) desired signal component.
  • the cost function may therefore particularly be constructed as a measure of the difference between the desired signal component and the model signal, for example the squared difference between the corresponding values integrated over a time interval.
  • the approaches of the first and the second aspect of the present invention may favorably be combined.
  • the usage of a common reference signal minimizes frequency and phase differences right at the source, while the tracking additionally compensates within the detector module any phase shifts that were introduced in the intermediate signal path.
  • the invention further relates to the use of the magnetic sensor devices described above for molecular diagnostics, biological sample analysis, or chemical sample analysis, particularly in body fluids (blood, saliva etc.) and cells.
  • Molecular diagnostics may for example be accomplished with the help of magnetic beads that are directly or indirectly attached to target molecules.
  • a third aspect of the invention relates to a method for the detection of at least one magnetic particle, for example a magnetic bead attached to a label molecule, the method comprising the following steps:
  • the method is characterized in that the mentioned input signals are derived from a common reference signal having a reference frequency f ref .
  • the method comprises in general form the steps that can be executed with a magnetic sensor device according to the first aspect of the invention as it was described above. Therefore, reference is made to the preceding description for more information on the details, advantages and improvements of that method.
  • the phase relation between the input signals are locked by a feedback control loop.
  • a fourth aspect of the invention relates to a method for the detection of at least one magnetic particle which comprises the following steps:
  • the method comprises in general form the steps that can be executed with a magnetic sensor device according to the second aspect of the invention as it was described above. Therefore, reference is made to the preceding description for more information on the details, advantages and improvements of that method.
  • the adjustment of the model phase and/or the model amplitude is done by an optimization of a cost function that is determined from the desired signal component and the model signal.
  • Figure 1 illustrates the principle of a biosensor with a magnetic sensor device according to the present invention
  • Figure 2 depicts a block diagram of the circuitry of a magnetic sensor device according to the present invention
  • Figure 3 shows a first, basic embodiment of a supply unit for the magnetic sensor device
  • Figure 4 shows a second embodiment of a supply unit that includes a phase feedback loop
  • Figure 5 illustrates the incorporation of the supply unit of Figure 4 into the circuit of Figure 3;
  • Figure 6 shows a third embodiment of a supply unit with a phase feedback loop
  • Figure 7 shows a fourth embodiment of a supply unit with digital means for a waveform generation
  • Figure 8 shows the general design of a detector module that comprises a tracking module for tracking the phase of a model signal used for the demodulation of the sensor output;
  • Figure 9 shows a particular realization of the tracking module in the design of Figure 8.
  • Figure 10 shows a variant of Figure 8, in which also the amplitude of the model signal is tracked
  • Figure 11 shows a particular realization of the tracking module in the design of Figure 10
  • Figure 12 shows a further variant of Figure 8, in which a cost function is used that is independent of the amplitude of the model signal;
  • Figure 13 shows schematically the cost function of Figure 12;
  • Figure 14 shows a particular realization of the tracking module in the design of Figure 12.
  • Like reference numbers in the Figures refer to identical or similar components.
  • Magneto-resistive biochips or biosensors have promising properties for bio-molecular diagnostics, in terms of sensitivity, specificity, integration, ease of use, and costs. Examples of such biochips are described in the WO 2003/054566, WO 2003/054523, WO 2005/010542 A2, WO 2005/010543 Al, and WO 2005/038911 Al, which are incorporated into the present application by reference.
  • FIG. 1 illustrates the principle of a single magnetic sensor device 10 for the detection of superparamagnetic beads 2.
  • a biosensor consisting of an array of (e.g. 100) such sensor devices 10 may be used to simultaneously measure the concentration of a large number of different biological target molecules 1 (e.g. protein, DNA, amino acids) in a solution (e.g. blood or saliva).
  • the so-called “sandwich assay” this is achieved by providing a binding surface 14 with first antibodies 3, to which the target molecules 1 may bind.
  • Superparamagnetic beads 2 carrying second antibodies may then attach to the bound target molecules 1.
  • a current flowing in the wires 11 and 13 of the sensor 10 generates a magnetic field B, which then magnetizes the superparamagnetic beads 2.
  • the stray field B' from the superparamagnetic beads 2 introduces an in-plane magnetization component in the Giant Magneto Resistance (GMR) 12 of the sensor device 10, which results in a measurable resistance change.
  • Giant Magneto Resistance GMR
  • FIG. 2 shows the schematic block diagram of a circuitry that can be used in connection with the magnetic sensor device 10 of Figure 1.
  • Said circuitry comprises a current source 22 that is coupled to the conductor wires 11, 13 to provide them with a generator current Ii .
  • the GMR 12 is coupled to a second current source 23 that provides the GMR 12 with a sensor current I 2 .
  • GMR 12 i.e. the voltage drop across its resistance, is sent via an optional high pass filter (capacitor 24), an amplifier 25, a demodulator 26, a low pass filter 27, and an analog-to-digital (AD) converter 28 to the output 30 of the sensor device for final processing (e.g. by a personal computer).
  • the demodulator 26 and the low pass filter 27 can be seen as one exemplary realization of a detector module 100 that selectively processes and/or separates a desired signal component in the (pre-processed) GMR output.
  • the generator current Ii is modulated with a first frequency fi
  • the sensor current I 2 is modulated with a second frequency f 2
  • the input to the demodulator 26 has a frequency f 3 .
  • the frequency fi of the magnetic excitation field is chosen such that the 1/f noise and instability regime of the GMR sensor 12 is avoided, for example f i > IMHz.
  • I 2 I 2 ,o-sin(2 ⁇ -f 2 t).
  • N is the number of magnetic beads 2 in the vicinity of the GMR 12
  • ⁇ is a proportionality factor
  • ⁇ is a constant related to the capacitive and inductive crosstalk between the wires 11, 13 and the GMR 12
  • ⁇ is a constant (the GMR resistance) related to the sensor voltage induced by the sensor current I 2 in the GMR 12.
  • the formula shows that multiplication of Ampl(t) with a signal cos 2 ⁇ -(fi ⁇ f 2 ) in the demodulator 26 will extract a DC signal proportional to the desired number N (i.e. the value for f 3 is fi+f 2 or fi-f 2 ).
  • a problem of the described approach is that phase noise on any of the input signals to the wires 11, 13, the GMR sensor 12, and the demodulator 26 at frequencies fi, f 2 , or f 3 decreases the detection SNR of the biosensor. Furthermore, as the received magnetic signal SNR may be low, frequency and phase locking of f 3 to the magnetic signal may introduce extra noise.
  • Generating said input signals of frequencies fi, f 2 , f 3 from Phase-Locked-Loop (PLL) circuitry requires three Voltage-Controlled- Oscillators (VCOs), which is complicated and difficult to integrate on an IC. Therefore, a magnetic sensor device 10 is required that has a high detection SNR, a high stability, and an easy adjustable (excitation) frequency fi, while being easy to realize in discrete components (minimal components) and on an IC.
  • the excitation-, sense- and detection input signals with frequencies fi, f 2 , and f 3 are derived in a supply unit 21 from the frequency f ref of a single reference generator 20, in such a way that the phase noise between said signals is minimized.
  • phase noise or frequency drift in the reference generator 20 does not affect the detection SNR.
  • low-bandwidth PLL or DLL (Delay-Locked-Loop) circuitry may be added to the supply unit 21 practically without SNR degradation to optimize only the phase and not the frequency of the signals; this measure is feasible because (1) the frequencies fi, f 2 , f 3 are well defined and (2) their phase relation varies only slowly, e.g. by temperature and component tolerances.
  • the amount of phase shifting components can be minimized by digitally generating the required waveforms, which avoids temperature and component tolerance dependent detector behavior and eases discrete and integrated implementation.
  • a first embodiment of a supply unit 21 comprising digital dividers and generating square wave signals is sketched in Figure 3.
  • the reference frequency f ref from the frequency generator 20 is divided by three synchronized digital frequency dividers 51, 52, and 53, which are realized by an M counter, P counter, and N counter, respectively.
  • the P counter 52 which generates the lowest frequency f 2 , synchronizes the phase of the two other frequency dividers 51 and 53.
  • N counter 53 By presetting the N counter 53 to 25, a 90-degrees phase shift is introduced for generating a cosine.
  • Band-pass filters may be added to remove DC and higher harmonics in the divider output signals (cf. components 61, 62, 63 in Figure 4). Due to unequal phase/delay in the band-pass filters and signal paths, the phase relation between the signals will deviate from optimal. This effect may be compensated by adjusting the counter preset mechanism accordingly, e.g. by changing the load values in the M and N counters.
  • FIG. 4 shows three drivers 61, 62, and 63 that are coupled to the counters 51, 52, and 53, respectively, for transforming their square wave outputs into other waveforms.
  • the drivers 61, 62, 63 typically comprise high order band-pass filtering means to generate non-square wave shaped signals, e.g. sine waves.
  • the optimum phase relation between the signals may vary due to temperature changes, drift and tolerances of electrical components.
  • drift of the reference frequency generator 20 may introduce un-equal phase shifts in the three driver blocks 61-63.
  • This effect may be compensated by adaptive feedback of said phase relation by using a Phase-Lock-Loop or Delay-Locked-Loop system and e.g. control the preset value of the counters 51-53.
  • the phase relations should be determined at a point as close as possible near the sensor, e.g. the phase corresponding to fi at the input of the field-generating wires 11, 13, the phase corresponding to f 2 across the GMR sensor 12, and the demodulation frequency f 3 at the input of the synchronous demodulator 26.
  • phase detectors PDl and PD2 are compared by phase detectors PDl and PD2, respectively, and adjusted to the optimal value by varying the "Preset" values of the accompanying dividers via loop filters LFl and LF2, respectively. It should be noted in this respect that a phase detector is assumed to determine a phase error with respect to the transitions of the lowest frequency if two different frequencies are compared.
  • FIG. 5 shows the integration of the supply unit 121 described above into a magnetic sensor device 10 according to Figure 2, wherein the counters 51-53 and the drivers 61-63 have been lumped together in one block.
  • the phase detector PDl compares the input signal of frequency f 3 right before the demodulator 26 with a signal 70 derived between amplifier 25 and demodulator 26.
  • a second embodiment may be chosen that uses the dotted connection 70' to frequency f 2 instead of line 70.
  • the phase detectors PDl and PD2 generate both zero for zero degrees phase difference. This may be avoided by using the third embodiment of a supply unit 221 shown in Figure 6.
  • both phase detectors PDl, PD2 output zero when the phase difference equals 90 degrees.
  • Figure 7 shows a fourth embodiment of a supply unit 321 with digital dividers 51-53 and a generation of non-square wave signals.
  • analog circuitry e.g. higher-order filters, waveform shaping non-linear circuits etc.
  • DACs digital-to-analog Converters
  • LUT 71, 72, 73 look-up tables
  • gates-arrays combinatorial networks
  • microprocessors generate the desired waveforms from the counter-bits.
  • a triangle wave may alternatively be generated without look-up tables; the resulting wave approaches a sine wave or may be converted to a sine wave by moderate order filtering.
  • low- order band-pass filters 91, 92, 93 may be added to remove DC and frequency components above half the reference frequency (the Nyquist frequency). As a consequence phase shifts due to temperature and component tolerances are avoided, so that phase adjusting means may be omitted.
  • reference frequency f ref By varying the reference frequency f ref , alternative excitation frequencies may be generated without the need to tune analog filters.
  • Figure 8 shows a principal sketch of a first kind of detector module 100.
  • the detector module 100 receives as one input the measurement signal from the GMR sensor, wherein it will be assumed in the following that, for example by a suited band pass filtering BPF, this input will effectively only comprise the "desired signal component" u(t) according to (cf. equation (I)):
  • a phase ⁇ was introduced in this formula to take account of an unknown, typically time- varying phase shift.
  • the task of the detector module 100 is to provide at its output the "model amplitude" A' that is an estimation for the amplitude A of the desired signal component u(t) according to equation (2).
  • the detector module 100 may perform its function by demodulating the desired signal component u(t) in a demodulator 26 with the basic model signal so(t) and a subsequent filtering in a low pass filter 27. In reality, there will however be a phase difference between the desired signal component u(t) and the basic model signal s o (t) due to the non-zero phase ⁇ ⁇ 0.
  • the DC component of the demodulator output is therefore directly proportional to the amplitude A (i.e. the amount N of beads one is interested in), but is also related to the phase difference ( ⁇ - ⁇ ) between the sensor signal u(t) and the model signal s( ⁇ ,t).
  • This phase difference changes with time and may in practice cause variations in the output signal A' of up to about 20%, which exceed the RMS voltage of the additive thermal noise in a 1 Hz bandwidth detection system. Obviously, phase variations give thus rise to uncorrectable measurement errors and degradation of the overall system performance.
  • the detector module 100 comprises a "tracking module" 200 which receives in the embodiment of Figure 8 as input: the desired signal component u(t), the phase shifted model signal s( ⁇ ,t), and the model amplitude A'.
  • Figure 9 shows a particular realization of the tracking module 200. This realization is based on a cost function P( ⁇ ,A') that is defined by the integrated squared "error signal" e( ⁇ , A', t), i.e. the difference between the desired signal component u(t) on the one hand side and the model signal s( ⁇ ,t) multiplied with the model amplitude A' on the other hand side:
  • phase tracking module 200 The implementation of this gradient-based phase tracking algorithm as a first order phase locked loop is depicted in Figure 9.
  • the tracking module 200 now comprises a second cost function Q(A', ⁇ ) that is, similar to the already introduced cost function P( ⁇ , A'), used for a gradient descent, but now with respect to the variable A'.
  • the model amplitude A' that is managed and provided by a unit 103 will then directly contain the output result of the detector module 100.
  • the updated model amplitude A' new is calculated according to
  • the left hand part of the tracking module in Figure 11, which calculates the model phase update ⁇ , is the same as in Figure 9 (besides a factor A').
  • model signal A'-s( ⁇ , t) accurately approximates the sensor signal u(t).
  • model signal s( ⁇ , t) and the sensor signal u(t) can also be synchronized by maximizing the model amplitude A' at the output of the detector module 100 that is calculated by the demodulation of the sensor signal u(t) with the model signal s( ⁇ , t). According to equation (3), this model amplitude can be expressed as
  • Figure 12 shows the general layout of a detector module 100 for the aforementioned approach, while Figure 14 depicts the associated tracking module 200.
  • the invention is however not limited to the embodiments explicitly mentioned. Alternative dividing ratios, signal shapes (square, triangle, sine etc.), frequencies and combinations of the shown embodiments are part of the invention. Furthermore this invention may be particularly used to detect bio-chemical molecules in blood, saliva, in body fluids and in cells.

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  • Measuring Magnetic Variables (AREA)

Abstract

L'invention concerne un dispositif de détection magnétique (10) comprenant des fils (11, 13) pour la génération d'un champ magnétique avec une première fréquence f1, un détecteur à magnétorésistance géante (12) fonctionnant avec un courant d'entrée d'une seconde fréquence f2, et un démodulateur (26) fonctionnant à une troisième fréquence f3. En vue d'éviter une corruption de signal par un bruit de phase et d'améliorer le rapport signal sur bruit, les première, seconde et troisième fréquences sont dérivées par un bloc d'alimentation (121) à partir d'une fréquence de référence commune fref. Ladite dérivation peut, par exemple, être réalisée à l'aide de diviseurs de fréquences numériques. En outre, des détecteurs de phases (PD1, PD2) peuvent être utilisés dans une boucle de commande à rétroaction, afin de garantir des relations prédéterminées entre les phases des trois fréquences. Dans une autre mode de réalisation de cette invention, la phase et/ou l'amplitude d'un signal de modèle qui sont utilisées pour traiter un composant de signal souhaité dans la sortie du détecteur, sont suivies par un algorithme d'adaptation, par exemple, un baisse de gradient.
EP06809445A 2005-10-12 2006-09-29 Dispositif de detection magnetique a diverses frequences de fonctionnement interne Withdrawn EP1938084A2 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
EP06809445A EP1938084A2 (fr) 2005-10-12 2006-09-29 Dispositif de detection magnetique a diverses frequences de fonctionnement interne

Applications Claiming Priority (3)

Application Number Priority Date Filing Date Title
EP05109456 2005-10-12
EP06809445A EP1938084A2 (fr) 2005-10-12 2006-09-29 Dispositif de detection magnetique a diverses frequences de fonctionnement interne
PCT/IB2006/053561 WO2007042959A2 (fr) 2005-10-12 2006-09-29 Dispositif de detection magnetique a diverses frequences de fonctionnement interne

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EP1938084A2 true EP1938084A2 (fr) 2008-07-02

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US (1) US20080246471A1 (fr)
EP (1) EP1938084A2 (fr)
JP (1) JP2009511895A (fr)
CN (1) CN101283263B (fr)
WO (1) WO2007042959A2 (fr)

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Publication number Priority date Publication date Assignee Title
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CN101283263B (zh) 2011-01-26
JP2009511895A (ja) 2009-03-19
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WO2007042959A2 (fr) 2007-04-19
CN101283263A (zh) 2008-10-08

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