EP1657726A1 - Steuerschaltung und Steuerungsverfahren für ein Proportionalmagnetventil, insbesondere für Kraftfahrzeuge - Google Patents

Steuerschaltung und Steuerungsverfahren für ein Proportionalmagnetventil, insbesondere für Kraftfahrzeuge Download PDF

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Publication number
EP1657726A1
EP1657726A1 EP05110774A EP05110774A EP1657726A1 EP 1657726 A1 EP1657726 A1 EP 1657726A1 EP 05110774 A EP05110774 A EP 05110774A EP 05110774 A EP05110774 A EP 05110774A EP 1657726 A1 EP1657726 A1 EP 1657726A1
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EP
European Patent Office
Prior art keywords
winding
current
control circuit
signal
control
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP05110774A
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English (en)
French (fr)
Inventor
Andrea Nepote
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Marelli Europe SpA
Original Assignee
Magneti Marelli Sistemi Elettronici SpA
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Filing date
Publication date
Application filed by Magneti Marelli Sistemi Elettronici SpA filed Critical Magneti Marelli Sistemi Elettronici SpA
Publication of EP1657726A1 publication Critical patent/EP1657726A1/de
Withdrawn legal-status Critical Current

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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F7/00—Magnets
    • H01F7/06—Electromagnets; Actuators including electromagnets
    • H01F7/08—Electromagnets; Actuators including electromagnets with armatures
    • H01F7/18—Circuit arrangements for obtaining desired operating characteristics, e.g. for slow operation, for sequential energisation of windings, for high-speed energisation of windings
    • H01F7/1844—Monitoring or fail-safe circuits

Definitions

  • the present invention relates to a control circuit and method for a proportional solenoid valve provided with a control winding.
  • Solenoid valves of the proportional type can be used on motor vehicles, for example for controlling actuator devices in automatic gearboxes or in attitude control systems.
  • the control action is proportional to the intensity of the electrical signal exciting the control winding.
  • the solenoid valve can be likened to an inductive load.
  • the solenoid valve In applications such as the control of an automatic gearbox, the solenoid valve is typically excited with a sinusoidal current signal having a frequency of the order of 250 Hz. This signal is controlled in PWM mode by a control and drive circuit which is required to maintain a low current ripple around the mean value and a good frequency response.
  • the control and drive circuit must also be designed to provide protection against short circuits to ground and/or battery, and preferably to provide diagnosis of short-circuit or open-circuit conditions on the load.
  • the solenoid valve identified in a general way below as an inductive load L, is connected to a continuous voltage power source V B (the vehicle battery, for example) and to a conductor placed at a reference potential (ground), through a first switch SW H and a second switch SW L respectively.
  • V B the vehicle battery
  • a diode D in antiparallel recirculates the current in the discharge phase, when the load is not supplied.
  • a current sensing resistor with a low ohmic value is conveniently connected in series with the load to detect the current flowing through it, by differential reading for example, thus enabling the mean or peak current to be controlled.
  • This solution has the drawback of being expensive, because of the use of two switches to protect the solenoid valve, in a floating configuration, from short circuits to ground and battery.
  • This configuration has the drawback of exposing the load to evident risks of excess current and to undesired actuation in case of a short circuit to ground.
  • the typical solution is to use a "Smart Power" switching device made in MOSFET technology in the form of a floating-source n-channel transistor with an integrated control circuit (current mirror circuit configuration for reading the current flowing in the MOSFET).
  • a first drawback of the aforementioned solution is the limited switching speed of the MOSFET drive switch.
  • the activation time of the MOSFET typically ranges from 30 to 150 microseconds, whereas it would be preferable to have switching times of the order of 1 microsecond or less, to avoid excessive power dissipation during the switching edges of the PWM control signal (which in some applications has a frequency of up to 20 kHz).
  • a further drawback is the low precision of current measurement (errors of the order of 10%), which is insufficient to meet the specifications of the desired application (errors of the order of 1%).
  • this measurement is performed directly in the MOSFET by means of an integrated current mirror, the result is a function of the operating temperature and of the absolute value of the current itself.
  • the best solution in terms of control technology is to provide a current sensing resistor in series with the solenoid valve and connected to ground, to detect the current flowing in the load both during the activation state (ON) and during the inactivation state (OFF) of the switching device.
  • the current sensing resistor is connected in series with the MOSFET switching device and upstream or downstream of it.
  • This configuration makes it possible to measure the current in the load during the activation (ON) phase of the switch, and therefore to implement direct control of the peak current value and evaluation of the mean current by using estimators.
  • this signal not being referred to ground, but being superimposed on a common-mode voltage corresponding substantially to the value of the voltage supplied by the power supply battery where the resistor is upstream of the MOSFET, or floating between the supply voltage and the ground reference potential where the resistor is downstream of the MOSFET.
  • the object of the present invention is to provide a solution which overcomes the problems described above, and specifically to provide a control circuit configuration for a proportional solenoid valve which is of relatively low cost, but which is suitable for generating an excitation signal for the control winding of the solenoid valve which has a low ripple about the mean value and a good frequency response, with protection from short circuits to ground and/or battery and diagnosis of short-circuit or open-circuit conditions on the load.
  • this object is achieved by means of a control circuit having the characteristics claimed in Claim 1 and a method having the characteristics claimed in Claim 9.
  • a proportional solenoid valve and an associated control circuit are shown schematically in Figure 2. Elements or components identical or functionally equivalent to those shown in Figures 1a, 1b and 1c relating to the prior art have been indicated with the same references.
  • the proportional solenoid valve is of the type comprising a control winding or solenoid and a moving element (not shown) for implementing the control, the position of this element being controllable as a function of the intensity of the excitation current supplied to the winding.
  • the valve is represented as an inductive load L, having a first terminal connected permanently to a conductor maintained at a reference potential (ground) and a second terminal connected to a connecting terminal (pin) of a control circuit advantageously incorporated in a single on-board electronic control unit (ECU).
  • the ground connection is made by direct connection to a point on the vehicle's body, or through a return line to the control unit (ECU) to a corresponding ground terminal, for which no protection is required against short circuits to the battery.
  • ECU control unit
  • the control circuit comprises a drive branch connected to the control winding of the solenoid valve.
  • the drive branch includes a MOSFET power switch device, indicated by M, connected to a source of continuous supply voltage (V B ) such as the on-board battery of a motor vehicle, and capable of being driven in ON-OFF mode at its gate terminal.
  • a current recirculation branch comprising a recirculation device such as a diode D is connected in parallel with the load L.
  • a current sensing resistor R S is advantageously connected in the recirculation branch, between the anode terminal of the recirculation diode D and the conductor at ground potential of the control unit (ECU). This arrangement provides intrinsic protection from short circuit to the battery for the terminal connected to the recirculation diode, owing to the presence of the aforesaid diode.
  • the terminal of the current resistor R S opposite the ground conductor is connected to a first input SENS of a circuit for modulating the drive signal of the switching device, indicated by MOD, the output of which is connected to the input of a circuit for driving this switch, indicated as a whole by DV.
  • the modulation circuit MOD is designed to generate at its output a control signal with a modulated pulse width V PWM which, after being processed suitably by the drive circuit DV (by having its voltage increased, for example), is applied to the gate terminal of the MOSFET M.
  • the modulator circuit MOD also has a second input, indicated by SET, for receiving a signal indicating the desired intensity of the excitation signal (in other words, the desired degree of activation of the solenoid valve), for example from an associated microcontroller which is preferably located in the said control unit.
  • a p-channel or an n-channel MOSFET can be used alternatively as the power MOSFET component.
  • the corresponding drive circuit DV includes a bootstrap circuit, as shown in the attached Figure 3.
  • This comprises a driver DR for the MOSFET, this driver receiving at its input the signal V PWM , conveniently processed by an interface circuit LS for the level shift.
  • a bootstrap capacitor C connected to the source of supply voltage V B through a diode D1, is connected across the supply terminals of the driver DR, and has one terminal connected to the source terminal of the MOSFET.
  • the bootstrap capacitor C When the MOSFET is switched off, the bootstrap capacitor C is charged by a current flowing from the power supply source through the diode D1 and the load L towards ground. When the MOSFET becomes conducting, it brings its source terminal to a voltage close to that of the source.
  • the bootstrap capacitor is designed so as to provide a sufficient voltage between gate and source to keep the MOSFET conducting when it receives a conduction control signal. In the next cycle, the capacitor recovers the charge lost by the transfer of the said charge to the gate, by dissipation across the resistive elements of the driver and by dispersion.
  • a signal amplifier block AMP is provided in series with the SENS input, this block being formed by a feedback-controlled operational amplifier whose output is connected to the inverting input of a first threshold comparator COMP1.
  • the non-inverting input of the comparator coincides with the second input SET of the modulating circuit.
  • the output of the aforesaid comparator is connected to a control input of a monostable device, indicated by MS, of the type which can generate a pulse of predetermined duration T ON on detection of a signal edge at its control input.
  • the output of the monostable device is also connected to its control input through a feedback path comprising a second threshold comparator COMP2, to whose inverting input the signal at the output of the monostable device is applied, and whose non-inverting input is associated with a reference voltage level below the level of the voltage signal emitted by the monostable device (for example because it is obtained from the source of common supply voltage (5V) by division).
  • COMP2 second threshold comparator
  • the feedback path is formed in such a way as to reduce the input signal of the monostable device to zero immediately after its activation, in order to ensure that it is reset even in the situation in which, in the subsequent off period of the switch, the objective current is greater than that present in the recirculation branch, a situation which can occur, for example, if rapid increases in the load current are required.
  • a train of pulses generated by the monostable device MS forms the modulated duty cycle control signal V PWM emitted by the modulator circuit MOD.
  • the monostable device MS can advantageously be replaced by a circuit device based on the use of two other comparator circuits (COMP2 and COMP4) in cascade, configured according to the prior art so as to reproduce the behaviour of a monostable device, generating a pulse of predetermined duration T ON on the detection of a signal edge at its input.
  • COMP2 comparator circuits
  • This architectural solution can be used to save the cost of the monostable component by using a commercial integrated component such as the LM2901 quadruple comparator made by National Semiconductor, all four of whose independent comparators can thus be used.
  • a commercial integrated component such as the LM2901 quadruple comparator made by National Semiconductor, all four of whose independent comparators can thus be used.
  • Figure 6 shows examples of the variation of the load current (indicated by I L ), in other words the solenoid valve excitation current, the variation of the voltage (V S ) across the terminals of the sensing resistor R S , the variation of the corresponding signal V COMP applied to the input of the comparator COMP1 of the modulation circuit, and the variation of the control signal V PWM supplied to the MOSFET drive circuit.
  • the switch device M of the drive branch is arranged to couple the load L to the voltage source V B when it receives an activation signal at its input (the interval T ON ), thus causing a flow of current in the load.
  • the load L is disconnected from the voltage source and progressively discharges the accumulated energy, thus allowing current to flow in the recirculation branch.
  • the excitation signal applied to the control winding of the solenoid valve has a triangular-wave variation around the desired nominal mean value (controlled by an associated microcontroller by means of the SET input of the control circuit), with rising edges in the activation (conduction) intervals T ON of the MOSFET and decreasing edges in the inactivation (cut-off) intervals TOFF of the MOSFET.
  • the total period of the triangular wave is of the order of 50 microseconds, in other words approximately two orders of magnitude less than the period of the desired excitation signal, whose overall variation is sinusoidal with a frequency of the order of 250 Hz.
  • the signal V S detected across the sensing resistor R S is therefore zero in the activation intervals, when the current flows from the supply source to the load, and has the variation plotted in the figure in the intervals of inactivation, when the current flows in the recirculation branch.
  • This signal shows a virtually trapezoidal variation, rising to a peak value corresponding to the inactivation edge of the control signal V PWM and reducing its intensity progressively during the inactivation interval T OFF .
  • the total current flowing through the load can be estimated from the information about the signal V S (and therefore about the current I S flowing through the sensing resistor).
  • the PWM control signal is generated by feedback control of the current in the load, using an innovative modulation procedure with a fixed activation interval and an inactivation interval with threshold.
  • the activation interval T ON is set according to the inductance characteristics of the load and the available supply voltage.
  • the voltage signal V S is supplied to the SENS input of the modulation circuit MOD and is converted, by amplification and inversion of sign, to the signal V COMP .
  • the operational amplifier on which the amplifier block is based should have a high slew rate in order to be compatible with the duration of the inactivation interval TOFF (in other words it should have a switching time which is shorter by at least one order of magnitude), so that the edges of the signal V S can be replicated promptly in this interval.
  • the signal V COMP is compared in the comparator circuit COMP1 with a threshold value V th supplied to the corresponding SET input. It shows a decreasing trend until the threshold level is reached, at which point the signal emitted by the comparator is switched.
  • the corresponding switching edge is interpreted by the monostable device MS, or by the equivalent circuit arrangement, as a trigger signal, resulting in the output of a new activation pulse with the predetermined duration T ON .
  • the control method according to the present invention is based on the adjustment of the inactivation interval, which is the only interval in which, due to the proposed circuit architecture, it is possible to have information on the current flowing through the load.
  • the current is controlled cycle by cycle (the switching frequency is variable, since the activation time T ON is fixed) and the current control loop does not introduce limitations in terms of bandwidth, so that the maximum operating frequency is limited only by the overall period of the PWM control signal, and therefore by the predetermined fixed duration of the activation period (T ON ) - which is possibly adjustable if the configuration of Figure 4 is used - and by the duration of the inactivation period (T OFF ), which depends on the value of the peak voltage of V COMP and the value of the threshold voltage V TH .
  • the bandwidth is determined by the physical characteristics of the load (L) and by the supply voltage (V B ), which affect the maximum derivative of the current with respect to time which can be obtained through the load.
  • the control method with a fixed activation period conveniently enables the MOSFET to be driven more easily, since the corresponding activation time is no longer changeable according to the temperature, the supply voltage V B or the desired current intensity, thus making it possible to optimize the capacitance of the bootstrap capacitor C of the MOSFET drive circuit DV and enabling the MOSFET to be controlled over the whole interval T ON , without making the size of this capacitor too large.
  • circuit configuration proposed by the invention can be used to meet all the following requirements listed in the introduction to the present description, namely:

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  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Electronic Switches (AREA)
EP05110774A 2004-11-16 2005-11-15 Steuerschaltung und Steuerungsverfahren für ein Proportionalmagnetventil, insbesondere für Kraftfahrzeuge Withdrawn EP1657726A1 (de)

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
ITTO20040804 ITTO20040804A1 (it) 2004-11-16 2004-11-16 Circuito e procedimento di controllo per un'elettrovalvola proporzionale, particolarmente per l'impiego a bordo di autoveicoli.

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EP1657726A1 true EP1657726A1 (de) 2006-05-17

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EP05110774A Withdrawn EP1657726A1 (de) 2004-11-16 2005-11-15 Steuerschaltung und Steuerungsverfahren für ein Proportionalmagnetventil, insbesondere für Kraftfahrzeuge

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IT (1) ITTO20040804A1 (de)

Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103325523A (zh) * 2013-06-29 2013-09-25 歌尔声学股份有限公司 电磁铁保护电路
WO2017203289A1 (en) * 2016-05-27 2017-11-30 Haldex Brake Products Aktiebolag A control circuit for inductive loads in vehicles, comprising current sense-, current comparator- and current recirculation circuits
CN111983373A (zh) * 2020-09-16 2020-11-24 湖南行必达网联科技有限公司 一种电磁阀主动诊断电路、主动诊断系统及主动诊断方法

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4680667A (en) * 1985-09-23 1987-07-14 Motorola, Inc. Solenoid driver control unit
EP1381060A1 (de) * 2001-04-20 2004-01-14 Sanken Electric Co., Ltd. Solenoid-ansteuervorrichtung und ansteuerverfahren

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4680667A (en) * 1985-09-23 1987-07-14 Motorola, Inc. Solenoid driver control unit
EP1381060A1 (de) * 2001-04-20 2004-01-14 Sanken Electric Co., Ltd. Solenoid-ansteuervorrichtung und ansteuerverfahren

Cited By (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103325523A (zh) * 2013-06-29 2013-09-25 歌尔声学股份有限公司 电磁铁保护电路
CN103325523B (zh) * 2013-06-29 2015-08-26 歌尔声学股份有限公司 电磁铁保护电路
WO2017203289A1 (en) * 2016-05-27 2017-11-30 Haldex Brake Products Aktiebolag A control circuit for inductive loads in vehicles, comprising current sense-, current comparator- and current recirculation circuits
US11529937B2 (en) 2016-05-27 2022-12-20 Haldex Brake Products Aktiebolag Control circuit for operating inductive load devices, a braking system, and a vehicle including a braking system
CN111983373A (zh) * 2020-09-16 2020-11-24 湖南行必达网联科技有限公司 一种电磁阀主动诊断电路、主动诊断系统及主动诊断方法
CN111983373B (zh) * 2020-09-16 2023-07-28 湖南行必达网联科技有限公司 一种电磁阀主动诊断电路、主动诊断系统及主动诊断方法

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