EP1227707B1 - Microcontrôleur, alimentation à découpage et ballast électronique pour la mise en oeuvre d'au moins une lampe - Google Patents

Microcontrôleur, alimentation à découpage et ballast électronique pour la mise en oeuvre d'au moins une lampe Download PDF

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Publication number
EP1227707B1
EP1227707B1 EP01129890A EP01129890A EP1227707B1 EP 1227707 B1 EP1227707 B1 EP 1227707B1 EP 01129890 A EP01129890 A EP 01129890A EP 01129890 A EP01129890 A EP 01129890A EP 1227707 B1 EP1227707 B1 EP 1227707B1
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EP
European Patent Office
Prior art keywords
microcontroller
current
charging
discharging
inverter
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EP01129890A
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German (de)
English (en)
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EP1227707A2 (fr
EP1227707A3 (fr
Inventor
Peter Krummel
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Osram GmbH
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Patent Treuhand Gesellschaft fuer Elektrische Gluehlampen mbH
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    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/36—Controlling
    • H05B41/38—Controlling the intensity of light
    • H05B41/39—Controlling the intensity of light continuously
    • H05B41/392—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor
    • H05B41/3921—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor with possibility of light intensity variations

Definitions

  • the invention relates to a microcontroller according to the preamble of patent claim 1, a switching power supply according to claim 10, a ballast for at least one electric lamp according to the preamble of claim 11 and a method for operating at least one electric lamp according to the preamble of claim 25.
  • the invention relates to a microcontroller, which is provided for driving the switching transistors of a switched-mode power supply, preferably a switching power supply for the operation of electric lamps.
  • the switching power supplies commonly used for the operation of electric lamps are inverters, in particular half-bridge, full-bridge and push-pull inverters, as well as boost converter and buck converter.
  • Modern electronic ballasts for operating electric lamps usually have an inverter for generating a high-frequency alternating current for lamp operation and often also a boost converter as a DC power supply for the inverter.
  • the control of the switching transistors of the inverter and the boost converter is carried out by means of driver circuits which are designed as designed in analog technology integrated circuits.
  • modern electronic ballasts for electric lamps also contain a microcontroller, which is usually used for communication with a control unit arranged outside of the ballast and for the evaluation of the control commands of this control unit for the lamp operation and for monitoring the lamp operation.
  • the European Patent Application EP 0 708 579 A1 discloses a circuit arrangement for operating a high pressure discharge lamp to an inverter whose switching transistors are acted upon by means of a microcontroller and a downstream integrated driver circuit with pulse width modulated control signals.
  • the pulse width modulated control signals are generated by means of the auto-reload timer implemented in the microcontroller. In principle, this is a counter that works with the operating clock frequency of the microcontroller. During counting, the reaching of a reference value and the overflow of the counter are monitored.
  • the output of the auto-reload timer is at the high logic level and during the time that the counter requires to count from the reference value to the counter overflow, the output of the auto-reload timer is high Auto reload timers at logic low level.
  • pulse width modulated control signals for the inverter are generated by means of the microcontroller in order to enable lamp operation with a frequency modulated voltage in a small frequency range with a comparatively small number of discrete frequencies.
  • a microcontroller with a working clock frequency of more than 100 MHz is required for this purpose.
  • Such microcontroller but can not be used for cost reasons in electronic ballasts for lamp operation.
  • Patent US 6,137,240 shows a ballast with a low-cost microcontroller according to the preamble of independent claims 1 and 14th
  • the device for alternately charging and discharging a charge storage device and its control means make it possible to carry out alternately controlled charging and discharging operations in a charge storage device and, with the aid of the evaluation means, to evaluate the periods of time required for the partial charging or discharging of the charge storage device and in dependence thereon to generate a pulse width modulation control signal and / or frequency control signal. Even if the microcontroller according to the invention has only a low operating clock frequency, a finely graded pulse width modulation control and / or frequency control of a switching power supply can be realized with him, because the device for alternately charging and discharging a charge storage operates independently of the operating clock frequency of the microcontroller.
  • the microcontroller according to the invention comprises a frequency divider or a pulse divider which serves to detect at its input the switching of the device for alternately charging and discharging a charge store from discharging to charging or from charging to discharging and the input signal into signals for alternating control split by alternately switching switching means of the switching power supply.
  • the frequency divider or pulse divider With the aid of the frequency divider or pulse divider, the oscillation of the state of charge of the charge storage for the generation of frequency control signals and / or pulse width modulation control signals for the switching means of a switching power supply can be evaluated with alternating switching means.
  • the device for alternately charging and discharging a charge store comprises a controllable current source for charging the charge store with an adjustable charge current and a controllable current sink for charging the charge store with an adjustable discharge current.
  • the individual loading and unloading operations can be controlled independently of each other.
  • the controllable current source and current sink can be produced in a known manner by means of semiconductor technology and integrated into the microcontroller.
  • the controllable current source and the controllable current sink are designed such that their settings with respect to a reference current level are each variable with a resolution of at least 8 bits.
  • the reference current level for the charging and discharging current is advantageously predetermined by means of an ohmic resistance.
  • a control means for the device for As a control means for the device for alternately charging and discharging a charge storage advantageously at least one read-write memory is provided.
  • the contents of the random access memory may be continuously updated, for example programmatically, and read to control the device for alternately charging and discharging a charge storage.
  • the control means advantageously comprises a switching means adapted to switch over the device for alternately charging and discharging a charge storage from charging to discharging the charge storage upon reaching a first voltage value and switching the device for alternately charging and discharging a charge storage from discharge to charge of the charge storage upon reaching a second, lower voltage value is used.
  • the device for alternately charging and discharging a charge store is easily forced to alternate charge and discharge operations, so that the charge state of the charge store is subjected to sustained oscillation, which is evaluable to generate frequency control signals and / or pulse width modulation control signals.
  • the first or the second voltage value are advantageously adjustable by means of a read-write memory. Thereby, the aforementioned oscillation of the state of charge of the charge storage device can be programmatically influenced.
  • the microcontroller according to the invention also advantageously has interfaces for the detection of external signals or data and has a device for evaluating the external signals or data and for program-controlled determination of control values for controlling the device for alternately charging and discharging a charge store.
  • a control loop for the oscillation of the state of charge of the charge accumulator in dependence on external operating parameters and the control values derived therefrom can be realized.
  • the switching power supply according to the invention is characterized by a microcontroller according to one or more of claims 1 to 9.
  • the signals for pulse width modulation or frequency control of the switching transistors of the switching power supply are generated by the microcontroller in the switching power supply according to the invention.
  • the corresponding control signals are forwarded by the microcontroller directly or optionally via driver circuits to the control electrodes of the switching transistors of the switched-mode power supply. As already mentioned above, these control signals are independent of the operating clock frequency of the microcontroller.
  • the charge storage-charge alternately charging and discharging apparatus, the charge storage means, and the charge storage alternately charging and discharging control means constitute an oscillator which operates independently of the operating clock frequency of the microcontroller.
  • the oscillations of the state of charge of the charge storage are evaluated by means of the evaluation means for generating frequency control signals and / or pulse width modulation control signals for the inverter.
  • ballast according to the invention it is possible to realize all essential control functions of a modern, dimmable ballast with the help of a relatively simple and inexpensive microcontroller.
  • these are power factor correction, inverter control, lamp electrode heater control, load circuit control, lamp brightness control and lamp operation monitoring.
  • the ballast according to the invention comes out with comparatively few additional components. Most functions are taken over by the microcontroller in the ballast according to the invention.
  • the ballast according to the invention it is particularly simple to realize, for example, the end-of-life monitoring of the lamp, which is very complicated and expensive in the case of the ballasts according to the prior art.
  • the means for pulse width modulation control and / or frequency control advantageously a frequency divider or a pulse divider, which at its input detecting the switching of the device for alternately charging and discharging a charge storage device from discharging to charging or charging to discharging the charge storage device, and dividing the input signal into signals for alternately controlling the switching means of the inverter.
  • the ballast according to the invention advantageously has a heating device equipped with a controllable switching means, and the microcontroller has a comparator which compares the charge state of the charge store with a reference value for the lamp electrode heating and which generates a control signal for pulse width modulation of the controllable switching means of the heater is used.
  • the oscillation of the above-explained oscillator can be evaluated not only for controlling the inverter, but also for controlling the heating current for the lamp electrodes.
  • the reference value for the lamp electrode heating is advantageously adjustable by means of a read-write memory in order to be able to adapt the heating current for the lamp electrodes to the different operating states of the lamp.
  • the microcontroller also advantageously has synchronization means for synchronizing the controllable switching means of the heating device with a switching means of the inverter. This simplifies the control of the switching means of the heater. In addition, this will positively influence the vibration behavior of the inverter.
  • the DC supply circuit of the inverter advantageously has a step-up converter for power factor correction and / or to achieve sinusoidal as possible system power take-off
  • the microcontroller is provided with a second device for alternately charging and discharging a second charge storage and second control means for this second device Control the loading and / or unloading operations.
  • the second device for alternately charging and discharging a charge storage, the second charge storage and the second control means for this second device form a second oscillator, which also operates independently of the operating clock frequency of the microcontroller.
  • the microcontroller is also equipped with second evaluation means which serve to evaluate the oscillations of the state of charge of the second charge storage for generating pulse width modulation control signals and / or frequency control signals for the controllable switching means of the boost converter. In particular, the time intervals required for transferring the second charge store between different charge states are evaluated for this purpose. The microcontroller thus also takes over the control of the boost converter.
  • the second evaluation means advantageously comprise a first comparator for comparing the state of charge of the second charge storage with a first voltage value and a second comparator for comparing the state of charge of the second charge storage with a second, lower voltage value
  • the second control means advantageously have switching means for switching the second device for alternately charging and discharging a charge storage from charging to discharging the second charge storage upon reaching the first voltage value and for switching the second device for alternately charging and discharging a charge storage of discharge to charge the second charge storage on reaching the second, lower voltage value are used.
  • the first or second voltage value is adjustable by means of a read-write memory. Thereby, the first or second voltage value can be changed, for example by means of a program executed by the microcontroller, and stored for controlling the second device for alternately charging and discharging a charge storage device.
  • the two devices for alternately charging and discharging a charge store advantageously each have a controllable current source for charging the charge store or the second charge store with an adjustable charge current and a controllable current sink for charging the charge store or the second charge store with an adjustable discharge current.
  • the controllable current sources and current sinks can be on known Establish way with the help of the semiconductor technology and integrate into the microcontroller.
  • the two devices for alternately charging and discharging a charge storage device can thereby be manufactured by simple means as part of the microcontroller.
  • the settings of the controllable current sources and current sinks with respect to a reference current level are each variable with a resolution of at least 8 bits.
  • the aforementioned reference current level for the charging current and the discharge current can advantageously be predetermined by means of an ohmic resistance. This makes it possible to adapt the control of the inverter by appropriate dimensioning of the ohmic resistance to different mains voltages. In order to save components, preferably only a single ohmic resistor is used to specify the same reference current level for the charging and discharging currents of the two charge storage.
  • the microcontroller of the ballast according to the invention advantageously has at least one settable and resettable status bit, via which at least one controllable switching means of the inverter can be activated and deactivated. With the help of this status bit can be easily realized a shutdown of the inverter with a defective lamp or an end-of-life monitoring of the lamp. Instead, of course, the controllable switching means of the boost converter and thus the power supply of the inverter can be deactivated by means of the status bit to easily realize a safety shutdown of the ballast.
  • the microcontroller advantageously has one or more further settable and resettable status bits in order to be able to selectively switch the pulse width modulation control of the boost converter or of the inverter on or off. This makes it possible to apply to the controllable switching means of the boost converter and the inverter either exclusively with frequency control signals or pulse width modulation control signals or with frequency control signals and pulse width modulation control signals.
  • the microcontroller of the ballast according to the invention is advantageously provided with interfaces for detecting operating parameters of the boost converter or the inverter or the at least one electric lamp to evaluate the operating parameters by means of a program-controlled device of the microcontroller and control values for the devices for alternately charging and discharging a charge storage or to determine the reference value for the Lampenelektrodenloomung or the first and second reference value for the control of the boost converter.
  • the microcontroller is provided with interfaces for detecting at least one operating parameter of the boost converter, the inverter and the load circuit or the at least one electric lamp.
  • the ballast according to the invention advantageously has connections and means for communication with an externally arranged control device, which in turn are coupled to interfaces of the microcontroller.
  • the ballast according to the invention is prepared for the reception and processing of control commands of an external control device and for the transmission of status messages to the external control device. These processes are also controlled by the microcontroller of the ballast according to the invention.
  • the inventive method for operating at least one electric lamp on a ballast having an inverter with a microcontroller containing control circuit for the switching means of the inverter and at least one coupled to the inverter load circuit with terminals for the at least one lamp, according to the invention is characterized by, that with the aid of the microcontroller, a charge storage is applied alternately with a charging current and a discharge current, and the duration of the alternating charging and discharging processes of the charge storage is evaluated and in response generates a frequency control signal and / or a pulse width modulation control signal for alternately controlling the switching means of the inverter becomes.
  • the method according to the invention makes it possible to generate control signals for frequency control and / or pulse width modulation of the inverter independently of the operating clock frequency of the microcontroller with the aid of the microcontroller.
  • a comparatively cost-effective microcontroller that is to say a microcontroller with a low operating clock frequency, can be used in the ballast according to the invention for realizing all essential control functions.
  • a frequency divider or pulse divider is advantageously used, which detects the switching of the device for alternately charging and discharging a charge storage from discharging to charging the charge storage or charging to discharging the charge storage.
  • the inventive method also allows heating of the lamp electrodes by the heating current for the lamp electrodes is controlled by means of a controllable switching means.
  • the signals for the pulse width modulated control of the controllable switching means of the heating device are generated by means of a comparator, which compares the state of charge of the charge storage device with a reference value for the lamp electrode heating.
  • a comparator which compares the state of charge of the charge storage device with a reference value for the lamp electrode heating.
  • the heating power can be adjusted programmatically by means of the microcontroller.
  • the controllable switching means for controlling the heating current is advantageously switched on in synchronism with a switching means of the inverter. This simplifies the control of the controllable switching means of the heater.
  • the duty cycle of the controllable switching means for regulation the heating current is preferably less than or equal to the duty cycle of the corresponding switching means of the inverter.
  • the DC voltage supply of the inverter is controlled by means of a boost converter in order to ensure a power factor correction and / or a sinusoidal AC current drain.
  • the pulse width modulation control signals and / or the frequency control signals for the controllable switching means of the boost converter are also generated by means of the microcontroller by a second charge storage between different charge states is reloaded and the periods for reloading the second charge storage for generating the pulse width modulation control signals and / or the frequency control signals for the controllable switching means of the boost converter are evaluated.
  • the same microcontroller that is used to control the inverter can also be used to control the boost converter.
  • the reloading of the second charge storage device can be detected and evaluated in a simple manner by means of two comparators by the first comparator comparing the charge state of the second charge store with a first voltage value and the second comparator comparing the charge state of the second charge store with a second, lower voltage value.
  • the charging process is ended and the discharging process of the second charge storage device is started, while when the second, lower voltage value is reached, the discharging process is ended and the charging process of the second charge storage device is restarted.
  • the first or second voltage value are advantageously set by means of a read-write memory. As a result, the corresponding voltage value can be varied programmatically.
  • the circuit arrangement of the preferred embodiment of the ballast according to the invention is shown schematically. Because of their size, the circuitry had to be displayed on two sheets. At the connection points designated J10 to J26, the two halves of the circuit arrangement shown in FIGS. 1 and 2 are linked together.
  • This ballast is a so-called electronic ballast for the operation of fluorescent lamps.
  • the ballast has two mains voltage terminals J1, J2, to which a capacitor circuit C1 and the transformer L1 existing filter circuit for radio suppression of the ballast is connected.
  • This filter circuit is connected to a bridge rectifier, which is formed by four rectifier diodes D1, D2, D3 and D4.
  • the bridge rectifier D1-D4 is followed by the capacitor C2, which forms the DC output of the bridge rectifier D1-D4.
  • a boost converter is connected, which comprises the field effect transistor V1, the inductor L2, the diode D5 and the resistor R13.
  • the voltage applied to the capacitor C2 DC voltage serves as a supply voltage for the boost converter.
  • the gate electrode of the transistor V1 is connected via the resistor R4 to the pin 4 of the microcontroller MC, which takes over the control of the transistor V1.
  • the voltage output of the boost converter is formed by the DC link capacitor C3.
  • the voltage at the intermediate circuit capacitor C3 is monitored by means of the voltage divider resistors R2, R5 at the pin 21 of the microcontroller MC.
  • the voltage at the capacitor C2 is detected by means of the voltage divider resistors R1, R18 at the pin 20 of the microcontroller MC.
  • the half-bridge inverter consists essentially of the field effect transistors V2, V3, the trapezoidal capacitors C10, C11, the inductor L4, the coupling capacitors C15, C16 and the ignition capacitor C12.
  • a load circuit is connected, the inductor L4, the ignition capacitor C12, the terminals X1 to X8 for the electrode coils E1, E2 and E3, E4 of the two parallel-connected fluorescent lamps LP1, LP2, the transformer L5 and the coupling capacitors C15, C16 comprises.
  • the ignition capacitor C12 is connected in parallel to both lamps LP1, LP2.
  • the coupling capacitors C15, C16 are each arranged in series with one of the lamps LP1, LP2.
  • the transformer L5 serves to symmetrize the currents in the lamp circuits. For this purpose, each one of the transformer windings in one of the lamp circuits, that is, arranged in series with one of the lamps LP1, LP2.
  • the two lamp circuits are brought together again at the connection X8 and at the two connections of the coupling capacitors C15, C16 connected to the circuit-internal ground GRD.
  • the gate electrodes of the transistors V2, V3 are controlled via the resistors R6 and R7, respectively, by the microcontroller MC with the aid of the integrated circuit IC, which essentially has only driver circuits for driving the inverter transistors and auxiliary voltage generating circuits for the microcontroller MC ,
  • the half-bridge inverter generates a high-frequency current in the load circuit for the lamps LP1, LP2 with a frequency between approximately 30 kHz and 100 kHz. After the ignition of the gas discharge in the lamps LP1, LP2 flow in both lamp circuits via the terminal X8, the discharge path of the lamp LP1 or LP2, the terminal X5 or X7 and the coupling capacitors C16 and C15 high-frequency lamp currents.
  • the reactor L4 and the ignition capacitor C12 are formed as a series resonant circuit.
  • the ignition voltage required to ignite the gas discharge in the fluorescent lamps is provided by means of the method of resonance peaking at the ignition capacitor C12, during the ignition phase, the switching frequency of the transistors V2, V3 of the half-bridge inverter of the resonant frequency of the series resonant circuit is approximated.
  • the center tap between the inductor L4 and the firing capacitor C12 is connected to the pin 18 of the microcontroller MC via the capacitor C22, the resistor R24 and the forward-biased diode D12.
  • a half-wave of the alternating current component of the load current is monitored by means of the resistors R24, R25, the diodes D12, D13 and the capacitors C22, C23.
  • the other half-wave of the alternating current component of the current flowing in the load circuit is clamped by the diode D13 to the internal circuit ground potential GRD.
  • the pin 19 of the microcontroller MC is connected via the resistor R27 to the source electrode of the transistor V3 and coupled via the capacitor C24 to the internal circuit ground potential GRD.
  • the resistor R9 connects the source of the transistor V3 to the in-circuit ground potential GRD. At pin 19, the current through the transistor V3 is monitored.
  • the ballast further comprises a heater for the electrodes E1-E4 of the two fluorescent lamps, which is connected to the center tap between the two field effect transistors V2, V3 of the half-bridge inverter.
  • This heater essentially consists of the field effect transistor V4 and the transformer L3.
  • the primary winding of the transformer L3 is connected on the one hand to the center tap between the transistors V2, V3 and on the other hand to the drain terminal of the transistor V4 and in DC forward direction via the diode D8 to the positive pole of the DC link capacitor C3.
  • the source of the transistor V4 is connected to the internal circuit ground potential GRD via the resistor R17.
  • the three secondary windings of the transformer L3 are, with the lamps LP1, LP2 connected, in each case arranged together with a rectifier diode D9 or D10 or D11 in a closed circuit for heating the electrode filaments E1 and E3 or the electrode coil E2 or E4.
  • the heating current in the three equipped with the secondary windings of the transformer L3 heating circuits is controlled by the switching clock of the transistor V4.
  • its gate electrode is connected via the resistor R26 to the pin 10 of the microcontroller MC.
  • the heating device serves on the one hand for preheating the electrode filaments E1-E4 before the gas discharge is ignited in the lamps LP1, LP2, and on the other hand for heating the electrode filaments E1-E4 during the dimming operation of the lamps LP1, LP2.
  • the heating current that is, the current through the primary winding of the transformer L3 and the transistor V4, is monitored by means of the RC element R23, C18 at the pin 17 of the microcontroller MC.
  • the pin 17 is connected via the resistor R23 to the source electrode of the transistor V4.
  • a direct current path is realized which, starting from the positive pole of the capacitor C3, via the resistor R10, the connection X3, the electrode coil E1, the connection X8, the electrode coil E3, the connection X2 and the resistors R14, R22 is led to the in-circuit ground potential GRD.
  • This DC path is interrupted if one of the lamps LP1 or LP2 is missing or one of the electrode coils E1 or E3 is defective.
  • the center tap between the resistors R14, R22 is connected to the pin 25 of the microcontroller MC to monitor the DC path.
  • the ballast also includes a communication device DS for communicating with an external control device (not shown).
  • This device DS has two terminals J3, J4, which are connectable to the external control device.
  • the terminals J3, J4 serve to receive digital or analog control signals from the external control device and to send information, for example about the operating state of the lamps, from the ballast to the external control device.
  • a bidirectional connection with the external control device is possible via the connections J3, J4.
  • An output of the communication device DS is connected to the internal circuit ground potential GRD.
  • the pin 6 of the microcontroller MC is connected for the transmission of data to the external control unit to the input of the communication device DS and the pin 5 of the microcontroller MC is connected to receive and evaluate control commands from the external control device to the output of the communication device DS.
  • the integrated circuit IC contains driver circuits for the transistors V2, V3, in particular a bootstrap circuit for the transistor V2 and level-shift circuits for the control of the transistors V2, V3.
  • the capacitor C9 and the pins 1, 2, 3 and 14 of the integrated circuit IC are these driver circuits associated with the transistors V2, V3.
  • the control signals for controlling the switching clock of the transistors V2, V3 or for frequency control of the half-bridge inverter are generated by the microcontroller MC and fed via the pin 24 and 23 to the pin 9 and 10 of the integrated circuit IC.
  • a detector is realized, which prevents too high a current load of the transistors V2, V3.
  • the pin 5 of the integrated circuit IC is connected to the positive pole of the capacitor C2.
  • a power supply of the integrated circuit IC is ensured during the start-up phase, that is, before the half-bridge inverter has recorded its oscillation.
  • auxiliary voltages of 5 V or 15 V for the microcontroller MC are provided by means of the capacitors C14 and C25.
  • the voltage for supplying the integrated circuit IC and the microcontroller MC by means of the connected to the pin 7 of the integrated circuit IC and to the center tap between the firing capacitor C12 and the inductor L4 capacitor C13 and by means of one in the integrated circuit IC integrated two-point controller derived from the load circuit.
  • microcontroller MC The structure of the microcontroller MC and the generation of the control signals for the transistors V1-V4 are explained in more detail below with the aid of the microcontroller MC.
  • the microcontroller MC has a clock that determines the operating cycle of the microcontroller, a central processing unit, a program memory, a data memory and a mathematical unit for performing simple mathematical operations.
  • the aforementioned parts of the microcontroller MC are represented in the block diagram of Figure 2 by the module A.
  • the module A the pins 1 and 2, 15 to 22 and 23 to 28 assigned.
  • At the pins 1 to 2 is the Quartz crystal B2 connected to control the clock.
  • the working clock frequency of the microcontroller is 8 MHz.
  • the module B is an interface that is used to prepare the digital or analog data for communication with the communication device DS.
  • the module B, the pins 5 and 6 of the microcontroller MC are assigned.
  • the module C is a 5V power supply, which is connected via the pins 11 and 12 of the microcontroller MC to the capacitor C14 and to the ground potential GRD. Through the address and data bus D all components of the microcontroller MC are interconnected.
  • the first control module E and its associated pins 3, 4 and 9 of the microcontroller MC is used to control the transistor V1 of the boost converter.
  • the second control module G and its associated pins 7, 8 and 10 of the microcontroller MC is used to control the transistors V2 and V3 of the half-bridge inverter and to control the transistor V4 of the heater. Both control modules E, G are connected to each other via the data bus F.
  • the module H is a 15V voltage source, which is connected via the pins 13, 14 of the microcontroller MC to the ground potential GRD or to the capacitor C25.
  • the construction of the control module G is shown schematically in the block diagram of FIG.
  • the control module G has to control the transistors V2, V3 of the half-bridge inverter, the controllable current source SQ1, the controllable current sink SS1, the read-write memory DR1, DR2, the switch US1 for alternately turning on and off the controllable current source and current sink, the frequency divider FT1 for halving the frequency of the switching signal of the switch US1, the data memory DR3 for storing the control signals for the transistors V2, V3, the reference current source IR to specify a constant reference current I Ref for the controllable current source SQ1 and current sink SS1 and logic circuit components O1-O3, U1 -U6 on.
  • a constant output voltage of 2 V is provided, which flows according to the ohmic law through the resistor R30 a constant reference current I Ref .
  • the value of this reference current I Ref is predetermined by the choice of the resistance value of the resistor R30.
  • the linear operating range of the reference current I Ref extends from 5 ⁇ A to 50 ⁇ A.
  • the capacitor C27 is connected, which serves as an electrical charge storage. With the help of the controllable current source SQ1, the capacitor C27 is charged.
  • the controllable current source SQ1 is switched off by the switch US 1 and the controllable current sink is switched on, discharging the capacitor C27. If the voltage drop across the capacitor C27 reaches the value of 1.5 V, the controllable current sink SS1 is switched off by the switch US1 and the controllable current source SQ1 is switched on again, which recharges the capacitor to a voltage value of 3 V. In this way, the capacitor C27 is alternately charged and discharged. The voltage drop across the capacitor C27 therefore continually oscillates between the values 1.5 V and 3 V.
  • the controllable current source SQ1 and the controllable current sink SS1 and the switch US1 form a device for alternately charging and discharging the capacitor C27.
  • the charging current for the capacitor C27 generated by the controllable current source SQ1 can be set by means of the read-write memory DR1.
  • the random access memory DR1 is a 16-bit data register, of which 12 bits are used to control the current source SQ1.
  • the charging current for the capacitor C27 is therefore adjustable with a resolution of 12 bits between the values I Ref / 256 and 32 I Ref , wherein the abbreviation I Ref stands for the reference current intensity of the reference current source IR.
  • the entry in the data register DR1 determines the charging current for the current or subsequent charging process on the capacitor C27 and thus the time required for this charging process.
  • the discharge current of the capacitor C27 generated by the controllable current sink SS1 can be set by means of the read-write memory DR2.
  • the random access memory DR2 is an 8-bit data register.
  • the discharge current of the capacitor C27 is therefore adjustable with an 8-bit resolution between the values 0.25 I Ref and 128 I Ref .
  • the entry in the data register DR2 determines the discharge current for the current or subsequent discharge process on the capacitor C27 and thus the time required for this discharge process.
  • the oscillations of the state of charge of the capacitor C27 and the voltage drop across the capacitor C27 are therefore independent of the operating clock frequency of the microcontroller MC.
  • the switching signals of the switch US1 are evaluated by the frequency divider FT1 and the AND gates U1, U2 for generating control signals for the transistors V2, V3 of the half-bridge inverter.
  • the frequency divider FT1 detects only the switching pulses of the switch US1, which start a new charging of the capacitor C27, and switches its two outputs, which are respectively connected to the input of an AND gate U1 and U2, alternately at each such switching pulse to "High "or” Low “.
  • the switching signals of the switch US1 are also fed directly to the input of the AND gates U1, U2.
  • the status register SR1 contains a status bit for activating and deactivating the control signals for the transistor V2 and a status bit for activating and deactivating the control signals for the transistor V3.
  • the state of the status bit for activating and deactivating the control signals for the transistor V2 is monitored by the AND gate U2, while the state of the status bit for the transistor V3 is detected by the AND gate U1.
  • the output states of the AND gate U1 and U2 are stored in each case in one bit of the data register DR3 and can be retrieved via the address and data bus D at the pins 23 and 24 of the microcontroller MC.
  • the frequency of the half-bridge inverter that is, the switching clock of its transistors V2, V3, is controlled by the duration of the individual charging and discharging operations of the capacitor C27.
  • the triangular curve of the diagram a) shows the time profile of the voltage drop across the capacitor C27.
  • the voltage drop across the capacitor C27 varies linearly with the time between the values 1.5 V and 3 V.
  • the diagram b) shows the time course of the charging current for the capacitor C27.
  • the charge current can assume different discrete values according to the above explanations for the controllable current source SQ1 4096.
  • the diagram c) the time profile of the discharge current for the capacitor C27 is shown.
  • the discharge current can according to the above explanations for the controllable current sink SS1 256 different discrete values.
  • Diagram d) shows the time profile of the control signal LG, which can be called up at the pin 23 of the microcontroller MC, for the driver circuit of the transistor V3.
  • the diagram e) shows the time course of the controllable on the pin 24 of the microcontroller MC control signal HG for the driver circuit of the transistor V2.
  • the duration for the individual charging operations on the capacitor C27 is determined by the magnitude of the charging current IL1.
  • the duration for the individual discharging operations on the capacitor C27 is determined by the magnitude of the discharge current IE1.
  • both control signals LG and HG assume the logic state "Low”. This means that the transistor V2 or V3 is turned on as long as the control signal HG or LG assigned to it has the status "high”.
  • the transistors V2, V3 of the half-bridge inverter are alternately turned on and off in this way.
  • both transistors V2, V3 are turned off. The evaluation of the voltage curve on the capacitor C27 thus enables a frequency-modulated control of the half-bridge inverter.
  • the values for the charging current IL1 and the discharging current IE1 are determined by the data stored in the data registers DR1 and DR2, respectively.
  • This data is programmed programmatically using module A, depending on the pin 18 of the Microcontroller MC detected half-wave of the AC component of the current in the load circuit and determined by the current detected by the pin 19 through the transistor V3.
  • the module A of the microcontroller MC calculates program-controlled from the comparison of the aforementioned operating parameters with predetermined setpoints control values for controlling the controllable current source SQ1 and the controllable current sink SS1, which are stored in the data registers DR1 and DR2.
  • a control loop is implemented for the frequency-modulated control of the half-bridge inverter as a function of its operating parameters and the predetermined setpoint values.
  • the setpoint values for the frequency-modulated control of the half-bridge inverter are determined programmatically by the module A of the microcontroller MC, for example as a function of external control commands for dimming the lamps LP1, LP2, which are communicated via the interfaces J3, J4 of the communication device DS and the pin 5 of the microcontroller MC be supplied.
  • the data registers DR1 to DR4 and the status register SR1 are connected to the address and data bus D.
  • the voltage curve on the capacitor C27 shown in the diagram a) of Figure 5 is also also evaluated for generating pulse width modulated control signals for the transistor V4 of the heater for the electrode filaments E1-E4 of the lamps LP1, LP2.
  • the read-write memory DR4 designed as an 8-bit data register
  • the comparator K1 whose inverting input detects the voltage drop across the capacitor C27 and whose non-inverting input is controlled by the data register DR4
  • the comparator K1 compares the voltage curve across the capacitor C27 with the control value of the heating current stored in the data register DR4.
  • the aforementioned manipulated variable can be varied with a resolution of 8 bits. Accordingly, the voltage at the non-inverting input of the comparator K1 with the same resolution in the range of 1.5 V to 3 V is variable.
  • the output signal of the comparator K1 is supplied via the OR gate O1 and the AND gate U3 to the OR gate 03, whose output is connected to the input of the driver circuit TR1, via the pin 10 of the microcontroller MC and the resistor R26 drives the gate of the transistor V4.
  • the output signal of the comparator K1 is additionally supplied to the OR gate 02, whose output is connected to the AND gates U1 and U2.
  • the output of the AND gate U1 is connected to the OR gate 03 via the AND gate U3.
  • the output of the AND gate U2 is connected to the OR gate 03 via the AND gate U4.
  • the 8-bit status register SR1 has a first status bit for activating and deactivating a maximum heating current, which is connected to the OR gate 03 via the OR gate O1, the AND gate U3. Maximum heating current means that the turn-on duration of the transistor V4 is equal to the turn-on duration of the transistor V2 or V3.
  • the second status bit of the status register SR1, which is connected to the OR gate 03 via the AND gate U3, serves to activate or deactivate the synchronous switching on of the transistors V3 and V4.
  • the third status bit of the status register SR1, which is connected to the OR gate 03 via the AND gate U4, serves to activate or deactivate the synchronous switching on of the transistors V2 and V4.
  • the fourth status bit of the status register SR1 is connected to the AND gate U6, the output of which is connected to the control module E via the data bus F. Since the output of the AND gate U1 is connected to the AND gate U6, the connection of the control signal LG to the control module E is activated or deactivated by the fourth status bit.
  • the fifth status bit of the status register SR1 is connected to the OR gate 03 via the AND gate U5.
  • the AND gate U5 receives via the data bus F also from the control module E an input signal. By the fifth status bit, the synchronization of the control signals for the transistors V1 and V4 can be activated or deactivated.
  • the sixth congestion bit of the status register SR1 which is connected to the OR gate 02, is used to enable or disable the pulse width modulation of the control signals LG and HG.
  • the seventh or eighth status bit which is connected to the AND gate U1 or U2, is used to activate or deactivate the control signals LG and HG for the transistors V3 and V2 and for the transistor V4.
  • the seventh or eighth status bit can easily be a shutdown of the half-bridge inverter and the heater for defective lamps LP1, LP2 be performed.
  • a direct current path is realized by means of the resistor R10, the diode D9 and the corresponding secondary winding of the transformer L3, in which the electrode filaments E1 and E3 are connected in series. If one of the lamps LP1, LP2 is missing, this DC current path is interrupted. Via the resistor R14, the current in this DC path is monitored at the pin 25 of the microcontroller MC. If the aforementioned direct current path is interrupted, the control signal LG or HG can be switched off by resetting the seventh or eighth status bit of the status register SR1 and the half-bridge inverter thereby shut down.
  • a breakage of the electrode coil E2 or E4 via the corresponding winding of the transformer L5 and the resistor R16 or R15 at the pin 16 or 15 is detected by the microcontroller MC.
  • the current through the lamp LP1 or LP2 or the voltage drop across the coupling capacitor C15 or C16 is monitored at the pins 15 or 16 of the microcontroller MC by means of the voltage divider resistors R15, R21 or R16, R20, at the end of the service life the lamp LP1 or LP2 occurring rectifying effect of the lamp LP1 or LP2 to detect.
  • the information is evaluated by the microcontroller MC and can be transmitted via the pin 6 and the communication device DS to an external control device or used to control the transistors V2, V3 and V4.
  • the transistor V4 is always switched in synchronism with the transistor V2 of the half-bridge inverter.
  • the switch-on or the switch-off of the transistor T4 and thus the pulse width of the control signal HTG depend on the output of the comparator K1, which compares the control value stored in the data register DR4 for controlling the heating current with the instantaneous voltage drop across the capacitor C27.
  • the control signal HTG changes from the state "high” to the state "low” . Since the signal present at the non-inverting input of the comparator K1 can only assume values between 1.5 V and 3 V, the pulse width of the control signal HTG is less than or equal to the pulse width of the control signal HG. This means that the turn-on of the transistor V4 is at most as long as the turn-on of the transistor V2. In this case, the largest possible heating current flows through the electrode coils E1-E4.
  • the current through the transistor T4 or through the primary winding of the transformer L3 is monitored via the RC element R23, C18 at the pin 17 and programmatically compared by means of the module A with a desired value and in dependence on the Comparison a control value for generating the control signal HTG stored in the data register DR4.
  • the required heating current depends on the operating status of the lamps LP1, LP2. During the preheating phase, a relatively high heating current is needed to allow a gentle ignition of the gas discharge. In addition, a heating current for the electrode coils for heavily dimmed lamps LP1, LP2 is necessary.
  • control module E for controlling the transistor V1 of the boost converter, which is used for DC voltage supply of the downstream half-bridge inverter, is shown schematically.
  • the control module E comprises the controllable current source SQ2, the controllable current sink SS2, the read-write memories DR5, DR6, DR7, the status registers SR1, SR2, SR3, the comparators K2, K3, K4, K5 and the driver circuit TR2 for the transistor V1 on.
  • the aforementioned components of the control module E are by logical circuit components networked with each other.
  • the status register SR1 is the same status register already described in connection with the control module G.
  • the controllable current source SQ2 serves to charge the capacitor C26 connected to the pin 9 of the microcontroller MC, and the controllable current sink SS2 serves to discharge the capacitor C26.
  • the controllable current source SQ2 and the controllable current sink SS2 are each coupled to the reference current source IR.
  • the charge current and the discharge current for the capacitor C26 are each adjustable with an 8-bit resolution between the values 0.25 I Ref and 128 I Ref .
  • each designed as 8-bit data register write-read memory DR5 and DR6 serve. The charge current is adjusted by means of the data register DR6 and the discharge current by means of DR5.
  • the capacitor C26 connected to the pin 9 of the microcontroller MC is charged to a predefinable upper voltage value, which lies in the range of 1.5 V to 3 V.
  • a predefinable upper voltage value which lies in the range of 1.5 V to 3 V.
  • the charging process is aborted and the discharging process of the capacitor C26 is started with the aid of the controllable current sink SS2. If the voltage at the capacitor reaches the lower voltage value of 1.5 V, the discharging process is interrupted and a new charging process is started on the capacitor C26.
  • the activation and deactivation of the controllable current source SQ2 and the controllable current sink SS2 for alternately charging and discharging the capacitor C26 is performed by means of the RS flip-flop FL1 and by means of the comparators K2 and K4 or alternatively by means of the comparators K3 and K4.
  • the comparator K2 compares the voltage across the capacitor C26 with the upper voltage value while the comparator K4 compares the voltage across the capacitor C26 with the lower voltage value of 1.5V.
  • the upper voltage value is adjustable by means of the 8-bit data register DR7, which is connected to the inverting input of the comparator K2. Instead of the comparator K2 but also the comparator K3 can be selected to compare the voltage across the capacitor C26 with the upper voltage value.
  • the upper voltage value is 3 V and can not be varied.
  • To control the controllable current source SQ2 and the controllable current sink SS2 for each other alternating charging and discharging operations on the capacitor C26 are the outputs of the comparators K2 and K3 via the positive edge generator FG1, the AND gate U7 and the OR gate 04 and via the positive edge generator FG2, the AND- Gate U8 and the OR gate 04 connected to the set input of the RS flip-flop FL1.
  • the output of the comparator K4 is connected via the positive edge generator FG3 to the reset input of the RS flip-flop FL1.
  • the two outputs of the RS flip-flop FL1 are connected to the controllable current source SQ2 and to the controllable current sink SS2.
  • the controllable current source SQ2, the controllable current sink SS2, the comparators K2 (or K3) and K2 and the RS flip-flop FL1 form a device for alternately charging and discharging a charge storage device, which alternately charges the capacitor C26 with a charging current and a discharging current applied.
  • the voltage on the capacitor C26 therefore oscillates continuously between the upper and lower voltage value. This oscillation is independent of the operating clock frequency of the microcontroller MC.
  • the comparator K2 (or K3), K4, the positive-edge generators FG1-FG3, the RS flip -Flops FL2 and the logic circuit components U9-U11, 05, 06 generates a frequency-modulated and pulse width modulated control signal PG for the input of the driver circuit TR2, which is supplied via the pin 4 of the microcontroller MC and the resistor R4 of the gate electrode of the transistor V1.
  • the control module E also has the comparator K5, the RS flip-flops FL3, FL4, the OR gate 07 and the status registers SR2, SR3.
  • the status registers SR1-SR3 and the data registers DR5-DR7 are connected to the address and data bus D.
  • the current through the transistor V1 is monitored at pin 3 of the microcontroller MC.
  • the comparator K5 the OR gate 07 and the RS flip-flop FL4
  • the transistor V1 is protected from high currents by the control signal PG is turned off for the transistor V1 in the presence of too high a current.
  • the pin 3 of the microcontroller MC is connected to the non-inverting input of the comparator K5, while at the inverting input of the comparator K5 a reference value is applied, which by means of the status register SR3 with a resolution of 4 bits is adjustable between the values 0 V to 2 V and defines the switch-off threshold for the control signal PG.
  • the first status bit of the status register SR2 is set by means of the RS flip-flop FL3.
  • the second status bit of the status register SR2 is set in response to the output of the OR gate 06, and indicates whether or not a control signal PG is present.
  • status register SR3 uses the first four bits to drive the inverting input of comparator K5.
  • the fifth bit of the status register SR3 allows additional control of the reference current source IR.
  • the sixth bit of status register SR3 is unused.
  • the control signal for the driver circuit TR2 and the transistor V1 can be activated or deactivated.
  • the output signal of the comparator K2 or of the comparator K3 can optionally be activated. As a result, two different operating modes of the boost converter are possible.
  • the step-up converter not only regulates the supply voltage of the half-bridge inverter, but also serves to correct the power factor correction.
  • This mode of operation is preferred for operating discharge lamps, in particular fluorescent lamps.
  • the other mode of operation of the boost converter is suitable for operating low-voltage halogen incandescent lamps on an electronic transformer, which has a step-up converter for regulating the supply voltage of the downstream inverter.
  • the output signal of the comparator K2 is active.
  • the control signal PG can also be made available via the AND gate U12, the data bus F and the AND gate U5 by means of the fifth status bit of the status register SR1 at the pin 10 of the microcontroller MC for controlling the transistor V4.
  • control signal LG of the control module G for controlling the transistor V3 via the AND gate U6, the data bus F and the OR gate 07 by means of the fourth status bit of the status register SR1 also made available at the pin 4 of the microcontroller MC to control the transistor V1 become.
  • the generation of the control signal PG for the transistor V1 is explained in more detail below with reference to FIG.
  • the triangular curve in the diagram a) of FIG. 7 represents the temporal voltage curve across the capacitor C26.
  • the step-shaped curve in the diagram a) of FIG. 7 represents the time profile of the memory contents of the data register DR7, which can assume values between 1.5 V and 3 V with a resolution of 8 bits.
  • the diagram b) shows the time profile of the control signal PG for the gate electrode of the transistor V1 which can be called up at the pin 4 of the microcontroller MC.
  • the diagram c) of FIG. 7 shows the time profile of the signal generated by means of the RC element R32, C28 at the pin 3 of the microcontroller MC for monitoring the current through the transistor V1.
  • Diagram d) shows the time profile of the charging current generated by the controllable current source SQ2 for the capacitor C26 and in the diagram e) of FIG. 7 the time profile of the discharge current generated by the controllable current sink SS2 for the capacitor C26.
  • the capacitor C26 is alternately charged to an upper voltage value determined by the memory content of the data register DR7 and discharged to a lower voltage value of 1.5V.
  • the duration of the individual charging operations of the capacitor C26 is therefore determined by the upper voltage value and by the charging current IL2 which can be set by means of the data register DR6. Accordingly, the duration of the individual discharge processes is determined by the upper voltage value and the discharge current IE2 which can be set by means of the data register DR5.
  • the periods of time required for alternately charging and discharging the capacitor C26 are evaluated by means of the above-described logic circuit components of the control module E for generating the frequency-modulated and pulse-width-modulated control signal PG.
  • the comparison of the voltage curve shown in the diagram a) on the capacitor C26 with the control signal PG shown in the diagram b) shows that the transistor V1 is turned off during the charging operations on the capacitor C26 and during the discharging operations on the capacitor C26 is turned on. If the signal IV1 (diagram c) of FIG. 7) detected at the pin 3 of the microcontroller reaches the threshold set at the inverting input of the comparator K5, then the control signal PG is deactivated.
  • the voltage at capacitor C2 and at pin 21 of microcontroller MC the voltage at capacitor C3 is monitored at pin 20 of microcontroller MC. From these values, the current through the boost converter inductor L2 can be calculated by means of the module A of the microcontroller MC and, in dependence on these operating parameters, the memory contents of the data registers DR5, DR6 and DR7 in the module A program can be used to generate the control signal PG for the transistor V1 can be determined. In this way, a control loop is realized for the control of the transistor V1.
  • the invention is not limited to the embodiments described in detail above.
  • the invention can also be used to control the switching transistors of ballasts for the operation of high-pressure discharge lamps and of electronic transformers for the operation of low-voltage halogen incandescent lamps.
  • the device according to the invention designed as part of a microcontroller for alternately charging and discharging a charge store, to generate the frequency- or pulse-width-modulated control signals for the switching transistors of a full-bridge inverter or a push-pull inverter.

Landscapes

  • Circuit Arrangements For Discharge Lamps (AREA)
  • Discharge-Lamp Control Circuits And Pulse- Feed Circuits (AREA)

Claims (23)

  1. Ballast pour faire fonctionner au moins une lampe (LP1, LP2) électrique, qui a un onduleur, au moins un circuit de charge couplé à l'onduleur et ayant des bornes (X1 à X8) pour la au moins une lampe (LP1, Lp2) électrique, un circuit de commande du moyen (V2, V3) de commutation de l'onduleur et un circuit d'alimentation en tension continue de l'onduleur, le circuit de commande comprenant un microcontrôleur (MC) ayant un dispositif (G) de commande de modulation en largeur d'impulsions et/ou de commande en fréquence du moyen (V2, V3) de commutation de l'onduleur, le dispositif (G) de commande de modulation en largeur d'impulsion et/ou de commande en fréquence ayant
    - un dispositif (SQ1, SS1) de charge et de décharge alternées d'un accumulateur (C27) de charge,
    - des moyens de commande de ce dispositif (SQ1, SS1) pour commander les opérations de charge et/ou les opérations de décharge et
    - des moyens d'évaluation qui servent à évaluer la durée des processus alternés de charge et de décharge de l'accumulateur (C27) de charge et à produire, en fonction de cela, un signal de commande en fréquence et/ou un signal de commande de modulation en largeur d'impulsion pour commander les moyens (V2, V3) de commutation de l'onduleur,
    caractérisé en ce qu'il est prévu un diviseur (FT1) de fréquence ou un diviseur d'impulsion qui détecte sur son entrée la commutation du dispositif (SQ1, SS1) pour la charge et la décharge alternées d'un accumulateur de charge de la décharge à la charge ou de la charge à la décharge et qui divise le signal d'entrée en signaux de commande alternés des moyens (V2, V3) de commutation de l'onduleur.
  2. Ballast suivant la revendication 1, caractérisé en ce que le ballast a un dispositif de chauffage équipé d'un moyen (V4) de commutation pouvant être commandé pour l'alimentation des électrodes (E1 à E4) de lampe de la au moins une lampe (LP1, LP2) électrique en un courant de chauffage et le microcontrôleur (MC) a un comparateur (K1) qui compare l'état de charge de l'accumulateur (C27) de charge à une valeur témoin pour le chauffage des électrodes de lampe et qui sert à produire un signal de commande pour la modulation en largeur d'impulsion du moyen (V4) de commutation pouvant être commandé du dispositif de chauffage.
  3. Ballast suivant la revendication 2, caractérisé en ce que la valeur témoin peut être réglée au moyen d'une mémoire (DR4) d'écriture-lecture.
  4. Ballast suivant la revendication 2, caractérisé en ce que le microcontrôleur (MC) a un moyen (SR1) de synchronisation du moyen (V4) de commutation pouvant être commandé du dispositif de chauffage avec un moyen (V2) de commutation de l'onduleur.
  5. Ballast suivant la revendication 1, caractérisé en ce que
    - le circuit d'alimentation en tension continue a un survolteur pour corriger le facteur de puissance et/ou pour obtenir une absorption de courant du secteur aussi sinusoïdale que possible,
    - le microcontrôleur (MC) a un deuxième dispositif (SD2, SS2) de charge et de décharge alternée d'un deuxième accumulateur (C26) de charge,
    - le microcontrôleur (MC) a un deuxième moyen de commande de ce deuxième dispositif (SQ2, SS2) pour commander le processus de charge et/ou le processus de décharge et
    - le microcontrôleur (MC) a deux moyens d'évaluation qui servent à évaluer les laps de temps nécessaires pour faire passer le deuxième accumulateur (SQ2, SS2) de charge entre des états de charge différents et pour produire en fonction de cela un signal de commande de modulation en largeur d'impulsion et/ou un signal de commande en fréquence du moyen (V1) de commutation pouvant être commandé du survolteur.
  6. Ballast suivant la revendication 5, caractérisé en ce que le deuxième moyen d'évaluation a un premier comparateur (K2, K3) de comparaison de l'état de charge du deuxième accumulateur (C26) de charge à une première valeur de tension et un deuxième comparateur 'K4) de comparaison de l'état de charge du deuxième accumulateur (C26) de charge à une deuxième valeur de tension plus basse et en ce que le deuxième moyen de commande du deuxième dispositif (SQ2, SS2) pour la charge et la décharge alternées d'un accumulateur de charge à un moyen (FL1) de commutation qui sert à commuter le deuxième dispositif (SQ1, SS1 ; SQ2, SS2) de la charge à la décharge du deuxième accumulateur (C26) de charge lorsque est atteinte la première valeur de tension et à commuter le deuxième dispositif (SQ2, SS2) de la décharge à la charge du deuxième accumulateur (C26) de charge lorsque est atteinte la deuxième valeur de tension plus basse.
  7. Ballast suivant la revendication 6, caractérisé en ce que la première valeur de tension et la deuxième valeur de tension peuvent être réglées au moyen d'une mémoire (DR7) d'écriture-lecture.
  8. Ballast suivant la revendication 1 ou 5, caractérisé en ce que les dispositifs (SQ1, SS1 ; SQ2, SS2) de charge et de décharge alternées d'un accumulateur de charge ont, respectivement, une source (SQ1 ; SQ2) de courant réglable pour alimenter l'accumulateur (C27) de charge ou le deuxième accumulateur (C26) de charge en un courant de charge réglable et, respectivement, un puits (SS1 ; SS2) de courant réglable pour alimenter l'accumulateur (C27) de charge ou le deuxième accumulateur (C26) de charge en un courant de décharge réglable.
  9. Ballast suivant la revendication 8, caractérisé en ce que les réglages des sources (SQ1 ; SQ2) de courant réglables et des puits (SS1, SS2) de courant réglables peuvent être modifiés par rapport à un niveau (IR) de courant témoin en ayant respectivement une résolution d'au moins 8 bits.
  10. Ballast suivant la revendication 9, caractérisé en ce que le niveau (IR) de courant témoin pour le courant de charge et le courant de décharge peut être prescrit au moyen d'une résistance (R30) ohmique.
  11. Ballast suivant l'une ou plusieurs des revendications 1 à 10, caractérisé en ce qu'au moins un moyen (V2, V3) de commutation pouvant être commandé de l'onduleur et/ou le moyen (V4) de commutation pouvant être commandé du dispositif de chauffage et/ou le moyen (V1) de commutation pouvant être commandé du survolteur peut être activé ou peuvent être activés par un bit de statut pouvant être appliqué et pouvant être remis à l'état initial.
  12. Ballast suivant l'une des revendications 1 à 11, caractérisé en ce que le microcontrôleur (MC) a des interfaces (18, 19 ; 15, 16 ; 20, 21, 3) de détection des paramètres de fonctionnement de l'onduleur ou/et de la au moins une lampe (LP1, LP2) électrique ou/et du survolteur et a un dispositif (A) commandé par programme qui sert à évaluer les paramètres de fonctionnement et à déterminer des valeurs de réglage pour la commande des dispositifs (SQ1, SS1 ; SQ2, SS2) de charge et de décharge alternées d'un accumulateur de charge et/ou de détermination de la valeur témoin pour le chauffage des électrodes de lampe et/ou de détermination de la première ou de la deuxième valeur de tension.
  13. Ballast suivant l'une ou plusieurs des revendications 1 à 12, caractérisé en ce que le ballast a des bornes (J3, J4) et un moyen (DS) de communication avec un dispositif extérieur de commande et le microcontrôleur (MC) a des interfaces (5, 6) qui sont couplées aux bornes (J3, J4).
  14. Procédé pour faire fonctionner au moins une lampe (LP1, LP2) électrique à l'aide d'un ballast qui a un onduleur ayant un circuit de commande comportant un microcontrôleur (MC) du moyen (V2, V3) de commande de l'onduleur et au moins un circuit de charge couplé à l'onduleur et ayant des bornes (X1 à X8) pour la au moins une lampe (LP1, LP2) électrique, dans lequel à l'aide du microcontrôleur (MC) on alimente un accumulateur (C27) de charge en alternance en un courant de charge et en un courant de décharge et on évalue la durée des processus alternés de charge et de décharge de l'accumulateur (C27) de charge et en fonction de cela on produit un signal de commande en fréquence ou/et un signal de commande de modulation en largeur d'impulsion pour commander de façon alternée le moyen (V2, V3) de commutation de l'onduleur,
    caractérisé en ce que l'on détecte la commutation de la décharge à la charge de l'accumulateur (C27) de charge ou de la charge à la décharge de l'accumulateur (C27) de charge et on produit au moyen d'un diviseur (FT1) de fréquence ou d'un diviseur d'impulsion des signaux de commande pour commander de façon alternée le moyen (V2, V3) de commutation de l'onduleur.
  15. Procédé suivant la revendication 14, caractérisé en ce que l'on alimente les électrodes (E1 à E4) de lampe de la au moins une lampe (LP1, LP2) électrique en un courant de chauffage, le courant de chauffage étant réglé au moyen d'un moyen (V4) de commutation pouvant être commandé, en produisant les signaux de commande modulés en largeur d'impulsion du moyen (V4) de commande pouvant être commandé à l'aide d'un comparateur (K1) qui compare l'état de charge de l'accumulateur (C27) de charge à une valeur témoin pour le chauffage des électrodes de lampe.
  16. Procédé suivant la revendication 15, caractérisé en ce que l'on règle la valeur témoin en fonction de la puissance de chauffage souhaitée et on la mémorise dans une mémoire (DR4) d'écriture-lecture du microcontrôleur (MC).
  17. Procédé suivant la revendication 15, caractérisé en ce que l'on ferme le moyen (V4) de commutation pour la régulation du courant en chauffage en synchronisme avec un moyen (V2) de commutation de l'onduleur et en ce que la durée de fermeture du moyen (V4) de commutation pouvant être commandé est, pour la régulation du courant de chauffage, inférieure ou égale à la durée de fermeture du moyen (V2) de commutation de l'onduleur.
  18. Procédé suivant la revendication 14, caractérisé en ce que l'on règle la tension continue d'alimentation de l'onduleur au moyen d'un survolteur pour assurer une correction du facteur de puissance ou/et un prélèvement sinusoïdal du courant du secteur des signaux de commande de modulation en largeur d'impulsion et/ou des signaux de commande en fréquence du moyen (V1) de commutation pouvant être commandé du survolteur étant produits à l'aide du microcontrôleur (MC) en faisant passer un deuxième accumulateur (C26) de charge entre des états de charge différents et en évaluant les laps de temps de changement de charge du deuxième accumulateur (C26) de charge pour la production des signaux de commande de modulation en largeur d'impulsion et/ou des signaux de commande en fréquence du moyen (V1) de commutation pouvant être commandé du survolteur.
  19. Procédé suivant la revendication 18, caractérisé en ce qu'à l'aide d'un premier comparateur (K2, K3) on compare l'état de charge du deuxième accumulateur (C26) de charge à une première valeur de tension et à l'aide d'un deuxième comparateur (K4) on compare l'état de charge du deuxième accumulateur (C26) de charge à une deuxième valeur de tension plus basse, dans lequel, lorsque la première valeur de tension est atteinte, on met fin au processus de charge du deuxième accumulateur (C26) de charge et on fait débuter le processus de décharge du deuxième accumulateur (C26) de charge et dans lequel, lorsque la deuxième valeur de tension plus basse est atteinte, on met fin au processus de décharge du deuxième accumulateur (C26) de charge et on fait débuter l'opération de charge.
  20. Procédé suivant la revendication 19, caractérisé en ce que l'on règle la première valeur de tension ou/et la deuxième valeur de tension au moyen d'une mémoire (DR7) d'écriture-lecture.
  21. Procédé suivant la revendication 14 ou 18, caractérisé en ce que l'on produit le courant de charge au moyen d'une source (SQ1 ; SQ2) de courant et on règle l'intensité du courant au moyen d'une mémoire (DR1 ; DR6) d'écriture-lecture.
  22. Procédé suivant la revendication 14 ou 18, caractérisé en ce que l'on produit le courant de décharge au moyen d'un puits (SS1 ; SS2) de courant et l'on règle l'intensité du courant au moyen d'une mémoire (DR2 ; DR5) d'écriture-lecture.
  23. Procédé suivant l'une ou plusieurs des revendications 14 à 22, caractérisé en ce qu'à l'aide du microcontrôleur (MC) on contrôle des valeurs réelles de paramètres de fonctionnement de l'onduleur et/ou de la au moins une lampe (LP1, LP2) électrique et/ou du circuit d'alimentation en tension continue de l'onduleur et on les exploite pour commander les processus de charge ou de décharge des accumulateurs (C27 ; C26) de charge ou/et pour déterminer la valeur témoin de chauffage d'électrodes de lampe ou/et pour déterminer la première ou/et la deuxième valeur de tension.
EP01129890A 2001-01-23 2001-12-14 Microcontrôleur, alimentation à découpage et ballast électronique pour la mise en oeuvre d'au moins une lampe Expired - Lifetime EP1227707B1 (fr)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
DE10102940A DE10102940A1 (de) 2001-01-23 2001-01-23 Mikrocontroller, Schaltnetzteil, Vorschaltgerät zum Betrieb mindestens einer elektrischen Lampe und Verfahren zum Betreiben mindestens einer elektrischen Lampe
DE10102940 2001-01-23

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EP1227707A2 EP1227707A2 (fr) 2002-07-31
EP1227707A3 EP1227707A3 (fr) 2005-01-12
EP1227707B1 true EP1227707B1 (fr) 2007-09-05

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US (1) US6717374B2 (fr)
EP (1) EP1227707B1 (fr)
AT (1) ATE372659T1 (fr)
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DE (2) DE10102940A1 (fr)

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Publication number Publication date
CA2368897A1 (fr) 2002-07-23
EP1227707A2 (fr) 2002-07-31
US20020097008A1 (en) 2002-07-25
EP1227707A3 (fr) 2005-01-12
DE10102940A1 (de) 2002-08-08
ATE372659T1 (de) 2007-09-15
DE50112958D1 (de) 2007-10-18
US6717374B2 (en) 2004-04-06

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