Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L25/00—Baseband systems
  • H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
  • H04L25/40—Transmitting circuits; Receiving circuits
  • H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
  • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
  • H04L27/36—Modulator circuits; Transmitter circuits
  • H04L27/366—Arrangements for compensating undesirable properties of the transmission path between the modulator and the demodulator
  • H04L27/367—Arrangements for compensating undesirable properties of the transmission path between the modulator and the demodulator using predistortion
  • H04L27/368—Arrangements for compensating undesirable properties of the transmission path between the modulator and the demodulator using predistortion adaptive predistortion

Definitions

• the present invention generally relates to the field of data communications. More specifically, the present invention is for a method and apparatus for adaptively predistorting a signal before it is transmitted from a transmitter to a receiver of a communication system.
• Digital data communications systems are commonly used to transmit and/or receive data between remote transmitting and receiving locations.
• a central facet of any data communications system is the reliability and integrity of the data which is being communicated.
• the data which is being transmitted from the transmitting location should be identical to the data which is being received at the receiving location.
• the data which is received at the receiving location has oftentimes been corrupted with respect to the original data that was transmitted from the transmitting location.
• Such data communication errors may be attributed in part to one or more of the transmission equipment, the transmission medium or the receiving equipment. With respect to the transmission medium, these types of data errors are usually attributed to the less than ideal conditions associated with the particular transmission medium.
• the transmission medium which is typically air
• the transmission medium often suffers from atmospheric and other effects that tend to degrade the data being transmitted.
• Some of these non-ideal conditions may be modeled and taken into account in order to compensate for and thereby reduce or possibly eliminate any deleterious effects resulting therefrom.
• signal attenuation is a function of the distance that the data signal must propagate through the atmosphere.
• Other types of non- idealities associated with an air or atmospheric transmission medium are often highly random events which may not be modeled a priori and thus may not be compensated for or eliminated.
• the transmission of data over interconnecting wires also suffers from several noise and attenuation phenomena. For example, wires exhibit frequency dependent attenuation characteristics. Additionally, wires are subject to thermal or other types of noise.
• error detection is the use of a parity bit associated with each block of data to indicate whether the particular block contains an odd or even number of 1 bits.
• This is a very simple scheme which has numerous disadvantages. It is a simple type of error detection scheme which is capable of accurately detecting up to one bit error per data block.
• the use of a parity bit cannot detect the occurrence of two bit errors in a data block, since this is not even detected as a parity violation.
• the use of a parity bit only detects errors; it cannot correct errors.
• the receiving location typically requests retransmission of the particular data block from the transmitting location.
• error correction scheme commonly used in data communications systems is the use of redundant data transmissions and a voting circuit at the receiving location.
• the data being transmitted is repeated a number of times, such as five.
• all five data blocks are received and processed by a voting circuit which compares the five received versions of each data bit and determines the bit to be a 1 or 0 based on the voting consensus.
• a voting circuit which compares the five received versions of each data bit and determines the bit to be a 1 or 0 based on the voting consensus.
• Trellis coded modulation involves the use of forward error correction in a transmitter and corresponding decoding in a receiver to correct data errors which occur during transmission.
• the signal transmitted by the transmitter includes some form of memory or history so that a current signal is dependent on previous signals. This is achieved by limiting the choices for a next possible signal based on the current and previous signals. In this way, by knowing a sequence of previously received signals, the receiver obtains some assistance in determining the authenticity of a currently received signal. At the receiver, the sequence information of the received signals is used to select a next signal most likely to have been transmitted.
• Both the transmitter and receiver know the predetermined, limited choices for a next signal based on the previous signals. Thus, even if a signal gets corrupted during transmission, the receiver may be able to still identify the correct, intended signal.
• TCM method is disclosed in U.S. Patent No. 4,980,897 to Decker et al., entitled MULTI- CHANNEL TRELLIS ENCODER/DECODER, the contents of which are incorporated herein by reference.
• an (n,k,d) binary block code is a set of 2 k binary codewords of block length n and minimum distance d (i.e., coding distance).
• the transmitted data is partitioned into binary blocks of length k, then each block is mapped into a binary codeword of length n, which is then modulated and transmitted through the channel.
• Narrowband transmission formats such as frequency shift keying (FSK) or amplitude shift keying (ASK) are somewhat immune to frequency dependent attenuation, and thus may suffer little or no distortion.
• FSK frequency shift keying
• ASK amplitude shift keying
• Wideband transmission formats such as spread spectrum are less susceptible to the signal degradation caused by a narrowband attenuation null.
• the spread spectrum signal experiences more distortion due to frequency dependent attenuation.
• a conventional narrowband signaling format is susceptible to attenuation while a conventional wideband signaling format is susceptible to distortion.
• Another type of signal enhancing approach is to compensate, by appropriate filtering of the received signal, for errors due to frequency dependent differences in attenuation, time delay, or both, introduced by either the transmitter, the transmission medium and/or the receiver. In this way, the deleterious effects, such as intersymbol interference which causes bit errors, are eliminated and the received signal is restored to its original level and quality.
• This approach is generally referred to as equalization.
• the equalizer In most practical cases, the channel characteristics are unknown and may vary with time. Hence, the equalizer must be updated with each new channel connection, as in the case of a voice-band modem operating in a switched network, and must also adapt the filter settings automatically to track changes in the channel with time.
• equalizers There are two general types of equalizers. In preset equalizers, a training sequence of data bits is transmitted and compared at the receiver with a locally generated sequence. The resulting error voltages are used to adjust the equalizer filter tap gains to optimum settings (which result in minimum distortion). In adaptive equalizers, the tap gain adjustments are derived directly from the transmitted data bits via decision directed feedback of the equalizer output errors, to minimize these output errors.
• a set of parameters is established and used to emulate the effects of the distorting communication channel.
• a system is disclosed in U.S. Patent No. 5,285,474 to Chow et al., entitled METHOD FOR EQUALIZING A MULTICARRIER SIGNAL IN A MULTICARRIER COMMUNICATION SYSTEM, the contents of which are incorporated herein by reference.
• the disclosed process is an iterative process which is repeated until a predetermined convergence condition is met. Convergence is determined by comparing the equalized received signal (using the estimated parameters) with a local replica (at the receiver) of the training sequence.
• An alternative to equalization is predistortion, in which the distortive characteristics of the transmitter and/or channel are measured, and opposite, compensating distortions are introduced into the data entering the transmitter, so that the net effect at the receiver is a signal that has no net distortions, thereby eliminating the message errors introduced by the naturally occurring distortions.
• HPA high power amplifier
• Most high power amplifiers generally have a linear operating region (at low power) which progresses into a nonlinear region (at higher output power).
• the transmitter is operated at a higher power level.
• operating at the higher power level introduces nonlinearities associated with the amplifier.
• One approach to eliminating the nonlinearities at higher power levels is to use a higher power amplifier; however, this results in a more complex and costly circuit design.
• a higher power amplifier consumes more power, generates more heat and radio frequency interference (RFI), and occupies more circuit board space.
• RFID radio frequency interference
• Karam and Sari describe a technique for predistorting the in-phase and quadrature components of the location of each quadrature amplitude modulated (QAM) data symbol that is input to a nonlinear power amplifier in order to compensate for the warping of the constellation by the nonlinearity, and also to compensate for the spreading of the transmitted symbols into a dispersed cluster at each ideal constellation point, caused by intersymbol interference due to filtering of each transmitted symbol within the transmitter.
• This filtering causes the apparent constellation coordinates of each symbol to be somewhat varied by the values of preceding and following symbols. Since the input amplitude of each pulse is thereby altered, the amount of nonlinear distortion for pulses at any ideal symbol point also varies with the values of the preceding and following symbols.
• each constellation point may take on M " ' different predistorted values.
• a memory must be provided to store the required predistortion for each of these values. Since the size of this memory becomes impractically large for large constellations, Karam and Sari recommend a technique in which quadrant symmetry and partitioning of the ideal symbol points into groups are used to reduce the total memory requirement. They show that significant predistortion processing gain can be achieved for up to 256-QAM (and possibly higher) modulations with reasonable memory requirements.
• Karam and Sari's technique functions to compensate for transmitter nonlinearities, including both AM-AM (amplitude input distortions cause amplitude output distortions) and AM-PM (amplitude input distortions cause phase output distortions) conversion, it does not at all address the many types of distortions that can occur in the transmission channel and the receiver, many of which are time-varying, and all of which can affect the output symbol error rate.
• These distortions include offset biases, in-phase/quadrature gain mismatches, lock angle errors, quad angle errors, nonlinear distortions, phase noise, multipath, interference, and crosstalk.
• the technique presumes that the nonlinear gain and phase characteristics of the HPA remain invariant, so they can be measured and modeled initially, and then the measured scaling coefficients can be stored in memory to be used to appropriately predistort the input data.
• HPA nonlinearity curves can change with time, due to the effects of component aging, voltage changes, and temperature and other environmental variations. Therefore, it is necessary to periodically check and recalibrate the HPA nonlinearity parameters. While this is possible at the originating transmitter, it is much more difficult, if not impossible, at remote repeaters and satellite transponders.
• the second limitation of the Karam-Sari technique is that it requires knowledge of the in-phase and quadrature amplitudes of the symbols in the data stream at the input to the predistorter preceding the HPA. While this information is readily available at the originating transmitter, it is not so readily available at intermediate repeaters and transponders. There, the signals must first be demodulated and the in-phase and quadrature amplitudes of the symbols must be measured for input to the predistorter memory. The predistorted signals must then be remodulated for input to the HPA. These steps can add a significant penalty in added equipment size, weight, power, and cost, especially in the case of satellite transponders.
• the present invention provides an improved method and apparatus for adaptively predistorting a signal before it is transmitted.
• the adaptive predistortion of the present invention is accomplished at reduced complexity and cost, and is thus especially valuable in situations where the complexity and cost of the receiver must be kept as low as possible.
• the adaptive predistortion compensates for voltage offsets (biases), gain mismatches, lock and quad angle errors, and channel and receiver nonlinearities. Multipath and frequency-dispersive channel effects may be compensated for by using an adaptive equalizer following the receiver. In some systems, interference and crosstalk may be minimized by tuning the transmitter to carrier frequencies where these phenomena are minimized.
• the receiver accumulates certain statistics regarding the data sample values in each received symbol cluster in the signal's amplitude-phase constellation.
• control parameters and symbol predistortion values are calculated.
• Distortion phenomena controllable by predistortion are then corrected in an adaptive predistorter following a memory-based predistorter at the transmitter.
• the control parameters which were computed using a data processor at the transmitter (to lower receiver cost), are sent to the receiver over a low data rate forward control channel.
• the calculations on the statistical information accumulated at the receiver may be carried out at the receiver itself.
• a two-dimensional constellation of unique symbols, each representing a unique sequence of bits, may be formed by quadrature amplitude modulation (QAM), in which waveforms with two orthogonal phases (denoted in-phase and quadrature) are assigned one of M possible amplitudes in each coordinate.
• QAM quadrature amplitude modulation
• M is usually taken as a power of two, and the symbol amplitudes in each coordinate are usually equally spaced and symmetric about the origin. If M is an even power of two, we obtain a square QAM constellation. If M is an odd power of two and some corner symbol positions are eliminated, we obtain what is called a cross QAM constellation. For quadrature partial response (QPR) coding, M is not a power of two, but is instead an odd integer.
• QPR quadrature partial response
• M is not a power of two, but is instead an odd integer.
• the constellations used in carrierless AM/PM (CAP) modulation look like those of the modulation schemes discussed above, however, they are formed at baseband without modulation onto a carrier wave. The invention described herein can be applied to all these types of signals. With minor modifications, the present invention can also be applied to circular phase-shift keyed constellations and other constellations with symbols arranged on a non-square grid, and extensions to these constellation types are incorporated within the scope of this invention.
• a demodulator is used to determine which symbol was transmitted by dividing the constellation region into cells with boundaries equally spaced between adjacent ideal symbol locations. Comparison of the coordinates of the received symbol with the cell boundaries identifies the received symbol. The receiver then translates the identified symbol into the corresponding bit sequence associated with that symbol. Noise, intersymbol interference (LSI), and multipath will cause the positions of the received symbols to be smeared into dispersed clusters centered on the ideal coordinates of the transmitted symbols. Other impairments, such as biases, gain imbalances, lock and quad angle errors, and nonlinear effects, will shift the centers of the received symbol clusters from their ideal locations in predictable manners.
• LSI intersymbol interference
• Other impairments such as biases, gain imbalances, lock and quad angle errors, and nonlinear effects, will shift the centers of the received symbol clusters from their ideal locations in predictable manners.
• the analysis of the received signal is performed using constellation analysis (discussed in detail below).
• the constellation analysis is performed on statistical data concerning the received signal, which is measured at the receiver.
• the constellation analysis which may be performed at either the transmitter or the receiver, yields the distortion characteristics of the received signal. This information is then provided to the transmitter in order for the transmitter to appropriately compensate the outgoing signal before it is transmitted. It is generally more advantageous to compensate the signal at the transmitter, rather than at the receiver, because by the time the signal is received at the receiver, the signal may not be recoverable.
• the calculations may be performed at either the transmitter or the reciever, when minimizing the cost of the receiver is important, it is better to process the statistical data for the constellation analysis at the transmitter.
• Figure 1 is a block diagram of the predistortion system of the present invention.
• Figures 2a, 2b, and 2c are illustrations of various modulation schemes.
• Figure 3a is an illustration of the effects of positive lock angle error.
• Figure 3b is an illustration of the effects of negative lock angle error.
• Figure 4a is an illustration of the effects of positive quad angle error.
• Figure 4b is an illustration of the effects of negative quad angle error.
• Figure 5 is an illustration of the effects of combined lock angle error, quad angle error, and gain mismatch.
• Figure 6 is a block diagram of an alternative embodiment of the predistortion system of the present invention, which globally predistorts the signal by adjustment of a small number of transmitter parameters, rather than individually modifying the location in the constellation of each transmitted symbol.
• Figure 7 is an illustration of the technique of constellation decomposition into a set of simple 4-QAM subconstellations, each of which can be individually predistorted in a quasi-linear manner.
• the symbol source, or modulator, 12 supplies the data to be transmitted. This data may be in QAM or any other of the above mentioned formats.
• the modulated symbol stream is first input to a predistorter with memory 14, which may be implemented in accordance with the teachings of Karam and Sari discussed above.
• the data output from the predistorter with memory 14 is then input to an adaptive predistorter 16.
• the output of the adaptive predistorter 16 is then input to a power amplifier 18.
• the predistorter with memory 14 is generally used if the power amplifier is operated in a nonlinear mode, in which case it functions to compensate for the nonlinear effects of the power amplifier 18.
• test sequence In the case of a nonlinear power amplifier 18, then from time to time a test sequence of symbols is sent directly to the power amplifier 18, bypassing the adaptive predistorter 16 by means of switch 20.
• the test sequence must be re-sent at a rate comparable with the rate at which the gain of the power amplifier changes due to temperature or voltage variations.
• the test sequence can either be sent as a continuous symbol stream that occasionally interrupts the data stream, or preferably as selected symbols that are part of the overhead bits that accompany each frame of data. The frequency at which the test sequence is utilized depends upon the particular operating environment.
• the adaptive predistorter 16 may be used to compensate for the nonlinear operation of the power amplifier 18.
• the signal propagates through the forward transmission channel 22 to the receiver 24.
• the statistics regarding the positions, sizes, and shapes of the symbol clusters are measured on a periodic basis, and are sent via the reverse control channel 26 to the parameter processor computer 28 at the transmitter.
• the statistics are measured at the receiver 24 because transmitting the raw in-phase and quadrature (I-Q) measurements on all symbols used in the statistical analysis over the narrowband reverse control channel would greatly exceed the capacity of that channel.
• I-Q in-phase and quadrature
• the processor 28 calculates the parameters of the constellation distortions that arise in the forward channel and receiver. These parameters are used to adjust the I- Q coordinates of the transmitted symbols in the adaptive predistorter 16 so as to minimize the resulting received distortions. For some types of distortions, such as linear constellation offsets due to voltage biases in the receiver or gain imbalances (giving unequal scales in the I and Q coordinates) in the receiver, the values of the corrective voltages may be sent to the receiver 24 in the overhead bits on the forward transmission channel 22 to reduce the effects of these constellation offsets and gain imbalances. Finally, an adaptive equalizer 30 mitigates the effects of multipath propagation and frequency dependent (dispersive) phenomena, such as Rayleigh fading, over the forward transmission channel 22. The corrected symbol data then goes to the demodulator 32 where the information bit stream is extracted.
• multipath propagation and frequency dependent (dispersive) phenomena such as Rayleigh fading
• FIGS. 2a, 2b and 2c therein are illustrated the configurations of 16-QAM (quadrature amplitude modulation), 32-Cross QAM, and 25- QPR (quadrature partial response) symbol constellations.
• the constellation point (i,j) will ideally be located at coordinates (x , y - (i,j), where
• x tJ and y are the estimated model coordinates for the center of cluster (i,j).
• a x and A y are constants which represent the constellation offsets in the x and y directions, respectively.
• B xx and B yy are the linear gains, or column and row spacings, respectively (i.e., the first order change in the JC and y directions).
• B xy is the linear run per unit rise, or horizontal skew, in the column position as a function of row (i.e., the first order change in the x direction as a function of the y coordinate value at that point).
• B yx is the linear rise per unit run, or vertical skew, in the row position as a function of column (i.e., the first order change in the y direction as a function of the x coordinate value at that point).
• C xx , C yy , D xx , and D yy are the corresponding nonlinear gains, or variations in the column and row spacings. Specifically, C xx is the second order change in the x direction; C yy is the second order change in the y direction; D xx is the third order change in the x direction; and D yy is the third order change in the y direction.
• C xy , C yx , D xy , and D yx are the corresponding nonlinear (second and third order) skews in the column and row positions.
• B xx B
• )y l
• all other model parameters are zero '
• k denotes the individual symbols within each cluster
• the problem of estimating the parameters of the equations that provide a best least-squares fit to the available data samples (measured constellation points in phase space) is one of multiple linear regression, since the estimates depend linearly on the parameters.
• the standard statistical procedure for estimating such parameters from an overdetermined set of data i.e., there are more data samples than parameters
• Matrix notation will be used to set up and solve the corresponding normal equations for the parameters in Equations (3).
• the parameter matrices P x and P y and the matrix of coefficients Cij for cluster (i,j) are defined as follows:
• the composite coefficient, data, and error matrices C, D x , D y , E x , and E y for the entire constellation are defined as the following partitioned matrices
• Equations (17), (18), and (19) are summed over the same range in i, in j, and in k, B x and By remain 1x7 matrices.
• the parameter matrices P x and Py are obtained by inserting Equations (17), (18), and (19) into (16a) and (16b), and evaluating the results with the actual data coordinate values This gives the coefficients in
• Equations (3) The components of P x and Py, as given by Equations (4), are then inserted into Equations (3) to evaluate the best-fit model coordinates, (x, J ,y lJ ), of each cluster in the constellation.
• all transmitted symbols may be pre-shifted by amounts -A x and -A at the transmitter to accomplish the same effect.
• translational offsets in the constellation (A x and A ⁇ ) are typically due to misadjustments in the reciever, it is better to correct for such receiver errors at their source, within the receiver 24.
• the linear gain ratio between the x and y coordinates is
• this gain ratio is equal to the net linear I-Q gain mismatch of the modulator (transmitter) and demodulator (receiver).
• the I-Q gain mismatch is not so simply related to the x-y gain ratio. If the gain imbalance is assumed to be all in the receiver 24 (i.e., no distortions are assumed to be introduced by the transmitter and the forward transmission channel 22) and R G > 1, then a command may be sent to the receiver 24 to either decrease the in-phase gain by a factor of R G , or increase the quadrature gain by the same factor. If R G ⁇ 1 then a command may be sent to the receiver 24 to either increase the in-phase gain by a factor of R G , or decrease the quadrature gain by the same factor.
• this gain ratio R G is defined to depend upon only the linear terms in Equations (3).
• B xx and B yy are the spacings between adjacent columns and rows, respectively, in the constellation, assuming that all columns and rows are uniformly spaced (negligible nonlinearities C xx , D xx , etc.). If the nonlinear terms are non- negligible, the column and row spacing will be a function of amplitude.
• the second and third order gain ratios may be defined as follows:
• FIGs 3a and 3b are illustrations of lock angle error in the presence of a gain mismatch for the simple case of QPSK (i.e., 4-QAM) modulation.
• the angle of phase rotation (the lock angle error) ⁇ is here defined as positive in the clockwise direction.
• Figure 3a shows the geometry for positive ⁇
• Figure 3b shows the geometry for negative ⁇ .
• B xx is the measured spacing between columns of constant I coordinate, as projected upon the x axis.
• B yy is the measured spacing between rows of constant Q coordinate, as projected upon the y axis.
• B xx is measured positive to the right, and B yy is measured positive upward.
• B xy is the linear run per unit rise measured between successive rows of clusters.
• B yx is the linear rise per unit run measured between successive columns of clusters. It should be noted that B xy is measured positive to the right in going from a lower to an upper row. Similarly, B yx is measured positive upward in going from a left to a right-hand column.
• FIG. 3a and 3b Other angles shown in Figures 3a and 3b include , the angle from the rightmost cluster in the rotated constellation to the positive x axis, and ⁇ , the angle between the diagonals of the rotated constellation, measured between the right hand clusters having a common I coordinate, and originating at the rightmost cluster.
• Quad angle error measures the degree of departure of the I and Q coordinates of the modulator and demodulator from orthogonality in the x-y plane, indicating that the two modulator or demodulator branches are not strictly in phase quadrature.
• the output of the threshold decision slicer in one coordinate at the demodulator will depend on the value of the other coordinate as well, leading to decision errors in determining which symbol was actually transmitted.
• the constellation is "stretched” along one diagonal direction and compressed along the other diagonal direction.
• lock angle error results in a fixed rotation of the entire constellation, without "stretching".
• Figures 4a and 4b are illustrations of quad angle error in the presence of a gain mismatch, but no lock angle rotation, in the simple case of QPSK modulation. Again, all results in this section will apply equally well to all QAM and QPR modulations of any order.
• the coordinate gains B xx and B yy and the skews B xy and B yx are defined with the same sign conventions as noted previously for Figures 3a and 3b. Also, the angles ⁇ x and ⁇ y , defined previously, have the same sign convention as before, with positive being in the clockwise.
• the constellation In the absence of quad angle error, the constellation is rectangular (square if there is no gain imbalance between the I and Q components). The corner angles of a rectangle are, of course 90 degrees. If there is a quad angle error, the constellation is no longer rectangular, and the corner angles are either greater or less than 90 degrees.
• ⁇ denote the corner angle about the principle diagonal (the diagonal oriented upward to the right).
• the quad angle error ⁇ is the amount by which /departs from 90 degrees.
• the quad angle error ⁇ is related to the angles ⁇ x and ⁇ by
• Figure 5 is an illustration of combined lock angle error, quad angle error, and gain mismatch which shows the effect of a quad angle error superimposed upon a rotated rectangular QPSK constellation with lock and gain mismatch errors.
• the x and y gains and skews, B mn derived from the regression analysis of the observed data in the constellation model of Equations (3), indicate the spacings of the clusters in the distorted constellation.
• the B mn are defined to be positive upward and to the right, so that in
• Equation (28) the lock angle error is obtained from Equation (28)
• the gain ratio is obtained from Equation (29)
• the quad angle error is obtained from Equation (26). Since a non-zero lock angle error generally indicates a phase misadjustment in the carrier- recovery circuit of the receiver demodulator 32, so that the demodulated symbol phases are rotated by ⁇ relative to the transmitted phases, this misadjustment can be corrected by sending a command to an adjustable phase shifter (located within the receiver 24) at the output of the carrier recovery circuit (located within the receiver 24) to shift the carrier phase by - ⁇ , thereby rotating the entire constellation by - ⁇ .
• an adjustable phase shifter located within the receiver 24
• the same effect could be achieved by shifting the phase of the carrier oscillator in the transmitter modulator by - ⁇ . This would rotate the constellation into alignment with the symbol decision slicing circuitry in the demodulator equally well as by doing it in the carrier recovery circuit.
• a non-zero quad angle error indicates a phase error in the quadrature phase shifter in either the modulator or demodulator, so that the two branches of either the modulator or demodulator are not in phase quadrature, i.e., 90° apart in phase.
• a quad angle error of ⁇ in the modulator can be corrected by changing the modulator quadrature phase shifter by - ⁇ .
• a quad angle error of ⁇ in the demodulator can be corrected by either sending a command to the demodulator to change the demodulator quadrature phase shifter by an angle of - ⁇ , or by predistorting the transmitted constellation by changing the modulator quadrature phase shifter by - ⁇ . If quad angle errors are present in both the modulator and demodulator with a combined error of ⁇ , then they can be simultaneously corrected by changing the modulator quadrature phase shifter by - ⁇ . In general, it cannot be determined which device is responsible for the quad angle error without conducting separate special tests on each piece of equipment. Consequently, it is recommended that all quad angle errors be corrected by rotation of the modulator quadrature phase shifter by - ⁇ .
• Gain imbalances may occur in the modulator 12 prior to combining the in-phase and quadrature channels, or in the demodulator 32 after separating these channels.
• Gain imbalance in the modulator 12 can be eliminated by providing an adjustable linear amplifier in one of the channels, e.g., the quadrature channel, prior to combining, and by varying this gain in the direction to bring the linear gain ratio R G to 1.
• gain imbalances can occur in either or both the modulator 12 and demodulator 32, and since the source of the imbalance cannot be determined without conducting special tests on each device, it is recommended that all gain imbalances be corrected at the modulator 12. Since the lock and quad angle errors and the gain imbalance are all due to different phenomena, and since all are calculated simultaneously from the constellation measurements, the corrections for all effects can be implemented directly and simultaneously. All measurements, calculations, and corrective adjustments should be repeated iteratively until the angle errors and gain imbalance are made acceptably small.
• the control output of the parameter processor 28 in Figure 1 can instead be routed to the modulator 12.
• compensation for the various errors described above is carried out in the modulator 12 instead of the separate adaptive predistorter 16, which is no longer necessary. This is shown in Figure 6.
• any symmetric constellation may be treated as a set of nested QPSK subconstellations, as shown in Figure 7 for a 16-QAM constellation with I-Q gain mismatch.
• the numbers in parentheses in Figure 7 denote the (i,j) coordinates of the symbol clusters.
• the single numbers denote the subconstellations.
• all symbol clusters have the same nominal distance from the I axis and from the Q axis.
• Each elemental QPSK subconstellation (numbered k in Figure 7) may thus be represented by a linear model:
• Equations 33a and 33b correspond to Equation 3, but with only the constant and first order terms.
• d' 3/2
• i' and j' take on values in the set ⁇ 1,2 ⁇
• (i' - d') and (j' - d') take on values in the set ⁇ — j - > • _ ⁇ ⁇ •
• the coordinates ( x /k >y'y k ) m tn e subconstellation of course correspond to a set of coordinates [x IJ ,y lj ) in the full constellation.
• Equations (3), (34) and (35) expressions for the quantities E mn may be obtained for each subconstellation using Equations 3. These E mn expressions are expressed using the coefficient parameters of Equations 3 to relate the E mn coefficients to the coefficient parameters in Equations 3. For example, for subconstellation 1 in Figure 7, the following relationship between the coefficient parameters is obtained:
• E mn are independent of the second-order coefficients C mn .
• the E mn depend on the B mn and D mn with the same subscripts m and n.
• Equations (40), (41 ) and (42) give, respectively, the lock angle error, quad angle error, and gain imbalance for each subconstellation for the entire transmission chain, including the transmitter, receiver, and all intermediate repeaters and transponders.
• the ideal gain ratio for a subconstellation depends on its relation to the total constellation.
• the ideal gain ratios for Subconstellations 1 and 2, which are both square, are both equal to 1
• the ideal gain ratios for Subconstellations 3 and 4 which are both rectangular, are 3 and 1/3, respectively. Since the ideal gain ratio of a symbol depends on its position in the constellation, the gain ratio cannot be corrected by adjusting an amplifier gain in the source transmitter modulator since this would apply the same gain to all transmitted symbols.
• the I and Q coordinates of each symbol must be scaled according to the desired gain ratio for the subconstellation, in the adaptive predistorter, to bring the gain ratio of the symbol (which is equivalent to the tangent of the vector from the origin to the symbol location) to its ideal value, R Q , where k denotes the particular subconstellation to which the symbol belongs.
• R Q the gain ratio of the symbol
• k denotes the particular subconstellation to which the symbol belongs.
• the position of the symbol must be rotated about the origin by an angle of — ⁇ k to correct for the subconstellation lock angle error and associated nonlinear AM-PM conversion distortion.
• the Q axis of the subconstellation to which the symbol belongs must be skewed by an angle of - ⁇ ⁇ to compensate for the subconstellation 's quad angle error.

Landscapes

• Engineering & Computer Science (AREA)
• Computer Networks & Wireless Communication (AREA)
• Signal Processing (AREA)
• Physics & Mathematics (AREA)
• Spectroscopy & Molecular Physics (AREA)
• Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
• Transmitters (AREA)
• Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
• Amplifiers (AREA)

Applications Claiming Priority (3)

Publications (1)

Family

ID=25100366

Family Applications (1)

Country Status (8)

Cited By (1)

Families Citing this family (23)

Family Cites Families (4)

• 1997
  • 1997-12-24 AU AU57280/98A patent/AU738870B2/en not_active Ceased
  • 1997-12-24 WO PCT/US1997/024194 patent/WO1998028888A1/en not_active Application Discontinuation
  • 1997-12-24 JP JP52910798A patent/JP3993640B2/ja not_active Expired - Fee Related
  • 1997-12-24 CA CA002274522A patent/CA2274522A1/en not_active Abandoned
  • 1997-12-24 EP EP97953556A patent/EP0951769A1/en not_active Withdrawn
  • 1997-12-24 KR KR1019997005673A patent/KR20000062279A/ko not_active Application Discontinuation
  • 1997-12-24 IL IL13026497A patent/IL130264A0/xx unknown
• 1999
  • 1999-06-23 NO NO993128A patent/NO993128L/no not_active Application Discontinuation

Non-Patent Citations (1)

Also Published As

Similar Documents

Legal Events