CN1457127A - Active power filtering method with inversing capacitor regulation and branch impendance controlled decoupling - Google Patents

Abstract

First, the decoupling controlled by frequency domain is realized by using a group of band-pass filtering. Next, the resistance in the filtering branch circuit is negative feedback controlled in order to adjust voltage of the inversion capacitance making it keep at setting value of voltage. Then, the method of parameter adaptation is utilized to make the reactance at concerned ripple frequency point of the filtering branch circuit become zero so as to restrain ripple. Finally, the decoupling of resistances and reactances in filtering branches including two phase-shift filtering circuits with 90-degree phase shift is controlled so as to obtain the reference wave for pulse-duration modulation. The invented method is no need of DC power source. The invention solves phase-shift problem caused in each step in controlling procedure.

Description

The active power filtering method of inversion capacitance adjustment and branch road impendance controlled decoupling

Technical field:

The present invention relates to the active power filtering method of a kind of inversion capacitance adjustment and branch road impendance controlled decoupling, this method can maintain the operating voltage in order to the inversion electric capacity that substitutes DC power supply of single-phase electricity die mould inverter circuit DC side, can control simultaneously the reactance of filter branch, and be zero to suppress and eliminate the low-frequency ripple noise by making the reactance value of filter branch in the ripple frequency band of being concerned about.The invention belongs to the direct current supply technical field.

Background technology:

The Ripple Noise that comprises a large amount of low-frequency ranges by direct-current transmission voltage that the AC signal rectification is obtained or DC power supply voltage.These Ripple Noise will increase facility load on the DC transmission line of electric power system, produce electric energy loss, and its electromagnetic radiation is also with interference communications equipment.And the useful signal in the meeting of the Ripple Noise in the DC power supply interference load equipment has a strong impact on the normal operation of equipment.The method of traditional inhibition and elimination Ripple Noise is to adopt the filter circuit that is made of passive devices such as some resistance, inductance and electric capacity, and this is called as the passive filtering method.Along with the development of power electronic technology, the various devices that comprise power electronic device are applied to electric filtering, and this filtering method is called as active power filtering.Active power filtering can overcome that the traditional passive filtering parameter is inaccurate, the variation that can't adapt to frequency and load, easily with shortcoming such as system generation vibration.And active power filtering method can reach better filtering performance.In the bigger occasion of power, adopt active electric filter device can also reduce floor space, reduce the filtering cost.

" the electric power system journal of Institute of Electrical and Electronics Engineers in May, 1998, (IEEE Transactions on PowerSystems) " the 13rd volume the 2nd is interim has delivered one piece of exercise question for " bank is proposed the control principle analysis of the active dc filter that forest-road nurse current conversion station is adopted in this bank HVDC (High Voltage Direct Current) transmission system, (Analysis on the Control Principle of the Active DCFilter in the Lindome Converter Station of the Konti-Skan HVDC Link) " article.In this piece article, the author has described a kind of active electric filter device and control method thereof of the low-frequency ripple noise that can the filtering rectifier produces, and its theory diagram as shown in Figure 1.In existing this active electric filter device and control method thereof, after the direct voltage that comprises Ripple Noise inserts the input port of filter circuit, at first arrive output port through the smoothing reactor that seals in the circuit, output port is connecting passive filtration unit and the former limit of the coupling transformer that is in series with it between smoothing reactor and loop electrode, DC power supply inserts the secondary of coupling transformer by single-phase electricity die mould inverter circuit.The control method of existing this active electric filter device at first will be measured the output port current i by current transformer _Out, filtering i then _OutIn flip-flop and noise, extract ripple current i _lRipple current i _lObtain M signal i through comb filtering or the filtering of groove shape again _rRejection frequency is f if desired _iThe ripple composition, the transfer function F of selected comb filtering or the filtering of groove shape _a(s) should have approaching ± j2 π f _iTwo limits.Thus, transfer function F _a(s) amplitude-frequency response is at frequency f _iLocate also will level off to infinity.Then, M signal i _rHandle the reference wave signal u that obtains pulse-width modulation through system balance _a, reference wave signal u _aProduce control impuls through pulse-width modulation, control wave by power amplification after in order to drive single-phase electricity die mould inverter circuit.So, obtain reference wave signal u on the former limit of coupling transformer _aIn the voltage signal of low-frequency ac composition amplitude after amplifying.If supposing the input voltage ripple of whole filter is 0, and with reference wave signal u _aAs unique driving source, again with filter in current i that output port obtained _LaIn response, then can obtain the transfer function G of this input-output system according to circuit parameter _hThe transfer function characteristics of designed system balance part need satisfy in the document $F_C=\frac{1}{G_h}.$ Existing this active power filtering system transter model is shown in Figure 2.Input signal i among Fig. 2 _LhBe as reference wave signal u _aBe 0, the ripple voltage of input port is done the time spent separately, at the ripple current of filter output port.The transfer function of whole filtering control is: $F_S\left(s\right)=\frac{i_l}{i_{\mathrm{lh}}}=\frac{1}{1+F_a}$ Because F _a(s) limit is F _s(s) so zero point is if the transfer function F of comb filtering or the filtering of groove shape _a(s) have and approach ± j2 π f _iLimit, then | F _s(± j2 π f _i) | will level off to 0, promptly filtering control system can blanketing frequency be f _iLow-frequency ripple.

Existing this active power filtering method need obtain system transter G _h, and need carry out system balance according to this and handle.The design of system balance processing section depends on circuit parameter, and particularly the system balance processing procedure also needs corresponding change when load changes.This has strengthened the difficulty that system balance is handled, and causes the control vibration easily, and can reduce the effect that ripple suppresses.Filtering Processing in the existing this active power filtering method has partly adopted comb filtering or groove shape filtering method, thereby makes the transfer function F of Filtering Processing _a(s) limit approaches ± j2 π f _iBut the limit of transfer function is more near the imaginary axis, and then Filtering Processing process itself just is difficult to stable the realization more.The measurement of output port ripple current need filtering occupies larger proportion from the output port electric current flip-flop.Particularly when ripple is necessarily suppressed, output port ripple current composition will reduce, and this just requires measure portion to have higher precision, just can make this method reach better control effect.Existing this active power filtering control method is in the Digital Realization process, there is the time-delay that is difficult to compensate in the process of measuring sampling, algorithm computation and pulse-width modulation, this can cause bigger phase shift to the ripple of being concerned about frequency band, and has a strong impact on the effect that ripple suppresses.

Except the shortcoming of control method, the Circuits System of existing active power filtering method also can further be improved.The operation of existing active power filtering method needs a DC power supply device to come to be the power supply of single-phase electricity die mould inverter circuit.But, theory analysis as can be known, if the passive filtration unit in the system mainly is made up of electric capacity and inductance, its energy loss is very low.And the energy loss of coupling transformer and single-phase electricity die mould inverter circuit also can be done lowlyer.Thereby the effect of DC power supply provides the energy except giving the minor losses of filtering system, and its topmost function just provides a galvanic current to press for the operation of inverter circuit.Thus, can adopt inversion electric capacity to substitute DC power supply, and utilize this electric capacity to come to provide operation required direct voltage for inverter circuit.Simultaneously, can utilize the energy of the Ripple Noise that needs filtering to compensate for the small energy loss of filtering system, thereby keep the stable of inversion capacitance voltage.So just can make whole filtering system reduce by a DC power supply device, simplify circuit design, reduce system cost.In addition, because filtering system need not extra power supply, this also helps saving energy and reduce the cost, and has long-term economy.

Summary of the invention:

The objective of the invention is to propose the active power filtering method of a kind of inversion capacitance adjustment and branch road impendance controlled decoupling, wish to overcome the deficiency of existing active power filtering method, make control principle not be subjected to the influence of load variations, reduce in design process requirement for restriction to precision, time-delay and the phase shift of each funtion part, and adopt inversion electric capacity to substitute DC power supply, reduce system cost and energy loss.

The inversion capacitance adjustment that the present invention proposes and the active power filtering method of branch road impendance controlled decoupling utilize one group of bandpass filtering to realize the decoupling zero of frequency domain control earlier; Again filter branch resistance is carried out negative feedback control, to regulate the inversion capacitance voltage and to make it to maintain setting voltage value; Then, making filter branch with the method for parameter regulation is zero in the reactance of the ripple frequency of being concerned about, thereby realizes the inhibition to this frequency ripple composition; At last, handle the decoupling zero of H1 and H2 formation filter branch resistance and reactance based on two phase-shift filterings of 90 ° of phase shift mutual deviations and control, to obtain the reference wave signal that pulse-width modulation is used; This method contains successively and has the following steps:

(1) measures current i in the filter branch in parallel _APF

(2) with the filter branch current signal I that measures _APFImport one group of passband frequency and be respectively f ₁, f ₂..., f _mBandpass filtering treatment, obtain one group of ripple current signal i that comprises different frequency ripple composition ₁, i ₂..., i _mWherein, f ₁, f ₂..., f _mBe m the component frequency that needs the low-frequency ripple of filtering, each bandpass filtering treatment is at its passband frequency f _i(i=1,2 ..., m) have the highest amplitude gain A _i, and it should realize the threshold limit value of frequency domain decoupling zero greater than control system to the attenuation rate of other ripple frequency signal;

(3) the ripple current signal i that filtering is obtained ₁, i ₂..., i _mMultiply by frequency domain decoupling zero weight coefficient a respectively ₁, a ₂..., a _m, summation obtains first weighted sum current signal i then _Sum1, i.e. i _Sum1=a ₁I ₁+ a ₂I ₂+ ... + a _mI _mSimultaneously, the ripple current signal i that filtering is obtained ₁, i ₂..., i _mMultiply by frequency domain decoupling zero weight coefficient b respectively ₁, b ₂..., b _m, summation obtains second weighted sum current signal i then _Sum2, i.e. i _Sum2=b ₁I ₁+ b ₂I ₂+ ... + b _mI _mThe calculation procedure of above-mentioned frequency domain decoupling zero weight coefficient is as follows:

1. measure the voltage U at inversion electric capacity two ends _d

2. with the capacitance voltage value U that sets _D0With the inversion capacitance voltage U that measures _dIt is poor to get, and obtains the error amount Δ U of capacitance voltage _d, i.e. Δ U _d=U _D0-U _d

3. with the error amount Δ U of the capacitance voltage that obtains _dBe input to the proportion integration differentiation control that is made of a series of ratios, integration, differential, inertia or summation processing procedure, the negative feedback of keeping the inversion capacitance voltage in order to formation controls, and output obtains electric capacity charge power Control Parameter P _C

4. with the electric capacity charge power Control Parameter P that obtains _CCarry out the linear compensation processing and obtain an intermediate controlled parameters R; The computing formula that linear compensation is handled is as follows: $R=\frac{1}{{\mathrm{<figure-callout\; id="1"\; label="w"\; filenames="A0314311000222.png,A0314311000281.png"\; state="\{\{state\}\}">w</figure-callout>}}_1\&CenterDot;I_1^2+{\mathrm{<figure-callout\; id="2"\; label="w"\; filenames="A0314311000231.png,A0314311000241.png"\; state="\{\{state\}\}">w</figure-callout>}}_2\&CenterDot;{\mathrm{<figure-callout\; id="2"\; label="I"\; filenames="A0314311000231.png,A0314311000241.png"\; state="\{\{state\}\}">I</figure-callout>}}_2^2+\&CenterDot;\&CenterDot;\&CenterDot;+w_m\&CenterDot;I_m^2}\&CenterDot;P_C$

Wherein, w _i(i=1,2 ..., be thereby that middle Control Parameter R weighting is obtained corresponding frequency f m) _i(i=1,2 ..., resistance Control Parameter P m) _i(i=1,2 ..., weight coefficient m) is got positive constant; I _i(i=1,2 ..., m) be filter branch current signal i _APFAt frequency f _i(i=1,2 ..., the effective value of ripple composition m), I _i(i=1,2 ..., m) can pass through current filter branch current signal i _APFThe analysis of carrying out the ripple composition obtains or it is set to fixed numbers;

5. the intermediate controlled parameters R that calculates be multiply by one group of weight coefficient w ₁, w ₂..., w _m, obtain one group of resistance Control Parameter p successively ₁, p ₂..., p _m, that is:

p _i＝w _i·R (i＝1，2，…，m)；

6. measure the voltage u at filter branch in parallel two ends _APFWith the current i in the filter branch in parallel _APF

7. by the voltage u of the filter branch in parallel that measures _APFAnd current i _APFCalculate filter branch respectively at frequency f ₁, f ₂..., f _mReactance value X ₁, X ₂..., X _m

8. according to the reactance of filter branch, the adjusting control of process parameter obtains one group of reactance Control Parameter q ₁, q ₂..., q _mThe adjusting of parameter control can adopt constant assignment or proportion integration differentiation control or time-delay to regulate a kind of in the control; Wherein, constant assignment control be during according to the coupling transformer secondary short circuit filter branch at frequency f ₁, f ₂..., f _mReactance value X ₁₀, X ₂₀..., X _M0, according to following formula to reactance Control Parameter q _iAssignment:

q _i＝-X _i0 (i＝1，2，…，m)；

7. proportion integration differentiation control will be aforementioned the go on foot the filter branch reactance X that calculates _i(i=1,2 ..., m) as input, through ratio, integration, differential, inertia or summation processing procedure, so that system is based on f _i(i=1,2 ..., m) the filter branch reactance of frequency constitutes negative feedback control, and output obtains reactance Control Parameter q _i(i=1,2 ..., m);

Time-delay is regulated control and will be aforementioned the be 7. gone on foot the filter branch reactance X that calculates _i(i=1,2 ..., m) as input, after discretization through ratio, difference, storage or summation processing procedure, in order to obtain corresponding f _i(i=1,2, m) one group of negative feedback regulating and controlling amount of the filter branch reactance of frequency, and when filter branch is concerned about that ratio that the ripple current of frequency accounts for total ripple quantity surpasses set point, the value of this set point in 0.0～1.0 scope, again with this group regulating and controlling amount at a certain time interval respectively with current reactance Control Parameter q _i(i=1,2 ..., m) summation obtains the new reactance Control Parameter q that exports _i(i=1,2 ..., m);

9. with the aforementioned the 5. resistance Control Parameter p that obtain of step _i(i=1,2 ..., m) and the aforementioned the 8. reactance Control Parameter q that obtain of step _i(i=1,2 ..., m) carry out decoupling zero and handle, and obtain frequency domain decoupling zero weight coefficient a according to following formula _i(i=1,2 ..., m) and b _i(i=1,2 ..., m): $\left[\begin{array}{c}{a}_i\\ b_i\end{array}\right]=\frac{A_{\mathrm{\&Delta;}}}{A_i\&CenterDot;A_H\&CenterDot;n\&CenterDot;{\mathrm{<figure-callout\; id="0"\; label="U\; d"\; filenames="A0314311000222.png,A0314311000271.png"\; state="\{\{state\}\}">U</figure-callout>}}_d}\&CenterDot;\left[\begin{array}{cc}\cos {\mathrm{\&theta;}}_i& \sin {\mathrm{\&theta;}}_i\\ -\sin {\mathrm{\&theta;}}_i& {\cos \mathrm{\&theta;}}_i\end{array}\right]\&CenterDot;\left[\begin{array}{c}{p}_i\\ q_i\end{array}\right]---(i=\mathrm{1,2},\&CenterDot;\&CenterDot;\&CenterDot;,m);$

Wherein, A _ΔBe that the modulation signal amplitude that adopts, A are handled in pulse-width modulation _i(i=1,2 ..., be that the passband frequency is f in aforementioned (2) step m) _i(i=1,2 ..., the highest amplitude gain of bandpass filtering treatment m), n is the no-load voltage ratio of coupling transformer, U _D0Be the inversion capacitance voltage of setting, θ _i(i=1,2 ..., be m) from the filter branch current i _APFMeasuring process, through bandpass filtering treatment, weighted sum, H1 phase-shift filtering, summation, pulse-width modulation, pulsed drive, voltage inversion and transformer-coupled a series of processing, to the whole process that obtains the former limit of coupling transformer controlled voltage at f _i(i=1,2 ..., m) total phase shift of frequency;

(4) first weighted sum current signal i that above-mentioned (3) step is obtained _Sum1Behind first group of phase-shift filtering H1, obtain first phase-shift filtering signal i _Hb1Simultaneously, second weighted sum current signal i that above-mentioned (3) step is obtained _Sum2Behind second group of phase-shift filtering H2, obtain second phase-shift filtering signal i _Hb2If the transfer function of two groups of phase-shift filtering H1 and H2 is used H respectively _H1(s) and H _H2(s) expression, then in the ripple frequency band of being concerned about, the transfer function characteristics of phase-shift filtering H1 and H2 should satisfy following relation:

Wherein, f is the frequency of being concerned about in the frequency band, and f＞0; A _HIt is arbitrarily positive constant.Under the condition that satisfies the control precision requirement, can there be certain error in above-mentioned constraint to transfer function characteristics;

(5) with above-mentioned first and second phase-shift filtering signal i _Hb1And i _Hb2Summation obtains the reference wave signal i that pulse-width modulation is used _Ref, i.e. i _Ref=i _Hb1+ i _Hb2

(6) with reference signal i _RefCarry out pulse-width modulation and handle back acquisition one prescription wave pulse signal; The way of square-wave pulse signal equals power electronic device number controlled in the single-phase electricity die mould inverter circuit, the pulse generation frequency that pulse-width modulation is handled should be greater than 2 times of the low-frequency ripple frequency band upper limit of being concerned about, and the duty ratio of the potential pulse of being exported by the single-phase electricity die mould inverter circuit of this prescription wave pulse signal control should satisfy following formula:

A wherein _ΔBe that the modulation signal amplitude that adopts is handled in pulse-width modulation, get the positive count value;

(7) square-wave pulse signal that above-mentioned (6) step is obtained carries out going to drive after the power amplification power electronic device in the single-phase electricity die mould inverter circuit, in order to changing the duty ratio of potential pulse that the single-phase electricity die mould inverter circuit output of voltage is provided by inversion electric capacity, and make the duty ratio of this potential pulse satisfy requirement in the step (6);

(8) above-mentioned potential pulse is loaded into the coupling transformer secondary, and act on the filter branch that is constituted with passive filtration unit series connection by the former limit of coupling transformer, in order to obtain the voltage and current output signal of low-frequency ripple after filtered at the circuit output end mouth that is in parallel with this filter branch.

The active power filtering system that proposes according to the active power filtering method of aforesaid inversion capacitance adjustment and branch road impendance controlled decoupling consists of:

Active power filter circuit, it contains: the smoothing reactor on the end line of going into that is serially connected in the direct voltage input signal that comprises Ripple Noise; Smoothing reactor that is connected in parallel on output port that is constituted by former limit series connection and the filter branch between the loop electrode in order to the passive filtration unit of bearing direct voltage and coupling transformer; Output is linked into the single-phase electricity die mould inverter circuit of coupling transformer secondary; Be attempted by the inversion capacitor of single-phase electricity die mould inverter circuit input;

Be installed in the current transformer on the filter branch;

Be installed in the voltage transformer between the inversion electric capacity two ends;

Be installed in the voltage transformer between the filter branch two ends;

The one group of band pass filter that is connected in series with the measurement output of current transformer, the weighted sum circuit, H1 and H2 all-pass filter, adder and the pulse-width modulation circuit that constitute by multiplier and adder successively;

Produce circuit in order to the adjustable direct current signal that produces the capacitance voltage signal of setting;

The subtracter that is connected in series with the voltage transformer at inversion electric capacity two ends, proportion integration differentiation control circuit, linear compensation circuit and realize the gain amplifying circuit of weighted successively;

Successively with one group of the measurement output serial connection of the voltage transformer summation current transformer of filter branch corresponding f respectively _i(i=1,2 ..., m) the branch road reactance counting circuit of frequency and parameter regulation control circuit;

With linear compensation is the decoupling zero treatment circuit of input through the resistance Control Parameter of weighting acquisition and the reactance Control Parameter of parameter regulation control circuit acquisition afterwards again;

Pulse driving circuit, its input is connected with the pulse signal that above-mentioned pulse-width modulation is partly exported, and its output then is connected with the trigger control end of single-phase electricity die mould inverter circuit;

In above-mentioned active power filtering system, the no-load voltage ratio of coupling transformer, the magnitude of voltage U of the inversion electric capacity of n and setting _D0Should satisfy n * U _D0The maximum amplitude of the ripple voltage of output port in being concerned about frequency range, n * U simultaneously during greater than the coupling transformer secondary short circuit _D0The maximum amplitude of the ripple voltage of output port in being concerned about frequency range in the time of must opening a way less than filter branch again.

Further the operation principle of the inventive method is illustrated below.Might as well establish actual filter branch electric current at f _k(k=1,2 ..., m) ripple component of frequency is

I wherein _kAnd _kBe respectively the effective value and the initial phase of this frequency ripple component, t is the time.If measuring process is at f _kThe phase shift of frequency is θ _K1So, measure the filter branch current i that obtains _APFIn the f that comprised _kThe signal of frequency is If each bandpass filtering treatment is enough big to the attenuation rate of the ripple frequency signal outside its passband frequency, then each bandpass filtering treatment is at f _kThe output of frequency is respectively:

Wherein, θ _K2Be that the passband frequency is f _kBand pass filter at f _kThe phase shift of frequency.So, i _o(i=1,2 ..., m) with coefficient a _i(i=1,2 ..., m) and b _i(i=1,2 ..., m) the output result who is weighted summation respectively is:

Wherein, θ _K3Be that the weighted sum processing procedure is to f _kThe phase shift of frequency signal.If the H1 phase-shift filtering is handled f _kThe phase shift of frequency signal is θ _K4, because $\frac{H_{H2}\left(\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="A0314311000231.png,A0314311000241.png"\; state="\{\{state\}\}">j</figure-callout>}2\mathrm{\&pi;f}\right)}{H_{H1}\left(\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="A0314311000231.png,A0314311000241.png"\; state="\{\{state\}\}">j</figure-callout>}2\mathrm{\&pi;f}\right)}=j,$ So the H2 phase-shift filtering is handled f _kThe phase shift of frequency signal is θ _K4+ 90 °.So weighted sum is i as a result _Sum1And i _Sum2After passing through H1 phase-shift filtering and H2 phase-shift filtering respectively, obtain:

i _Sum1And i _Sum2Summation process is to f _kIf the phase shift of frequency signal is made as θ _K5, then the reference wave signal used of pulse-width modulation is:

Reference wave signal i _RefAfter pulse-width modulation, pulsed drive, control single-phase electricity die mould inverter circuit produces inverter voltage, and this voltage acts on filter branch by coupling transformer.If this process is at f _kThe phase shift of frequency is θ _K6, then the controlled voltage that produces on the former limit of coupling transformer is:

θ wherein _kRepresentative is from measuring the filter branch electric current to the total phase shift in the whole process of the output-controlled voltage in the former limit of coupling transformer, that is:

θ _k＝θ _k1+θ _k2+θ _k3+θ _k4+θ _k5+θ _k6

According to weighting parameters a _kAnd b _kWith resistance Control Parameter p _kWith reactance Control Parameter q _kDecoupling zero deal with relationship: $\left[\begin{array}{c}{a}_k\\ b_k\end{array}\right]=\frac{A_{\mathrm{\&Delta;}}}{A_k\&CenterDot;A_H\&CenterDot;n\&CenterDot;{\mathrm{<figure-callout\; id="0"\; label="U\; d"\; filenames="A0314311000222.png,A0314311000271.png"\; state="\{\{state\}\}">U</figure-callout>}}_d}\&CenterDot;\left[\begin{array}{cc}\cos {\mathrm{\&theta;}}_k& \sin {\mathrm{\&theta;}}_k\\ -\sin {\mathrm{\&theta;}}_k& \cos {\mathrm{\&theta;}}_k\end{array}\right]\&CenterDot;\left[\begin{array}{c}{p}_k\\ q_k\end{array}\right]$

So, the controlled voltage u on the former limit of coupling transformer _COVCan be expressed as:

The input of this controlled voltage source is the electric current of filter branch, so controlled voltage source can equivalence be an impedance component also.Generate impedance at f if establish this equivalence _kThe resistance value of frequency is R _Eqk+ jX _Eqk, then as can be known according to following formula: $R_{\mathrm{eqk}}=\frac{U_d}{U_{d0}}\&CenterDot;p_k$ $X_{\mathrm{eqk}}=\frac{U_d}{U_{d0}}\&CenterDot;q_k$

Analyze as seen thus, handle by the frequency domain decoupling zero processing of the inventive method and the decoupling zero of resistance and reactance control, the most at last resistance Control Parameter p _kGenerate impedance at f with equivalence _kThe resistance value R of frequency _EqkCorresponding, and reactance Control Parameter q _kGenerate impedance at f with equivalence _kThe reactance value X of frequency _EqkCorresponding.Particularly as inversion capacitance voltage U _dWith setting voltage U _D0When identical, then have:

R _eqk＝p _k

X _eqk＝q _k

The resistance value size of equivalence generation impedance has determined the size of single-phase electricity die mould inverter circuit and inversion electric capacity absorbed power from the ripple signal.If it is bigger that equivalence generates the resistance value of impedance, make the power that absorbed power is in operation and consumes greater than inverter circuit and inversion electric capacity, then inversion electric capacity will be recharged, and capacitance voltage raises.On the contrary, less if equivalence generates the resistance value of impedance, make absorbed power be not enough to compensate for the power that inverter circuit and inversion electric capacity are in operation and consume, then inversion electric capacity will be in discharge condition, and capacitance voltage reduces.So, just can control and keep the magnitude of voltage of inversion capacitance voltage by regulating the resistance value that equivalence generates impedance, thereby provide stable operating voltage for inverter circuit for setting.On the other hand, if make filter branch at f _kFrequency comprises that the total impedance of resistance and reactance is 0, then f _kThe ripple current of the frequency filter branch of must flowing through, thus ripple voltage and ripple current in the load are suppressed.The active component that equivalence generates impedance is used to control the inversion capacitance voltage, its resistance value was inevitable greater than zero when capacitance voltage was stablized, so it is inevitable greater than zero at main ripple frequency to generate the filter branch resistance that resistance, passive filtration unit resistance and the coupling transformer resistance of impedance constitutes by equivalence, also just can't be in order to the inhibition ripple.And the reaction component of equivalence generation impedance can be in order to suppress ripple.That is to say, can assign to offset passive filtration unit and coupling transformer at f by regulating this reactive part _kThe reactance X of frequency _K0, make filter branch at f _kThe total reactance of frequency is 0, thereby suppresses the ripple voltage and the ripple current of output port.Though the inventive method can't reach optimum in the resistance control of filter branch, but owing to comprise that the total energy consumption of the filter branch of inversion electric capacity, inverter circuit, coupling transformer and passive filtration unit can design lowlyer in the actual engineering, so keep the inversion capacitance voltage stable after, filter branch can be smaller in the resistance value of being concerned about the ripple frequency.In the ordinary course of things, simple passive filtration unit is bigger than its resistance influence to the influence of filtering at the reactance value of ripple frequency, so making the filter branch reactance by active power filtering method is 0 still can play the effect that suppresses ripple, and the ripple situation is reduced in electric power standard or the actual allowance scope that requires.

The present invention is based on by the implementation procedure of regulating equivalence and generating the resistance value control inversion capacitance voltage of impedance that a FEEDBACK CONTROL constitutes.With the inversion capacitance voltage U that measures _dWith the capacitance voltage U that sets _D0Compare.When the inversion capacitance voltage is lower than setting voltage, the error amount Δ U of capacitance voltage _dFor just, this error amount is handled through proportion integration differentiation control and linear compensation, and will increase equivalence and generate the resistance value of impedance, thereby increase absorbed power at each ripple frequency, be that inversion electric capacity charges.In like manner, when the inversion capacitance voltage is higher than setting voltage, the error amount Δ U of capacitance voltage _dFor negative, control will reduce equivalence and generate the resistance value of impedance at each ripple frequency, thereby reduce absorbed power, be the inversion capacitor discharge.For the resistance that can make equivalence generate impedance has different resistance values at different ripple frequencies, Cai handle the intermediate controlled parameters R of acquisition at first through being given corresponding f by assignment after the different weightings through linear compensation _k(k=1,2 ..., m) the resistance Control Parameter p of frequency _k(k=1,2 ..., m).If ignore the error in the control, then the ripple power of inverter circuit and inversion electric capacity absorption is: $P_{\mathrm{COV}}=\frac{U_d}{U_{d0}}\&CenterDot;(w_1\&CenterDot;I_1^2+w_2\&CenterDot;I_2^2+\&CenterDot;\&CenterDot;\&CenterDot;+w_m\&CenterDot;I_m^2)\&CenterDot;R$

Because generating the variation of impedance, equivalence can cause filter branch ripple current effective value I _k(k=1,2 ..., variation m), this is disadvantageous for constituting stable proportion integration differentiation control, offsets this variation item so the present invention has adopted linear compensation to handle, that is: $R=\frac{1}{{\mathrm{<figure-callout\; id="1"\; label="w"\; filenames="A0314311000222.png,A0314311000281.png"\; state="\{\{state\}\}">w</figure-callout>}}_1\&CenterDot;I_1^2+{\mathrm{<figure-callout\; id="2"\; label="w"\; filenames="A0314311000231.png,A0314311000241.png"\; state="\{\{state\}\}">w</figure-callout>}}_2\&CenterDot;I_2^2+\&CenterDot;\&CenterDot;\&CenterDot;+w_m\&CenterDot;I_m^2}\&CenterDot;P_C$ So have: $P_{\mathrm{COV}}=\frac{U_d}{U_{d0}}\&CenterDot;P_C$

If the capacitance of inversion electric capacity is C _d, the power loss of inverter circuit and inversion electric capacity is P _LossSo the rate of change of inversion capacitance voltage can be expressed as: $\frac{{\mathrm{dU}}_d}{\mathrm{dt}}=\frac{1}{C_d\&CenterDot;U_d}\&CenterDot;(P_{\mathrm{COV}}-P_{\mathrm{loss}})$

Thus, at corresponding certain capacitance voltage value U _dThe control neighborhood near, can carry out the local linearization analysis of system.This analysis can obtain from electric capacity charge power Control Parameter P _CTo inversion capacitance voltage U _dTransfer function be: $H_C\left(s\right)=\frac{1}{C_d\&CenterDot;{\mathrm{<figure-callout\; id="0"\; label="U\; d"\; filenames="A0314311000222.png,A0314311000271.png"\; state="\{\{state\}\}">U</figure-callout>}}_d\&CenterDot;s}$

So, utilize linear compensation to handle and simplified from Control Parameter P _CTo the transfer function form of inversion capacitance voltage, be convenient to the control of design proportion integral differential, thereby constitute negative feedback control stable, inversion capacitance voltage fast.

After in case the inversion capacitance voltage is stable, i.e. U _d=U _D0, just can adopt the constant assignment method directly with reactance Control Parameter q _kBe set to-X _K0Thereby, at f _kFrequency makes equivalence generate the reactance value X of impedance _Eqk=-X _K0Reactance X with passive filtration unit and coupling transformer _K0Offset, promptly filter branch is at f _kTotal reactance of frequency is 0.But the control method of constant assignment is a kind of control mode of open loop.If there is error in control system, the perhaps filter branch reactance X during the coupling transformer secondary short circuit _K0Be difficult to accurate acquisition, then need the feedback closed loop form that adopts proportion integration differentiation control or time-delay to regulate control.If regulating control, proportion integration differentiation control or time-delay constituted stable negative feedback control loop, because by the direct input variable of the current filter branch reactance of measuring and calculating acquisition as control, the integral action of proportion integration differentiation control in addition, perhaps the accumulative action of control to regulated quantity regulated in time-delay, and the reactance value that makes the stable state of control system reach filter branch is 0 state.So at f _kFrequency, the closed-loop control process will be regulated reactance Control Parameter q automatically _kMake equivalence generate the reactance value X of impedance _EqkApproach-X _K0, the ripple voltage and the ripple current of output also will be inhibited simultaneously.Certainly, the initial condition of closed-loop control is with q _kBe set to approaching-X _K0Value help filtering control procedure fast and stable.

In sum, active power filtering method of the present invention and system thereof need not to provide DC power supply for inverter circuit, by control filter branch impedance, realize simultaneously to the constant control of inversion capacitance voltage and based on the filter branch reactance to the low-frequency ripple Noise Suppression.This method control principle is clear, and its control procedure is not subjected to the influence of load variations.Though filter branch resistance can't reach zero, this also makes ripple suppress to reach optimum.As long as but the ripple situation is reduced in the allowance scope, the inventive method is owing to saved a DC power supply device, and its economy is significant.If the energy loss of active power filtering of the present invention system itself is low more, its filter branch resistance is just more little, and ripple suppresses effect and just can improve, and so also helps saving energy and reduce the cost, and reduces operating energy loss.

The present invention handles by decoupling zero and has realized control system from measuring the compensation of filter branch electric current to total phase shift of the output-controlled voltage in the former limit of coupling transformer, thereby has solved the influence that the phase shift that no doubt exists in measurement, bandpass filtering, weighted sum, H1 phase-shift filtering, read group total, pulse-width modulation, pulsed drive, voltage inversion and transformer coupled these processing procedures suppresses ripple.And compensation also solved sampling and the digitlization phase shift problem that time-delay caused in calculating, and makes in the control system some processing procedures can consider to adopt directly perceived, stable, reliable, exact figure mode to realize.

Emulation experiment shows that the feasibility of active power filtering method of the present invention and system thereof and ripple rejection all reach practical requirement.

Description of drawings:

Fig. 1 is the theory diagram of existing a kind of active electric filter device and control method thereof.

Fig. 2 is the transfer function model of existing a kind of active electric filter device and control method thereof.

Fig. 3 is the theory diagram of the active power filtering method of inversion capacitance adjustment of the present invention and branch road impendance controlled decoupling.

Fig. 4 is the way circuit block diagram of active power filtering system when adopting the control of constant assignment to calculate the reactance Control Parameter of inversion capacitance adjustment of the present invention and branch road impendance controlled decoupling.

Fig. 5 is the decoupling zero treatment circuit block diagram in the active power filtering system of inversion capacitance adjustment of the present invention and branch road impendance controlled decoupling.

Fig. 6 is the way circuit block diagram of active power filtering system when adopting proportion integration differentiation control or time-delay adjusting control to calculate the reactance Control Parameter of inversion capacitance adjustment of the present invention and branch road impendance controlled decoupling.

Fig. 7 is the rectification circuit that is used for producing 500 kilovolts of DC transmission system of the direct voltage input signal that comprises Ripple Noise in the embodiment of the invention.

Fig. 8 is as the circuit of active power filtering system output loading in the embodiment of the invention.

Fig. 9 is the circuit of the single tuning passive filtration unit that adopts in the embodiment of the invention.

Figure 10 is the H that adopts in the embodiment of the invention _F1, H _F2, H _F3And H _F4The amplitude-frequency response of bandpass filtering treatment.

Figure 11 is the amplitude-frequency response and the phase-frequency response curve of the H1 phase-shift filtering that adopts in the embodiment of the invention.

Figure 12 is the amplitude-frequency response and the phase-frequency response curve of the H2 phase-shift filtering that adopts in the embodiment of the invention.

Figure 13 is in the embodiment of the invention $\frac{H_{H2}\left(\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="A0314311000231.png,A0314311000241.png"\; state="\{\{state\}\}">j</figure-callout>}2\mathrm{\&pi;f}\right)}{H_{H1}\left(\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="A0314311000231.png,A0314311000241.png"\; state="\{\{state\}\}">j</figure-callout>}2\mathrm{\&pi;f}\right)}$ The phase-frequency response curve.

Figure 14 is the embodiment of the invention when adopting time-delay to regulate control to calculate the reactance Control Parameter, the reference waveform signal that the input pulse-width modulation when the coupling transformer secondary inserts inverter voltage from being shorted to is handled.

Figure 15 is the embodiment of the invention when adopting time-delay to regulate control to calculate the reactance Control Parameter, the variation waveform of the intermediate controlled parameters R when the coupling transformer secondary inserts inverter voltage from being shorted to.

Figure 16 is the embodiment of the invention when adopting time-delay to regulate control to calculate the reactance Control Parameter, the inversion capacitance voltage waveform when the coupling transformer secondary inserts inverter voltage from being shorted to.

Figure 17 is the embodiment of the invention when adopting time-delay to regulate control to calculate the reactance Control Parameter, the output end voltage waveform when the coupling transformer secondary inserts inverter voltage from being shorted to.

Figure 18 is the embodiment of the invention when adopting time-delay to regulate control to calculate the reactance Control Parameter, the output end current waveform when the coupling transformer secondary inserts inverter voltage from being shorted to.

Figure 19 is the embodiment of the invention when adopting time-delay to regulate control to calculate the reactance Control Parameter, the effective value change curve of 600 hertz of ripple compositions the output end current when the coupling transformer secondary inserts inverter voltage from being shorted to.

Embodiment:

A specific embodiment of the present invention is to design for being reduced in the low-frequency ripple noise that 500 kilovolts of rectification circuits in the DC transmission system are loaded on the transmission line.With AC rectification be direct current circuit as shown in Figure 7, wherein three-phase alternating current bus frequency is 50 hertz, line voltage is 382.9 kilovolts; The no-load voltage ratio of the transformer T2 of the transformer T1 of star/triangle type and star/star type is 345.0 kilovolts: 213.5 kilovolts, and rated capacity 603.7 megavolt-amperes, the per unit value of short-circuit impedance is 0.18; Adopt a rectifier bridge T who is made of 12 controllable silicons to realize 12 impulse commutation circuit, the conducting pilot angle is got 5 degree, about 500 kilovolts of the direct voltage of output.The voltage that this rectification circuit produces mainly comprises 600,1200,1800,2400 hertz low-frequency ripple, and this voltage will be imported active power filtering of the present invention system.

The output loading of the embodiment of active power filtering method of the present invention and system thereof as shown in Figure 8.Comprise one 200 kilometers transmission line in Fig. 8, power transmission line adopts the double bundle conductor apart from 50 meters on ground, 0.46 meter of spacing, directly over high 8 meters are lightning protection ground wires, the earth is as loop electrode.The power transmission line other end has the smoothing reactor of L1=0.597 henry, R1=250 Europe load resistance, and by three electric capacity (C1=0.84 microfarads, the C2=3.0 microfarad, the C3=0.209 microfarad), two inductance (L2=0.84 milihenries, the L3=0.336 henry) and the passive filter circuit that constitutes of two resistance (R2=10 kilo-ohm, R3=6.3 Europe).

The embodiment of active power filtering method of the present invention and system thereof is when adopting the constant assignment to calculate reactance Control Parameter q _kThe time circuit block diagram as shown in Figure 4, calculate reactance Control Parameter q when adopting proportion integration differentiation control or time-delay to regulate control _kThe time circuit block diagram as shown in Figure 6.Because the ripple frequency of being concerned about is 600,1200,1800,2400 hertz, thus m=4, f ₁=600Hz, f ₂=1200Hz, f ₃=1800Hz, f ₄=2400Hz.Among Fig. 4 and Fig. 6, the smoothing reactor L that incoming line one end seals in _sGet 0.2 henry; After coupling transformer former limit series connection, be connected in parallel on the passive filtration unit employing single-tuned circuit as shown in Figure 6 of output port again, wherein capacitor C _C=1.0 microfarads, inductance L _C=0.0312 henry, resonance frequency are 900 hertz, and obviously the relative direct current of passive filtration unit is an open-circuit condition, and direct voltage will be loaded into capacitor C _COn; The no-load voltage ratio of coupling transformer is 20 kilovolts: 10 kilovolts, i.e. n=2, the short-circuit impedance of coupling transformer other circuit parameter relatively can ignore; A C _dThe inversion electric capacity of=20.0 microfarads is connected on the direct-flow input end of the single-phase electricity die mould inverter circuit that constitutes by 4 IGBT and with 4 diodes of IGBT reverse parallel connection, and the ac output end of single-phase electricity die mould inverter circuit inserts the secondary of coupling transformer.

In embodiments of the present invention, the first step of control method is measured filter branch electric current in parallel by current transformer earlier, and kilo-ampere is got by unit.Because obtaining the process of pulse-width modulation reference wave, bandpass filtering, weighted sum, phase-shift filtering and summation among the embodiment all adopt digital form to realize, so measuring process also comprises the processing that the continuous measurement signal that obtains is sampled, sample frequency is got 20 KHz, the filter branch current signal i that obtains dispersing _APF(n).Then, i _APF(n) be input to corresponding 600,1200,1800,2400 hertz the logical digital filtering processing of four bands respectively, the transfer function form that their Z-transformation is represented is respectively: $H_{f1}\left(z\right)=\frac{0.0055900155\&CenterDot;(1-z^{-2})}{1-1.9535925\&CenterDot;z^{-1}+0.98881997\&CenterDot;z^{-2}}$ $H_{f2}\left(z\right)=\frac{0.0073086810\&CenterDot;(1-z^{-2})}{1-1.8459621\&CenterDot;z^{-1}+0.98538264\&CenterDot;z^{-2}}$ $H_{f3}\left(z\right)=\frac{0.0053297099\&CenterDot;(1-z^{-2})}{1-1.6796558\&CenterDot;z^{-1}+0.98934058\&CenterDot;z^{-2}}$ $H_{f4}\left(z\right)=\frac{0.006798292\&CenterDot;(1-z^{-2})}{1-1.4480248\&CenterDot;z^{-1}+0.98640214\&CenterDot;z^{-2}}$ H _F1, H _F2, H _F3And H _F4The amplitude-frequency response of four bandpass filtering treatment is respectively as (a) and (b) among Fig. 7, (c) with (d).Ripple component i to 600,1200,1800,2400 hertz of the difference correspondences that obtain after the Filtering Processing ₁, i ₂, i ₃, and i ₄Carry out two groups of weighted sums, obtain first weighted sum current signal i _Sum1=a ₁I ₁+ a ₂I ₂+ a ₃I ₃+ a ₄I ₄With second weighted sum current signal i _Sum2=b ₁I ₁+ b _iI ₂+ b ₃I ₃+ b ₄I ₄

Frequency domain decoupling zero weight coefficient a ₁, a ₂, a ₃, a ₄, b ₁, b ₂, b ₃And b ₄Calculating at first need the negative feedback control by the inversion capacitance voltage to obtain resistance Control Parameter p ₁, p ₂, p ₃And p ₄The negative feedback control of inversion capacitance voltage obtains the voltage U at inversion electric capacity two ends by voltage measuring transformer _d, kilovolt is got by unit.Then, the inversion capacitance voltage (U of setting _D0=5.0 kilovolts) with the capacitance voltage U that measures _dIt is poor to get, and obtains error amount Δ U _d=U _D0-U _dThe inversion capacitance voltage U that sets _D0With the product of the no-load voltage ratio n of coupling transformer be nU _D0=10.0 kilovolts, the maximum amplitude (about 3 kilovolts) of the ripple voltage of output port in being concerned about frequency range of this product value filter circuit during, and the maximum amplitude (about 12 kilovolts) of this product value output port ripple voltage in the care frequency range when also open a way less than filter branch greater than the coupling transformer secondary short circuit.To inversion capacitance voltage error amount Δ U _dThe transfer function form of the proportion integration differentiation control procedure of carrying out is: $H_{\mathrm{PID}4p}\left(s\right)=\frac{1}{1+0.001\&CenterDot;s}\&CenterDot;(0.04+\frac{6.4}{s})$ Obtain electric capacity charge power Control Parameter P through proportion integration differentiation control output _CThe weight coefficient that is adopted when middle Control Parameter R being weighted calculating is got:

w ₁=w ₂=w ₃=w ₄So=1.0, the formula that calculates the linear compensation processing of intermediate controlled parameters R is: $R=\frac{1}{I_1^2+I_2^2+I_3^2+I_4^2}\&CenterDot;P_C$ Filter branch electric current wherein is at the effective value I of 600,1200,1800,2400 hertz ripple composition ₁, I ₂, I ₃And I ₄Employing is to filter branch current signal i _APFThe method of carrying out Fourier transform obtains, and the unit of fetching data is a kilo-ampere.Based on proportion integration differentiation control middle Control Parameter R has been carried out amplitude limit among the embodiment, made its variation be no more than-50～200 number range.At last, by:

p ₁=p ₂=p ₃=p ₄=R obtains corresponding 600,1200,1800,2400 hertz resistance Control Parameter p respectively ₁, p ₂, p ₃And p ₄

Calculate reactance Control Parameter q when adopting the control of constant assignment _kThe time, ignore the impedance of coupling transformer, according to impedance j (147.64), j102.61, j264.44, the j404.17 of passive filter, produce circuit with reactance Control Parameter q by the adjustable direct current signal among Fig. 4 at 600,1200,1800,2400 hertz _kBe set to following constant:

q ₁＝147.64；q ₂＝-102.61；q ₃＝-264.44；q ₄＝-404.17。

Adopt proportion integration differentiation control or time-delay to regulate control and obtain reactance Control Parameter q _kThe time, at first need to measure the voltage u of filter branch in parallel _APFWith filter branch current i in parallel _APFAgain to u _APFCarry out Fourier transform, obtain filter branch voltage u _APFRipple effective value U at 600,1200,1800,2400 hertz ₁, U ₂, U ₃, U ₄With ripple phase place _U1, _U2, _U3, _U4Simultaneously, to i _APFCarry out Fourier transform, obtain the filter branch current i _APFRipple effective value I at 600,1200,1800,2400 hertz ₁, I ₂, I ₃, I ₄With ripple phase place _I1, _I2, _I3, _I4Then according to following formula calculation of filtered branch road respectively 600,1200,1800,2400 hertz branch road reactance: Wherein, k=1,2,3,4.

Control can be with X if embodiment adopts proportion integration differentiation _kThe proportion integration differentiation control circuit that input has following transfer function form: $H_{\mathrm{PID}4q}\left(s\right)=-(0.5+\frac{90}{s})\&CenterDot;\frac{1}{1+0.01s}$ And output obtains reactance Control Parameter q _k

If embodiment adopts time-delay to regulate control, then control system accounts for 90% when above of total ripple current when 600,1200,1800,2400 hertz of ripple current compositions of detection filter branch road, and the judgement main circuit is not in transient process.The filter branch impedance of measuring is through after the discretization, can adopt currency or currency and the differential data of the last group of centrifugal pump of being stored passes through the result of weighted sum as the FEEDBACK CONTROL regulated quantity.But present embodiment for the sake of simplicity, adopted form to be-0.8X _kA proportional of (k=1,2,3,4) is as the FEEDBACK CONTROL regulated quantity.When main circuit and filter branch are not in transient process, every 0.01 second this regulated quantity and current reactance Control Parameter q _K0Summation obtains new reactance Control Parameter q _k, that is:

q _k=q _K0-0.8X _k(k=1,2,3,4) are if t ₀Be the control starting moment, then initial condition is with reactance Control Parameter q _kBe set to the estimated parameters result of institute in the control of constant assignment, that is:

q ₁(t ₀)＝147.64；q ₂(t ₀)＝-102.61；q ₃(t ₀)＝-264.44；q ₄(t ₀)＝-404.17。

When the FEEDBACK CONTROL that adopts the inversion capacitance voltage obtains resistance Control Parameter p _k, and adopt control of constant assignment or proportion integration differentiation control or time-delay adjusting control to obtain reactance Control Parameter q _kAfter, be achieved as follows the decoupling zero processing of expression formula again according to circuit shown in Figure 5, thereby obtain frequency domain decoupling zero weight coefficient a _kAnd b _k(k=1,2,3,4): $\left[\begin{array}{c}{a}_k\\ b_k\end{array}\right]=\frac{A_{\mathrm{\&Delta;}}}{A_k\&CenterDot;A_H\&CenterDot;n\&CenterDot;U_{d0}}\&CenterDot;\left[\begin{array}{cc}\cos {\mathrm{\&theta;}}_k& \sin {\mathrm{\&theta;}}_k\\ -{\sin \mathrm{\&theta;}}_k& \cos {\mathrm{\&theta;}}_k\end{array}\right]\&CenterDot;\left[\begin{array}{c}{p}_k\\ q_k\end{array}\right]---(k=\mathrm{1,2,3,4})$ Wherein, modulated triangular wave amplitude A _Δ=5.0; The bandpass filtering gain A _k=1.0 (k=1,2,3,4); The phase-shift filtering gain A _H=1.0; Coupling transformer no-load voltage ratio n=2.0; The inversion capacitance voltage U that sets _D0=5.0; Total phase shift from measurement filter branch electric current to the output-controlled voltage in the former limit of coupling transformer is respectively at 600,1200,1800,2400 hertz: θ ₁=-0.36828 radian, θ ₂=-1.89014 radians, θ ₃=-2.84317 radians, θ ₄=-3.54499 radians.So, according to above-mentioned parameter, be by the adjustable direct current signal generation circuit generation numerical value among Fig. 5 $\frac{A_{\mathrm{\&Delta;}}}{A_k\&CenterDot;A_H\&CenterDot;n\&CenterDot;U_d}\&CenterDot;\cos {\mathrm{\&theta;}}_k$ With $\frac{A_{\mathrm{\&Delta;}}}{A_k\&CenterDot;A_H\&CenterDot;n\&CenterDot;U_d}\&CenterDot;\sin {\mathrm{\&theta;}}_k$ Two regular signals, the p of input _kAnd q _kBy four multipliers, an adder and a subtracter, output at last obtains frequency domain decoupling zero weight coefficient a again _kAnd b _k

Weighted sum current signal i _Sum1And i _Sum2Respectively through obtaining phase-shift filtering signal i behind H1 and the H2 phase-shift filtering _Hb1And i _Hb2The transfer function form that the Z-transformation of H1 and H2 phase-shift filtering is represented is respectively: $H_{H1}\left(z\right)=\frac{0.14677965-1.00149913\&CenterDot;z^{-1}+1.83187525\&CenterDot;z^{-2}-z^{-3}}{1-1.83187525\&CenterDot;z^{-1}+1.00149913\&CenterDot;z^{-2}-0.14677965\&CenterDot;z^{-3}}$ $H_{H2}\left(z\right)=\frac{-0.13884066+0.17122666\&CenterDot;z^{-1}+0.93626531\&CenterDot;z^{-2}-1.96496093\&CenterDot;z^{-3}+z^{-4}}{1-1.96496093\&CenterDot;z^{-1}+0.93626531\&CenterDot;z^{-2}+0.17122666\&CenterDot;z^{-3}-0.13884066\&CenterDot;z^{-4}}$ By the amplitude-frequency response of the H1 of Figure 11 and Figure 12 and H2 phase-shift filtering and phase-frequency response curve as seen, H1 and H2 phase-shift filtering are A to the amplitude-frequency gain of input signal _h=1.0.By Figure 13's $\frac{H_{H2}\left(j2\mathrm{\&pi;f}\right)}{H_{H1}\left(j2\mathrm{\&pi;f}\right)}$ The phase-frequency response curve as seen, in 500～3000 hertz of frequency domain scopes being concerned about, $\frac{H_{H2}\left(j2\mathrm{\&pi;f}\right)}{H_{H1}\left(j2\mathrm{\&pi;f}\right)}$ Phase place be 90.00 ± 0.02 the degree.Again to phase-shift filtering signal i _Hb1And i _Hb2Summation obtains the reference wave signal i that pulse-width modulation is used _Ref, i.e. i _Ref=i _Hb1+ i _Hb2Pulse-width modulation adopts the comparison circuit of reference wave and modulated triangular wave to realize.It is 10 KHz that modulated triangular wave generation circuit produces frequency, and maximum amplitude is A _Δ=5.0 modulated triangular wave.If reference wave signal i _RefAmplitude is greater than the current amplitude of modulated triangular wave, and then pulse-width modulated output signal will be controlled inverter circuit at coupling transformer secondary loading+U through behind the pulse driving circuit _dCapacitance voltage (to be labeled as the positive voltage direction among Fig. 4 and Fig. 6); If reference wave signal i _RefAmplitude is less than the current amplitude of modulated triangular wave, and then pulse-width modulated output signal will be controlled inverter circuit at coupling transformer secondary loading-U through behind the pulse driving circuit _dCapacitance voltage.

Embodiment based on active power filtering method of the present invention and system thereof has carried out simulation calculation.Here only provide based on time-delay and regulate the simulation result that is obtained when the reactance Control Parameter is calculated in control.In 0.4 second of beginning, the coupling transformer secondary short circuit is controlled inoperative.After 0.4 second, single-phase electricity die mould inverter circuit inserts the coupling transformer secondary, and the inventive method is started working.Figure 14 has provided the reference signal i that handles from 0.38 second to 0.56 second input pulse-width modulation _RefWaveform, reference signal i _RefAmplitude does not surpass the maximum amplitude A of modulated triangular wave _Δ=5.0.Figure 15 is the variation waveform from 0.38 second to 0.56 second intermediate controlled parameters R, because the initial voltage of inversion electric capacity is 0, the intermediate controlled parameters R is got maximum limit amplitude 200 at system's start-up period, thereby makes the charging of inversion electric capacity.The variation waveform of inversion capacitance voltage between 0.38 second to 0.56 second as shown in figure 16.Through after 0.06 second starting time, the inversion capacitance voltage is basicly stable at 5.0 kilovolts again.In the time of the control of inversion capacitance voltage, the filter branch reactance also is conditioned and makes it is zero.Figure 17 and Figure 18 are respectively filter circuit output end voltage waveform and output end current waveform.Output end voltage and electric current comprise tangible low-frequency ripple when the coupling transformer secondary short circuit, and after method of the present invention started 0.1 second, low-frequency ripple obviously was inhibited.Further spectrum analysis can be illustrated more clearly in the repressed effect of low-frequency ripple.Figure 19 has provided the change curve of the effective value of 600 hertz of ripple compositions in the output end current, though this ripple composition is not eliminated fully, has also reduced more than 4 times.1200, the filter effect of 1800 and 2400 hertz of ripple compositions is similar with it.

It is worthy of note that at last active power filtering method of the present invention and system thereof are intended to suppress low-frequency ripple, but can introduce the noise of the modulating frequency that the power electronic device switch causes simultaneously, Figure 17 and 18 also reflects this point.Yet after having suppressed low-frequency ripple, high-frequency noise can be eliminated by some simple filter circuits.

Claims (5)

1. The active power filtering method of 1, inversion capacitance adjustment and branch road impendance controlled decoupling contains the step that one group of bandpass filtering treatment realizes the frequency domain control decoupling, it is characterized in that: this method utilizes one group of bandpass filtering to realize the decoupling zero of frequency domain control earlier; Again filter branch resistance is carried out negative feedback control, to regulate the inversion capacitance voltage and to make it to maintain setting voltage value; Then, making filter branch with the method for parameter regulation is zero in the reactance of the ripple frequency of being concerned about, thereby realizes the inhibition to this frequency ripple composition; At last, handle the decoupling zero of H1 and H2 formation filter branch resistance and reactance based on two phase-shift filterings of 90 ° of phase shift mutual deviations and control, to obtain the reference wave signal that pulse-width modulation is used; This method contains successively and has the following steps:

   (1) measures current i in the filter branch in parallel _APF

   (2) with the filter branch current signal i that measures _APFImport one group of passband frequency and be respectively f ₁, f ₂..., f _mBandpass filtering treatment, obtain one group of ripple current signal i that comprises different frequency ripple composition ₁, i ₂..., i _mWherein, f ₁, f ₂..., f _mBe m the component frequency that needs the low-frequency ripple of filtering, each bandpass filtering treatment is at its passband frequency f _i(i=1,2 ..., m) have the highest amplitude gain A _i, and it should realize the threshold limit value of frequency domain decoupling zero greater than control system to the attenuation rate of other ripple frequency signal;

   (3) the ripple current signal i that filtering is obtained ₁, i ₂..., i _mMultiply by frequency domain decoupling zero weight coefficient a respectively ₁, a ₂..., a _m, summation obtains first weighted sum current signal i then _Sum1, i.e. i _Sum1=a ₁I ₁+ a ₂I ₂+ ... + a _mI _mSimultaneously, the ripple current signal i that filtering is obtained ₁, i ₂..., i _mMultiply by frequency domain decoupling zero weight coefficient b respectively ₁, b ₂..., b _m, summation obtains second weighted sum current signal i then _Sum2, i.e. i _Sum2=b ₁I ₁+ b ₂I ₂+ ... + b _mI _mThe calculation procedure of above-mentioned frequency domain decoupling zero weight coefficient is as follows:

   1. measure the voltage U at inversion electric capacity two ends _d

   2. with the capacitance voltage value U that sets _D0With the inversion capacitance voltage U that measures _dIt is poor to get, and obtains the error amount Δ U of capacitance voltage _d, i.e. Δ U _d=U _D0-U _d

   3. with the error amount Δ U of the capacitance voltage that obtains _dBe input to the proportion integration differentiation control that is made of a series of ratios, integration, differential, inertia or summation processing procedure, the negative feedback of keeping the inversion capacitance voltage in order to formation controls, and output obtains electric capacity charge power Control Parameter P _C

   4. with the electric capacity charge power Control Parameter P that obtains _CCarry out the linear compensation processing and obtain an intermediate controlled parameters R; The computing formula that linear compensation is handled is as follows: $R=\frac{1}{w_1\&CenterDot;I_1^2+w_2\&CenterDot;I_2^2+\&CenterDot;\&CenterDot;\&CenterDot;+w_m\&CenterDot;I_m^2}\&CenterDot;P_C$

   Wherein, w _i(i=1,2 ..., be thereby that middle Control Parameter R weighting is obtained corresponding frequency f m) _i(i=1,2 ..., resistance Control Parameter P m) _i(i=1,2 ..., weight coefficient m) is got positive constant; I _i(i=1,2 ..., m) be filter branch current signal i _APFAt frequency f _i(i=1,2 ..., the effective value of ripple composition m), I _i(i=1,2 ..., m) can pass through current filter branch current signal i _APFThe analysis of carrying out the ripple composition obtains or it is set to fixed numbers;

   5. the intermediate controlled parameters R that calculates be multiply by one group of weight coefficient w ₁, w ₂..., w _m, obtain one group of resistance Control Parameter p successively ₁, p ₂..., p _m, that is:

   p _i＝w _i·R (i＝1，2，…，m)；

   6. measure the voltage u at filter branch in parallel two ends _APFWith the current i in the filter branch in parallel _APF

   7. by the voltage u of the filter branch in parallel that measures _APFAnd current i _APFCalculate filter branch respectively at frequency f ₁, f ₂..., f _mReactance value X ₁, X ₂..., X _m

   8. according to the reactance of filter branch, the adjusting control of process parameter obtains one group of reactance Control Parameter q ₁, q ₂..., q _m

   9. with the aforementioned the 5. resistance Control Parameter P that obtain of step _i(i=1,2 ..., m) and the aforementioned the 8. reactance Control Parameter q that obtain of step _i(i=1,2 ..., m) carry out decoupling zero and handle, and obtain frequency domain decoupling zero weight coefficient q according to following formula _i(i=1,2 ..., m) and b _i(i=1,2 ..., m): $\left[\begin{array}{c}{a}_i\\ b_i\end{array}\right]=\frac{A_{\mathrm{\&Delta;}}}{A_i\&CenterDot;A_H\&CenterDot;n\&CenterDot;U_{d0}}\&CenterDot;\left[\begin{array}{cc}\cos {\mathrm{\&theta;}}_i& \sin {\mathrm{\&theta;}}_i\\ -{\sin \mathrm{\&theta;}}_i& \cos {\mathrm{\&theta;}}_i\end{array}\right]\&CenterDot;\left[\begin{array}{c}{p}_i\\ q_i\end{array}\right]---(i=\mathrm{1,2},\&CenterDot;\&CenterDot;\&CenterDot;,m);$

   Wherein, A _ΔBe that the modulation signal amplitude that adopts, A are handled in pulse-width modulation _i(i=1,2 ..., be that the passband frequency is f in aforementioned (2) step m) _i(i=1,2 ..., the highest amplitude gain of bandpass filtering treatment m), n is the no-load voltage ratio of coupling transformer, U _D0Be the inversion capacitance voltage of setting, θ _i(i=1,2 ..., be m) from the filter branch current i _APFMeasuring process, through bandpass filtering treatment, weighted sum, H1 phase-shift filtering, summation, pulse-width modulation, pulsed drive, voltage inversion and transformer-coupled a series of processing, to the whole process that obtains the former limit of coupling transformer controlled voltage at f _i(i=1,2 ..., m) total phase shift of frequency;

   (4) first weighted sum current signal i that above-mentioned (3) step is obtained _Sum1Behind first group of phase-shift filtering H1, obtain first phase-shift filtering signal i _Hb1Simultaneously, second weighted sum current signal i that above-mentioned (3) step is obtained _Sum2Behind second group of phase-shift filtering H2, obtain second phase-shift filtering signal i _Hb2If the transfer function of two groups of phase-shift filtering H1 and H2 is used H respectively _H1(s) and H _H2(s) expression, then in the ripple frequency band of being concerned about, the transfer function characteristics of phase-shift filtering H1 and H2 should satisfy following relation:

   

   Wherein, f is the frequency of being concerned about in the frequency band, and f＞0; A _HIt is arbitrarily positive constant.Under the condition that satisfies the control precision requirement, can there be certain error in above-mentioned constraint to transfer function characteristics;

   (5) with above-mentioned first and second phase-shift filtering signal i _Hb1And i _Hb2Summation obtains the reference wave signal i that pulse-width modulation is used _Ref, i.e. i _Ref=i _Hb1+ i _Hb2

   (6) with reference signal i _RefCarry out pulse-width modulation and handle back acquisition one prescription wave pulse signal; The way of square-wave pulse signal equals power electronic device number controlled in the single-phase electricity die mould inverter circuit, the pulse generation frequency that pulse-width modulation is handled should be greater than 2 times of the low-frequency ripple frequency band upper limit of being concerned about, and the duty ratio of the potential pulse of being exported by the single-phase electricity die mould inverter circuit of this prescription wave pulse signal control should satisfy following formula:

   

   A wherein _ΔBe that the modulation signal amplitude that adopts is handled in pulse-width modulation, get the positive count value;

   (7) square-wave pulse signal that above-mentioned (6) step is obtained carries out going to drive after the power amplification power electronic device in the single-phase electricity die mould inverter circuit, in order to changing the duty ratio of potential pulse that the single-phase electricity die mould inverter circuit output of voltage is provided by inversion electric capacity, and make the duty ratio of this potential pulse satisfy requirement in the step (6);

   (8) above-mentioned potential pulse is loaded into the coupling transformer secondary, and act on the filter branch that is constituted with passive filtration unit series connection by the former limit of coupling transformer, in order to obtain the voltage and current output signal of low-frequency ripple after filtered at the circuit output end mouth that is in parallel with this filter branch.
2. 2, the active power filtering method of inversion capacitance adjustment according to claim 1 and branch road impendance controlled decoupling, it is characterized in that: in the 8. in the step of the computational process of the frequency domain decoupling zero weight coefficient of described step (3), the adjusting of parameter control realizes by the control of constant assignment, it during according to the coupling transformer secondary short circuit filter branch at frequency f ₁, f ₂..., f _mReactance value X ₁₀, X ₂₀..., X _M0, according to following formula to reactance Control Parameter q _iAssignment:

   q _i＝-X _i0 (i＝1，2，…，m)。
3. 3, the active power filtering method of inversion capacitance adjustment according to claim 1 and branch road impendance controlled decoupling, it is characterized in that: in the 8. in the step of the computational process of the frequency domain decoupling zero weight coefficient of described step (3), the adjusting control of parameter is that the control of passing ratio integral differential realizes that it is with the filter branch reactance X that calculates _i(i=1,2 ..., m) as input, through ratio, integration, differential, inertia or summation processing procedure, so that system is based on f _i(i=1,2 ..., m) the filter branch reactance of frequency constitutes negative feedback control, and output obtains reactance Control Parameter q _i(i=1,2 ..., m).
4. 4, the active power filtering method of inversion capacitance adjustment according to claim 1 and branch road impendance controlled decoupling, it is characterized in that: in the 8. in the step of the computational process of the frequency domain decoupling zero weight coefficient of described step (3), the adjusting control of parameter is regulated control by time-delay and is realized that it is with the filter branch reactance X that calculates _i(i=1,2 ..., m) as input, after discretization through ratio, difference, storage or summation processing procedure, in order to obtain corresponding f _i(i=1,2, m) one group of negative feedback regulating and controlling amount of the filter branch reactance of frequency, and when filter branch is concerned about that ratio that the ripple current of frequency accounts for total ripple quantity surpasses set point, the value of this set point in 0.0～1.0 scope, again with this group regulating and controlling amount at a certain time interval respectively with current reactance Control Parameter q _i(i=1,2 ..., m) summation obtains the new reactance Control Parameter q that exports _i(i=1,2 ..., m).
5. 5, the active power filtering method of inversion capacitance adjustment according to claim 1 and branch road impendance controlled decoupling and the active power filtering system that proposes contains active power filter circuit is characterized in that this system comprises as the lower part:

   Active power filter circuit, it contains: the smoothing reactor on the end line of going into that is serially connected in the direct voltage input signal that comprises Ripple Noise; Smoothing reactor that is connected in parallel on output port that is constituted by former limit series connection and the filter branch between the loop electrode in order to the passive filtration unit of bearing direct voltage and coupling transformer; Output is linked into the single-phase electricity die mould inverter circuit of coupling transformer secondary; Be attempted by the inversion capacitor of single-phase electricity die mould inverter circuit input;

   Be installed in the current transformer on the filter branch;

   Be installed in the voltage transformer between the inversion electric capacity two ends;

   Be installed in the voltage transformer between the filter branch two ends;

   The one group of band pass filter that is connected in series with the measurement output of current transformer, the weighted sum circuit, H1 and H2 all-pass filter, adder and the pulse-width modulation circuit that constitute by multiplier and adder successively;

   Produce circuit in order to the adjustable direct current signal that produces the capacitance voltage signal of setting;

   The subtracter that is connected in series with the voltage transformer at inversion electric capacity two ends, proportion integration differentiation control circuit, linear compensation circuit and realize the gain amplifying circuit of weighted successively;

   Successively with one group of the measurement output serial connection of the voltage transformer summation current transformer of filter branch corresponding f respectively _i(i=1,2 ..., m) the branch road reactance counting circuit of frequency and parameter regulation control circuit;

   With linear compensation is the decoupling zero treatment circuit of input through the resistance Control Parameter of weighting acquisition and the reactance Control Parameter of parameter regulation control circuit acquisition afterwards again;

   Pulse driving circuit, its input is connected with the pulse signal that above-mentioned pulse-width modulation is partly exported, and its output then is connected with the trigger control end of single-phase electricity die mould inverter circuit;

   In above-mentioned active power filtering system, the magnitude of voltage U of the inversion electric capacity of the no-load voltage ratio n of coupling transformer and setting _D0Should satisfy n * U _D0The maximum amplitude of the ripple voltage of output port in being concerned about frequency range, n * U simultaneously during greater than the coupling transformer secondary short circuit _D0The maximum amplitude of the ripple voltage of output port in being concerned about frequency range in the time of must opening a way less than filter branch again.

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