CN112234808B - Double-frequency ripple suppression circuit and suppression method of single-phase inverter - Google Patents

Double-frequency ripple suppression circuit and suppression method of single-phase inverter Download PDF

Info

Publication number
CN112234808B
CN112234808B CN202010943514.6A CN202010943514A CN112234808B CN 112234808 B CN112234808 B CN 112234808B CN 202010943514 A CN202010943514 A CN 202010943514A CN 112234808 B CN112234808 B CN 112234808B
Authority
CN
China
Prior art keywords
capacitor
ripple
current
voltage
inductor
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Active
Application number
CN202010943514.6A
Other languages
Chinese (zh)
Other versions
CN112234808A (en
Inventor
张岩
黄燕飞
高晓阳
高远
刘进军
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Xian Jiaotong University
Original Assignee
Xian Jiaotong University
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Xian Jiaotong University filed Critical Xian Jiaotong University
Priority to CN202010943514.6A priority Critical patent/CN112234808B/en
Publication of CN112234808A publication Critical patent/CN112234808A/en
Application granted granted Critical
Publication of CN112234808B publication Critical patent/CN112234808B/en
Active legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/157—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators with digital control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/007—Plural converter units in cascade

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Rectifiers (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention provides a double-frequency ripple suppression circuit and a suppression method of a single-phase inverter, wherein the suppression circuit comprises: input DC power supply VdcInductor L, MOSFET power tube SdMOSFET power tube SqDiode D1Diode D2Capacitor C1Capacitor C2Full-control inverter bridge and filter inductor LfFilter capacitor CfAnd a load resistance R; the control method is that the double frequency ripple wave on the input inductor is controlled to be 0, the capacitor voltage ripple wave is controlled to be in a complementary state, and the DC bus capacitor absorbs the double frequency ripple wave energy. The method can realize energy conversion between the direct current side and the alternating current side, avoid the adverse effect of low-frequency ripples on the direct current side and the alternating current side, and greatly reduce the requirements of the system on passive elements such as inductance and capacitance.

Description

Double-frequency ripple suppression circuit and suppression method of single-phase inverter
Technical Field
The invention belongs to the technical field of electric power, and particularly relates to a double-frequency ripple suppression circuit and a suppression method based on a three-level boost two-stage single-phase inverter.
Background
The single-phase DC-AC converter is used as an interface device between a DC power supply and an AC load or a single-phase AC power grid, bears the responsibility of single-phase DC-AC electric energy conversion, provides stable energy for electric equipment, and has wide application in modern social life. However, there is an inherent problem with single-phase dc-ac energy conversion systems: the power input into the converter by the direct current power supply is constant, the power required by an alternating current load or a power grid is power with double power frequency (100Hz) fluctuation, the unbalance of instantaneous power at the alternating current side and the direct current side can generate current ripples with double power frequency on the inductor at the direct current side, and voltage ripples with double power frequency are generated on the capacitor at the direct current side, so that the problem of double frequency ripples of a single-phase inverter system is solved. In a specific application scenario, the problem has many adverse effects: if the direct current side power supply is a fuel cell or a photovoltaic cell panel, the current fluctuating on the inductor influences the service life of the fuel cell and influences the MPPT efficiency of the photovoltaic cell panel; if the ac side is connected to a single-phase network or is used to drive an electric motor, the fluctuating voltage on the dc bus affects the quality of the power incorporated into the network and the drive of the motor.
In order to solve the problem of power fluctuation of a single-phase direct current-alternating current system, a traditional solution is to adopt a passive power decoupling strategy, namely, the value of capacitance and inductance inside a converter is increased, and double-frequency ripple waves are suppressed within a reasonable range, so that the work of a direct current side power supply and an alternating current side load is not influenced. This method has proven effective but also causes various problems such as excessive system size, high cost, poor reliability, and the like.
In order to overcome the defects of the passive power decoupling scheme, the active power decoupling scheme has attracted wide attention in recent years. The implementation method is that a buffer circuit is added on the basis of the original converter, and the energy storage device in the buffer circuit is used for absorbing power fluctuation, so that the adverse effect of low-frequency ripples in the converter is eliminated, and the use of an electrolytic capacitor and a large inductor is avoided. However, most of these methods require additional circuits, which are costly and complicated to control.
Disclosure of Invention
The invention aims to provide a double-frequency ripple suppression circuit and a suppression method of a single-phase inverter, which realize energy conversion between a direct current side and an alternating current side, avoid the adverse effect of low-frequency ripples on the direct current side and the alternating current side, and greatly reduce the requirements of a system on passive elements such as an inductance and a capacitance.
In order to achieve the purpose, the invention adopts the following technical means:
double frequency ripple suppression circuit of single-phase inverter includes: input DC power supply VdcInductor L, MOSFET power tube SdMOSFET power tube SqDiode D1Diode D2Capacitor C1Capacitor C2Full-control inverter bridge Sap、San、Sbp、SbnFilter inductor LfFilter capacitor CfAnd a load resistance R;
the input DC power supply VdcThe anode of the inductor is connected with one end of an input inductor L, and the other end of the inductor L is connected with a diode D1Anode of and MOSFET power tube SdA drain electrode of (1); diode D1Cathode and capacitor C1The positive electrodes of the two electrodes are connected; MOSFET power tube SdSource electrode of and MOSFET power tube SqThe drain electrodes of the two electrodes are connected; MOSFET power tube SqSource electrode of the transistor is connected with a direct current power supply VdcCathode and diode D2A cathode of (a); capacitor C1Negative pole of the capacitor C2Anode and MOSFET power tube SdA source electrode of (a); capacitor C2Negative electrode of (D) is connected with diode2The anode of (1); capacitor C1And a capacitor C2The input end of the full-control inverter bridge is connected in series; the positive output of the full-control inverter bridge is connected with an inductor LfThe output of the negative pole of the full-control inverter bridge is connected with the common ground at the AC side; capacitor CfConnected in parallel with the load resistor R and having one end connected with an inductor LfAnd the other end of the anode is connected to a common ground on the AC side.
Preferably, the fully-controlled inverter bridge consists of four MOSFET power tubes Sap、San、Sbp、SbnAnd (4) forming.
Preferably, the relationship between the double frequency power of the suppression circuit satisfies:
PL_2ω+2PC_2ω＝Pin_2ω-Po_2ω
wherein, PL_2ωIs the double frequency ripple power on the inductor L; pC_2ωIs a single capacitor C1Or C2The second-order frequency ripple power; pin_2ωRepresenting the double frequency power of the input DC power supply; po_2ωIs the double frequency power output by the AC side.
Preferably, the capacitance C1Is smaller than the capacitance C2The capacity value of (c).
The control method of double-frequency ripple suppression circuit of single-phase inverter controls the double-frequency ripple on the input inductor L to 0 and utilizes the capacitor C1And C2The double frequency wave energy in the system is absorbed, and the capacitor voltage wave is controlled to be in a complementary state.
As a further improvement of the invention, the device comprises a direct current component and a frequency doubling ripple component which are respectively controlled;
the control of the direct current component adopts a voltage-current double closed loop structure: real-time sampling direct-current bus voltage signal Vdc_linkFiltering the ripple signal with fixed frequency to obtain DC component
Figure BDA0002674462420000031
Command signal corresponding to given steady-state average value of DC bus voltage
Figure BDA0002674462420000032
Differential pass voltage loop regulator GvObtaining a reference value of a DC component of an input inductor current
Figure BDA0002674462420000033
Real-time sampled inductive current signal iLFiltering out double power frequency component after passing through a trap to obtain the DC component of the inductive current and the reference value of the current signal
Figure BDA0002674462420000034
Differential input current loop regulator GIObtaining a controlled quantity D of a DC componentdq；
Suppression of double frequency ripple on inductor current: inductive current i through real-time samplingLAfter passing through a trap, the second harmonic component is removed, and the residual DC component is mixed with the sampled signal iLObtaining low-frequency ripple i by differencerippleThe reference value of the current ripple is 0, and the difference is obtained by the current ripple and the reference value of the current ripple through a current ripple regulator GRObtaining the disturbance control quantity ddq_2ω；
Complementary control of capacitor voltage ripple: is to bedqAnd ddq_2ωAnd the input capacitance voltage complementation calculation module calculates to obtain the control quantity ddAnd dqThen, a corresponding modulation wave C is generated by the amplitude limiterdAnd Cq(ii) a Generation of switching signal S by means of modulated wavewdAnd SwqImplementation for MOSFET power tube SdAnd MOSFET power transistor SqAnd (4) controlling.
As a further improvement of the present invention, the formula of the capacitance-voltage complementation calculation module is as follows:
Figure BDA0002674462420000035
wherein d isd,dqRespectively representing MOSFET power tubes SdAnd SqDuty ratio of individual conduction, ddqRepresentative of MOSFET power transistors SdAnd SqA duty cycle of common conduction; θ is a phase difference between the ac side voltage and the current, and is determined by the ac side load.
As a further improvement of the present invention, two capacitance values of the capacitor are selected to satisfy the following requirements:
Figure BDA0002674462420000036
wherein: k, selecting a capacitor voltage ratio according to the relation between the ripple ratio and the capacitor voltage ratio; vm、ImVoltage current peak value, K, for the AC output of the converterVIs the direct current bus voltage ripple coefficient, omega is the power frequency angular frequency,
Figure BDA0002674462420000041
is the dc component of the dc bus voltage.
Compared with the prior art, the invention has the following advantages:
the control method of the invention does not need to add extra circuit devices at first, and only needs to optimize the circuit parameters and the control method on the basis of the original circuit; then, the relation of the double-frequency ripple power in the system is clarified, the state of the lowest double-frequency energy of the system is obtained, the control idea that the double-frequency ripple of the input inductive current is controlled to be 0 is provided, and the direct current bus capacitor is utilized to absorb the double-frequency ripple of the system, so that the double-frequency ripple on the input inductive current is inhibited, and the capacitor voltage ripple on the direct current bus is controlled. The control method provided based on the control thought separately adjusts the direct current component and the frequency doubling ripple component of the system, controls the suppression of the inductor current ripple to reduce the frequency doubling ripple on the inductor by 94.7%, adopts the complementary control of the direct current bus capacitor voltage ripple, reduces the low frequency ripple of the direct current bus capacitor voltage by 76.7% by modifying the modulation wave, and is simple and effective.
Furthermore, the control and modulation method for suppressing the double-frequency ripple comprises double control of an inductive current ripple and a capacitor voltage ripple in the system. The state that the double-frequency power is the lowest in the system is obtained through modeling and analyzing the double-frequency ripple power of the system, namely the inductance current ripple is controlled to be close to 0, and the capacitance current ripple is controlled to be in a complementary state, so that the adverse effect of low-frequency ripple on a direct current side and an alternating current side is avoided while energy conversion of the direct current side and the alternating current side is realized, and the requirements of the system on passive elements such as inductance and capacitance are greatly reduced.
Drawings
FIG. 1 is a circuit topology employed by the present invention;
FIG. 2a is a circuit topology of a converter used in the present invention in a mode one case;
FIG. 2b is a circuit topology diagram of a converter used in the present invention in the case of mode two;
FIG. 2c is a circuit topology diagram of a converter used in the present invention in the case of the three modes;
FIG. 3a is a vector diagram of the double-frequency ripple power in the circuit topology analyzed by the present invention;
FIG. 3b is a vector diagram of the double frequency power in the system when the circuit topology analyzed by the present invention has the minimum double frequency power.
FIG. 4 is a graph showing the relationship between the non-complementation interval of the capacitor voltage ripple complementation and the capacitance-to-capacitance ratio of two DC buses;
FIG. 5 is a diagram of the main waveforms inside the circuit when the existence of the non-complementary sections is considered by the control method of the present invention;
FIG. 6 is a reference diagram of the capacitance value of the DC bus capacitor according to the present invention;
FIG. 7 is a block diagram of a control method of the present invention;
FIG. 8a shows a control method of a capacitor C without the present invention1And C2Voltage waveform of (1), inductor current waveform iL；
FIG. 8b is a DC bus voltage waveform without the control method of the present invention; the AC output voltage current waveform and the modulation wave.
FIG. 8C shows a capacitor C using the control method of the present invention1And C2Voltage waveform, DC bus voltage waveform and input inductance current waveform;
FIG. 8d is a graph of current waveform using AC output voltage; control quantity ddqA waveform; modulated wave CdAnd CqThe waveform of (a);
Detailed Description
The present invention will be described in detail below with reference to the accompanying drawings and specific embodiments.
The key point of the invention is that the relationship between the double-frequency energy in the system is obtained according to the double-frequency ripple model analysis of the system, thereby obtaining the basic idea of controlling the double-frequency ripple; and a method for separately controlling the direct current component and the frequency doubling ripple component is adopted, and a method for complementarily controlling the inductive current ripple suppression and the direct current bus capacitor voltage ripple is designed.
As shown in FIG. 1, the circuit diagram adopted by the control method of the present invention includes an input DC power supply VdcAn inductor L, two MOSFET power transistors SdAnd SqTwo diodes D1And D2Two capacitors C of DC bus1And C2And a set of fully-controlled inverter bridges. AC output filter inductor LfAnd a filter capacitor CfAnd a load resistor R.
Input DC power supply VdcThe anode of the inductor is connected with one end of an input inductor L, and the other end of the inductor L is connected with a diode D1Anode of and MOSFET power tube SdOf the substrate. Diode D1Cathode and capacitor C1The positive electrodes of (a) and (b) are connected. MOSFET power tube SdSource electrode of and MOSFET power tube SqAre connected. MOSFET power tube SqSource electrode ofCurrent source VdcCathode and diode D2The cathode of (1). Capacitor C1Negative pole of the capacitor C2Anode and MOSFET power tube SdOf the substrate. Capacitor C2Negative electrode of (D) is connected with diode2Of (2) an anode. Capacitor C1And a capacitor C2The input of the full-control inverter bridge is connected in series. The positive output of the full-controlled inverter bridge is connected with an inductor LfThe output of the negative pole of the full-control inverter bridge is connected with the common ground at the AC side. Capacitor CfConnected in parallel with the load resistor R and having one end connected with an inductor LfAnd the other end of the anode is connected to a common ground on the AC side.
The system frequency doubling ripple control method comprises the following steps:
firstly, establishing a double-frequency ripple model of a system to obtain the relation between double-frequency power in the system:
PL_2ω+2PC_2ω＝Pin_2ω-Po_2ω
wherein P isL_2ωIs the double frequency ripple power on the inductor L; pC_2ωIs a single capacitor C1Or C2The second-order frequency ripple power; pin_2ωRepresenting the double frequency power of the input DC power supply; po_2ωIs the double frequency power output by the AC side.
According to the analysis result of the relation vector diagram between the double frequency powers, the following results are obtained: when the double-frequency ripple power does not exist on the inductor, the double-frequency ripple power cannot be generated on the input power supply, and the double-frequency power required to be stored by the energy storage element in the system is the minimum, namely Po_2ω. In order to reduce the requirement of a system on passive elements, the idea of controlling the frequency doubling ripple of the system is as follows: the double frequency ripple on the inductor L is controlled to 0 by the capacitor C1And C2Absorbing the double frequency ripple energy in the system. In order to reduce the requirement of the system on the direct current bus capacitance, the complementary control of capacitor voltage ripples is adopted, and a capacitor C with a small capacitance value is selected1And a relatively large capacitance value capacitor C2。
The concrete description is as follows:
according to MOSFET power transistor SdAnd MOSFET power transistor SqIn different states, this applicationThree modes of operation of the transducer are contemplated, as shown in fig. 2a to 2 c.
Memory MOSFET power tube SdConducting MOSFET power tube SqThe conducted equivalent circuit is a converter mode I; at this time, the diode D1And a diode D2Are all reverse biased and cannot be conducted; inductor L charging, DC bus capacitor C1And C2In series, as shown in fig. 2 a.
Memory MOSFET power tube SdConducting MOSFET power tube SqThe switched-off equivalent circuit is a converter mode II; at this time, the diode D1Is reverse biased and is not conducted, diode D2Conduction, inductor L discharge, capacitor C1Discharge, capacitance C2And (6) charging. As shown in fig. 2 b.
Memory MOSFET power tube SdOff, MOSFET power transistor SqThe conducted equivalent circuit is a converter mode III; at this time, the diode D2Is reverse biased and is not conducted, diode D1Conduction, inductor L discharge, capacitor C1Charging, capacitance C2And (4) discharging. As shown in fig. 2 c.
From the above analysis, an average model of the transducer can be built:
Figure BDA0002674462420000071
wherein L is inductance of the inductor, C1、C2Is the capacitance value of the DC bus capacitor iLIs an inductive current, vC1Is a capacitor C1Upper voltage, vC2Is a capacitor C2Upper voltage, ddIs a MOSFET power tube SdConduction, SqDuty cycle of turn-off, dqIs a MOSFET power tube SdOff, SqDuty ratio of conduction idc_linkIs the current on the dc bus.
The conventional three-level boost circuit generally selects two identical DC bus capacitance values, namely C1＝C2And the capacitance is large enough to make the two DC capacitors equally divide the DC bus voltage, and dd＝dqThe averaging model can be simplified as:
Figure BDA0002674462420000072
wherein v isCRepresenting the capacitance of a single capacitor on the DC bus, ddqIs a MOSFET power tube SdConduction, SqThe duty ratio of switching on simultaneously satisfies: ddq＝1-dd-dq。
The expression of the output voltage and the current at the AC side is as follows:
Figure BDA0002674462420000073
wherein Vm、ImThe peak values of the output voltage current on the ac side, and θ is the phase angle at which the ac current lags behind the ac voltage.
The dc bus current expression may be further expressed as:
idc_link＝Idc_link+i2ω(t)＝Idc_link+I2ωcos(2ωt-α) (4)
wherein I2ωRepresents the peak value of the double frequency ripple of the direct current side current, and alpha is the phase angle of the double frequency ripple. From the conservation of energy, one can obtain:
Figure BDA0002674462420000081
differentiating the average model obtained in the step (3), and substituting the average model obtained in the step (4) into the average modelLAnd vCTwo second order differential equations of (1):
Figure BDA0002674462420000082
and (3) solving the two differential equations in the step (6) to obtain a double-frequency ripple model of the system:
Figure BDA0002674462420000083
further, equation (7) is simplified according to the frequency doubling of the system
Figure BDA0002674462420000084
VCFor the DC component of the voltage on a single capacitor of the DC bus, vC_2ωIs the double frequency ripple component of the voltage on the capacitor. I isLIs the direct component of the current in the inductor L, iL_2ωIs a double frequency ripple component of the current on the inductor L. VC_2ωIs the amplitude of a double frequency ripple of the voltage on the capacitor, IL_2ωIs twice the amplitude of the frequency ripple of the inductive current.
The double frequency power of the input power supply is:
Pin_2ω＝VdciL_2ω＝VdcIL_2ωcos(2ωt-α) (9)
the output power at the AC side is as follows:
Figure BDA0002674462420000091
from the average model of the circuit, the equation of state for the second harmonic component can be derived:
Figure BDA0002674462420000092
according to equation (11), the double frequency power stored on the inductor and capacitor is:
Figure BDA0002674462420000093
Figure BDA0002674462420000094
in combination with (9) (10) (12) (13), the system stores a double frequency power having the following relationship:
PL_2ω+2PC_2ω＝Pin_2ω-Po_2ω (14)
and expressing the double frequency power of each part in a vector form:
Figure BDA0002674462420000095
the corresponding vector diagram is shown in figure 3 a. Since the output power cannot be changed depending on the load, when the double frequency power inputted from the dc power supply is 0, the sum of the double frequency power stored in the capacitor and the inductor is the minimum, as shown in fig. 3 b. The double frequency power stored in the inductor is also 0, and the double frequency power stored in the system is completely stored in the capacitor. In this case, α ═ θ is satisfied.
The invention also discloses a single-phase inverter frequency-doubled ripple control method based on the three-level boost cascaded full bridge, which specifically comprises the following steps:
the control of the direct current component and the frequency doubling ripple component in the system is separately carried out. The control of the direct current component adopts a voltage-current double closed loop structure, and the direct current bus voltage signal V is sampled in real timedc_linkAfter passing through a wave trap, the ripple signal with fixed frequency (here, double frequency component) is filtered out to obtain a DC component
Figure BDA0002674462420000096
Command signal corresponding to given steady-state average value of DC bus voltage
Figure BDA0002674462420000101
Differential pass voltage loop regulator GvObtaining a reference value of a DC component of an input inductor current
Figure BDA0002674462420000102
Real-time sampled inductive current signal iLFiltering out a twofold work after passing through a trapThe frequency component obtains the DC component of the inductive current and the reference value of the current signal
Figure BDA0002674462420000103
Differential input current loop regulator GIThe controlled quantity D of the DC component can be obtaineddq. Wherein the voltage loop regulator GvCurrent loop regulator GIAre all PI (proportional integral) regulators.
Inductive current ripple suppression inductive current i requiring real-time samplingLAfter passing through a trap filter, the second harmonic component is removed, and the residual DC component is mixed with the real-time sampling signal iLObtaining real-time low-frequency ripple i by differencerippleThe reference value of the current ripple is 0, and the difference is input into the current ripple regulator GRObtaining the disturbance control quantity ddq_2ω. Will DdqAnd ddq_2ωThe sum is transmitted to a capacitance-voltage complementary calculation module. Wherein G isRIs a PI regulator.
The control design related to the voltage ripple complementation of the direct current bus capacitor is as follows:
one switching period TsInner two DC bus capacitors C1And C2The respective output charges are:
Qout＝idc_linkTs (16)
when the circuit works in a mode 3, the capacitor C1Charging, the charge added in one switching cycle is:
QC1＝dqiLTs (17)
when the circuit works in a mode 2, the capacitor C2Charging, the charge added in one switching cycle is:
QC2＝ddiLTs (18)
the capacitor C in one switching cycle1The increase in the upper voltage is:
Figure BDA0002674462420000104
on the capacitor C2The increase in voltage is:
Figure BDA0002674462420000105
capacitor C1The decrement of the upper voltage is:
Figure BDA0002674462420000106
capacitor C2The decrement of the upper voltage is:
Figure BDA0002674462420000111
in order to keep the voltage of the dc link constant, it is desirable that the voltage variation of the two capacitors in each switching period is 0, and then:
Figure BDA0002674462420000112
substituting (22) into (18) to (21) simplifies:
Figure BDA0002674462420000113
since the input power at the dc side is equal to the dc component of the output power, the fluctuating power is absorbed by the capacitor, giving:
Figure BDA0002674462420000114
s in inverter bridgeapThe duty cycle of the conduction is:
Figure BDA0002674462420000115
although the capacitance values of the two capacitors are different, the double frequency ripple is controlled to be in a complementary state, but the voltage direct current components on the capacitors are the same and are half of the voltage on the direct current side. According to the working principle of the circuit, the current variation on the input inductor in one switching period is 0:
to obtain:
Figure BDA0002674462420000116
the relationship between the current of the dc bus and the output ac current can be expressed as:
Figure BDA0002674462420000117
d can be solved by substituting (25), (27) and (28) into (24)d，dq：
Figure BDA0002674462420000118
Wherein: d is not less than 0d≤1-ddq，0≤dq≤1-ddq。
Due to dd、dqThe control method of the complementary capacitor voltage has a non-complementary interval:
let d d0, according to equation (29), the first non-complementary segment is expressed angularly as:
Figure BDA0002674462420000121
its length is expressed in degrees as:
Figure BDA0002674462420000122
let dd＝1-ddqAnd in the same way, the second non-complementary interval is:
Figure BDA0002674462420000123
its length is expressed in degrees as:
Figure BDA0002674462420000124
FIG. 4 shows the length of two non-complementary sections and the capacitance C of the DC bus1、C2The relationship between the ratios. The calculation result shows that the two non-complementary sections are symmetrically distributed and have the same length.
MOSFET power transistor S for designing parameters of semiconductor devicedDiode D1Reverse voltage greater than capacitance C1A peak value of the upper voltage; MOSFET power Sq，D2Reverse voltage greater than capacitance C2The peak value of the upper voltage.
Regarding the design of the inductor L parameters, the ripple Δ i of the inductor current over one switching cycle is notedLThe size satisfies the following conditions:
Figure BDA0002674462420000125
in the above formula, ddqIs a MOSFET power tube Sd、SqSimultaneous on duty cycle, TsIs the switching period of the MOSFET power tube, and L is the inductance value of the inductor, wherein the MOSFET power tube SdAnd MOSFET power transistor SqThe switching period and the switching frequency of (2) are the same.
Recording the ripple factor K of the inductive currentIComprises the following steps:
Figure BDA0002674462420000126
in the above formula，iLFor inductor current, Δ iLIs the inductor current ripple.
The inductance value L obtained by combining the formula (34) and the formula (35) satisfies the following condition:
Figure BDA0002674462420000131
in order to control the voltage complementation of the DC bus capacitor, the DC bus capacitor C needs to be designed1(capacitance of small capacitance) and C2(large capacitance) parameters. Two DC bus capacitors C of traditional circuit1And C2The values of (A) are the same and are all C. First assume that the capacity values satisfy: c1+C2＝2C。
If the traditional circuit adopts the suppression of the inductive current ripple, only the DC bus capacitor C is used for absorbing the double frequency ripple, and the voltage ripple of the DC bus is as follows:
Figure BDA0002674462420000132
wherein Vm、ImThe peak values of the ac output voltage and current. ω is the angular frequency of the ac output. C is the capacitance value of a single capacitor of the direct current bus. U shape+Representing the maximum value of the DC bus capacitor voltage in steady state, U-Representing the minimum value of the dc bus capacitor voltage at steady state.
In order to realize the capacitance-voltage complementation, according to fig. 5, the ripple wave is generated because the uncontrollable interval exists, so that the energy of the part cannot be balanced, and assuming that the output voltage and the current are in the same phase, the two uncontrollable intervals are symmetrical and have the same length, and the generated ripple waves are also the same:
Figure BDA0002674462420000133
the dc bus voltage ripple ratio of the two methods is:
Figure BDA0002674462420000134
recording direct current bus voltage ripple factor KVComprises the following steps:
Figure BDA0002674462420000135
the value of the direct current bus capacitor C meets the following conditions:
Figure BDA0002674462420000141
wherein: vm、ImVoltage current peak value, K, for the AC output of the converterVIs the ripple coefficient of DC bus voltage, omega is the angular frequency of power frequency, Vdc_LinkThe average value of the dc bus voltage.
According to fig. 6, a proper capacitor voltage ratio is selected according to the relation between the ripple ratio and the capacitor voltage ratio, and the ratio K is generally selected to be between 0.04 and 0.1, so that the ripple of the dc bus voltage can be reduced by 70% to 80%. The value of the direct current bus capacitor meets the following requirements:
Figure BDA0002674462420000142
thereby establishing a control block diagram of the converter as shown in fig. 7. The control of the direct current component and the frequency doubling ripple component in the system is separately carried out, so that the energy conversion of the converter and the suppression of the ripple are independent.
Figure BDA0002674462420000143
The direct current bus voltage steady-state average value instruction signal is directly given by a digital controller. Vdc_linkFiltering the direct current bus voltage signal obtained by real-time sampling to obtain a double-frequency ripple signal after passing through a wave trap to obtain a real-time direct current bus voltage component, and
Figure BDA0002674462420000144
difference is input to PI regulator GVObtaining the reference value of the DC component of the input inductive current
Figure BDA0002674462420000145
Real-time sampled inductive current signal iLFiltering out double power frequency component after passing through a trap to obtain the DC component of the inductive current and the reference value of the current signal
Figure BDA0002674462420000146
Difference is input to PI regulator GIThe control variable D of the system can be obtaineddq. The traditional double-loop control method of the voltage outer loop and the current inner loop is adopted to accurately control the input inductive current and the direct current component of the direct current bus voltage.
Real-time sampled inductive current iLAfter passing through a trap filter, the second harmonic component is removed, and the residual DC component is mixed with the real-time sampling signal iLObtaining the inductance current frequency doubling ripple i by differencerippleThe reference value of the current ripple is 0, and the difference is input into a PI regulator G of the regulatorRObtaining the disturbance control quantity ddq_2ω。
Will DdqAnd ddq_2ωAnd ddqAn input capacitance voltage complementary calculation module for calculating the control quantity d according to the formula (29)d、dqThen, a corresponding modulation wave C is generated by the amplitude limiterdAnd Cq. Generation of switching signal S by means of modulated wavewdAnd SwqImplementation for MOSFET power tube SdAnd MOSFET power transistor SqAnd (4) controlling.
The direct control of the current ripple of the inductor at the direct current side and the voltage ripple of the direct current bus basically eliminates the double frequency ripple of the current on the inductor, and reduces the requirement of low pass on the inductor. The voltage ripple on the capacitor is controlled to be in a complementary state, and the requirements of the direct current bus voltage ripple and the direct current bus capacitor are reduced while the power conversion of the converter is realized.
In order to verify the theoretical analysis of the converter, the invention provides a design example.
The converter parameters are as follows: vdc＝80V,Pin＝1.25kW,Vdc_link＝200V,M＝0.8,fs＝10kHz,L＝2mH,C1＝100uF,C2＝2000uF,C＝1050uF,Lf＝2mH,Cf＝10uF,R＝10.67Ω,fline50Hz, wherein flineAt power frequency, C1、C2Two capacitance values of a direct current bus designed for a capacitance-voltage complementation method; c is two same capacitance values of the direct current bus in the traditional method.
FIG. 8a shows the voltage waveforms v on two DC bus capacitors without using the control method proposed by the present inventionC1And vC2And the waveform i of the input inductor currentL. According to the modeling result of the frequency doubling ripple, the theoretical voltage ripple and the theoretical current ripple can be solved as follows: vC_2ω＝14.6517V；IL_2ω9.3276 a; the simulation result is basically consistent with the model calculation result. And the phase difference between the capacitor voltage and the inductor current is 90 deg. Both prove the correctness of the frequency doubling ripple model.
FIG. 8b shows the DC bus voltage waveform V without the control method proposed by the present inventiondc_Linkd.C. output voltage vacCurrent iLfWaveform, and modulated wave CdAnd Cq. The output voltage THD is 7.467%, and the output current THD is 9.269%.
FIG. 8c shows the voltage waveforms v on two DC bus capacitors using the control method of the present inventionC1And vC2DC bus voltage waveform Vdc_linkAnd the waveform i of the input inductor currentL. The direct current bus voltage ripple is reduced to about 7V, and is reduced by 76.6% compared with 30V of the traditional control method. The inductor current ripple is controlled to be substantially constant within 1V.
FIG. 8d shows the AC side output voltage v using the control method of the present inventionacAnd current iLfThe waveform of (a); control variable ddqWave form of (2), modulated wave CdAnd CqThe waveform of (2). The voltage THD was 1.61% and the current THD was 5.611%. The quality of the output voltage and current waveform is remarkably improvedIt is good.
Finally, it should be noted that the above examples are only for illustrating the technical solutions of the present invention, and are not intended to limit the embodiments. It will be apparent to those skilled in the art that various other changes and modifications can be made in the above-described embodiments without departing from the spirit and scope of the invention, and it is intended that all such changes and modifications be within the scope of the invention as defined by the appended claims. The scope of the invention is defined by the appended claims and equivalents thereof.

Claims (6)

1. The control method of the double-frequency ripple suppression circuit of the single-phase inverter is characterized in that the circuit comprises the following steps: input DC power supply VdcInductor L, MOSFET power tube SdMOSFET power tube SqDiode D1Diode D2Capacitor C1Capacitor C2Full-control inverter bridge Sap、San、Sbp、SbnFilter inductor LfFilter capacitor CfAnd a load resistance R;
the input DC power supply VdcThe anode of the inductor is connected with one end of an input inductor L, and the other end of the inductor L is connected with a diode D1Anode of and MOSFET power tube SdA drain electrode of (1); diode D1Cathode and capacitor C1The positive electrodes of the two electrodes are connected; MOSFET power tube SdSource electrode of and MOSFET power tube SqThe drain electrodes of the two electrodes are connected; MOSFET power tube SqSource electrode of the transistor is connected with a direct current power supply VdcCathode and diode D2A cathode of (a); capacitor C1Negative pole of the capacitor C2Anode and MOSFET power tube SdA source electrode of (a); capacitor C2Negative electrode of (D) is connected with diode2The anode of (1); capacitor C1And a capacitor C2The input end of the full-control inverter bridge is connected in series; the positive output of the full-control inverter bridge is connected with an inductor LfThe output of the negative pole of the full-control inverter bridge is connected with the common ground at the AC side; capacitor CfConnected in parallel with the load resistor R and having one end connected with an inductor LfThe other end of the anode is connected with the common ground at the AC side;
control methodThe method comprises the following steps: the double frequency ripple on the input inductor L is controlled to 0 by the capacitor C1And C2Absorbing the double frequency ripple energy in the system, and controlling the capacitor voltage ripple to be in a complementary state;
the method specifically comprises a direct current component and a frequency doubling ripple component which are respectively controlled;
the control of the direct current component adopts a voltage-current double closed loop structure: real-time sampling direct-current bus voltage signal Vdc_linkFiltering the ripple signal with fixed frequency to obtain DC component
Figure FDA0003168853430000011
Command signal corresponding to given steady-state average value of DC bus voltage
Figure FDA0003168853430000012
Differential pass voltage loop regulator GvObtaining a reference value of a DC component of an input inductor current
Figure FDA0003168853430000013
Real-time sampled inductive current signal iLFiltering out double power frequency component after passing through a trap to obtain the DC component of the inductive current and the reference value of the current signal
Figure FDA0003168853430000014
Differential input current loop regulator GIThe controlled quantity D of the DC component can be obtaineddq；
Suppression of double frequency ripple on inductor current: inductive current i through real-time samplingLAfter passing through a trap, the second harmonic component is removed, and the residual DC component is mixed with the sampled signal iLObtaining low-frequency ripple i by differencerippleThe reference value of the current ripple is 0, and the difference is obtained by the current ripple and the reference value of the current ripple through a current ripple regulator GRObtaining the disturbance control quantity ddq_2ω；
Complementary control of capacitor voltage ripple: is to bedqAnd ddq_2ωAnd the input capacitor voltage complementation calculation module calculates to obtainControl quantity ddAnd dqThen, a corresponding modulation wave C is generated by the amplitude limiterdAnd Cq(ii) a Generation of switching signal S by means of modulated wavewdAnd SwqImplementation for MOSFET power tube SdAnd MOSFET power transistor SqAnd (4) controlling.
2. The method of claim 1, wherein the fully-controlled inverter bridge is formed by four MOSFET power transistors Sap、San、Sbp、SbnAnd (4) forming.
3. The method of claim 1, wherein the relationship between the double frequency power of the suppression circuit satisfies:
PL_2ω+2PC_2ω＝Pin_2ω-Po_2ω
wherein, PL_2ωIs the double frequency ripple power on the inductor L; pC_2ωIs a single capacitor C1Or C2The second-order frequency ripple power; pin_2ωRepresenting the double frequency power of the input DC power supply; po_2ωIs the double frequency power output by the AC side.
4. The method of claim 1, wherein the capacitance C is1Is smaller than the capacitance C2The capacity value of (c).
5. The control method of claim 1, wherein the capacitance-voltage complementation calculation module has the formula:
Figure FDA0003168853430000021
wherein d isd,dqRespectively representing MOSFET power tubes SdAnd SqDuty ratio of individual conduction, ddqRepresentative of MOSFET power transistors SdAnd SqA duty cycle of common conduction; theta is the phase between the voltage and the current on the AC sideThe potential difference is determined by the load on the AC side.
6. The control method according to claim 1, wherein two capacitance values of the capacitor are selected to satisfy the following requirements:
Figure FDA0003168853430000022
wherein: k, selecting a capacitor voltage ratio according to the relation between the ripple ratio and the capacitor voltage ratio; vm、ImVoltage current peak value, K, for the AC output of the converterVIs the direct current bus voltage ripple coefficient, omega is the power frequency angular frequency,
Figure FDA0003168853430000023
is the dc component of the dc bus voltage.
CN202010943514.6A 2020-09-09 2020-09-09 Double-frequency ripple suppression circuit and suppression method of single-phase inverter Active CN112234808B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN202010943514.6A CN112234808B (en) 2020-09-09 2020-09-09 Double-frequency ripple suppression circuit and suppression method of single-phase inverter

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN202010943514.6A CN112234808B (en) 2020-09-09 2020-09-09 Double-frequency ripple suppression circuit and suppression method of single-phase inverter

Publications (2)

Publication Number Publication Date
CN112234808A CN112234808A (en) 2021-01-15
CN112234808B true CN112234808B (en) 2021-09-03

Family

ID=74116256

Family Applications (1)

Application Number Title Priority Date Filing Date
CN202010943514.6A Active CN112234808B (en) 2020-09-09 2020-09-09 Double-frequency ripple suppression circuit and suppression method of single-phase inverter

Country Status (1)

Country Link
CN (1) CN112234808B (en)

Families Citing this family (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN113659860B (en) * 2021-07-27 2023-05-30 广东志成冠军集团有限公司 Switching power amplifier, control method and control system thereof
CN113726210B (en) * 2021-07-30 2024-01-12 西安交通大学 Low-frequency ripple suppression circuit and method for direct-current bus of two-stage double-active-bridge grid-connected inverter
CN114142453B (en) * 2021-11-18 2024-02-13 厦门大学 Secondary current ripple suppression method based on active large capacitance
CN115051565B (en) * 2022-07-12 2024-07-16 西安交通大学 Grid-connected inverter of bidirectional half-bridge direct-current converter and ripple control method
CN115296552B (en) * 2022-08-16 2024-05-28 易事特储能科技有限公司 Low-interference high-efficiency TL-Boost control method
CN116937742B (en) * 2023-07-25 2024-02-13 浙江大学 Double frequency ripple current suppression circuit and method based on reconfigurable battery system
CN116760270B (en) * 2023-08-11 2023-11-07 西南交通大学 Boost-PFC converter for stabilizing voltage secondary ripple

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103023360A (en) * 2012-07-03 2013-04-03 中南大学 Single-phase alternating current (AC)/ direct current (DC) converter with secondary fluctuating power decoupling and control method thereof
CN109194113A (en) * 2018-08-02 2019-01-11 西安交通大学 The power factor corrector and its control method for having active power decoupling function

Family Cites Families (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN105141116B (en) * 2015-08-07 2017-12-26 华东交通大学 The method for reducing single-phase inverter intermediate DC link low-frequency ripple voltage
CN107196506A (en) * 2017-07-07 2017-09-22 华南理工大学 A kind of three-level Boost converter repeats dead beat Compound Control Strategy
CN111245238B (en) * 2020-03-25 2020-12-08 中车青岛四方车辆研究所有限公司 Three-level Boost circuit control method and system

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103023360A (en) * 2012-07-03 2013-04-03 中南大学 Single-phase alternating current (AC)/ direct current (DC) converter with secondary fluctuating power decoupling and control method thereof
CN109194113A (en) * 2018-08-02 2019-01-11 西安交通大学 The power factor corrector and its control method for having active power decoupling function

Also Published As

Publication number Publication date
CN112234808A (en) 2021-01-15

Similar Documents

Publication Publication Date Title
CN112234808B (en) Double-frequency ripple suppression circuit and suppression method of single-phase inverter
CN109194113B (en) Power factor corrector with active power decoupling function and control method thereof
CN113726210B (en) Low-frequency ripple suppression circuit and method for direct-current bus of two-stage double-active-bridge grid-connected inverter
CN107834886A (en) A kind of single-stage boost inverter and its control method
CN115051565A (en) Bidirectional half-bridge direct-current converter grid-connected inverter and ripple wave control method
WO2023226317A1 (en) Control method and system for vienna rectifier
CN115622424A (en) Secondary ripple voltage suppression method for direct-current bus of two-stage three-level AC/DC converter
CN112003463A (en) Single-phase PWM rectification DC side voltage secondary ripple suppression method
CN116683750A (en) IPOS-DC/DC converter cascading single-phase inverter and method
CN111181420B (en) Single-phase Vienna rectifier and control method thereof
CN117039976A (en) CLLC bidirectional resonant converter cascading grid-connected inverter and inhibition method thereof
CN205249078U (en) Z-source three-level inverter and air conditioning system
CN114123203A (en) Direct-current bus voltage ripple suppression strategy during voltage unbalance of alternating-current power grid
CN111049201B (en) Coordination control method for AC/DC power grid hybrid high-power interface converter
CN116317499A (en) Single-phase inverter based on flying capacitor type three-level boost and control method
CN112350590A (en) Uncontrolled rectifier harmonic compensation circuit and control method
CN104300820A (en) Digital control method of two-stage three-phase three-level photovoltaic grid-connected inverter
US11695322B2 (en) AC-side symmetrically-split single-phase inverter for decoupling
CN112117925B (en) DCM single-bridge-arm integrated split-source inverter control method for photovoltaic grid-connected occasions
CN106655738A (en) Electrolytic capacitor-free quasi single stage inverter and control method therefor
CN113541186A (en) Double closed-loop control method and system for single-phase LC type grid-connected inverter
CN109842317B (en) Differential converter based on Boost and Buck-Boost circuits and application thereof
CN104836465B (en) LC serial-type three-phase PWM rectifier current iterative learning control method
CN113824129B (en) Power compensation control for improving bidirectional power stability of grid-connected converter system
Ajmeera et al. A Novel Reduced Capacitance with Quasi-Z-Source Inverter for RES Application

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
GR01 Patent grant
GR01 Patent grant