CN1240177C - Active power filter method with D.C. source and branch impedance decoupling control and its system - Google Patents

Abstract

The present invention relates to an active power filtering method with a direct current source and branch impedance decoupling control and a system thereof, which belongs to the field of direct current power transmission. The method is characterized in that the method uses band-pass filtering for realizing the decoupling of frequency domain control; decoupling control over a branch circuit resistor and a reactor is formed on the basis of the filtering treatment of two shift phases with the phase shift mutual difference of 90 DEG; the total impedance of a filtering branch circuit on a concerned ripple frequency point is equal to zero or approaches to zero by adjustment; consequently, the ripple of the frequency point is filtered. The system is characterized in that after band-pass filtering and weighting summation are carried out on current on the filtering branch circuit, phase shift filtering, the summation, pulse width modulation and pulse drive are carried out on the filtering branch circuit for controlling the existing active power filter with a direct current source for supplying power. The method does not need system compensation for filtering low-frequency ripple noise. Simultaneously, the present invention solves a phase shift problem due to sampling and digital time delay, and has the advantages of clear principle and easy realization.

Description

Active power filtering method and system that the branch impedance decoupling zero control of DC source is arranged

Technical field

The present invention relates to a kind of active power filtering method and system that the branch impedance decoupling zero control of DC source is arranged, this method and system thereof can realize the control to the impedance of filter branch, and are zero to suppress and eliminate the low-frequency ripple noise by making the resistance value of filter branch in the ripple frequency band of being concerned about.The invention belongs to the direct current supply technical field.

Background technology

The Ripple Noise that comprises a large amount of low-frequency ranges by direct-current transmission voltage that the AC signal rectification is obtained or DC power supply voltage.These Ripple Noise will increase facility load on the DC transmission line of electric power system, produce electric energy loss, and its electromagnetic radiation is also with interference communications equipment.And the useful signal in the meeting of the Ripple Noise in the DC power supply interference load equipment has a strong impact on the normal operation of equipment.The method of traditional inhibition and elimination Ripple Noise is to adopt the filter circuit that is made of passive devices such as some resistance, inductance and electric capacity, and this is called as the passive filtering method.Along with the development of power electronic technology, the various devices that comprise power electronic device are applied to electric filtering, and this filtering method is called as active power filtering.Active power filtering can overcome that the traditional passive filtering parameter is inaccurate, the variation that can't adapt to frequency and load, easily with shortcoming such as system generation vibration.And active power filtering method can reach better filtering performance.In the bigger occasion of power, adopt active electric filter device can also reduce floor space, reduce the filtering cost.

" the electric power system journal of Institute of Electrical and Electronics Engineers in May, 1998, (IEEE Transactions on PowerSystems) " the 13rd volume the 2nd interim delivered one piece of exercise question for " bank carries-this bank HVDC (High Voltage Direct Current) transmission system in the control principle analysis of the active dc filter that adopted of forest-road nurse current conversion station, (Analysis on the Control Principle of the Active DCFilter in the Lindome Converter Station of the Konti-Skan HVDC Link) " article.In this piece article, the author has described a kind of active electric filter device and control method thereof of the low-frequency ripple noise that can the filtering rectifier produces, and its theory diagram as shown in Figure 1.In existing this active electric filter device and control method thereof, after the direct voltage that comprises Ripple Noise inserts the input port of filter circuit, at first arrive output port through the smoothing reactor that seals in the circuit, output port is connecting passive filtration unit and the former limit of the coupling transformer that is in series with it between smoothing reactor and loop electrode, DC power supply inserts the secondary of coupling transformer by single-phase electricity die mould inverter circuit.The control method of existing this active electric filter device at first will be measured the output port current i by current transformer _Out, filtering i then _OutIn flip-flop and noise, extract ripple current i _lRipple current i _lObtain M signal i through comb filtering or the filtering of groove shape again _rRejection frequency is f if desired _iThe ripple composition, the transfer function F of selected comb filtering or the filtering of groove shape _a(s) should have approaching ± j2 π f _iTwo limits.Thus, transfer function F _a(s) amplitude-frequency response is at frequency f _iLocate also will level off to infinity.Then, M signal i _rHandle the reference wave signal u that obtains pulse-width modulation through system balance _a, reference wave signal u _aProduce control impuls through pulse-width modulation, control wave by power amplification after in order to drive single-phase electricity die mould inverter circuit.So, obtain reference wave signal u on the former limit of coupling transformer _aIn the voltage signal of low-frequency ac composition amplitude after amplifying.If supposing the input voltage ripple of whole filter is 0, and with reference wave signal u _aAs unique driving source, again with filter in current i that output port obtained _LaIn response, then can obtain the transfer function G of this input-output system according to circuit parameter _hThe transfer function characteristics of designed system balance part need satisfy in the document $F_c=\frac{1}{G_h}.$ Existing this active power filtering system transter model is shown in Figure 2.Input signal i among Fig. 2 _LhBe as reference wave signal u _aBe 0, the ripple voltage of input port is done the time spent separately, at the ripple current of filter output port.The transfer function of whole filtering control is:

$F_s\left(s\right)=\frac{i_l}{i_{\mathrm{lh}}}=\frac{1}{1+F_a}$

Because F _a(s) limit is F _s(s) so zero point is if the transfer function F of comb filtering or the filtering of groove shape _a(s) have and approach ± j2 π f _iLimit, then | F _s(± j2 π f _i) | will level off to 0, promptly filtering control system can blanketing frequency be f _iLow-frequency ripple.

Existing this active power filtering method need obtain system transter G _h, and need carry out system balance according to this and handle.The design of system balance processing section depends on circuit parameter, and particularly the system balance processing procedure also needs corresponding change when load changes.This has strengthened the difficulty that system balance is handled, and causes the control vibration easily, and can reduce the effect that ripple suppresses.Filtering Processing in the existing this active power filtering method has partly adopted comb filtering or groove shape filtering method, thereby makes the transfer function F of Filtering Processing _a(s) limit approaches ± j2 π f _iBut the limit of transfer function is more near the imaginary axis, and then Filtering Processing process itself just is difficult to stable the realization more.And, owing to can not make F _a(s) limit is equal to ± j2 π f _iSo existing method can't reach optimum ripple and suppress.In theory, rely on the output port ripple current all can't reach optimum ripple inhibition, and can only make the ripple voltage of output port and electric current maintain zero the less level that approaches as the active power filtering method of control input variable.The measurement of output port ripple current need filtering occupies larger proportion from the output port electric current flip-flop.Particularly when ripple is necessarily suppressed, output port ripple current composition will reduce, and this just requires measure portion to have higher precision, just can make this method reach better control effect.Existing this active power filtering control method is in the Digital Realization process, there is the time-delay that is difficult to compensate in the process of measuring sampling, algorithm computation and pulse-width modulation, this can cause bigger phase shift to the ripple of being concerned about frequency band, and has a strong impact on the effect that ripple suppresses.

Summary of the invention

The objective of the invention is to propose a kind of active power filtering method and system that the branch impedance decoupling zero control of DC source is arranged, wish to overcome the deficiency of existing active power filtering method, make control principle not be subjected to the influence of load variations, can realize that optimum ripple suppresses, reduce in design process requirement for restriction to precision, time-delay and the phase shift of each funtion part, and make control more stable, finally obtain better ripple and suppress effect.

The active power filtering method of the branch impedance decoupling zero control that DC source is arranged that the present invention proposes is a kind of decoupling zero that utilizes bandpass filtering to realize frequency domain control, handle the decoupling zero control that constitutes filter branch resistance and reactance based on two phase-shift filterings of 90 ° of phase shift mutual deviations again, equal zero or level off to zero in the total impedance of the ripple frequency of being concerned about by regulating filter branch, thereby realize the method to this frequency ripple filtering, this method contains successively and has the following steps:

The 1st step: measure the current signal i in the filter branch in parallel _APF

The 2nd step: with the current signal i that measures _APFImport one group of passband frequency and be respectively f ₁, f ₂..., f _mBandpass filtering treatment, obtain one group of ripple current signal i that comprises different frequency ripple composition ₁, i ₂..., i _mWherein, f ₁, f ₂..., f _mBe m the component frequency that needs the low-frequency ripple of filtering, m is a positive integer, and each bandpass filtering treatment is at its passband frequency f _iHas the highest amplitude gain A _i, i=1 wherein, 2 ..., m, and this bandpass filtering treatment realizes the threshold limit value of frequency domain decoupling zero greater than control system to the attenuation rate of other ripple frequency signal;

The 3rd step: the ripple current signal i that filtering is obtained ₁, i ₂..., i _mMultiply by frequency domain decoupling zero weight coefficient a respectively ₁, a ₂..., a _m, summation obtains first weighted sum current signal i then _Sum1, i.e. i _Sum1=a ₁I ₁+ a ₂I ₂+ ... + a _mI _mSimultaneously, the ripple current signal i that filtering is obtained ₁, i ₂..., i _mMultiply by frequency domain decoupling zero weight coefficient b respectively ₁, b ₂..., b _m, summation obtains second weighted sum current signal i then _Sum2, i.e. i _Sum2=b ₁I ₁+ b ₂I ₂+ ... + b _mI _mThe calculation procedure of above-mentioned frequency domain decoupling zero weight coefficient is as follows:

The 3-1 step: the voltage signal u that measures filter branch in parallel two ends _APFWith the current signal i in the filter branch in parallel _APF

The 3-2 step: by the voltage signal u that measures _APFWith current signal i _APFCalculate filter branch respectively at frequency f ₁, f ₂..., f _mImpedance R ₁+ jX ₁, R ₂+ jX ₂..., R _m+ jX _mWherein, R ₁, R ₂..., R _mBe the active component of branch impedance, X ₁, X ₂..., X _mIt is the reaction component of branch impedance;

The 3-3 step: according to the impedance of filter branch, the adjusting control of process parameter obtains one group of resistance Control Parameter p ₁, p ₂..., p _mWith one group of reactance Control Parameter q ₁, q ₂..., q _m

The 3-4 step: with the resistance Control Parameter p that obtains _iWith reactance Control Parameter q _iCarry out decoupling zero according to following formula and handle, and obtain frequency domain decoupling zero weight coefficient a _iAnd b _i:

$\left(\begin{array}{c}{a}_i\\ b_i\end{array}\right)=\frac{A_{\mathrm{\Δ}}}{A_i{\·A}_H\·n\·U_d}\·\left(\begin{array}{cc}\cos {\mathrm{\θ}}_i& \sin {\mathrm{\θ}}_i\\ -\sin {\mathrm{\θ}}_i& \cos {\mathrm{\θ}}_i\end{array}\right)\·\left(\begin{array}{c}{p}_i\\ q_i\end{array}\right);$

Wherein, i=1,2 ..., m, A _ΔBe that the modulation signal amplitude that adopts, A are handled in pulse-width modulation _iBe that the passband frequency is f in aforementioned the 2nd step _iThe highest amplitude gain of bandpass filtering treatment, n is the no-load voltage ratio of coupling transformer, U _dBe output voltage for single-phase electricity die mould inverter circuit direct current power source supplying power, θ _iBe from current signal i _APFMeasuring process, through bandpass filtering treatment, weighted sum, first phase-shift filtering processing, summation, pulse-width modulation, pulsed drive, voltage inversion and transformer-coupled a series of processing, to the whole process that obtains the former limit of coupling transformer controlled voltage at f _iTotal phase shift of frequency;

The 4th step: with first weighted sum current signal i of above-mentioned the 3rd step acquisition _Sum1After first phase-shift filtering processing, obtain first phase-shift filtering signal i _Hb1Simultaneously, second weighted sum current signal i that above-mentioned the 3rd step is obtained _Sum2After second phase-shift filtering processing, obtain second phase-shift filtering signal i _Hb2If the transfer function of first and second phase-shift filtering processing is used H respectively _H1(s) and H _H2(s) expression, then in needing the ripple frequency band of filtering, the transfer function characteristics that these two phase-shift filterings are handled satisfies following relation:

$\left\{\begin{array}{c}\left|H_{H1}\left(\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="C0313763300232.png,C0313763300241.png"\; state="\{\{state\}\}">j</figure-callout>}2\mathrm{\&pi;f}\right)\right|=\left|H_{H2}\left(\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="C0313763300232.png,C0313763300241.png"\; state="\{\{state\}\}">j</figure-callout>}2\mathrm{\&pi;f}\right)\right|=A_H\\ \frac{H_{H2}\left(\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="C0313763300232.png,C0313763300241.png"\; state="\{\{state\}\}">j</figure-callout>}2\mathrm{\&pi;f}\right)}{H_{H1}\left(\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="C0313763300232.png,C0313763300241.png"\; state="\{\{state\}\}">j</figure-callout>}2\mathrm{\&pi;f}\right)}=j\end{array}\right.$

Wherein, f is the interior frequency of ripple frequency band that needs filtering, and f＞0; A _HIt is arbitrarily positive constant.The error threshold that the error of above-mentioned transfer function characteristics sets less than the requirement of foundation control precision;

The 5th step: with above-mentioned first and second phase-shift filtering signal i _Hb1And i _Hb2Summation obtains the reference wave signal that pulse-width modulation is used

i _Ref, i.e. i _Ref=i _Hb1+ i _Hb2

The 6th step of gadolinium: with reference signal i _RefCarry out pulse-width modulation and handle back acquisition one prescription wave pulse signal; The way of square-wave pulse signal equals power electronic device number controlled in the single-phase electricity die mould inverter circuit, the pulse generation frequency that pulse-width modulation is handled is greater than 2 times of the ripple frequency band upper limit that needs filtering, and the duty ratio of the potential pulse of being exported by the single-phase electricity die mould inverter circuit of this prescription wave pulse signal control satisfies following formula:

A wherein _ΔBe that the modulation signal amplitude that adopts is handled in pulse-width modulation, get the positive count value;

The 7th step: will be above-mentioned the square-wave pulse signal of the 6th step acquisition carry out going to drive after the power amplification power electronic device in the single-phase electricity die mould inverter circuit, have the duty ratio of potential pulse of the single-phase electricity die mould inverter circuit output of DC power supply in order to change, and make the duty ratio of this potential pulse satisfy the requirement in the 6th step;

The 8th step: above-mentioned potential pulse is loaded into the coupling transformer secondary, and act on the filter branch that is constituted with passive filtration unit series connection by the former limit of coupling transformer, in order to obtain the voltage and current output signal of low-frequency ripple after filtered at the circuit output end mouth that is in parallel with this filter branch.

The active power filtering system that proposes according to the active power filtering method of the aforesaid branch impedance decoupling zero control that DC source arranged consists of:

Active power filter circuit, it contains: the smoothing reactor on the end line of going into that is serially connected in the direct voltage input signal that comprises Ripple Noise; Smoothing reactor that is connected in parallel on output port that is made of the former limit series connection in order to the passive filtration unit of bearing direct voltage and coupling transformer and the filter branch between the loop electrode: output is linked into the single-phase electricity die mould inverter circuit of coupling transformer secondary: output is linked into the DC power supply of single-phase electricity die mould inverter circuit input:

Be installed in the current transformer on the filter branch;

Be installed in the voltage transformer between the filter branch two ends;

The one group of band pass filter that is connected in series with the measurement output of current transformer, weighted sum circuit, first and second all-pass filters, adder and pulse-width modulation circuits of constituting by multiplier and adder successively;

Frequency domain decoupling zero weight coefficient counting circuit comprises: measure the one group difference corresponding f of the filter branch voltage and current signal of acquisition as input with the voltage transformer summation current transformer ₁, f ₂..., f _mThe branch road resistance calculations circuit of frequency and branch road reactance counting circuit, and the adjusting control circuit of parameter and decoupling zero treatment circuit;

Pulse driving circuit, its input is connected with the pulse signal that above-mentioned pulse-width modulation is partly exported, and its output then is connected with the trigger control end of single-phase electricity die mould inverter circuit;

In above-mentioned active power filtering system, the no-load voltage ratio n of coupling transformer and DC power supply output voltage U _dSatisfy n * U _dThe maximum amplitude of the ripple voltage of output port in needing the ripple frequency band of filtering during greater than the coupling transformer secondary short circuit,

Wherein, n＞0, U _d＞0.

Further the operation principle of the inventive method is illustrated below.Might as well establish actual filter branch electric current at f _k(k=1,2 ..., m) ripple component of frequency is

I wherein _WkAnd _kBe respectively the effective value and the initial phase of this frequency ripple component, t is the time.If measuring process is at f _kThe phase shift of frequency is θ _K1So, measure the filter branch current i that obtains _APFIn the f that comprised _kThe signal of frequency is

If each bandpass filtering treatment is enough big to the attenuation rate of the ripple frequency signal outside its passband frequency, then each bandpass filtering treatment is at f _kThe output of frequency is respectively:

Wherein, θ _K2Be that the passband frequency is f _kBand pass filter at f _kThe phase shift of frequency.So, i _i(i=1,2 ..., m) with coefficient a _i(i=1,2 ..., m) and b _i(i=1,2 ..., m) the output result who is weighted summation respectively is:

Wherein, θ _K3Be that the weighted sum processing procedure is to f _kThe phase shift of frequency signal.If first phase-shift filtering is handled f _kThe phase shift of frequency signal is θ _K4, because $\frac{H_{H2}\left(\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="C0313763300232.png,C0313763300241.png"\; state="\{\{state\}\}">j</figure-callout>}2\mathrm{\&pi;f}\right)}{H_{H1}\left(\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="C0313763300232.png,C0313763300241.png"\; state="\{\{state\}\}">j</figure-callout>}2\mathrm{\&pi;f}\right)}=j,$ So second phase-shift filtering handled f _kThe phase shift of frequency signal is θ _K4+ 90 °.So weighted sum is i as a result _Sum1And i _Sum2After handling through first and second phase-shift filterings respectively, obtain:

i _Sum1And i _Sum2Summation process is to f _kIf the phase shift of frequency signal is made as θ _K5, then the reference wave signal used of pulse-width modulation is:

Reference wave signal i _RefAfter pulse-width modulation, pulsed drive, control single-phase electricity die mould inverter circuit produces inverter voltage, and this voltage acts on filter branch by coupling transformer.If this process is at f _kThe phase shift of frequency is θ _K6, then the controlled voltage that produces on the former limit of coupling transformer is:

θ wherein _kRepresentative is from measuring the filter branch electric current to the total phase shift in the whole process of the output-controlled voltage in the former limit of coupling transformer, that is:

θ _k＝θ _k1+θ _k2+θ _k3+θ _k4+θ _k5+θ _k6

According to weighting parameters a _kAnd b _kWith resistance Control Parameter p _kWith reactance Control Parameter q _kDecoupling zero deal with relationship:

$\left(\begin{array}{c}{a}_k\\ b_k\end{array}\right)=\frac{A_{\mathrm{\&Delta;}}}{A_k\&CenterDot;A_H\&CenterDot;n\&CenterDot;U_d}\&CenterDot;\left(\begin{array}{cc}\cos {\mathrm{\&theta;}}_k& \sin {\mathrm{\&theta;}}_k\\ -\sin {\mathrm{\&theta;}}_k& \cos {\mathrm{\&theta;}}_k\end{array}\right)\&CenterDot;\left(\begin{array}{c}{p}_k\\ q_k\end{array}\right)$

So, the controlled voltage u on the former limit of coupling transformer _COVCan be expressed as:

The input of this controlled voltage source is the electric current of filter branch, so controlled voltage source can equivalence be an impedance component also.Generate impedance at f if establish this equivalence _kThe resistance value of frequency is R _Eqk+ jX _Eqk, then as can be known according to following formula:

R _eqk＝p _k

X _eqk＝q _k

Analyze as seen thus, handle by the frequency domain decoupling zero processing of the inventive method and the decoupling zero of resistance and reactance control, the most at last resistance Control Parameter p _kGenerate impedance at f with equivalence _kThe resistance value R of frequency _EqkCorresponding, and reactance Control Parameter q _kGenerate impedance at f with equivalence _kThe reactance value X of frequency _EqkCorresponding.

If the control of constant assignment is adopted in the calculating of frequency domain decoupling zero weight coefficient, then

R _eqk＝p _k＝-R _k0

X _eqk＝q _k＝-X _k0

If R _K0+ jX _K0Filter branch is at f when equaling the coupling transformer secondary short circuit _kThe impedance of frequency, then when active power filtering of the present invention system operate as normal, the total impedance of filter branch equals R _K0+ jX _K0Generate impedance R with equivalence _Eqk+ jX _Eqk=-R _K0-jX _K0Sum is so filter branch is at f _kThe total impedance of frequency is 0.If obviously filter branch is at f _kThe total impedance of frequency is 0 or levels off to 0, then whole or most f _kThe ripple current of the frequency filter branch of must flowing through, thus make ripple voltage in the load and ripple current by filtering or inhibition.

The control method of constant assignment is a kind of control mode of open loop.If there is error in control system, the perhaps filter branch impedance R during the coupling transformer secondary short circuit _K0+ jX _K0Be difficult to accurate acquisition, the feedback closed loop form that then can adopt proportion integration differentiation control or time-delay to regulate control.If regulating control, proportion integration differentiation control or time-delay constituted stable negative feedback control loop, because by the direct input variable of the current filter branch total impedance of measuring and calculating acquisition as control, the integral action of proportion integration differentiation control in addition, perhaps the accumulative action of control to regulated quantity regulated in time-delay, and the stable state of control system just filter branch total impedance is tending towards 0 state.So at f _kFrequency, the closed-loop control process will be regulated resistance Control Parameter p automatically _kWith reactance Control Parameter q _kMake equivalence generate impedance R _Eqk+ jX _EqkApproach-R _K0-jX _K0, and ripple suppresses also will approach optimal effectiveness.Certainly, the initial condition of closed-loop control is with p _kAnd q _kBe set to approaching-R _K0With-X _K0Value help filtering control procedure fast and stable.

In sum, active power filtering method of the present invention and system thereof directly control the filter branch impedance, thereby realize low-frequency ripple Noise Suppression and elimination.This method need not system balance, and control procedure is not subjected to the influence of load variations.The controlled target of system of the present invention is to make filter branch at f _k(k=1,2 ..., m) the frequency impedance is 0, this also is the optimal solution of this active power filtering system filtering ripple.Because the feedback closed loop form that can adopt proportion integration differentiation control or time-delay to regulate control makes the control system automatic optimal,, active power filtering method of the present invention and system thereof suppress effect again so having better ripple.The present invention handles by decoupling zero and has realized control system from measuring the compensation of filter branch electric current to total phase shift of the output-controlled voltage in the former limit of coupling transformer, thereby has solved the influence that the phase shift that no doubt exists in measurement, bandpass filtering, weighted sum, first phase-shift filtering processing, read group total, pulse-width modulation, pulsed drive, voltage inversion and transformer coupled these processing procedures suppresses ripple.And compensation also solved sampling and the digitlization phase shift problem that time-delay caused in calculating, and makes in the control system some processing procedures can select to adopt directly perceived, stable, reliable, exact figure mode to realize.

Emulation experiment shows that the feasibility of active power filtering method of the present invention and system thereof and ripple rejection all reach practical requirement.

Description of drawings

Fig. 1 is the theory diagram of existing a kind of active electric filter device and control method thereof.

Fig. 2 is the transfer function model of existing a kind of active electric filter device and control method thereof.

Fig. 3 is the theory diagram of the active power filtering method of the branch impedance decoupling zero control that DC source arranged of the present invention.

Fig. 4 is the way circuit block diagram of active power filtering system when adopting the control of constant assignment that the branch impedance decoupling zero control of DC source is arranged of the present invention.

Fig. 5 is the decoupling zero treatment circuit block diagram in the active power filtering system of the branch impedance decoupling zero control that DC source arranged of the present invention.

Fig. 6 is the way circuit block diagram of active power filtering system when adopting proportion integration differentiation control or time-delay to regulate control that the branch impedance decoupling zero control of DC source is arranged of the present invention.

Fig. 7 is the rectification circuit that is used for producing 500 kilovolts of DC transmission system of the direct voltage input signal that comprises Ripple Noise in the embodiment of the invention.

Fig. 8 is as the circuit of active power filtering system output loading in the embodiment of the invention.

The circuit of the single tuning passive filtration unit that Fig. 9 is in the embodiment of the invention to be adopted.

Figure 10 is the H that adopts in the embodiment of the invention _F1(z), H _F2(z), H _F3(z) and H _F4(z) amplitude-frequency response of bandpass filtering treatment.

Figure 11 is that first phase-shift filtering that adopts in the embodiment of the invention is handled H _H1(z) amplitude-frequency response and phase-frequency response curve.

Figure 12 is that second phase-shift filtering that adopts in the embodiment of the invention handled H _H2(z) amplitude-frequency response and phase-frequency response curve.

Figure 13 is in the embodiment of the invention The phase-frequency response curve.

Figure 14 is the reference waveform signal that the input pulse-width modulation when the coupling transformer secondary that emulation obtained that the embodiment of the invention is controlled based on the time-delay adjusting is inserted inverter voltage from being shorted to is handled.

Figure 15 is the output end voltage waveform when the coupling transformer secondary that emulation obtained that the embodiment of the invention is controlled based on the time-delay adjusting is inserted inverter voltage from being shorted to.

Figure 16 is the output end current waveform when the coupling transformer secondary that emulation obtained that the embodiment of the invention is controlled based on the time-delay adjusting is inserted inverter voltage from being shorted to.

Figure 17 is the effective value change curve of 600 hertz of ripple compositions the output end current when embodiment of the invention is regulated control based on time-delay the coupling transformer secondary that emulation obtained is inserted inverter voltage from being shorted to.

Embodiment

A specific embodiment of the present invention is to design for being reduced in the low-frequency ripple noise that 500 kilovolts of rectification circuits in the DC transmission system are loaded on the transmission line.With AC rectification be direct current circuit as shown in Figure 7, wherein three-phase alternating current bus frequency is 50 hertz, line voltage is 382.9 kilovolts; The no-load voltage ratio of the transformer T2 of the transformer T1 of star/triangle type and star/star type is 345.0 kilovolts: 213.5 kilovolts, and rated capacity 603.7 megavolt-amperes, the per unit value of short-circuit impedance is 0.18; Adopt a rectifier bridge T who is made of 12 controllable silicons to realize 12 impulse commutation circuit, the conducting pilot angle is got 5 degree, about 500 kilovolts of the direct voltage of output.The voltage that this rectification circuit produces mainly comprises 600,1200,1800,2400 hertz low-frequency ripple, and this voltage will be imported active power filtering of the present invention system.

The output loading of the embodiment of active power filtering method of the present invention and system thereof as shown in Figure 8.Comprise one 200 kilometers transmission line in Fig. 8, power transmission line adopts the double bundle conductor apart from 50 meters on ground, 0.46 meter of spacing, directly over high 8 meters are lightning protection ground wires, the earth is as loop electrode.The power transmission line other end has the smoothing reactor of L1=0.597 henry, R1=250 Europe load resistance, and by three electric capacity (C1=0.84 microfarads, the C2=3.0 microfarad, the C3=0.209 microfarad), two inductance (L2=0.84 milihenries, the L3=0.336 henry) and the passive filter circuit that constitutes of two resistance (R2=10 kilo-ohm, R3=6.3 Europe).

The embodiment of active power filtering method of the present invention and system thereof is when adopting the constant assignment to obtain resistance Control Parameter p _kWith reactance Control Parameter q _kThe time circuit block diagram as shown in Figure 4, the circuit block diagram when adopting proportion integration differentiation control or time-delay to regulate control is as shown in Figure 6.Because the ripple frequency of being concerned about is 600,1200,1800,2400 hertz, thus m=4, f ₁=600Hz, f ₂=1200Hz, f ₃=1800Hz, f ₄=2400Hz.Among Fig. 4 and Fig. 6, the smoothing reactor L that incoming line one end seals in _sGet 0.2 henry; After coupling transformer former limit series connection, be connected in parallel on the passive filtration unit employing single-tuned circuit as shown in Figure 9 of output port again, wherein capacitor C _C=1.0 microfarads, inductance L _C=0.0312 henry, resonance frequency are 900 hertz, and obviously the relative direct current of passive filtration unit is an open-circuit condition, and direct voltage will be loaded into capacitor C _COn; The no-load voltage ratio of coupling transformer is 20 to do volt: 10 kilovolts, i.e. and n=2, the short-circuit impedance of coupling transformer other circuit parameter relatively can be ignored; An output U _d=5.0 kilovolts DC power supply is through inserting the secondary of coupling transformer by 4 IGBT and with the single-phase electricity die mould inverter circuit that 4 diodes of IGBT reverse parallel connection constitute.The no-load voltage ratio n of coupling transformer and DC power supply voltage U _dProduct be nU _d=10.0 kilovolts, the maximum amplitude (about 3 kilovolts) of the ripple voltage of filter circuit output port in being concerned about frequency range during greater than the coupling transformer secondary short circuit.

In embodiments of the present invention, the first step of control method is measured filter branch electric current in parallel by current transformer earlier, and kilo-ampere is got by unit.Because the bandpass filtering among the embodiment, weighted sum, phase-shift filtering are handled and the process of obtaining the pulse-width modulation reference wave of suing for peace all adopts digital form to realize, so measuring process also comprises the processing that the continuous measurement signal that obtains is sampled, sample frequency is got 20 KHz, the filter branch current signal i that obtains dispersing _APF(n).Then, i _APF(n) be input to four bandpass digital filters of corresponding 600,1200,1800,2400 hertz respectively, the transfer function form that their Z-transformation is represented is respectively:

$H_{f1}\left(z\right)=\frac{0.0055900155\&CenterDot;(1-z^{-2})}{1-1.9535925\&CenterDot;z^{-1}+0.98881997.z^{-2}}$

$H_{f2}\left(z\right)=\frac{0.0073086810\&CenterDot;(1-z^{-2})}{1-1.8459621\&CenterDot;z^{-1}+0.98538264\&CenterDot;z^{-2}}$

$H_{f3}\left(z\right)=\frac{0.0053297099\&CenterDot;(1-z^{-2})}{1-1.6796558\&CenterDot;z^{-1}+0.98934058\&CenterDot;z^{-2}}$

$H_{f4}\left(z\right)=\frac{0.0067989292\&CenterDot;(1-z^{-2})}{1-1.4480248\&CenterDot;z^{-1}+0.98640214\&CenterDot;z^{-2}}$

H _F1(z), H _F2(z), H _F3(z) and H _F4(z) amplitude-frequency response of four band pass filters is respectively as (a) and (b) among Figure 10, (c) with (d).Ripple component i to 600,1200,1800,2400 hertz of the difference correspondences that obtain after the Filtering Processing ₁, i ₂, i ₃And i ₄Carry out two groups of weighted sums, obtain first weighted sum current signal i _Sum1=a ₁I ₁+ a ₂I ₂+ a ₃I ₃+ a ₄I ₄With second weighted sum current signal i _Sum2=b ₁I ₁+ b ₂I ₂+ b ₃I ₃+ b ₄I ₄

When adopting the control of constant assignment to obtain resistance Control Parameter p _kWith reactance Control Parameter q _kThe time, ignore the impedance of coupling transformer, according to impedance j (147.64), j102.61, j264.44, the j404.17 of passive filter, produce circuit with resistance Control Parameter p by the adjustable direct current signal among Fig. 4 at 600,1200,1800,2400 hertz _kWith reactance Control Parameter q _kBe set to following constant:

p ₁＝p ₂＝p ₃＝p ₄＝0.0：

q ₁＝147.64；q ₂＝-102.61；q ₃＝-264.44；q ₄＝-404.17。

Adopt proportion integration differentiation control or time-delay to regulate control and obtain resistance Control Parameter p _kWith reactance Control Parameter q _kThe time, at first need to measure the voltage u of filter branch in parallel _APFWith filter branch current i in parallel _APFAgain to u _APFCarry out Fourier transform, obtain filter branch voltage u _APFRipple effective value U at 600,1200,1800,2400 hertz _F1, U _F2, U _F3, U _F4With ripple phase place _Uf1, _Uf2, _Uf3, _Uf4Simultaneously, to i _APFCarry out Fourier transform, obtain the filter branch current i _APFRipple effective value I at 600,1200,1800,2400 hertz _F1, I _F2, I _F3, I _F4With ripple phase place _If1, _If2, _If3, _If4Then according to following formula calculation of filtered branch road respectively 600,1200,1800,2400 hertz branch impedance:

Wherein, k=1,2,3,4.

Control can be with R if embodiment adopts proportion integration differentiation _kAnd X _kThe proportion integration differentiation control circuit that input has following transfer function form:

$H_{\mathrm{PlD}}\left(s\right)=-(0.5+\frac{90}{s})\&CenterDot;\frac{1}{1+0.01s}$

And output obtains resistance Control Parameter p _kWith reactance Control Parameter q _k

If embodiment adopts time-delay to regulate control, then control system accounts for 80% when above of total ripple current when 600,1200,1800,2400 hertz of ripple current compositions of detection filter branch road, and the judgement main circuit is not in transient process.The filter branch impedance of measuring is through after the discretization, can adopt currency or currency and the differential data of the last group of centrifugal pump of being stored passes through the result of weighted sum as the FEEDBACK CONTROL regulated quantity.But present embodiment for the sake of simplicity, adopted form to be-0.8R _kWith-0.8X _kA proportional of (k=1,2,3,4) is as the FEEDBACK CONTROL regulated quantity.When main circuit and filter branch are not in transient process, every 0.01 second this regulated quantity and current resistance Control Parameter p _K0With reactance Control Parameter q _K0Summation, the resistance Control Parameter p that is newly exported _kWith reactance Control Parameter q _k, that is:

$\left\{\begin{array}{c}{p}_k=p_{k0}-0.8\&CenterDot;R_k\\ q_k=q_{k0}-0.8\&CenterDot;X_k\end{array},(k=\mathrm{1,2,3,4})\right.$

If t ₀Be the control starting moment, then initial condition is with resistance Control Parameter p _kWith reactance Control Parameter q _kBe set to the estimated parameters result of institute in the control of constant assignment, that is:

p ₁(t ₀)＝p ₂(t ₀)＝p ₃(t ₀)＝p ₄(t ₀)＝0.0；

q ₁(t ₀)＝147.64；q ₂(t ₀)＝-102.61；q ₃(t ₀)＝-264.44；q ₄(t ₀)＝-404.17。

Adopt control of constant assignment or proportion integration differentiation control or time-delay to regulate control and obtain resistance Control Parameter p _kWith reactance Control Parameter q _kAfter, be achieved as follows the decoupling zero processing of expression formula again according to circuit shown in Figure 5, thereby obtain frequency domain decoupling zero weight coefficient a _kAnd b _k(k=1,2,3,4):

$\left(\begin{array}{c}{a}_k\\ b_k\end{array}\right)=\frac{A_{\mathrm{\&Delta;}}}{A_k\&CenterDot;A_H\&CenterDot;n\&CenterDot;U_d}\&CenterDot;\begin{array}{c}\left(\begin{array}{cc}\cos {\mathrm{\&theta;}}_k& \sin {\mathrm{\&theta;}}_k\\ -\sin {\mathrm{\&theta;}}_k& \cos {\mathrm{\&theta;}}_k\end{array}\right)\&CenterDot;\left(\begin{array}{c}{p}_k\\ q_k\end{array}\right),(k=\mathrm{1,2,3,4})\end{array}$

Wherein, modulated triangular wave amplitude A _Δ=5.0; The bandpass filtering gain A _k=1.0 (k=1,2,3,4); The gain A that phase-shift filtering is handled _H=1.0; Coupling transformer no-load voltage ratio n=2.0; The DC power supply voltage U _d=5.0; Get respectively at 600,1200,1800,2400 hertz to total phase shift of the output-controlled voltage in the former limit of coupling transformer from measuring the filter branch electric current: θ ₁=-0.36828 radian, θ ₂=-1.89014 radians, θ ₃=-2.84317 radians, θ ₄=-3.54499 radians.So, according to above-mentioned parameter, be by the adjustable direct current signal generation circuit generation numerical value among Fig. 5 $\frac{A_{\mathrm{\&Delta;}}}{A_k\&CenterDot;A_H{\&CenterDot;n\&CenterDot;U}_d}\&CenterDot;\cos {\mathrm{\&theta;}}_k$ With $\frac{A_{\mathrm{\&Delta;}}}{A_k\&CenterDot;A_H{\&CenterDot;n\&CenterDot;U}_d}\&CenterDot;\sin {\mathrm{\&theta;}}_k$ Two regular signals, the p of input _kAnd q _kBy four multipliers, an adder and a subtracter, output at last obtains frequency domain decoupling zero weight coefficient a again _kAnd b _k

Weighted sum current signal i _Sum1And i _Sum2Be respectively H through transfer function respectively _H1(z) and H _H2(z) after handling, two phase-shift filterings obtain phase-shift filtering signal i _Hb1And i _Hb2H _H1(z) and H _H2(z) form is respectively:

$H_{H1}\left(z\right)=\frac{0.14677965-1.00149913\&CenterDot;z^{-1}+1.83187525\&CenterDot;z^{-2}-z^{-3}}{1-1.83187525\&CenterDot;z^{-1}+1.00149913\&CenterDot;z^{-2}-0.14677965\&CenterDot;z^{-3}}$

$H_{H2}\left(z\right)=\frac{-0.13884066+0.17122666\&CenterDot;z^{-1}+0.93626531\&CenterDot;z^{-2}-1.96496093\&CenterDot;z^{-3}+z^{-4}}{1-1.96496093\&CenterDot;z^{-1}+0.93626531\&CenterDot;z^{-2}+0.17122666\&CenterDot;z^{-3}-0.13884066\&CenterDot;z^{-4}}$

Phase-shift filtering by Figure 11 and Figure 12 is handled H _H1(z) and H _H2(z) amplitude-frequency response and phase-frequency response curve as seen, the amplitude-frequency gain that two phase-shift filterings are handled input signal is A _H=1.0.By Figure 13's

The phase-frequency response curve as seen, in 500～3000 hertz of frequency domain scopes being concerned about, Phase place be 90.00 ± 0.02 the degree.Again to phase-shift filtering signal i _Hb1And i _Hb2Summation obtains the reference wave signal i that pulse-width modulation is used _Ref, i.e. i _Ref=i _Hb1+ i _Hb2Pulse-width modulation adopts the comparison circuit of reference wave and modulated triangular wave to realize.It is 10 KHz that modulated triangular wave generation circuit produces frequency, and maximum amplitude is A _Δ=5.0 modulated triangular wave.If reference wave signal i _RefAmplitude is greater than the current amplitude of modulated triangular wave, and then pulse-width modulated output signal will be controlled inverter circuit at coupling transformer secondary loading+U through behind the pulse driving circuit _dVoltage (to be labeled as the positive voltage direction among Fig. 4 and Fig. 6); If reference wave signal i _RefAmplitude is less than the current amplitude of modulated triangular wave, and then pulse-width modulated output signal will be controlled inverter circuit at coupling transformer secondary loading-U through behind the pulse driving circuit _dVoltage.

Embodiment based on active power filtering method of the present invention and system thereof has carried out simulation calculation.Here only provide based on time-delay and regulate the simulation result that control is obtained.In 0.4 second of beginning, the coupling transformer secondary short circuit is controlled inoperative.After 0.4 second, single-phase electricity die mould inverter circuit inserts the coupling transformer secondary, and the inventive method is started working.Figure 14 has provided the reference signal i from 0.38 second to 0.50 second input pulse-width modulation _RefWaveform, reference signal i _RefAmplitude does not surpass the maximum amplitude A of modulated triangular wave _ΔFrom the output end current waveform of the output end voltage waveform of Figure 15 and Figure 16 as seen, output end voltage and electric current comprise tangible low-frequency ripple during the coupling transformer secondary short circuit, and after method of the present invention started 0.05 second, low-frequency ripple was obviously eliminated.Further spectrum analysis can be illustrated more clearly in the repressed effect of low-frequency ripple.Figure 17 has provided the change curve of the effective value of 600 hertz of ripple compositions in the output end current, and the filtration result of 1200,1800 and 2400 hertz of ripple compositions is similar with it.It is worthy of note that active power filtering method of the present invention and system thereof are intended to eliminate low-frequency ripple, but can introduce the noise of the modulating frequency that the power electronic device switch causes simultaneously, Figure 15 and 16 also reflects this point.Yet after having suppressed low-frequency ripple, high-frequency noise can be eliminated by some simple filter circuits.

Claims (5)

1, there is the active power filtering method of the branch impedance decoupling zero control of DC source to contain the decoupling method that bandpass filtering treatment realizes frequency domain control, it is characterized in that: this method is a kind of decoupling zero that utilizes bandpass filtering to realize frequency domain control, handle the decoupling zero control that constitutes filter branch resistance and reactance based on two phase-shift filterings of 90 ° of phase shift mutual deviations again, equal zero or level off to zero in the total impedance of the ripple frequency that needs filtering by regulating filter branch, thereby realize the method to this frequency ripple filtering, this method contains successively and has the following steps:

The 1st step: measure the current signal i in the filter branch in parallel _APF

The 2nd step: with the current signal i that measures _APFImport one group of passband frequency and be respectively f ₁, f ₂..., f _mBandpass filtering treatment, obtain one group of ripple current signal i that comprises different frequency ripple composition ₁, i ₂..., i _mWherein, f ₁, f ₂..., f _mBe m the component frequency that needs the low-frequency ripple of filtering, m is a positive integer, and each bandpass filtering treatment is at its passband frequency f _iHas the highest amplitude gain A _i, i=1 wherein, 2 ..., m, and this bandpass filtering treatment realizes the threshold limit value of frequency domain decoupling zero greater than control system to the attenuation rate of other ripple frequency signal;

The 3rd step: the ripple current signal i that filtering is obtained ₁, i ₂..., i _mMultiply by frequency domain decoupling zero weight coefficient a respectively ₁, a ₂..., a _m, summation obtains first weighted sum current signal i then _Sum1, i.e. i _Sum1=a ₁I ₁+ a ₂I ₂+ ... + a _mI _mSimultaneously, the ripple current signal i that filtering is obtained ₁, i ₂..., i _mMultiply by frequency domain decoupling zero weight coefficient b respectively ₁, b ₂..., b _m, summation obtains second weighted sum current signal i then _Sum2, i.e. i _Sum2=b ₁I ₁+ b ₂I ₂+ ... + b _mI _mThe calculation procedure of above-mentioned frequency domain decoupling zero weight coefficient is as follows:

The 3-1 step: the voltage signal u that measures filter branch in parallel two ends _APFWith the current signal i in the filter branch in parallel _APF

The 3-2 step: by the voltage signal u that measures _APFWith current signal i _APFCalculate filter branch respectively at frequency f ₁, f ₂..., f _mImpedance R ₁+ jX ₁, R ₂+ jX ₂..., R _m+ jX _mWherein, R ₁, R ₂..., R _mBe the active component of branch impedance, X ₁, X ₂..., X _mIt is the reaction component of branch impedance;

The 3-3 step: according to the impedance of filter branch, the adjusting control of process parameter obtains one group of resistance Control Parameter p ₁, p ₂..., p _mWith one group of reactance Control Parameter q ₁, q ₂..., q _m

The 3-4 step: with the resistance Control Parameter p that obtains _iWith reactance Control Parameter q _iCarry out decoupling zero according to following formula and handle, and obtain frequency domain decoupling zero weight coefficient a _iAnd b _i:

$\left(\begin{array}{c}{a}_i\\ b_i\end{array}\right)=\frac{A_A}{A_i\&CenterDot;A_H\&CenterDot;\&CenterDot;U_d}\&CenterDot;\left(\begin{array}{cc}\cos {\mathrm{\&theta;}}_i& \sin {\mathrm{\&theta;}}_i\\ -\sin {\mathrm{\&theta;}}_i& \cos {\mathrm{\&theta;}}_i\end{array}\right)\&CenterDot;\left(\begin{array}{c}{p}_i\\ q_i\end{array}\right);$

Wherein, i=1,2 ..., m, A _ΔBe that the modulation signal amplitude that adopts, A are handled in pulse-width modulation _iBe that the passband frequency is f in aforementioned the 2nd step _iThe highest amplitude gain of bandpass filtering treatment, n is the no-load voltage ratio of coupling transformer, U _dBe output voltage for single-phase electricity die mould inverter circuit direct current power source supplying power, θ _iBe from current signal i _APFMeasuring process, through bandpass filtering treatment, weighted sum, first phase-shift filtering processing, summation, pulse-width modulation, pulsed drive, voltage inversion and transformer-coupled a series of processing, to the whole process that obtains the former limit of coupling transformer controlled voltage at f _iTotal phase shift of frequency;

The 4th step: with first weighted sum current signal i of above-mentioned the 3rd step acquisition _Sum1After first phase-shift filtering processing, obtain first phase-shift filtering signal i _Hb1Simultaneously, second weighted sum current signal i that above-mentioned the 3rd step is obtained _Sum2After second phase-shift filtering processing, obtain second phase-shift filtering signal i _Hb2If the transfer function of first and second phase-shift filtering processing is used H respectively _H1(s) and H _H2(s) expression, then in needing the ripple frequency band of filtering, the transfer function characteristics that these two phase-shift filterings are handled satisfies following relation:

$\left\{\begin{array}{c}\left|H_{H1}\left(j2\mathrm{\&pi;f}\right)\right|=\left|H_{H2}\left(j2\mathrm{\&pi;f}\right)\right|=A\\ \frac{H_{H2}\left(j2\mathrm{\&pi;f}\right)}{H_{H2}\left(j2\mathrm{\&pi;f}\right)}=j\end{array}\right.$

Wherein, f is the interior frequency of ripple frequency band that needs filtering, and f＞0; A _HIt is arbitrarily positive constant.The error threshold that the error of above-mentioned transfer function characteristics sets less than the requirement of foundation control precision;

The 5th step: with above-mentioned first and second phase-shift filtering signal i _Hb1And i _Hb2Summation obtains the reference wave signal i that pulse-width modulation is used _Ref, i.e. i _Ref=i _Hb1+ i _Hb2

The 6th step: with reference signal i _RefCarry out pulse-width modulation and handle back acquisition one prescription wave pulse signal; The way of square-wave pulse signal equals power electronic device number controlled in the single-phase electricity die mould inverter circuit, the pulse generation frequency that pulse-width modulation is handled is greater than 2 times of the ripple frequency band upper limit that needs filtering, and the duty ratio of the potential pulse of being exported by the single-phase electricity die mould inverter circuit of this prescription wave pulse signal control satisfies following formula:

A wherein _ΔBe that the modulation signal amplitude that adopts is handled in pulse-width modulation, get the positive count value;

The 7th step: will be above-mentioned the square-wave pulse signal of the 6th step acquisition carry out going to drive after the power amplification power electronic device in the single-phase electricity die mould inverter circuit, have the duty ratio of potential pulse of the single-phase electricity die mould inverter circuit output of DC power supply in order to change, and make the duty ratio of this potential pulse satisfy the requirement in the 6th step;

The 8th step: above-mentioned potential pulse is loaded into the coupling transformer secondary, and act on the filter branch that is constituted with passive filtration unit series connection by the former limit of coupling transformer, in order to obtain the voltage and current output signal of low-frequency ripple after filtered at the circuit output end mouth that is in parallel with this filter branch.

2, the active power filtering method that the branch impedance decoupling zero control of DC source is arranged according to claim 1, it is characterized in that: in 3-3 step of the computational process of the frequency domain decoupling zero weight coefficient in described the 3rd step, the adjusting of parameter control realizes by the control of constant assignment, it during according to the coupling transformer secondary short circuit filter branch at frequency f ₁, f ₂..., f _mResistance value R ₁₀+ jX ₁₀, R ₂₀+ jX ₂₀..., R _M0+ jX _M0, according to following formula to resistance Control Parameter p _iWith reactance Control Parameter q _iAssignment:

$\left\{\begin{array}{c}{p}_i=-R_{i0}\\ q_i=-X_{i0}\end{array}\right.$

Wherein, i=1,2 ..., m.

3, the active power filtering method that the branch impedance decoupling zero control of DC source is arranged according to claim 1, it is characterized in that: in 3-3 step of the computational process of the frequency domain decoupling zero weight coefficient in described the 3rd step, the adjusting control of parameter is that the control of passing ratio integral differential realizes that it is with the filter branch resistance R that calculates _iWith reactance X _iAs input, through ratio, integration, differential, inertia or summation processing procedure, so that system is based on f _iThe filter branch resistance of frequency and reactance constitute negative feedback control, and output obtains resistance Control Parameter p _iWith reactance Control Parameter q _i, i=1 wherein, 2 ..., m.

4, the active power filtering method that the branch impedance decoupling zero control of DC source is arranged according to claim 1, it is characterized in that: in 3-3 step of the computational process of the frequency domain decoupling zero weight coefficient in described the 3rd step, the adjusting control of parameter is regulated control by time-delay and is realized that it is with the filter branch resistance R that calculates _iWith reactance X _iAs input, after discretization,, distinguish corresponding f in order to acquisition through ratio, difference, storage or summation processing procedure _iOne group of negative feedback regulating and controlling amount of the filter branch resistance of frequency and reactance, and the ratio that needs the ripple current of the ripple frequency of filtering to account for total ripple quantity when filter branch is when surpassing set point, the value of this set point in 0.0～1.0 scope, again with this group regulating and controlling amount at a certain time interval respectively with current resistance Control Parameter pi and reactance Control Parameter q _iSummation obtains the new resistance Control Parameter p that exports _iWith reactance Control Parameter q _i, i=1 wherein, 2 ..., m.

5, the active power filtering method of the branch impedance decoupling zero control that DC source the arranged according to claim 1 and active power filtering system that proposes contains active power filter circuit is characterized in that this system comprises as the lower part:

Active power filter circuit, it contains: the smoothing reactor on the end line of going into that is serially connected in the direct voltage input signal that comprises Ripple Noise; Smoothing reactor that is connected in parallel on output port that is constituted by former limit series connection and the filter branch between the loop electrode in order to the passive filtration unit of bearing direct voltage and coupling transformer; Output is linked into the single-phase electricity die mould inverter circuit of coupling transformer secondary; Output is linked into the DC power supply of single-phase electricity die mould inverter circuit input;

Be installed in the current transformer on the filter branch;

Be installed in the voltage transformer between the filter branch two ends;

The one group of band pass filter that is connected in series with the measurement output of current transformer, weighted sum circuit, first and second all-pass filters, adder and pulse-width modulation circuits of constituting by multiplier and adder successively;

Frequency domain decoupling zero weight coefficient counting circuit comprises: measure the one group difference corresponding f of the filter branch voltage and current signal of acquisition as input with the voltage transformer summation current transformer ₁, f ₂..., f _mThe branch road resistance calculations circuit of frequency and branch road reactance counting circuit, and the adjusting control circuit of parameter and decoupling zero treatment circuit;

Pulse driving circuit, its input is connected with the pulse signal that above-mentioned pulse-width modulation is partly exported, and its output then is connected with the trigger control end of single-phase electricity die mould inverter circuit;

In above-mentioned active power filtering system, the no-load voltage ratio n of coupling transformer and DC power supply output voltage U _dSatisfy n * U _dThe maximum amplitude of the ripple voltage of output port in needing the ripple frequency band of filtering during greater than the coupling transformer secondary short circuit, n＞0 wherein, U _d＞0.

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