CN115276207A - Wide-range high-adaptability power supply conversion circuit - Google Patents

Wide-range high-adaptability power supply conversion circuit Download PDF

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Publication number
CN115276207A
CN115276207A CN202210943472.5A CN202210943472A CN115276207A CN 115276207 A CN115276207 A CN 115276207A CN 202210943472 A CN202210943472 A CN 202210943472A CN 115276207 A CN115276207 A CN 115276207A
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voltage
resistor
triode
terminal
power supply
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CN202210943472.5A
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Inventor
漆星宇
张明
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Jiangsu Runic Technology Co ltd
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Jiangsu Runic Technology Co ltd
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Priority to CN202210943472.5A priority Critical patent/CN115276207A/en
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J9/00—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting
    • H02J9/04—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source
    • H02J9/06—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source with automatic change-over, e.g. UPS systems
    • H02J9/068—Electronic means for switching from one power supply to another power supply, e.g. to avoid parallel connection
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J9/00—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting
    • H02J9/04—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source
    • H02J9/06—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source with automatic change-over, e.g. UPS systems
    • H02J9/061—Circuit arrangements for emergency or stand-by power supply, e.g. for emergency lighting in which the distribution system is disconnected from the normal source and connected to a standby source with automatic change-over, e.g. UPS systems for DC powered loads
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0083—Converters characterised by their input or output configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/10—Arrangements incorporating converting means for enabling loads to be operated at will from different kinds of power supplies, e.g. from ac or dc
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only

Abstract

The invention relates to a wide-range high-adaptability power supply conversion circuit. The circuit comprises a current domain comparator, a voltage comparison selection signal and a voltage comparison selection signal, wherein when an input common mode voltage VCM is greater than a power supply voltage VDD, the voltage comparison selection signal is in a first level state; outputting a voltage comparison selection signal in a second level state when the input common-mode voltage VCM is not greater than the power supply voltage VDD; the voltage selection driving circuit receives a voltage comparison selection signal output by the current domain comparator, wherein when the voltage comparison selection signal is in a first level state, the voltage selection driving circuitThe voltage selection driving circuit selects the input common-mode voltage VCM as the conversion output voltage V power (ii) a When the voltage comparison selection signal is in a second level state, the voltage selection driving circuit selects the power supply voltage VDD as the conversion output voltage V power . The invention can effectively realize the processing of the input common-mode voltage and the power supply voltage so as to adapt to the power supply requirements of current detection amplifiers and the like on different input common-mode voltages, and has wide application range.

Description

Wide-range high-adaptability power supply conversion circuit
Technical Field
The invention relates to a conversion circuit, in particular to a wide-range high-adaptability power conversion circuit.
Background
In CMOS integrated circuit design, the operational amplifier using PMOS as input pair tube has the requirement of input common mode working voltage range from (VSS-0.2V) to (VDD-1.5V); for an operational amplifier using NMOS as input pair transistors, there is a common mode operating voltage range requirement of (VSS + 1.5V) to (VDD + 0.2V). In practical application, for an operational amplifier with better performance, an input common mode working voltage range from rail to rail, namely (VSS-0.2V) to (VDD + 0.2V), is generally required; VSS is a negative power supply when the power is supplied by a double power supply, and VDD is a positive power supply when the power is supplied by the double power supply.
The current detection amplifier applied to scenes such as high-voltage short-circuit protection, a DC/DC converter and the like is different from a general operational amplifier which can only process signals with input common-mode voltage within a power supply voltage range and is required to have a wide common-mode input range expanded beyond power supply voltage. In addition, for the current detection amplifier, when the input common-mode voltage is lower than the power supply voltage, the common operational amplifier with the rail-to-rail input characteristic can process the input signal normally; when the input common-mode voltage is higher than the power supply voltage, if the input stage circuit of the operational amplifier is still powered by the power supply voltage, the input signal cannot be processed normally, and at this time, a power supply not lower than the input common-mode voltage is required to power the input stage operational amplifier, but in a determined application scenario, it is impractical to additionally switch a higher power supply.
Therefore, in practical application scenarios such as a current detection amplifier, how to effectively implement processing of the input common mode voltage so as to adapt to the input common mode voltage and the power supply voltage under different voltage conditions to achieve stable operation meeting different application scenarios is a technical problem which is urgently needed to be solved at present.
Disclosure of Invention
The invention aims to overcome the defects in the prior art, and provides a wide-range high-adaptability power conversion circuit which can effectively process input common-mode voltage and power voltage so as to adapt to the power supply requirements of current detection amplifiers and the like on different input common-mode voltages, and is wide in application range, safe and reliable.
According to the technical scheme provided by the invention, the wide-range high-adaptability power supply conversion circuit comprises:
the current domain comparator receives and extracts the input common-mode voltage VCM through the VIN + connecting end and the VIN-connecting end, and compares the extracted input common-mode voltage VCM with the power supply voltage VDD in a current domain, wherein when the input common-mode voltage VCM is greater than the power supply voltage VDD, a voltage comparison selection signal in a first level state is output; when the input common-mode voltage VCM is not more than the power supply voltage VDD, outputting a voltage comparison selection signal in a second level state;
the voltage selection driving circuit is adaptively connected with the current domain comparator and receives a voltage comparison selection signal output by the current domain comparator, wherein when the voltage comparison selection signal is in a first level state, the voltage selection driving circuit selects an input common-mode voltage VCM as a conversion output voltage V power (ii) a When the voltage comparison selection signal is in a second level state, the voltage selection driving circuit selects the power supply voltage VDD as the conversion output voltage V power 。
The current domain comparator comprises a common mode voltage extracting part for extracting and obtaining an input common mode voltage VCM, a common mode voltage converting part for converting the input common mode voltage VCM into a required current, a power supply voltage converting part for converting a power supply voltage VDD into the required current, and a voltage comparison selection output part for outputting a voltage comparison selection signal,
the common mode voltage conversion part and the power supply voltage conversion part are in adaptive connection with the voltage comparison selection output part, and when the current converted by the common mode voltage conversion part is greater than the current converted by the power supply voltage conversion part, a voltage comparison selection signal with a first level state being a high level is output through the voltage comparison selection output part;
when the current converted by the common mode voltage conversion part is less than or equal to the current converted by the power supply voltage conversion part, the voltage comparison selection signal with the second level state being the low level is output through the voltage comparison selection output part.
The common-mode voltage extraction part comprises a resistor R10 and a resistor R20, one end of the resistor R10 is connected with the VIN + connecting end, one end of the resistor R20 is connected with the VIN-connecting end, and the other end of the resistor R10 and the other end of the resistor R20 are connected with each other to form a common-mode voltage extraction node VCM;
the common-mode voltage conversion part comprises an inverting tube N40, one end of the inverting tube N40 is connected with a common-mode voltage extraction node VCM, and the other end of the inverting tube N40 is connected with the voltage comparison selection output part through a common-mode amplification current mirror, wherein the common-mode amplification current mirror comprises a triode Q1 and a triode Q2;
the collector terminal of the triode Q1, the base terminal of the triode Q1 and the base terminal of the triode Q2 are all connected with the other end of the inverse ratio tube N40, the emitter terminal of the triode Q1 and the emitter terminal of the triode Q2 are all grounded, and the collector terminal of the triode Q2 is in adaptive connection with the voltage comparison selection output part.
The power supply voltage conversion part comprises a inverting tube N30 and a power supply amplifying current mirror, the power supply amplifying current mirror comprises a triode Q3 and a triode Q4, wherein one end of the inverting tube N30 is connected with a power supply voltage VDD, the other end of the inverting tube N30 is connected with a base end of the triode Q3, a collector end of the triode Q4 is connected with a base end of the triode Q4, an emitter end of the triode Q3 and an emitter end of the triode Q4 are grounded, and the collector end of the triode Q3 is connected with the voltage comparison selection output part in an adaptive mode.
The resistor R10 and the resistor R20 are resistors with the same resistance, and the resistance formed by the inverse proportion tube N30 is the same as the resistance formed by the inverse proportion tube N40;
the amplification factor of the common mode amplification current mirror is the same as that of the power supply amplification current mirror.
The voltage comparison selection output part comprises a current comparator U1 and a Schmidt trigger SCT1, a collector electrode end of a triode Q2 is connected with the in-phase end of the current comparator U1, a collector electrode end of a triode Q3 is connected with the anti-phase end of the current comparator U1, an output end of the current comparator U1 is connected with an input end of the Schmidt trigger SCT1, and an output end of the Schmidt trigger SCT1 is connected with a voltage selection driving circuit;
the power supply terminal of the current comparator U1 is connected to the positive terminal of the internal low voltage source VCC _ inner, and the negative terminal of the internal voltage source VCC _ inner is grounded.
The voltage selection driving circuit comprises a high-voltage NMOS tube HVN1 for receiving a voltage comparison selection signal, a high-voltage PMOS tube HVP1 for being adaptively connected with a power supply voltage VDD and a high-voltage PMOS tube HVP2 for being adaptively connected with an input common-mode voltage VCM, wherein,
the source terminal of the high-voltage NMOS transistor HVN1 is connected with the source terminal of the high-voltage NMOS transistor HVN2, the drain terminal of the high-voltage NMOS transistor HVN1 is connected with the grid terminal of the PMOS transistor P2 and one end of the resistor R30, and the source terminal of the high-voltage NMOS transistor HVN1 is grounded through a current source I3;
the drain terminal of the high-voltage PMOS tube HVP1 is connected with the power supply voltage VDD, the source terminal of the high-voltage PMOS tube HVP1 is connected with the other end of the resistor R30, the source terminal of the PMOS tube P2, the emitter terminal of the triode T1, the source terminal of the high-voltage PMOS tube HVP2 and the emitter terminal of the triode T4 to form a voltage source for outputting the conversion output voltage V power A converted voltage output terminal of (a);
the grid terminal of the high-voltage PMOS tube HVP1 is connected with the drain terminal of the high-voltage NMOS tube HVN2, the drain terminal of the PMOS tube P2, the input terminal of the inverter INV1, the base terminal of the triode T3 and the collector terminal of the triode T3;
the output of the inverter INV1 is connected with the grid terminal of a high-voltage PMOS tube HVP2, the emitter terminal of the triode T3 is connected with the base terminal of the triode T2 and the collector terminal of the triode T2, and the emitter terminal of the triode T2 is connected with the base terminal of the triode T1 and the collector terminal of the triode T1;
a base electrode end of the triode T4 and a collector electrode end of the triode T4 are connected with an emitter electrode end of the triode T5, the base electrode end of the triode T5 and the collector electrode end of the triode T5 are connected with an emitter electrode end of the triode T6, the base electrode end of the triode T6 and the collector electrode end of the triode T6 are connected with a first power supply end of the inverter INV1 and a drain end of the high-voltage NMOS tube HVN3, and a second power supply end of the inverter INV1 is connected with a conversion voltage output end;
the grid end of the high-voltage NMOS tube HVN3 and the grid end of the high-voltage NMOS tube HVN2 are both connected with a bias voltage NBIAS1, and the source end of the high-voltage NMOS tube HVN3 is grounded through a current source I4.
Further comprises a current detection amplifier comprising an operational amplifier A1, wherein the voltage selection drive circuit outputs a converted output voltage V power As a supply voltage for the operational amplifier A1;
the non-inverting end of the operational amplifier A1 is connected with one end of a resistor R3 and one end of a resistor R4, the inverting end of the operational amplifier A1 is connected with one end of the resistor R1 and one end of a resistor R2, the other end of the resistor R3 is connected with a VIN + connecting end, the other end of the resistor R1 is connected with a VIN-connecting end, the other end of the resistor R2 is connected with the output end of the operational amplifier A1 to form a current detection amplification output end OUT of the current detection amplifier, and the other end of the resistor R4 is connected with a reference connecting end REF.
The resistor R4 is configured as a reference resistor R0, a tuning resistor string and a pre-trimming resistor R6, wherein,
the adjusting resistor string comprises m +1 adjusting resistors which are sequentially connected in series, one end corresponding to the first adjusting resistor in series connection forms a first end of the adjusting resistor string, one end corresponding to the tail resistor in series connection forms a second end of the adjusting resistor string, the resistance values of the m +1 adjusting resistors are increased in equal proportion along the direction of the first adjusting resistor in series connection and the tail resistor in series connection, and each adjusting resistor is connected with a trimming fuse in parallel;
one end of the reference resistor R0 is connected with one end of the pre-trimming resistor R6 and the first end of the adjusting resistor string, and the other end of the reference resistor R0 is in adaptive connection with the resistor R3 and the non-inverting end of the operational amplifier A1; the other end of the pre-trimming resistor R6 trims and trims the test reference connection end REF _ test, and the second end of the adjusting resistor string is connected with the reference connection end REF;
when the common mode rejection ratio of the current detection amplifier is configured, the pre-trimming resistor R6 is connected in, and the pre-trimming offset voltage under the connection of the pre-trimming resistor R6 is measured;
and determining a target adjusting resistance value accessed by the adjusting resistor string according to the pre-adjusting offset voltage and the target offset voltage, and fusing a corresponding adjusting fuse in the adjusting resistor string to obtain an actual adjusting resistance value matched with the target adjusting resistance value.
The common ratio of the resistance values of the adjacent adjusting resistors in the adjusting resistor string is two, the reference resistor R0 is smaller than the resistor R2, and the resistance value of the pre-trimming adjusting resistor R6 is configured to be the resistance value corresponding to the adjusting resistor located in the middle in the adjusting resistor string.
The invention has the advantages that: comparing the input common-mode voltage VCM with the power supply voltage VDD in a current domain through a current domain comparator, and outputting a corresponding voltage comparison selection signal according to the magnitude of the input common-mode voltage VCM and the power supply voltage VDD; when the voltage comparison selection signal is in the first level state, the voltage selection driving circuit selects the input common-mode voltage VCM as the conversion output voltage V power (ii) a When the voltage comparison selection signal is in a second level state, the voltage selection driving circuit selects the power supply voltage VDD as the conversion output voltage V power . The conversion output voltage V output by the voltage selection drive circuit power As a power supply voltage in a practical application scene, the processing of the input common-mode voltage VCM and the power supply voltage VDD can be effectively realized, so that the power supply requirements of a current detection amplifier and the like at different input common-mode voltages VCM can be met, the adaptation range is improved, and the power supply voltage is safe and reliable.
Drawings
Fig. 1 is a circuit block diagram of the present invention.
Fig. 2 is a circuit schematic of the current domain comparator of the present invention.
FIG. 3 is a schematic circuit diagram of the voltage selection driving circuit of the present invention.
FIG. 4 is a schematic diagram of the present invention in conjunction with a current sense amplifier.
Fig. 5 is a schematic diagram of the current sense amplifier configured with the common mode rejection ratio according to the present invention.
Description of reference numerals: the circuit comprises a 1-current domain comparator, a 2-voltage selection driving circuit, a 3-common mode voltage extraction part, a 4-common mode voltage conversion part, a 5-power voltage conversion part and a 6-voltage comparison selection output part.
Detailed Description
The invention is further illustrated by the following specific figures and examples.
In order to effectively realize the processing of the input common-mode voltage VCM and the power voltage VDD so as to be capable of adapting to the power supply requirements of the current detection amplifier and the like at different input common-mode voltages VCM, the wide-range high-adaptability power conversion circuit of the invention specifically comprises:
the current domain comparator 1 receives and extracts the input common-mode voltage VCM through the VIN + connecting end and the VIN-connecting end, and compares the extracted input common-mode voltage VCM with the power supply voltage VDD in a current domain, wherein when the input common-mode voltage VCM is greater than the power supply voltage VDD, a voltage comparison selection signal in a first level state is output; outputting a voltage comparison selection signal in a second level state when the input common-mode voltage VCM is not greater than the power supply voltage VDD;
a voltage selection driving circuit 2 adaptively connected to the current domain comparator 1 and receiving a voltage comparison selection signal output by the current domain comparator 1, wherein when the voltage comparison selection signal is in a first level state, the voltage selection driving circuit 2 selects the input common-mode voltage VCM as the conversion output voltage V power (ii) a When the voltage comparison selection signal is in the second level state, the voltage selection driving circuit 2 selects the power supply voltage VDD as the conversion output voltage V power 。
As can be seen from the above description, for the input common-mode voltage VCM in a specific application scenario, the processing capability of the application scenario is expanded to a wide common-mode input range beyond the power supply voltage VDD, that is, the processing capability of the specific application scenario for the input common-mode voltage is satisfied without replacing the power supply voltage VDD.
In order to meet the requirements of the application scenarios, in the wide-range high-adaptability power conversion circuit of the present invention, the current domain comparator 1 is used to compare the input common-mode voltage VCM with the power voltage VDD in the current domain, that is, the input common-mode voltage VCM and the power voltage VDD are converted into corresponding currents and then compared. Specifically, when the input common-mode voltage VCM is directly compared with the power supply voltage VDD by using a voltage comparator or the like, a problem that the power supply of the voltage comparator is difficult to determine is encountered. This is because the function of the voltage comparator is to compare the input common-mode voltage VCM with the power supply voltage VDD, but the correctness of the comparison result can be ensured only by using the higher voltage of the input common-mode voltage VCM and the power supply voltage VDD as the power supply of the voltage comparator, which may have the logic deadlock problem of high voltage determination.
In the embodiment of the invention, the current domain comparator 1 can be powered by an internal low-voltage power supply, so that the accuracy of voltage judgment can be ensured. In specific implementation, the input common-mode voltage VCM is received through the VIN + connecting end and the VIN-connecting end, and when the received input common-mode voltage VCM is compared with the power supply voltage VDD and the input common-mode voltage VCM is greater than the power supply voltage VDD, a voltage comparison selection signal in a first level state is output; and outputting a voltage comparison selection signal in a second level state when the input common-mode voltage VCM is not greater than the power supply voltage VDD. When the input common-mode voltage VCM is less than or equal to the power supply voltage VDD, the current-domain comparator 1 outputs a voltage comparison selection signal in a second level state, where the first level state and the second level state are two opposite level states, and if the first level state is a high level, the second level state is a low level; similarly, when the first level state is a low level, the second level state is a high level state. The specific conditions of the first level state and the second level state can be selected according to requirements so as to meet the actual application requirements.
The voltage selection driving circuit 2 is adaptively connected with the current domain comparator 1, that is, the voltage selection driving circuit 2 receives the voltage comparison selection signal output by the current domain comparator 1. In specific operation, when the voltage comparison selection signal is in the first level state, the voltage selection driving circuit 2 selects the input common-mode voltage VCM as the conversion output voltage V power (ii) a When the voltage comparison selection signal is in the second level state, the voltage selection driving circuit 2 selects the power supply voltage VDD as the conversion output voltage V power . Converting the output voltage V outputted from the voltage selection drive circuit 2 power As a power supply voltage in an actual application scenario, it can be known from the above description that processing on the input common-mode voltage VCM and the power supply voltage VDD can be effectively implemented, so as to adapt to power supply requirements of the current detection amplifier and the like at different input common-mode voltages VCM, and improve an adaptation range.
Further, the current domain comparator 1 includes a common mode voltage extracting section 3 for extracting the input common mode voltage VCM, a common mode voltage converting section 4 for converting the input common mode voltage VCM into a desired current, a power supply voltage converting section 5 for converting the power supply voltage VDD into the desired current, and a voltage comparison selection output section 6 for outputting a voltage comparison selection signal,
the common mode voltage conversion part 4 and the power supply voltage conversion part 5 are in adaptive connection with the voltage comparison selection output part 6, and when the current converted by the common mode voltage conversion part 4 is larger than the current converted by the power supply voltage conversion part 5, a voltage comparison selection signal with a first level state being a high level is output through the voltage comparison selection output part 6;
when the current converted by the common mode voltage conversion unit 4 is equal to or less than the current converted by the power supply voltage conversion unit 5, the voltage comparison selection signal whose second level state is a low level is output by the voltage comparison selection output unit 6.
In the embodiment of the invention, the common mode voltage extraction part 3 is in adaptive connection with the VIN + connection end and the VIN-connection end, and extracts the input common mode voltage VCM. The conversion of the input common-mode voltage VCM into a corresponding current can be realized by the common-mode voltage converting part 4, and the conversion of the supply voltage VDD into a corresponding current can be realized by the supply voltage converting part 5, so that a direct comparison of the currents can be performed.
The voltage comparison selection signal is output by the voltage comparison selection output unit 6, and in specific implementation, when the current converted by the common mode voltage conversion unit 4 is greater than the current converted by the power supply voltage conversion unit 5, the voltage comparison selection signal with the first level state being high level is output by the voltage comparison selection output unit 6; when the current converted by the common mode voltage conversion unit 4 is equal to or less than the current converted by the power supply voltage conversion unit 5, the voltage comparison selection signal whose second level state is a low level is output by the voltage comparison selection output unit 6.
Fig. 2 shows a specific implementation of the current-domain comparator 1, specifically, the common-mode voltage extracting unit 3 includes a resistor R10 and a resistor R20, one end of the resistor R10 is connected to the VIN + connection terminal, one end of the resistor R20 is connected to the VIN-connection terminal, and the other end of the resistor R10 and the other end of the resistor R20 are connected to form a common-mode voltage extracting node VCM;
the common-mode voltage conversion part 4 comprises an inverting tube N40, one end of the inverting tube N40 is connected with a common-mode voltage extraction node VCM, and the other end of the inverting tube N40 is connected with the voltage comparison selection output part 6 through a common-mode amplification current mirror, wherein the common-mode amplification current mirror comprises a triode Q1 and a triode Q2;
the collector terminal of the triode Q1, the base terminal of the triode Q1 and the base terminal of the triode Q2 are all connected with the other end of the inverse ratio tube N40, the emitter terminal of the triode Q1 and the emitter terminal of the triode Q2 are all grounded, and the collector terminal of the triode Q2 is in adaptive connection with the voltage comparison selection output part 6.
The power supply voltage conversion part 5 comprises a inverting tube N30 and a power supply amplifying current mirror, the power supply amplifying current mirror comprises a triode Q3 and a triode Q4, wherein one end of the inverting tube N30 is connected with a power supply voltage VDD, the other end of the inverting tube N30 is connected with the base end of the triode Q3, the collector end of the triode Q4 is connected with the base end of the triode Q4, the emitter end of the triode Q3 and the emitter end of the triode Q4 are grounded, and the collector end of the triode Q3 is connected with the voltage comparison selection output part 6 in an adaptive mode.
In specific implementation, the resistor R10 and the resistor R20 are resistors with the same resistance, that is, R10= R20. When the wide-range high-adaptability power supply conversion circuit is applied to a scene of a current detection amplifier, a small sensing resistor needs to be externally hung for differential voltage acquisition when the current detection amplifier is used, the sensing resistor is connected in parallel with an equivalent resistor formed by connecting a resistor R10 and a resistor R20 in series, and the resistor R10 and the resistor R20 are large enough to reduce the influence on the effective value of the sensing resistor and the current detection precision.
In addition, if the input common-mode voltage VCM is higher than the power voltage VDD, the input common-mode voltage VCM supplies power to the input stage of the current detection amplifier, and then currents must flow through the resistors R10 and R2 to generate additional voltage drop. In order to ensure that no excessive voltage drop is generated on the terminal R10 and the resistor R20, the input stage supply voltage of the current detection amplifier is ensured to meet the requirement of handling a high common mode input voltage VCM, and the resistance values of the resistor R10 and the resistor R20 cannot be too large. In specific implementation, the corresponding resistance values of the resistor R10 and the resistor R20 are determined according to the current consumed by the input stage of the current detection amplifier in the specific implementation, so as to meet the requirements of practical application.
In specific implementation, the inverting transistor N40 includes a plurality of NMOS transistors, and the gate terminals of all the NMOS transistors in the inverting transistor N40 are connected to each other and are connected to the common-mode voltage extraction node VCM. For the adjacent NMOS tubes, the source terminal of one NMOS tube is connected with the drain terminal of the other NMOS tube, and the inverse ratio tube N40 can form the function of large resistance. The triodes Q1 to Q4 can adopt NPN triodes.
In specific implementation, the form of the inversion tube N30 is exactly the same as that of the inversion tube N40, and the inversion tube N30 can be referred to the above description and fig. 2, and will not be described again here. In the embodiment of the present invention, the resistance value formed by the inverter tube N30 is the same as the resistance value formed by the inverter tube N40.
The amplification factor of the common mode amplification current mirror is the same as that of the power supply amplification current mirror. When the method is implemented in concrete time, the corresponding current amplification factors of the common-mode amplification current mirror and the power supply amplification current mirror are both 1: and 8, amplifying the current converted from the input common-mode voltage VCM and the current converted from the power supply voltage VDD by 8 times, and then comparing the amplified currents.
During specific work, for an input common-mode voltage VCM, the input common-mode voltage VCM is converted into current through the inverting tube N40, and the current flows into the voltage comparison selection output part 6 after being amplified through a common-mode amplification current mirror formed by the triode Q1 and the triode Q2.
For the power supply voltage VDD, the power supply voltage VDD is converted into a current through the inverting transistor N30, and a power supply amplifying current mirror formed by the triode Q3 and the triode Q4 is amplified to the triode Q3, that is, finally loaded to the voltage comparison selection output part 6, so that the two converted currents are compared by the voltage comparison selection output part 6 to obtain a voltage comparison selection signal.
In specific operation, if the input common-mode voltage VCM is higher than the power supply voltage VDD, the current converted by the common-mode voltage converting part 4 is larger than the current converted by the power supply voltage converting part 5, and a high-level voltage comparison selection signal is obtained through the voltage comparison selection output part 6; if the input common mode voltage VCM is not greater than the power supply voltage VDD, the current converted by the common mode voltage conversion unit 4 is equal to or less than the current converted by the power supply voltage conversion unit 5, and a low-level voltage comparison selection signal is obtained by the voltage comparison selection output unit 6.
In fig. 2, the voltage comparison selection output unit 6 includes a current comparator U1 and a schmitt trigger SCT1, the collector terminal of the transistor Q2 is connected to the in-phase terminal of the current comparator U1, the collector terminal of the transistor Q3 is connected to the inverting terminal of the current comparator U1, the output terminal of the current comparator U1 is connected to the input terminal of the schmitt trigger SCT1, and the output terminal of the schmitt trigger SCT1 is connected to the voltage selection driving circuit 2;
the power supply terminal of the current comparator U1 is connected to the positive terminal of the internal low voltage source VCC _ inner, and the negative terminal of the internal voltage source VCC _ inner is grounded.
In fig. 2, the current converted by the common mode voltage conversion unit 4 is I1, the current converted by the power supply voltage conversion unit 5 is I2, and the current comparator U1 compares the current I1 with the current I2 and outputs a voltage comparison selection signal through the schmitt trigger SCT 1. The current comparator U1 may adopt an existing common form, and may be specifically selected as needed so as to satisfy the comparison of the current I1 and the current I2. Of course, in specific implementation, the voltage comparison selection output unit 6 may also take other forms, and may be specifically selected according to needs so as to meet actual requirements, and the description is not repeated here.
The internal low voltage source VCC _ inner may output the internal low voltage VCC _ inner, i.e. the power supply to the current domain comparator 1 is realized by the internal low voltage source VCC _ inner. The details of the internal low voltage VCC _ inner can be selected from the above description according to practical application scenarios. In specific implementation, a voltage comparison selection signal output by the current domain comparator 1 is used as a grid control signal of a high-voltage NMOS switch tube HVN4 in the voltage selection driving circuit 2 after passing through the Schmitt trigger SCT1 in the internal low-voltage Vcc _ inner; i.e., the voltage range of the voltage comparison selection signal is 0-Vcc _ inner. In specific implementation, the supply voltage of the schmitt trigger SCT1 is also the internal low voltage Vcc _ inner.
One implementation of the voltage selection driving circuit 2 is shown in fig. 3, the voltage selection driving circuit 2 includes a high voltage NMOS HVN1 for receiving the voltage comparison selection signal, a high voltage PMOS HVP1 for adapting the connection to the supply voltage VDD, and a high voltage PMOS HVP2 for adapting the connection to the input common-mode voltage VCM, wherein,
the source terminal of the high-voltage NMOS tube HVN1 is connected with the source terminal of the high-voltage NMOS tube HVN2, the drain terminal of the high-voltage NMOS tube HVN1 is connected with the grid terminal of the PMOS tube P2 and one end of the resistor R30, and the source terminal of the high-voltage NMOS tube HVN1 is grounded through a current source I3;
the drain terminal of the high-voltage PMOS tube HVP1 is connected with the power supply voltage VDD, the source terminal of the high-voltage PMOS tube HVP1 is connected with the other end of the resistor R30, the source terminal of the PMOS tube P2, the emitter terminal of the triode T1, the source terminal of the high-voltage PMOS tube HVP2 and the emitter terminal of the triode T4 to form a voltage source for outputting the conversion output voltage V power A converted voltage output terminal of (a);
the grid terminal of the high-voltage PMOS tube HVP1 is connected with the drain terminal of the high-voltage NMOS tube HVN2, the drain terminal of the PMOS tube P2, the input terminal of the inverter INV1, the base terminal of the triode T3 and the collector terminal of the triode T3;
the output of the inverter INV1 is connected with the grid terminal of a high-voltage PMOS tube HVP2, the emitter terminal of the triode T3 is connected with the base terminal of the triode T2 and the collector terminal of the triode T2, and the emitter terminal of the triode T2 is connected with the base terminal of the triode T1 and the collector terminal of the triode T1;
a base electrode end of the triode T4 and a collector electrode end of the triode T4 are connected with an emitter electrode end of the triode T5, the base electrode end of the triode T5 and the collector electrode end of the triode T5 are connected with an emitter electrode end of the triode T6, the base electrode end of the triode T6 and the collector electrode end of the triode T6 are connected with a first power supply end of the inverter INV1 and a drain end of the high-voltage NMOS tube HVN3, and a second power supply end of the inverter INV1 is connected with a conversion voltage output end;
the grid end of the high-voltage NMOS tube HVN3 and the grid end of the high-voltage NMOS tube HVN2 are both connected with a bias voltage NBIAS1, and the source end of the high-voltage NMOS tube HVN3 is grounded through a current source I4.
Specifically, the gate terminal of the high voltage NMOMS tube HVN4 is connected to the output terminal of the schmitt trigger SCT 1. The current source I3 and the current source I4 are used for providing static current, and the specific conditions of the current source I3 and the current source I4 for providing the static circuit can be selected and determined according to the actual application requirements. In specific implementation, an input end of the inverter INV1 is connected to a drain end of the PMOS transistor P2, a gate end of the high-voltage PMOS transistor HVP1, a drain end of the high-voltage NMOS transistor HVN2, a base end of the transistor T3 and a collector end of the transistor T3 to form a node a, and an output end of the inverter INV1 is connected to a gate end of the high-voltage PMOS transistor HVP2 to form a node B.
In specific implementation, the triodes T1 to T6 all adopt PNP triodes; of course, other forms may also be adopted, which may be specifically selected according to needs and are not described herein again. The specific magnitude of the bias voltage NBIAS1 can be selected as required to meet the actual application requirements.
When the voltage comparison selection signal loaded to the grid terminal of the high-voltage NMOS tube HVN1 is at a high level, the high-voltage NMOS tube HVN1 is turned on. After the high voltage NMOS transistor HVN1 is turned on, current flows through the resistor R30 and provides bias voltage for the PMOS transistor P2 to turn on the PMOS transistor P2. At this time, node A is pulled up to a high voltage level, and node B is pulled down to a low voltage level, which is three times lower than the threshold voltage of the PMOS transistor. Based on the voltages of the node A and the node B, the HVP1 of the high-voltage PMOS tube is turned off, the HVP2 of the high-voltage PMOS tube is turned on, and the conversion output voltage V output by the conversion voltage output end power Is the input common mode voltage VCM.
Corresponding to the above state, when the voltage comparison selection signal applied to the gate terminal of the high voltage NMOS transistor HVN1 is at a low level, the high voltage NMOS transistor HVN1 is turned off, and a current flows through the transistor T1, the transistor T2, and the transistor T3 to clamp the potential of the node a to the conversion output voltage V power Low triple V be Low potential of (c). Wherein, V be Is the triode BE junction (base-emitter junction) voltage. To ensureThe reliability of the circuit is ensured, and the voltage reduction clamping voltage formed by the serial connection of the triode T1, the triode T2 and the triode T3 is slightly lower than the voltage reduction clamping voltage formed by the transmission of the triode T4, the triode T5 and the triode T6. The slightly lower value is generally lower than a smaller difference value, and the specific condition of the slightly lower value can be selected according to actual application scenes and the like so as to meet the actual application requirements. At this time, the node A is at a low potential and the node B is at a high potential, so that the high voltage PMOS transistor HVP1 is turned on, the high voltage PMOS transistor HVP2 is turned off, and the converted output voltage V output by the converted voltage output terminal is power Is the supply voltage VDD.
In specific implementation, when the voltage of the gate terminal of the high-voltage PMOS transistor is high, the high level of the gate terminal is equal to the source potential of the high-voltage PMOS transistor, for any one of the high-voltage PMOS transistor HVP1 and the high-voltage PMOS transistor HVP 2; when the grid voltage of the PMOS tube is low level, the low level potential is equal to the source potential of the high-voltage PMOS tube HVP1 or HVP2 minus three times of voltage V be . Specifically, the source potentials of the high-voltage PMOS tube HVP1 and the high-voltage POMS tube HVP4 are the conversion output voltage V power Therefore, the gate-source voltages corresponding to the high-voltage PMOS tube HVP1 and the high-voltage PMOS tube HVP2 are always within the 5V voltage range, so that the high-voltage PMOS tube ensures that the source and the drain can resist high voltage, and the high-voltage PMOS tube is not damaged.
In summary, for a wide-range high-adaptability power conversion circuit, there are:
Figure BDA0003786714620000101
specifically, when the input common mode voltage VCM is higher than the power supply voltage VDD, the voltage comparison selection signal outputted from the current domain comparator 1 is at a high level, the high voltage NMOS HVN1 is turned on, and the conversion output voltage V outputted through the conversion voltage output terminal is converted power Is the input common mode voltage VCM; when the input common mode voltage VCM is less than or equal to the power supply voltage VDD, the voltage comparison selection signal output by the current domain comparator is at a low level, the high-voltage NMOS tube HVN1 is switched off, and the conversion output voltage V output by the conversion voltage output end is converted power The function of self-adaptive power supply voltage conversion of the power supply voltage is realized for the power supply voltage VDD.
As shown in fig. 4 and 5, a current detection amplifier is further included, and the current detection amplifier includes an operational amplifier A1, wherein the voltage selection driving circuit outputs a converted output voltage V power As a supply voltage for the operational amplifier A1;
the non-inverting end of the operational amplifier A1 is connected with one end of a resistor R3 and one end of a resistor R4, the inverting end of the operational amplifier A1 is connected with one end of the resistor R1 and one end of a resistor R2, the other end of the resistor R3 is connected with a VIN + connecting end, the other end of the resistor R1 is connected with a VIN-connecting end, the other end of the resistor R2 is connected with the output end of the operational amplifier A1 to form a current detection amplification output end OUT of the current detection amplifier, and the other end of the resistor R4 is connected with a reference connecting end REF.
Specifically, the adaptive power conversion circuit in fig. 4 and 5 is in the form of the circuit in fig. 1, and when the wide-range high-adaptive power conversion circuit in fig. 1 is used to provide the power supply voltage for the operational amplifier A1 in the current detection amplifier, as can be seen from the above description, the current detection amplifier can be extended to a wide common-mode input range beyond the power supply voltage VDD. The operational amplifier A1 in fig. 4 and 5 may be a multi-stage operational amplifier, and the wide-range high-adaptability power conversion circuit supplies power to the input stage of the operational amplifier A1. That is, when the input common-mode voltage VCM is lower than the power voltage VDD, the input stage in the operational amplifier A1 is powered by the power voltage VDD; when the input common-mode voltage VCM is higher than the supply voltage VDD, the input stage within the operational amplifier A1 is powered by the input common-mode voltage VCM.
As known to those skilled in the art, a current detection amplifier is an amplifier that detects current by measuring a voltage drop across a resistor in a current path, and outputs a voltage or current proportional to the measured current, and is widely used in the fields of automobiles, power management, battery chargers, and the like.
In the current sense amplifier shown in fig. 4, ideally, the gain configuration by the resistors R1 to R4 should satisfy the condition of the formula (1), and an ideal output voltage represented by the formula (2) can be obtained.
Figure BDA0003786714620000102
V OUT ＝V REF +V sense ×Gain (2)
Wherein Gain is the Gain of the current sense amplifier, V REF Is a reference voltage, V OUT Is the output voltage of the current sense amplifier. Reference voltage V REF Typically to the reference connection REF.
The common mode rejection ratio of the operational amplifier is an important index, which reflects the capability of a differential amplifier circuit to amplify differential signals and the capability of inhibiting the amplification of common mode signals, and the calculation formula of the common mode rejection ratio is
Figure BDA0003786714620000111
Wherein CMRR1 is the common mode rejection ratio of the operational amplifier, A d Is the voltage amplification of the differential mode signal, A c Is the voltage amplification of the common mode signal.
If the operational amplifier A1 has a sufficiently high common-mode rejection ratio, then the common-mode rejection ratio of the entire current sense amplifier is determined by the matching accuracy between the resistors. According to the definition of the common mode rejection ratio, the common mode rejection ratio CMRR2 of the current detection amplifier is obtained as follows:
Figure BDA0003786714620000112
assuming a target gain of 1, ideally all resistors should be equal, and if there is a 0.1% mismatch in one of the resistors, assuming R1= R2= R3= R, and R4=1.001R, then there is:
Figure BDA0003786714620000113
therefore, the common mode rejection ratio of the current sense amplifier is always affected by the mismatch of the resistors, and even if there is a mismatch of 0.1% in one of the resistors, the common mode rejection ratio of the current sense amplifier is only 66dB. As is apparent from the above equation, if a high common mode rejection ratio is desired, it is required that the ratio of the resistor R2 to the resistor R1 is infinitely close to the ratio of the resistor R4 to the resistor R3, and if the ratios are equal, an infinite common mode rejection ratio is obtained.
If the common mode rejection ratio of the current detection amplifier is expected to reach more than 80dB, the resistance matching precision needs to reach 0.01 percent, and the resistance matching precision is difficult to reach about 0.3 percent due to the limitation of the domestic semiconductor process. Therefore, in the prior art, the matching precision of the device guaranteed by only the process cannot be realized when the current detection amplifier with high common mode rejection ratio is required to be manufactured.
In order to realize the common mode rejection ratio configuration of the current detection amplifier, so as to satisfy the condition that the current detection amplifier has a higher common mode rejection ratio, in the embodiment of the invention, the resistor R4 is configured as a reference resistor R0, a regulating resistor string and a pre-trimming resistor R6, wherein,
the adjusting resistor string comprises m +1 adjusting resistors which are sequentially connected in series, one end corresponding to the first adjusting resistor in series connection forms a first end of the adjusting resistor string, one end corresponding to the tail resistor in series connection forms a second end of the adjusting resistor string, the resistance values of the m +1 adjusting resistors are increased in equal proportion along the direction of the first adjusting resistor in series connection and the tail resistor in series connection, and each adjusting resistor is connected with a trimming fuse in parallel;
one end of the reference resistor R0 is connected with one end of the pre-trimming resistor R6 and the first end of the adjusting resistor string, and the other end of the reference resistor R0 is in adaptive connection with the resistor R3 and the non-inverting end of the operational amplifier A1; the other end of the pre-trimming resistor R6 trims and trims the test reference connection terminal REF _ test, and the second end of the adjusting resistor string is connected with the reference connection terminal REF;
when the common mode rejection ratio of the current detection amplifier is configured, the pre-trimming resistor R6 is connected in, and the pre-trimming offset voltage under the condition that the pre-trimming resistor R6 is connected in is measured;
and determining a target adjusting resistance value accessed by the adjusting resistor string according to the pre-adjusting offset voltage and the target offset voltage, and fusing corresponding adjusting fuses in the adjusting resistor string to obtain an actual adjusting resistance value matched with the target adjusting resistance value.
In order to make the current sense amplifier have a high common mode rejection ratio, as can be seen from the above description, the ratio of the resistor R2 to the resistor R1 should be infinitely close to the ratio of the resistor R4 to the resistor R3, in the embodiment of the present invention, the ratio of the resistor R4 to the resistor R3 is changed by adjusting the size of the resistor R4, and the resistance value approaches the ratio of the resistor R2 to the resistor R1. During circuit design, the resistor R0 is designed to be smaller than the resistor R2, and after chip production is completed, part of adjusting resistors in the adjusting resistor string (Res Block) is connected into the circuit, so that the purpose of changing the ratio of the resistor R4 to the resistor R3 and approaching the ratio of the resistor R2 to the resistor R1 is achieved.
In fig. 5, m +1 adjusting resistors in the adjusting resistor string are respectively: the resistance relation of the adjusting resistors R30, R51, R52, 8230, R5 (m-1), R5m and m +1 can be as follows: 1:2:4:2 m-1 ：2 m Namely, on the basis of the resistance value of the resistor R30, the corresponding resistance values of adjacent adjusting resistors are in an equal ratio relationship, and the common ratio of the resistance values of the adjacent adjusting resistors in the adjusting resistor string is 2, wherein the resistance value of the resistor R30 determines the configurable accuracy, and the smaller the resistance value of the resistor R30 is, the higher the configurable common mode rejection ratio is. In specific implementation, the adjusting resistors in the adjusting resistor string are all connected in parallel with a trimming fuse, that is, in an initial state, the default resistor R30 to the resistor R5m are all shorted by the corresponding trimming fuse. In fig. 5, trimming fuses are Fuese0, fuse1, \8230 \ 8230;, and Fusem, which correspond to adjustable resistors R30, R51, \8230;, and R5m, respectively, one to one.
The resistor R6 is used as a pre-trimming resistor, and the resistance of the pre-trimming resistor R6 is preferably the middle value between the adjusting resistor R30 and the adjusting resistor R5m, that is, the resistance of the pre-trimming resistor R6 is configured to the resistance corresponding to the middle adjusting resistor in the adjusting resistor string.
When the pre-trimming resistor R6 is connected into the circuit, the pre-trimming offset voltage output by the current detection amplifier can be measured by adopting the technical means commonly used in the technical field. For the determined current detection amplifier, the target offset voltage can be determined according to an application scene and the like, so that the actual resistance value of the configuration resistor R4 can be determined according to the relation between the target offset voltage and the pre-correction adjustment offset voltage, and the adjustable resistor which needs to be connected into the adjusting resistor string can be determined. After the adjustable resistor needing to be accessed is determined, the corresponding trimming fuse is fused by a common technical means in the technical field, and finally the resistance value configuration of the resistor R4 is realized, namely the common mode rejection ratio configuration of the current detection amplifier can be realized.
In specific implementation, the common-mode rejection ratio of the current detection amplifier can be equivalently determined as follows:
Figure BDA0003786714620000121
wherein Gain is the Gain of the current sense amplifier, vos out1 For receiving the output offset voltage Vos when the common-mode voltage is VCM1 out2 The output offset voltage when the input common-mode voltage is VCM2, VCM1 is the lowest input common-mode voltage of the current detection amplifier, and VCM2 is the highest input common-mode voltage of the current detection amplifier. When Vos out2 -Vos out1 Near 0, CMMR2 has a large value, and therefore Vos can be reduced out2 -Vos out1 The value of (b) is used as an index for observing the common mode rejection ratio of the current detection amplifier.
Therefore, as can be seen from the above description, the adjustable resistor to be connected in the adjusting resistor string is determined according to the relationship between the target offset voltage and the pre-trimming offset voltage. In specific implementation, the person skilled in the art can obtain:
Figure BDA0003786714620000131
when the resistance value of the configuration resistor R4 is changed to be delta R, then
Figure BDA0003786714620000132
When the resistance value of the configuration resistor R4 is changed to 2 Δ R, then
Figure BDA0003786714620000133
When Δ R < R3+ R4,
Figure BDA0003786714620000134
in order to make the current sense amplifier have a high common mode rejection ratio CMMR2, it is required that
Figure BDA0003786714620000135
When the common mode rejection ratio CMMR2 of the current detection amplifier is 100dB, the deviation of any resistance value of the resistors R1-R4 is required to be only
Figure BDA0003786714620000136
That is, the minimum precision of the resistance change Δ R of the configuration resistor R4 is
Figure BDA0003786714620000137
ΔR＜＜R3+R4。
Therefore, as can be seen from the above description, it is considered that the amount of change Δ R of the resistance R4 corresponds to Vos out2 The amount of change is proportional. At this time, the variation Δ R and Vos according to the resistance R4 out2 And the variable quantity is measured to obtain the pre-correction adjustment offset voltage, and then the adjustable resistor needing to be connected in the adjusting resistor string can be determined according to the target offset voltage.

Claims (10)

1. A wide-range high-adaptability power conversion circuit is characterized by comprising:
the current domain comparator (1) receives and extracts an input common-mode voltage VCM through a VIN + connecting end and a VIN-connecting end, and compares the extracted input common-mode voltage VCM with a power supply voltage VDD in a current domain, wherein when the input common-mode voltage VCM is greater than the power supply voltage VDD, a voltage comparison selection signal in a first level state is output; outputting a voltage comparison selection signal in a second level state when the input common-mode voltage VCM is not greater than the power supply voltage VDD;
the voltage selection driving circuit (2) is adaptively connected with the current domain comparator (1) and receives a voltage comparison selection signal output by the current domain comparator (1), wherein when the voltage comparison selection signal is in a first level state, the voltage selection driving circuit (2) selects the input common-mode voltage VCM as the conversion output voltage V power (ii) a When the voltage comparison selection signal is in a second level state, the voltage selection driving circuit (2) selects the power supply voltage VDD as the conversion output voltage V power 。
2. The wide-range high-adaptability power conversion circuit according to claim 1, wherein: the current domain comparator (1) comprises a common mode voltage extracting part (3) for extracting and obtaining an input common mode voltage VCM, a common mode voltage converting part (4) for converting the input common mode voltage VCM into a required current, a power supply voltage converting part (5) for converting a power supply voltage VDD into the required current, and a voltage comparison selection output part (6) for outputting a voltage comparison selection signal,
the common mode voltage conversion part (4) and the power supply voltage conversion part (5) are in adaptive connection with the voltage comparison selection output part (6), and when the current converted by the common mode voltage conversion part (4) is larger than the current converted by the power supply voltage conversion part (5), the voltage comparison selection signal with the first level state being high level is output through the voltage comparison selection output part (6);
when the current converted by the common mode voltage conversion part (4) is less than or equal to the current converted by the power supply voltage conversion part (5), a voltage comparison selection signal with the second level state being a low level is output by a voltage comparison selection output part (6).
3. The wide-range high-adaptability power conversion circuit according to claim 2, wherein: the common-mode voltage extraction part (3) comprises a resistor R10 and a resistor R20, one end of the resistor R10 is connected with the VIN + connecting end, one end of the resistor R20 is connected with the VIN-connecting end, and the other end of the resistor R10 and the other end of the resistor R20 are mutually connected to form a common-mode voltage extraction node VCM;
the common-mode voltage conversion part (4) comprises an inverting tube N40, one end of the inverting tube N40 is connected with a common-mode voltage extraction node VCM, and the other end of the inverting tube N40 is connected with the voltage comparison selection output part (6) through a common-mode amplification current mirror, wherein the common-mode amplification current mirror comprises a triode Q1 and a triode Q2;
the collector end of the triode Q1, the base end of the triode Q1 and the base end of the triode Q2 are connected with the other end of the inverse ratio tube N40, the emitter end of the triode Q1 and the emitter end of the triode Q2 are grounded, and the collector end of the triode Q2 is in adaptive connection with the voltage comparison selection output part (6).
4. The wide-range high-adaptability power conversion circuit according to claim 3, wherein: supply voltage conversion part (5) are including comparing pipe N30 and power amplification current mirror, power amplification current mirror includes triode Q3 and triode Q4, wherein, the one end of comparing pipe N30 is connected with supply voltage VDD, the other end of comparing pipe N30 and triode Q3's base extreme, triode Q4's collection electrode end and triode Q4's base extreme are connected, triode Q3's emitter extreme and triode Q4's emitter extreme all ground connection, triode Q3's collection electrode end and voltage comparison select output part (6) adaptation connection.
5. The wide-range high-adaptability power conversion circuit of claim 4, wherein: the resistor R10 and the resistor R20 are resistors with the same resistance, and the resistance formed by the inverse proportion tube N30 is the same as the resistance formed by the inverse proportion tube N40;
the amplification factor of the common mode amplification current mirror is the same as that of the power supply amplification current mirror.
6. The wide-range high-adaptability power conversion circuit of claim 4, wherein: the voltage comparison selection output part (6) comprises a current comparator U1 and a Schmidt trigger SCT1, a collector terminal of a triode Q2 is connected with a same-phase terminal of the current comparator U1, a collector terminal of a triode Q3 is connected with an opposite-phase terminal of the current comparator U1, an output terminal of the current comparator U1 is connected with an input terminal of the Schmidt trigger SCT1, and an output terminal of the Schmidt trigger SCT1 is connected with the voltage selection driving circuit (2);
the power supply terminal of the current comparator U1 is connected to the positive terminal of the internal low voltage source VCC _ inner, and the negative terminal of the internal voltage source VCC _ inner is grounded.
7. The wide-range high-adaptability power conversion circuit according to any one of claims 2 to 6, characterized in that: the voltage selection driving circuit (2) comprises a high-voltage NMOS transistor HVN1 for receiving a voltage comparison selection signal, a high-voltage PMOS transistor HVP1 for adapting and connecting a power supply voltage VDD and a high-voltage PMOS transistor HVP2 for adapting and connecting an input common-mode voltage VCM, wherein,
the source terminal of the high-voltage NMOS tube HVN1 is connected with the source terminal of the high-voltage NMOS tube HVN2, the drain terminal of the high-voltage NMOS tube HVN1 is connected with the grid terminal of the PMOS tube P2 and one end of the resistor R30, and the source terminal of the high-voltage NMOS tube HVN1 is grounded through a current source I3;
the drain terminal of the high-voltage PMOS tube HVP1 is connected with the power supply voltage VDD, the source terminal of the high-voltage PMOS tube HVP1 is connected with the other end of the resistor R30, the source terminal of the PMOS tube P2, the emitter terminal of the triode T1, the source terminal of the high-voltage PMOS tube HVP2 and the emitter terminal of the triode T4 to form a voltage source for outputting the conversion output voltage V power A converted voltage output terminal of (a);
the grid terminal of the high-voltage PMOS tube HVP1 is connected with the drain terminal of the high-voltage NMOS tube HVN2, the drain terminal of the PMOS tube P2, the input terminal of the inverter INV1, the base terminal of the triode T3 and the collector terminal of the triode T3;
the output of the inverter INV1 is connected with the grid terminal of a high-voltage PMOS tube HVP2, the emitter terminal of the triode T3 is connected with the base terminal of the triode T2 and the collector terminal of the triode T2, and the emitter terminal of the triode T2 is connected with the base terminal of the triode T1 and the collector terminal of the triode T1;
the base electrode end of the triode T4 and the collector electrode end of the triode T4 are connected with the emitter electrode end of the triode T5, the base electrode end of the triode T5 and the collector electrode end of the triode T5 are connected with the emitter electrode end of the triode T6, the base electrode end of the triode T6 and the collector electrode end of the triode T6 are connected with a first power supply end of the inverter INV1 and a drain electrode end of the high-voltage NMOS tube HVN3, and a second power supply end of the inverter INV1 is connected with a conversion voltage output end;
the grid end of the high-voltage NMOS tube HVN3 and the grid end of the high-voltage NMOS tube HVN2 are both connected with a bias voltage NBIAS1, and the source end of the high-voltage NMOS tube HVN3 is grounded through a current source I4.
8. The wide-range high-adaptability power conversion circuit according to any one of claims 1 to 6, characterized in that: further comprises a current detection amplifier comprising an operational amplifier A1, wherein the voltage selection drive circuit outputs a converted output voltage V power As the supply voltage for the operational amplifier A1;
the non-inverting end of the operational amplifier A1 is connected with one end of a resistor R3 and one end of a resistor R4, the inverting end of the operational amplifier A1 is connected with one end of the resistor R1 and one end of a resistor R2, the other end of the resistor R3 is connected with a VIN + connecting end, the other end of the resistor R1 is connected with a VIN-connecting end, the other end of the resistor R2 is connected with the output end of the operational amplifier A1 to form a current detection amplification output end OUT of the current detection amplifier, and the other end of the resistor R4 is connected with a reference connecting end REF.
9. The wide-range high-adaptability power conversion circuit of claim 8, wherein: the resistor R4 is configured as a reference resistor R0, a tuning resistor string and a pre-trimming resistor R6, wherein,
the adjusting resistor string comprises m +1 adjusting resistors which are sequentially connected in series, one end corresponding to the first adjusting resistor in series connection forms a first end of the adjusting resistor string, one end corresponding to the tail resistor in series connection forms a second end of the adjusting resistor string, the resistance values of the m +1 adjusting resistors are increased in equal proportion along the direction of the first adjusting resistor in series connection and the tail resistor in series connection, and each adjusting resistor is connected with a trimming fuse in parallel;
one end of the reference resistor R0 is connected with one end of the pre-trimming resistor R6 and the first end of the adjusting resistor string, and the other end of the reference resistor R0 is in adaptive connection with the resistor R3 and the non-inverting end of the operational amplifier A1; the other end of the pre-trimming resistor R6 trims and trims the test reference connection terminal REF _ test, and the second end of the adjusting resistor string is connected with the reference connection terminal REF;
when the common mode rejection ratio of the current detection amplifier is configured, the pre-trimming resistor R6 is connected in, and the pre-trimming offset voltage under the condition that the pre-trimming resistor R6 is connected in is measured;
and determining a target adjusting resistance value accessed by the adjusting resistor string according to the pre-adjusting offset voltage and the target offset voltage, and fusing corresponding adjusting fuses in the adjusting resistor string to obtain an actual adjusting resistance value matched with the target adjusting resistance value.
10. The wide-range high-adaptability power conversion circuit of claim 9, wherein: the common ratio of the resistance values of the adjacent adjusting resistors in the adjusting resistor string is two, the reference resistor R0 is smaller than the resistor R2, and the resistance value of the pre-trimming adjusting resistor R6 is configured to be the resistance value corresponding to the adjusting resistor located in the middle in the adjusting resistor string.
CN202210943472.5A 2022-08-08 2022-08-08 Wide-range high-adaptability power supply conversion circuit Pending CN115276207A (en)

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN116827320A (en) * 2023-07-27 2023-09-29 江苏润石科技有限公司 Fast-response self-adaptive power supply conversion circuit

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN116827320A (en) * 2023-07-27 2023-09-29 江苏润石科技有限公司 Fast-response self-adaptive power supply conversion circuit
CN116827320B (en) * 2023-07-27 2024-01-26 江苏润石科技有限公司 Fast-response self-adaptive power supply conversion circuit

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