WO2024103442A1 - 射频功率放大器 - Google Patents

射频功率放大器 Download PDF

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WO2024103442A1
WO2024103442A1 PCT/CN2022/135078 CN2022135078W WO2024103442A1 WO 2024103442 A1 WO2024103442 A1 WO 2024103442A1 CN 2022135078 W CN2022135078 W CN 2022135078W WO 2024103442 A1 WO2024103442 A1 WO 2024103442A1
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Prior art keywords
resistor
analog switch
filter capacitor
power amplifier
module
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PCT/CN2022/135078
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English (en)
French (fr)
Inventor
肜新伟
李烨
郑海荣
陈基锋
王佳升
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深圳先进技术研究院
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Publication of WO2024103442A1 publication Critical patent/WO2024103442A1/zh

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32—Modifications of amplifiers to reduce non-linear distortion
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/52—Circuit arrangements for protecting such amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/56—Modifications of input or output impedances, not otherwise provided for
    • H03F1/565—Modifications of input or output impedances, not otherwise provided for using inductive elements
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/189—High-frequency amplifiers, e.g. radio frequency amplifiers
    • H03F3/19—High-frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only
    • H03F3/195—High-frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only in integrated circuits
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H03F3/213—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only in integrated circuits
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/451—Indexing scheme relating to amplifiers the amplifier being a radio frequency amplifier
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02D—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN INFORMATION AND COMMUNICATION TECHNOLOGIES [ICT], I.E. INFORMATION AND COMMUNICATION TECHNOLOGIES AIMING AT THE REDUCTION OF THEIR OWN ENERGY USE
    • Y02D30/00—Reducing energy consumption in communication networks
    • Y02D30/70—Reducing energy consumption in communication networks in wireless communication networks

Definitions

  • the present application relates to the technical field of medical magnetic resonance imaging, for example, to a radio frequency power amplifier.
  • RF power amplifiers are widely used in medical magnetic resonance imaging, wireless communication base stations, radar, electronic countermeasures, and other fields.
  • the ultra short echo time (UTE) in the magnetic resonance system is generally required to be in the range of 8uS-500uS, and its echo time is limited by the opening or closing time of the RF power amplifier.
  • RF power amplifiers and their bias circuits mainly inject low-frequency intermodulation distortion into the input part of the operational amplifier connected to the gate of the field effect transistor to replace the decoupling capacitor and achieve low-power intermodulation distortion decoupling.
  • the operational amplifier has a weak driving capability for capacitors, which easily causes ringing; the stabilization time increases, affecting the transient response time of the RF power amplifier; the ability to inject intermodulation distortion generated by the high-power amplifier becomes poor, causing the signal output by the RF power amplifier to have serious intermodulation distortion problems.
  • the present application provides a radio frequency power amplifier.
  • the present application provides a radio frequency power amplifier, comprising: a direct current power supply module, the direct current power supply module comprising a first filter capacitor, the first filter capacitor being connected between the output end and the ground end of the direct current power supply module, and the output end of the direct current power supply module outputs a direct current voltage; an analog switch module, comprising an analog switch and a second filter capacitor; the input end of the analog switch is connected to the output end of the direct current power supply module, and the second filter capacitor is connected between the input end and the ground end of the analog switch; wherein the capacitance value of the second filter capacitor is less than the capacitance value of the first filter capacitor; a gate direct current channel module, connected between the output end of the analog switch and the bias input end of the radio frequency power amplifier module, and configured to output a bias voltage to the radio frequency power amplifier module after processing the signal output by the analog switch module; the gate direct current channel module and the analog switch module are arranged close to the radio frequency power amplifier module; the radio frequency power amplifier module is configured to amplify the
  • FIG1 is a schematic diagram of the structure of a radio frequency power amplifier provided by the present application.
  • FIG2 is a schematic diagram of the structure of another radio frequency power amplifier provided by the present application.
  • FIG3 is a schematic diagram of the structure of another radio frequency power amplifier provided by the present application.
  • FIG4 is a schematic diagram of the structure of another radio frequency power amplifier provided in the present application.
  • FIG1 is a schematic diagram of the structure of a radio frequency power amplifier provided in the present application. This embodiment is applicable to the case of radio frequency power amplification.
  • the radio frequency power amplifier 1 comprises:
  • the DC power supply module 10 includes a first filter capacitor C1, which is connected between the output terminal of the DC power supply module 10 and the ground terminal GND.
  • the first filter capacitor C1 is the output filter capacitor of the DC power supply module 10, and the output terminal of the DC power supply module 10 outputs a DC voltage.
  • the DC power supply module 10 can be a DC regulated power supply module with temperature compensation function, adjustable voltage and low noise.
  • the temperature compensation function of the DC power supply module 10 can be used to compensate for the drift caused by the temperature change of the RF power amplifier module 40.
  • the noise range of the DC power supply module 10 can be between 0.8uVrms and 20uVrms.
  • the model of the DC power supply module 10 can be various. Exemplarily, the DC power supply module 10 can be LP38798, or LT3045, etc.
  • the analog switch module 20 includes an analog switch U1 and a second filter capacitor C2; the input end of the analog switch is connected to the output end of the DC power supply module 10, and the second filter capacitor C2 is connected between the input end of the analog switch U1 and the ground terminal GND; wherein the capacitance value of the second filter capacitor C2 is less than the capacitance value of the first filter capacitor C1.
  • the second filter capacitor C1 is arranged closer to the input end of the analog switch U1 relative to the first filter capacitor C1.
  • the second filter capacitor C1 is arranged in parallel with the first filter capacitor C1, and the input end of the analog switch U1 and the DC power supply module 10 share the second filter capacitor C2. Due to the different self-resonance frequencies of actual capacitors, the first filter capacitor C1 and the second filter capacitor C2 are connected in parallel, achieving a low impedance from hundreds of hertz to several megahertz, thereby improving the filtering effect.
  • the enable terminal of the analog switch U1 is used to control the on or off of the output voltage signal of the DC power supply module 10.
  • the type of the analog switch U1 can be an integrated circuit with a normally open and normally closed function, or a circuit with a normally open and normally closed function built by separate transistors.
  • the analog switch U1 can be TMU6219, or it can be a switch with similar functions built by ADG1419, ADG1459, ADG849 or transistors.
  • the on or off time of the analog switch U1 affects the on or off time of the RF power amplifier 1, and the on-resistance and peak current of the analog switch U1 affect the charging transient response time of the RF power amplifier 1.
  • the analog switch U1 has good RF performance and low insertion loss, providing a good channel to the second filter capacitor C2 for spurious and low-frequency intermodulation distortion.
  • the embodiment of the present application can achieve low-frequency decoupling of intermodulation distortion by setting a new second filter capacitor C2.
  • capacitors all have parasitic inductance, the larger the capacity of the capacitor, the larger the parasitic inductance, and the parasitic inductance is connected in series with the capacitor to form an LCR resonant circuit, where L is the inductance related to the lead length, R is the lead resistance, and C is the capacitor.
  • L is the inductance related to the lead length
  • R is the lead resistance
  • C the capacitor.
  • There is a resonant frequency point in the resonant circuit and the impedance at the resonant frequency point is the lowest.
  • the characteristics of the capacitor also change.
  • the capacitor When the operating frequency is lower than the resonant frequency, the capacitor is generally capacitive, and when the operating frequency is higher than the resonant frequency, the capacitor is generally inductive, and at this time the capacitor loses the function of decoupling.
  • the capacitance value selection of the capacitor generally depends on the resonant frequency of the capacitor. Capacitors with different capacitance values and different packages can filter out signals of different frequencies.
  • the first filter capacitor C1 can be used to filter out waves of about tens of Hz (hertz) to hundreds of KHz (kilohertz); the second filter capacitor C2 is used to filter out waveforms near or below one-tenth of the central operating frequency of the RF pulse signal.
  • the central operating frequency of the RF pulse signal is 210MHz (megahertz)
  • the second filter capacitor C2 is used to filter out waveforms of about 1MHz to 20MHz.
  • the main charging and discharging real constants of the transient response of the RF power amplifier 1 and the switching speed of the analog switch U1 determine the turn-on or turn-off time of the RF power amplifier 1.
  • the second filter capacitor C2 is set at the output end of the analog switch, since the analog switch U1 has an on-resistance, the charging time after passing through the analog switch U1 increases, and the main charging real constant of the transient response of the RF power amplifier 1 becomes larger.
  • the gate DC channel module 30 is connected between the output end of the analog switch U1 and the bias input end of the RF power amplifier module 40, and is used to process the signal output by the analog switch module 20 and output the bias voltage to the RF power amplifier module 40.
  • the RF power amplifier module 40 is used to amplify the RF pulse signal and output it.
  • the gate DC channel module 30 is used to ensure that the RF power amplifier 1 does not self-oscillate during operation and reduce the ringing phenomenon of the RF power amplifier module 40.
  • the RF pulse signal includes a fixed sequence of pulses and a variable sequence of pulses.
  • the DC power supply module 10, the analog switch module 20 and the gate DC channel module 30 constitute a DC bias circuit of the RF power amplifier 1, which can be applied not only to RF power amplifiers modulated by narrow pulses of a fixed sequence, but also to RF power amplifiers modulated by pulses of a variable sequence with variable pulse width and variable period, especially the ultra-short echo time UTE sequence in the magnetic resonance system, and can also be used in common RF power amplifiers.
  • the working process of the RF power amplifier 1 is as follows: the DC power supply module 10 adjusts the output DC voltage according to the needs of the RF power amplifier module 40, provides the RF power amplifier module 40 with a suitable static working current, and determines the DC voltage of the appropriate static working point.
  • the first filter capacitor C1 and the second filter capacitor C2 filter out the RF pulse signals of different working frequencies and input them into the analog switch U1.
  • the pulse signal at the enable end of the analog switch U1 controls the on and off of the analog switch U1, thereby controlling the on and off of the RF pulse signal.
  • the gate DC channel module 30 processes the signal output by the analog switch module 20 and outputs a bias voltage to the RF power amplifier module 40; the gate DC channel module 30 and the analog switch module 20 are arranged close to the RF power amplifier module 40, and the RF power amplifier module 40 is used to amplify the RF pulse signal and output it.
  • the intermodulation distortion and spurious generated by the nonlinear effect and memory effect enter the analog switch through the gate DC channel module 30, and the low-frequency band is decoupled through the first filter capacitor C1 and the second filter capacitor C2.
  • the output power is close to the P1dB compression point to produce nonlinear effect and memory effect, its process is equivalent to AM modulation, the modulation signal is the baseband pulse signal.
  • AM modulation will make the single sideband spectrum of the baseband signal move to the RF carrier, and have two symmetrical sidebands on the left and right.
  • RF pulse modulation produces more sidebands and spurious.
  • the Fourier transform of the RF pulse modulated signal can be written as:
  • ⁇ and T are the pulse width and pulse period of the RF pulse signal respectively
  • n is an arbitrary integer.
  • the discrete multiple spectrums pass through the nonlinear effect and memory effect of the RF high power amplifier, generating more frequency harmonics, intermodulation distortion and spurious signals, which are all unwanted interference signals.
  • the variable pulse width and period cause the frequency band of the intermodulation distortion and spurious signals generated by the RF power amplifier to change.
  • the decoupling of the first filter capacitor C1 and the second filter capacitor C2 can reduce the intermodulation distortion and spurious signals of the output RF pulse signal.
  • the decoupling capacitor at the output end of the analog switch can be reduced, thereby shortening the transient response time of the RF power amplifier and achieving fast turn-on or fast turn-off of the RF power amplifier; by setting the capacitance value of the second filter capacitor to be smaller than the capacitance value of the first filter capacitor; the first filter capacitor and the second filter capacitor can filter out waveforms of different frequencies, thereby reducing intermodulation distortion and spurious of the RF output signal.
  • FIG2 is a schematic diagram of the structure of another radio frequency power amplifier provided in the present application.
  • the capacitance value of the first filter capacitor C1 is 10-100 times the capacitance value of the second filter capacitor C2.
  • the filtering effect can be improved.
  • the DC power supply module 10 includes a linear voltage regulator chip U2, an input filter capacitor C4, a third resistor R3, a fourth resistor R4, a fifth resistor R5 and a sixth resistor R6; the input terminal IN of the linear voltage regulator chip U2 is connected to the first power supply VCC, the output terminal OUT of the linear voltage regulator chip U2 is connected to the first filter capacitor C1; the input filter capacitor C4 is connected to the input terminal of the linear voltage regulator chip U2; the third resistor R3, the fourth resistor R4, the fifth resistor R5 and the sixth resistor R6 are connected in series between the output terminal OUT of the linear voltage regulator chip U2 and the ground terminal GND, the common connection terminal of the fifth resistor R5 and the sixth resistor R6 is connected to the feedback input terminal FB of the linear voltage regulator chip U2, the fourth resistor R4 is an adjustable resistor, and the fifth resistor R5 is a thermistor.
  • the ground terminal GND1 of the linear voltage regulator chip U2 is grounded.
  • the output voltage Vout of the linear voltage regulator chip U2 Vref*(1+(R2+R4+R5)/R6), and Vref is the internal reference voltage value of the linear voltage regulator chip U2.
  • the third resistor R3 is a fixed resistor
  • the fourth resistor R4 is an adjustable resistor, which adjusts the DC output voltage to adjust the output voltage to meet the bias voltage requirement of the field effect tube in the RF power amplifier module 40.
  • the fifth resistor R5 is a thermistor. According to the temperature coefficient of the RF power amplifier 1, the use of an inverted thermistor can reduce the change in current caused by temperature change, thereby ensuring the stability of the output current and meeting the stability of the static working current of the field effect tube.
  • the analog switch module 20 also includes a first resistor R1, including: the input end of the analog switch U1 includes a first input end IN1 and a second input end IN2, the first input end IN1 of the analog switch U1 is connected to the output end of the DC power supply module 10, the second filter capacitor C2 is connected between the first input end IN1 of the analog switch U1 and the ground end GND, and the first resistor R1 is connected between the second input end IN2 of the analog switch U1 and the ground end GND.
  • the first resistor R1 is a discharge resistor.
  • the second end of the first resistor R1 can be grounded or connected to a negative power supply.
  • the RF power amplifier 1 is an RF power amplifier 1 composed of gallium nitride (GaN)
  • the second end of the first resistor R1 is connected to a negative power supply to shut down.
  • the pulse signal at the enable end of the analog switch U1 controls the analog switch U1 to connect the DC power supply module 10 to turn on the RF pulse signal, or the pulse signal at the enable end of the analog switch U1 controls the analog switch U1 to connect the first resistor R1 to turn off the RF pulse signal.
  • the on or off time of the analog switch U1 affects the on or off time of the RF power amplifier 1, and the on resistance and peak current of the analog switch U1 affect the transient response time of the charging of the input capacitor of the field effect tube.
  • the discharge of the input capacitor of the field effect tube is realized through the first resistor R1, and the transient response time of the discharge of the input capacitor affects the closing time of the field effect tube.
  • the opening time of the analog switch U1 and the charging time of the field effect tube input capacitor are the opening time of the field effect RF power amplifier 1;
  • the closing time of the analog switch U1 and the discharging time of the field effect tube input capacitor are the closing time of the RF power amplifier 1.
  • the gate DC channel module 30 includes a second resistor R2, a first inductor L1, a decoupling capacitor C3, and an overvoltage protection diode TVS1; the first end of the first inductor L1 is connected to the output end of the analog switch U1, the second end of the first inductor L1 is connected to the first end of the second resistor R2, and the second end of the second resistor R2 is connected to the bias input end; the overvoltage protection diode TVS1 is connected to the first end of the first inductor L1, the first end of the decoupling capacitor C3 is connected between the overvoltage protection diode TVS1 and the first end of the first inductor L1, and the second end of the decoupling capacitor C3 is connected to the ground end GND.
  • the second resistor R2 is a stable resistor.
  • the second resistor R2, the first inductor L1 and the decoupling capacitor C3 are used together to ensure that the RF power amplifier 1 does not self-oscillate when working; in addition, the second resistor R2 plays a damping role in the transient response process of the field effect tube, reducing the ringing phenomenon of the gate during the conduction process of the field effect tube.
  • the first inductor L1 attenuates part of the energy at the central operating frequency of the RF pulse signal, and the remaining RF energy is decoupled through the decoupling capacitor C3.
  • the intermodulation distortion and harmonics generated by the nonlinear effect and memory effect of the RF power amplifier 1, and the high-frequency band energy are also decoupled through the decoupling capacitor C3.
  • the decoupling capacitor C3 is a decoupling capacitor selected for the central operating frequency of the RF pulse signal.
  • the capacitance value is relatively low, generally in the pico-farad (pF) level.
  • the capacitance value of the decoupling capacitor C3 is less than the capacitance value of the first filter capacitor C1, which is mainly determined by the central operating frequency of the RF pulse signal.
  • the analog switch U1 is decoupled by the second filter capacitor C2.
  • the capacitance value of the second filter capacitor C2 is relatively large, and the commonly used value is in the microfarad (uF) level, thereby reducing the intermodulation distortion of the RF pulse signal as a whole. Due to the bandwidth (3dB) limitation of the analog switch U1, the center operating frequency of the RF pulse signal is usually higher than the bandwidth of the analog switch U1; when the center operating frequency of the RF pulse signal is within the bandwidth of the analog switch U1, the decoupling capacitor C3 can be used. At this time, the first filter capacitor C1 and the second filter capacitor C2 are used for decoupling, and the transient response time is further improved.
  • the second end of the second resistor R2 is also connected to the third power source VBiss.
  • the RF power amplifier module 40 includes an input matching and DC isolation circuit M1, an output matching and DC isolation circuit M2, a field effect transistor Q1, and a second inductor L2; the input matching and DC isolation circuit M1 is connected between the RF pulse signal input terminal RF_in and the gate of the field effect transistor Q1, the second inductor L2 is connected between the first electrode of the field effect transistor Q1 and the second power supply VDD, the output matching and DC isolation circuit M2 is connected between the RF pulse signal output terminal RF_out and the first electrode of the field effect transistor Q1, the second electrode of the field effect transistor Q1 is connected to the ground terminal GND, and the gate of the field effect transistor Q1 is used as a bias input terminal.
  • the second inductor L2 is a choke inductor.
  • the RF power amplifier module 40 further includes a fifth filter capacitor C5, which is used to filter out the noise of the second power supply VDD to make the output of the second power supply VDD more stable.
  • the fifth filter capacitor is connected between the second power supply VDD and the ground terminal GND.
  • the input matching and DC isolation circuit M1 matches the RF input signal with the field effect transistor Q1 in the working frequency band and isolates the DC bias of the gate of the field effect transistor Q1; the output matching and DC isolation circuit M2 matches the RF output signal with the output of the field effect transistor Q1 and isolates the second power supply VDD connected to the second pole of the field effect transistor Q1.
  • the second power supply VDD of the field effect tube is usually high, and the overvoltage protection diode TVS1 keeps the gate of the field effect tube at a safe voltage.
  • the parasitic capacitance of the field effect tube in a multi-kilowatt RF power amplifier cannot be ignored.
  • the input capacitance (Ciss) value is large, and the generated spurious and harmonic waves have high power. It is not suitable to use an operational amplifier to decouple the gate input. Usually, capacitors are used for decoupling to reduce the spurious and harmonic waves after the output.
  • the input capacitance Ciss of the field effect tube Q1 is Cgs+Cgd.
  • Cgs is the parasitic capacitance between the gate and the source
  • Cgd is the parasitic capacitance between the gate and the drain.
  • Ru2 is the on-resistance of the analog switch U1, which is usually less than a few ohms.
  • the charging and discharging real constants and the switching speed of the analog switch U1 determine the turn-on or turn-off time of the RF power amplifier 1.
  • the total value of the first filter capacitor C1 and the second filter capacitor C2 is much larger than the field effect tube input capacitance and the decoupling capacitor C3.
  • the envelope amplitude of the stray AM modulation signal is reduced by adding the second filter capacitor C2 before the analog switch U1 and without the second filter capacitor C2; or by using a spectrum analyzer and an attenuator, the power of the stray signal is reduced by observing the RF pulse signal output terminal RF_out.
  • the short on or off time of the RF power amplifier is beneficial to the transmission of RF hard pulses.
  • the on or off time of a conventional RF power amplifier is 10uS to 20uS, which cannot meet the requirement of the ultra-short echo time of 8uS-500uS in the magnetic resonance system; the on or off time of the RF power amplifier of the present application is 40nS-300nS, which meets the requirements of the UTE sequence RF hard pulse RF transmission and meets the pulse sequences of other functions.
  • the second filter capacitor C2 is disposed close to the field effect transistor Q1. By disposing the second filter capacitor C2 close to the field effect transistor Q1, the influence of the PCB routing can be reduced and the decoupling performance of the second filter capacitor C2 can be improved.
  • the RF high power amplifier used is a MRFX1K80N field effect tube.
  • the capacitance value of the decoupling capacitor C3 is generally 510pF
  • the capacitance value of the second filter capacitor C2 is generally 1uF
  • the capacitance value of the first filter capacitor C1 is generally 10uF.
  • the second resistor R2 is 6.2 ohms
  • the first inductor L1 is 22nH
  • the first resistor R1 is 100 ohms
  • the second power supply VDD is 75V
  • the saturated output power of the RF pulse signal is 2400W
  • the opening or closing time of the RF power amplifier is 40nS-300nS, which meets the requirements of the UTE sequence RF hard pulse RF transmission.
  • the embodiment of the present application is verified by experiments.
  • the maximum low-frequency band maximum amplitude in the output is 2.9MH AM modulation signal
  • the AM modulation signal with a period of 1S and a pulse width of 1mS and a maximum intermodulation frequency of 7.3MHz the output is lower than -51dBc after adding the second filter capacitor.
  • the intermodulation signals in the low-frequency band are also different, and the use of this example has an improvement of more than 6dB.
  • the embodiment of the present application can be applied to a variable sequence pulse power amplifier to realize the rapid opening or closing of the variable sequence pulse power amplifier, especially the ultra-short echo time sequence in the magnetic resonance system. Since the on-resistance and peak current of the analog switch affect the transient response time of the RF power amplifier, By setting the second filter capacitor and setting the position of the second filter capacitor at the input end of the analog switch, the decoupling capacitor at the output end of the analog switch can be reduced, thereby shortening the transient response time of the RF power amplifier and realizing fast opening or fast shutting down of the RF power amplifier; by setting the capacitance value of the second filter capacitor to be smaller than the capacitance value of the first filter capacitor; the first filter capacitor and the second filter capacitor can filter out waveforms of different frequencies and reduce the intermodulation distortion of the RF output signal; the decoupling capacitor can filter out the remaining RF energy of the central working frequency of the RF pulse signal, and can also reduce the intermodulation distortion and high-frequency band energy of the harmonics caused by
  • FIG3 is a schematic diagram of the structure of another radio frequency power amplifier provided in the present application.
  • the gate DC channel module 30 includes a second resistor R2, a first inductor L1, a decoupling capacitor C3, and an overvoltage protection diode TVS1; the first end of the first inductor L1 is connected to the output end of the analog switch U1, the second end of the first inductor L1 is connected to the first end of the second resistor R2, and the second end of the second resistor R2 is connected to the bias input end; the overvoltage protection diode TVS1 is connected to the first end of the first inductor L1; the decoupling capacitor C3 is connected between the ground end GND and the first end of the second resistor R2, and the second end of the second resistor R2 is connected to the bias input end to achieve stable reduction of coupling.
  • the DC power supply module 10, the analog switch module 20 and the RF power amplifier module 40 in Fig. 3 are shown as an example including the structure shown in Fig. 2.
  • the RF power amplifier 1 has the beneficial effects of any of the above embodiments of the present application.
  • FIG4 is a schematic diagram of the structure of another radio frequency power amplifier provided in the present application.
  • the DC power supply module 10 includes an operational amplifier U3, a first input filter capacitor C6, a transistor Q2, a seventh resistor R7, an eighth resistor R8, a ninth resistor R9, a tenth resistor R10 and an eleventh resistor R11;
  • the first input filter capacitor C6 is connected to the in-phase input terminal IN+ of the operational amplifier U3;
  • the seventh resistor R7, the eighth resistor R8 and the ninth resistor R9 are connected in series between the first power supply VCC and the ground terminal GND, and the common end of the eighth resistor R8 and the ninth resistor R9 is electrically connected to the in-phase input terminal IN+ of the operational amplifier U3 to provide a reference voltage for the in-phase input terminal IN+ of the operational amplifier U3;
  • the eighth resistor R8 is an adjustable resistor, and the ninth resistor R9 is a thermistor.
  • the tenth resistor R10 is connected between the inverting input terminal IN- of the operational amplifier U3 and the ground terminal GND; the eleventh resistor R11 is connected between the collector of the transistor Q2 and the first end of the tenth resistor R10; the base of the transistor Q2 is connected to the output end of the operational amplifier U3, the collector of the transistor Q2 is connected to the first power supply VCC, and the emitter of the transistor Q2 is connected to the first filter capacitor C1.
  • a transistor is added to an operational amplifier (without requiring conversion rate and bandwidth) to increase the driving current, thereby forming a low-noise, low-cost DC power supply module 10.
  • the analog switch module 20, the gate DC channel module 30 and the RF power amplifier module 40 in Fig. 4 are shown as an example including the structure shown in Fig. 2.
  • the RF power amplifier 1 has the beneficial effects of any of the above embodiments of the present application.

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Abstract

本申请公开了一种射频功率放大器，包括：直流电源模块，直流电源模块包括第一滤波电容，第一滤波电容连接于直流电源模块的输出端和接地端之间；模拟开关模块，包括模拟开关和第二滤波电容；模拟开关的输入端连接直流电源模块的输出端，第二滤波电容连接于模拟开关的输入端和接地端之间；栅极直流通道模块，连接于模拟开关的输出端和射频功率放大模块的偏置输入端之间。

Description

射频功率放大器
本申请要求在2022年11月18日提交中国专利局、申请号为202211445087.4的中国专利申请的优先权，以上申请的全部内容通过引用结合在本申请中。
技术领域
本申请涉及医疗磁共振成像技术领域，例如涉及一种射频功率放大器。
背景技术
随着无线技术的发展，射频功率放大器广泛的应用到医疗磁共振成像、无线通讯的基站、雷达、电子对抗、等领域。其中磁共振系统中的超短回波时间(ultra short echo time，UTE)，UTE的时间范围一般要求在8uS-500uS，其回波时间受到射频功率放大器的开启或关闭时间的限制。
当前，射频功率放大器及其偏置电路，主要是把低频的互调失真灌入场效应管栅极连接的运算放大器的输入部分，来替代去耦合电容，实现功率较低的互调失真去耦合。
运算放大器对于电容的驱动能力较弱，容易造成振铃现象；稳定时间增加，影响射频功率放大器的瞬态响应时间；对高功率放大器产生的互调失真灌入能力变差，使得射频功率放大器输出的信号存在较严重的互调失真问题。
发明内容
本申请提供了一种射频功率放大器。
本申请的提供了一种射频功率放大器，包括：直流电源模块，直流电源模块包括第一滤波电容，第一滤波电容连接于直流电源模块的输出端和接地端之间，直流电源模块的输出端输出直流电压；模拟开关模块，包括模拟开关和第二滤波电容；模拟开关的输入端连接直流电源模块的输出端，第二滤波电容连接于模拟开关的输入端和接地端之间；其中，第二滤波电容的电容值小于第一滤波电容的电容值；栅极直流通道模块，连接于模拟开关的输出端和射频功率放大模块的偏置输入端之间，设置为对模拟开关模块输出的信号处理后输出偏置电压至射频功率放大模块；栅极直流通道模块和模拟开关模块，靠近射频功率放大模块设置；射频功率放大模块设置为将射频脉冲信号进行放大后输出。
应当理解，本部分所描述的内容并非旨在标识本申请的实施例的关键或重 要特征，也不用于限制本申请的范围。本申请的其它特征将通过以下的说明书而变得容易理解。
附图说明
图1是本申请提供的一种射频功率放大器的结构示意图；
图2是本申请提供的又一种射频功率放大器的结构示意图；
图3是本申请提供的又一种射频功率放大器的结构示意图；
图4是本申请提供的又一种射频功率放大器的结构示意图。
具体实施方式
需要说明的是，本申请的说明书和权利要求书及上述附图中的术语“第一”、“第二”等是用于区别类似的对象，而不必用于描述特定的顺序或先后次序。应该理解这样使用的数据在适当情况下可以互换，以便这里描述的本申请的实施例能够以除了在这里图示或描述的那些以外的顺序实施。此外，术语“包括”和“具有”以及他们的任何变形，意图在于覆盖不排他的包含，例如，包含了一系列步骤或单元的过程、方法、系统、产品或设备不必限于清楚地列出的那些步骤或单元，而是可包括没有清楚地列出的或对于这些过程、方法、产品或设备固有的其它步骤或单元。
图1为本申请提供的一种射频功率放大器的结构示意图，本实施例可适用于射频功率放大的情况。
如图1所示，该射频功率放大器1包括：
直流电源模块10，直流电源模块10包括第一滤波电容C1，第一滤波电容C1连接于直流电源模块10的输出端和接地端GND之间，第一滤波电容C1为直流电源模块10的输出滤波电容，直流电源模块10的输出端输出直流电压。
其中，直流电源模块10可以是具有温度补偿功能的、电压可调的和低噪声的直流稳压电源模块。在一实施例中，直流电源模块10的温度补偿功能，可以用来补偿射频功率放大模块40温度变换造成的漂移。直流电源模块10噪声范围可以在0.8uVrms到20uVrms之间。直流电源模块10的型号可以是各种的。示例性的，直流电源模块10可以是LP38798，还可以是LT3045等。
模拟开关模块20，包括模拟开关U1和第二滤波电容C2；模拟开关的输入端连接直流电源模块10的输出端，第二滤波电容C2连接于模拟开关U1的输入端和接地端GND之间；其中，第二滤波电容C2的电容值小于第一滤波电容C1 的电容值。第二滤波电容C1相对于第一滤波电容C1更靠近模拟开关U1的输入端设置。换句话说，第二滤波电容C1与第一滤波电容C1并联设置，模拟开关U1的输入端和直流电源模块10共享第二滤波电容C2。由于实际电容的自谐振频点不同，第一滤波电容C1和第二滤波电容C2并联，实现了从数百赫兹到数兆赫兹的低阻抗，提高了滤波效果。
模拟开关U1的使能端，用于控制直流电源模块10输出电压信号的接通或断开。模拟开关U1的类型可以是具有常开常闭功能的集成电路，还可以是分离的晶体管搭建的具有常开常闭功能的电路。示例性的，模拟开关U1可以是TMU6219，还可以是ADG1419、ADG1459、ADG849或者晶体管搭建的类似功能的开关。模拟开关U1的导通或关断时间，影响射频功率放大器1的开启或关断的时间，模拟开关U1的导通电阻和峰值电流，影响射频功率放大器1的充电瞬态响应时间。模拟开关U1具有良好的射频性能，插入损耗小，为杂散和低频段的互调失真提供良好通道到第二滤波电容C2。
本申请实施例通过设置新的第二滤波电容C2,可以实现对互调失真的低频段去耦合。例如，由于电容都存在寄生电感，容量越大的电容一般寄生电感就越大，而且寄生电感与电容串联，形成LCR谐振电路，这里L是和引线长度有关的电感、R是引线电阻、C是电容。谐振电路存在一个谐振频率点，在谐振频率点处的阻抗最低，随着射频功率放大器工作频率不同，电容的特性也随之变化，在工作频率低于谐振频率时，电容总体呈容性，在工作频率高于谐振频率时，电容总体呈感性，此时电容就失去了去耦的作用。电容的容值选择一般取决于电容的谐振频率，不同容值和不同封装的电容可以滤除不同频率的信号，通过设置第二滤波电容C2的电容值小于第一滤波电容C1的电容值，可以使得第一滤波电容C1用于滤除几十Hz(赫兹)到几百KHz(千赫兹)左右的波；第二滤波电容C2用于滤除射频脉冲信号中心工作频率十分之一频率附近及其以下的波形。示例性的，当射频脉冲信号的中心工作频率为210MHz(兆赫兹)时，第二滤波电容C2用于滤除1MHz到20MHz左右的波形。
由于射频功率放大器1瞬态响应的主要充放电实常数和模拟开关U1的切换速度，决定了射频功率放大器1的开启或关闭时间。当第二滤波电容C2设置在模拟开关的输出端时，由于模拟开关U1存在导通电阻，因此经过模拟开关U1后充电时间增加，射频功率放大器1瞬态响应的主要充电实常数变大，通过设置第二滤波电容的位置在模拟开关U1的输入端，可以缩短射频功率放大器1的瞬态响应时间，实现射频功率放大器1的快速开启或快速关断。
栅极直流通道模块30，连接于模拟开关U1的输出端和射频功率放大模块40的偏置输入端之间，用于对模拟开关模块20输出的信号处理后输出偏置电压至射频功率放大模块40。射频功率放大模块40用于将射频脉冲信号进行放大后输出。栅极直流通道模块30用于确保射频功率放大器1在工作的时候不发生自激振荡，减少射频功率放大模块40的振铃现象。
射频脉冲信号包括固定序列的脉冲和可变序列的脉冲。换句话说，直流电源模块10、模拟开关模块20和栅极直流通道模块30构成射频功率放大器1的直流偏置电路，该直流偏置电路不仅能够应用于固定序列的窄脉冲调制的射频功率放大器，也可以应用到可变脉冲宽度和可变周期的可变序列的脉冲调制的射频功率放大器，尤其是磁共振系统中的超短回波时间UTE序列，也可以用于常用的射频功率放大器。
射频功率放大器1的工作过程为：直流电源模块10根据射频功率放大模块40的需求调节输出直流电压，为射频功率放大模块40提供合适的静态工作电流，确定合适的静态工作点的直流电压。第一滤波电容C1和第二滤波电容C2滤除不同工作频率的射频脉冲信号后输入到模拟开关U1，通过模拟开关U1使能端的脉冲信号控制模拟开关U1的导通与关断，进而控制射频脉冲信号开启与关断。当模拟开关U1导通时，栅极直流通道模块30对模拟开关模块20输出的信号处理后输出偏置电压至射频功率放大模块40；栅极直流通道模块30和模拟开关模块20，靠近射频功率放大模块40设置，射频功率放大模块40用于将射频脉冲信号进行放大后输出。
射频脉冲信号经过射频功率放大模块40后，由非线性效应和记忆效应产生的互调失真和杂散，经栅极直流通道模块30进入模拟开关，低频段的经第一滤波电容C1和第二滤波电容C2去耦合。
产生射频脉冲调制信号，输出功率靠近P1dB压缩点产生非线性效应和记忆效应，它的过程相当于AM调制，调制信号为基带脉冲信号。AM调制将使得基带信号的单边带频谱搬移至射频载波上，同时具有左右两个对称边带，射频脉冲调制产生更多的边带和杂散。
射频脉冲调制信号的傅里叶变换可以写为：
Figure PCTCN2022135078-appb-000001
式中，τ和T分别为射频脉冲信号的脉冲宽度和脉冲周期，Ω＝2π/T和ω c分 别为射频脉冲信号的重复频率和载波频率，n为任意整数。
由此傅里叶变换可知，射频脉冲信号包含的频率分量为ω c+nΩ，理论上具有无穷多频率分量。正因为信号是周期性的，所以其频谱是离散的，相邻谱线频间距为Ω，Sa(x)函数决定了各个谱线的幅度。
离散的多个频谱经过射频高功率放大器的非线性效应和记忆效应，产生更多频点的谐波、互调失真和杂散，这些都是不需要的干扰信号。可变的脉宽和周期，造成射频功率放大器产生的互调失真和杂散的频段发生变化。经过第一滤波电容C1和第二滤波电容C2的去耦合，可以降低输出射频脉冲信号的互调失真和杂散。
本申请实施例，由于模拟开关的导通电阻和峰值电流，影响射频功率放大器的瞬态响应时间，通过设置第二滤波电容，并且设置第二滤波电容的位置在模拟开关的输入端，可以减少模拟开关输出端的去耦合电容，从而可以缩短射频功率放大器的瞬态响应时间，实现射频功率放大器的快速开启或快速关断；通过设置第二滤波电容的电容值小于第一滤波电容的电容值；可以使得第一滤波电容和第二滤波电容滤除不同频率的波形，减少射频输出信号的互调失真和杂散。
图2为本申请提供的又一种射频功率放大器的结构示意图。
如图2所示，在一实施例中，第一滤波电容C1的电容值为第二滤波电容C2电容值的10-100倍。通过设置第一滤波电容C1的电容值为第二滤波电容C2电容值的10-100倍，可以提高滤波效果。
直流电源模块10包括线性稳压芯片U2、输入滤波电容C4、第三电阻R3、第四电阻R4、第五电阻R5和第六电阻R6；线性稳压芯片U2的输入端IN连接第一电源VCC、线性稳压芯片U2的输出端OUT连接第一滤波电容C1；输入滤波电容C4连接线性稳压芯片U2的输入端；第三电阻R3、第四电阻R4、第五电阻R5和第六电阻R6串联于线性稳压芯片U2的输出端OUT和接地端GND之间，第五电阻R5和第六电阻R6的公共连接端连接线性稳压芯片U2的反馈输入端FB，第四电阻R4为可调电阻，第五电阻R5为热敏电阻。线性稳压芯片U2的接地端GND1接地。
其中，线性稳压芯片U2的输出电压Vout＝Vref*(1+(R2+R4+R5)/R6)，Vref为线性稳压芯片U2的内部参考电压值。第三电阻R3为固定电阻，第四电阻R4为可调电阻，调节直流输出电压，用来调节输出的电压，满足射频功率放大模块40中场效应管偏置电压的需求，第五电阻R5为热敏电阻，根据射频功率放 大器1的温度系数，使用反相的热敏电阻，可以降低温度变化引起电流的变化，从而保证输出电流的稳定性，满足场效应管静态工作电流的稳定性。
模拟开关模块20还包括第一电阻R1，包括：模拟开关U1的输入端包括第一输入端IN1和第二输入端IN2，模拟开关U1的第一输入端IN1连接直流电源模块10的输出端，第二滤波电容C2连接于模拟开关U1的第一输入端IN1和接地端GND之间，第一电阻R1连接于模拟开关U1的第二输入端IN2和接地端GND之间。其中，第一电阻R1为放电电阻。
在一实施例中，针对不同型号的射频功率放大器1，第一电阻R1的第二端可以接地，还可以接负电源。示例性的，当射频功率放大器1为氮化镓(GaN)构成的射频功率放大器1时，第一电阻R1的第二端接负电源进行关闭。模拟开关U1使能端的脉冲信号控制模拟开关U1接通直流电源模块10开启射频脉冲信号或者模拟开关U1使能端的脉冲信号控制模拟开关U1接通第一电阻R1经过地关闭射频脉冲信号。模拟开关U1的导通或关闭时间，影响射频功率放大器1的开启或关闭的时间，模拟开关U1的导通电阻和峰值电流，影响场效应管输入电容充电的瞬态响应时间。模拟开关U1关闭后，通过第一电阻R1实现场效应管的输入电容的放电，输入电容放电的瞬态响应时间影响场效应管的关闭时间。其中，模拟开关U1开启时间和场效应管输入电容的充电时间，即为场效应的射频功率放大器1的开启时间；模拟开关U1的关闭时间和场效应管输入电容的放电时间，即为射频功率放大器1的关闭时间。
栅极直流通道模块30包括第二电阻R2、第一电感L1、去耦电容C3、和过压保护二极管TVS1；第一电感L1的第一端连接模拟开关U1的输出端，第一电感L1的第二端连接第二电阻R2的第一端，第二电阻R2的第二端连接偏置输入端；过压保护二极管TVS1连接第一电感L1的第一端，去耦电容C3的第一端连接于过压保护二极管TVS1和第一电感L1的第一端之间，去耦电容C3的第二端连接接地端GND。
其中，第二电阻R2为稳定电阻，第二电阻R2、第一电感L1和去耦电容C3三者配合使用，可以确保射频功率放大器1在工作的时候不发生自激振荡；另外第二电阻R2在场效应管的瞬态响应过程中，起到阻尼作用，减少场效应管导通过程中栅极的振铃现象。第一电感L1在射频脉冲信号的中心工作频率，衰减部分能量，剩余的射频能量，经过去耦电容C3去耦合。射频功率放大器1由于非线性效应和记忆效应产生的互调失真及谐波，高频段能量，也经过去耦电容C3去耦合。去耦电容C3是针对射频脉冲信号的中心工作频率选取的去耦电 容，电容值比较低，一般为皮法(pF)级，去耦电容C3的电容值小于第一滤波电容C1的电容值，主要由射频脉冲信号的中心工作频率决定。对于非线性效应和记忆效应产生的较低频段，经过模拟开关U1，由第二滤波电容C2去耦合，第二滤波电容C2的电容值比较大，常用的为微法(uF)级，从而整体降低射频脉冲信号的互调失真。由于模拟开关U1的带宽(3dB)限制，通常射频脉冲信号中心工作频率高于模拟开关U1的带宽；当射频脉冲信号的中心工作频率处于模拟开关U1的带宽内，去耦合电容C3可以不用，此时由第一滤波电容C1和第二滤波电容C2去耦合，并且进一步提高瞬态响应时间。
第二电阻R2的第二端还与第三电源VBiss连接。
射频功率放大模块40包括输入匹配及直流隔离电路M1、输出匹配及直流隔离电路M2、场效应管Q1、和第二电感L2；输入匹配及直流隔离电路M1连接于射频脉冲信号输入端RF_in和场效应管Q1的栅极之间，第二电感L2连接于场效应管Q1的第一极和第二电源VDD之间，输出匹配及直流隔离电路M2连接于射频脉冲信号输出端RF_out和场效应管Q1的第一极之间，场效应管Q1的第二极连接接地端GND，场效应管Q1的栅极作为偏置输入端。其中，第二电感L2是扼流电感。
射频功率放大模块40还包括第五滤波电容C5，第五滤波电容C5用于滤除第二电源VDD的杂波，使第二电源VDD的输出更稳定。第五滤波电容连接于第二电源VDD和接地端GND之间。
输入匹配及直流隔离电路M1使射频输入信号和场效应管Q1在工作频段匹配，并隔离场效应管Q1栅极的直流偏置；输出匹配及直流隔离电路M2使射频输出信号和场效应管Q1输出匹配，并隔离场效应管Q1第二极连接的第二电源VDD。
场效应管的第二电源VDD通常较高，过压保护二极管TVS1，使得场效应管的栅极处于一个安全电压下。数千瓦级别的射频功率放大器中场效应管的寄生电容不可以忽视，输入电容(Ciss)值较大，产生的杂散和谐波，功率较大，不适合使用运算放大器在栅极输入灌流去耦合，通常使用电容来去耦合，降低输出后的杂散和谐波。
场效应管Q1的输入电容Ciss＝Cgs+Cgd。其中，Cgs为栅极与源极之间的寄生电容，Cgd为栅极与漏极之间的寄生电容。射频功率放大器1瞬态响应的主要充电实常数为：τ1＝R2*Ciss，τ4＝Ru2*C2。其中，Ru2为模拟开关U1的导通电阻，通常小于几欧姆。射频功率放大器1瞬态响应的主要放电实常数为τ 2＝(R1+R2)*Ciss，τ3＝R1*C3。充放电实常数和模拟开关U1的切换速度，决定了射频功率放大器1的开启或关闭时间。第一滤波电容C1和第二滤波电容C2的总值远大于场效应管输入电容和去耦电容C3。通过将第二滤波电容C2放在模拟开关U1之前，对比通常放在模拟开关U1之后的滤波电容，对开启或关闭时间，都有极大的改善，主要充电实常数，从Ru2*(C2+C3)改变到Ru2*C2,示波器上可以观察瞬态响应时间的差异。在示波器上，从第五滤波电容C5端，能够清楚的观察到模拟开关U1前增加第二滤波电容C2和没有第二滤波电容C2，产生杂散的AM调制信号包络幅度的降低；或者使用频谱分析仪和衰减器，观察射频脉冲信号输出端RF_out，杂散信号的功率降低。射频功率放大器短的开启或关闭时间，有益于射频硬脉冲的发射，常规的射频功率放大器的开启或关闭时间为10uS到20uS，满足不了磁共振系统中的超短回波时间8uS—500uS的要求；本申请的射频功率放大器的开启或关闭时间为40nS—300nS，满足UTE序列射频硬脉冲射频发射的需求，而且满足其它功能的脉冲序列。
在一实施例中，第二滤波电容C2靠近场效应管Q1设置。通过设置第二滤波电容C2靠近场效应管Q1，可以降低PCB走线的影响，提高第二滤波电容C2的去耦合性能。
本申请实施例中，使用的射频高功率放大器为MRFX1K80N的场效应管，当射频脉冲信号的中心工作频率为210MHz时，去耦电容C3的电容值一般为510pF，第二滤波电容C2的电容值一般为1uF，第一滤波电容C1的电容值一般为10uF。第二电阻R2为6.2欧姆，第一电感L1为22nH,第一电阻R1为100欧姆，第二电源VDD为75V，射频脉冲信号的饱和输出功率为2400W，射频功率放大器的开启或关闭时间为40nS—300nS，满足UTE序列射频硬脉冲射频发射的需求。本申请实施例通过实验验证，在测试中心工作频率为210MHz，P1dB输出功率为2KW的射频功率放大器时，对于周期为100mS脉宽为200uS，输出中最大的低频段最大幅度为2.9MH的AM调制信号，通过设置模拟开关输入端的第二滤波电容(1uF)后，从输出端使用频谱分析仪检测到，AM调制信号输出幅度从-43dBc降到-50dBc。对于周期为1S脉宽为1mS，互调频率最大幅度的为7.3MHz的AM调制信号,添加第二滤波电容后输出低于-51dBc。对于周期和脉宽不同，低频段的互调信号也不同，使用本示例都有6dB以上的改善。
本申请实施例，能够应用于可变序列的脉冲功率放大器，实现可变序列脉冲功率放大器的快速开启或关闭，尤其是磁共振系统中的超短回波时间序列，由于模拟开关的导通电阻和峰值电流，影响射频功率放大器的瞬态响应时间， 通过设置第二滤波电容，并且设置第二滤波电容的位置在模拟开关的输入端，可以减少模拟开关输出端的去耦合电容，从而可以缩短射频功率放大器的瞬态响应时间，实现射频功率放大器的快速开启或快速关断；通过设置第二滤波电容的电容值小于第一滤波电容的电容值；可以使得第一滤波电容和第二滤波电容滤除不同频率的波形，减少射频输出信号的互调失真；通过去耦电容，可以滤除射频脉冲信号中心工作频率剩余的射频能量，还可以减少非线性效应和记忆效应产生的互调失真及谐波的高频段能量，通过设置第四电阻为可调电阻，可以使得直流输出模块输出电压可调，进而使得输出的直流电压满足场效应管的栅极的偏置电压的需求；通过设置第五电阻为热敏电阻，可以降低场效应管的温度变化引起的漂移，从而保证输出电流的稳定性；通过设置第二电阻第一电感和去耦电容，可以确保射频功率放大器在全频段稳定工作，不发生自激振荡，减少场效应管栅极的振铃现象。本申请的射频功率放大器的开启或关闭时间为40nS—300nS，能够满足磁共振中超短回波时间的序列的需求，也可以用于常用的射频功率放大器。
图3为本申请实提供的又一种射频功率放大器的结构示意图。
如图3所示，栅极直流通道模块30包括第二电阻R2、第一电感L1、去耦电容C3、和过压保护二极管TVS1；第一电感L1的第一端连接模拟开关U1的输出端，第一电感L1的第二端连接第二电阻R2的第一端，第二电阻R2的第二端连接偏置输入端；过压保护二极管TVS1连接第一电感L1的第一端；去耦电容C3连接于接地端GND和第二电阻R2的第一端之间，第二电阻R2的第二端连接偏置输入端，实现稳定降低耦合。
在一实施例中，图3的直流电源模块10、模拟开关模块20和射频功率放大模块40以包括图2所示的结构为例进行示出。该射频功率放大器1具备上述本申请任一实施例的有益效果。
图4为本申请实提供的又一种射频功率放大器的结构示意图。
如图4所示，直流电源模块10包括运算放大器U3、第一输入滤波电容C6、三极管Q2、第七电阻R7、第八电阻R8、第九电阻R9、第十电阻R10和第十一电阻R11；第一输入滤波电容C6连接运算放大器U3的同相输入端IN+；第七电阻R7、第八电阻R8和第九电阻R9串联于第一电源VCC和接地端GND之间，第八电阻R8和第九电阻R9的公共端与运算放大器U3的同相输入端IN+电连接，以为运算放大器U3的同相输入端IN+提供参考电压；第八电阻R8为可调电阻，第九电阻R9为热敏电阻。第十电阻R10连接于运算放大器 U3的反相输入端IN-和接地端GND之间；第十一电阻R11连接于三极管Q2的集电极和第十电阻R10的第一端之间；三极管Q2的基极连接运算放大器U3的输出端，三极管Q2的集电极连接第一电源VCC，三极管Q2的发射极连接第一滤波电容C1。
本申请实施例，通过运算放大器(不要求转换速率和带宽)添加晶体管提高驱动电流，构成低噪声、低成本的直流电源模块10。
在一实施例中，图4的模拟开关模块20、栅极直流通道模块30和射频功率放大模块40以包括图2所示的结构为例进行示出。该射频功率放大器1具备上述本申请任一实施例的有益效果。
本领域技术人员应该明白的是，根据设计要求和其他因素，可以进行各种修改、组合、子组合和替代。

Claims (10)

  1. 一种射频功率放大器，包括：
    直流电源模块(10)，所述直流电源模块(10)包括第一滤波电容(C1)，第一滤波电容(C1)连接于所述直流电源模块(10)的输出端和接地端(GND)之间，所述直流电源模块(10)的输出端输出直流电压；
    模拟开关模块(20)，包括模拟开关(U1)和第二滤波电容(C2)；所述模拟开关(U1)的输入端连接所述直流电源模块(10)的输出端，所述第二滤波电容(C2)连接于所述模拟开关(U1)的输入端和接地端(GND)之间；其中，所述第二滤波电容(C2)的电容值小于所述第一滤波电容(C1)的电容值；
    栅极直流通道模块(30)，连接于所述模拟开关(U1)的输出端和射频功率放大模块(40)的偏置输入端之间，设置为对所述模拟开关模块(20)输出的信号处理后输出偏置电压至所述射频功率放大模块(40)；
    所述栅极直流通道模块(30)和所述模拟开关模块(20)，靠近所述射频功率放大模块(40)设置；所述射频功率放大模块(40)设置为将射频脉冲信号进行放大后输出。
  2. 根据权利要求1所述的射频功率放大器，其中，所述第一滤波电容(C1)的电容值为所述第二滤波电容(C2)的电容值的10-100倍。
  3. 根据权利要求1所述的射频功率放大器，其中，所述模拟开关模块(20)还包括第一电阻(R1)，包括：
    所述模拟开关(U1)的输入端包括第一输入端(IN1)和第二输入端(IN2)，所述模拟开关(U1)的第一输入端(IN1)连接所述直流电源模块(10)的输出端，所述第二滤波电容(C2)连接于所述模拟开关(U1)的第一输入端(IN1)和接地端(GND)之间，所述第一电阻(R1)连接于所述模拟开关(U1)的第二输入端(IN2)和接地端(GND)之间。
  4. 根据权利要求1所述的射频功率放大器，其中，所述栅极直流通道模块(30)包括第二电阻(R2)、第一电感(L1)、去耦电容(C3)、和过压保护二极管(TVS1)；
    所述第一电感(L1)的第一端连接所述模拟开关(U1)的输出端，所述第一电感(L1)的第二端连接所述第二电阻(R2)的第一端，所述第二电阻(R2)的第二端连接所述偏置输入端；
    所述过压保护二极管(TVS1)连接所述第一电感(L1)的第一端，所述去耦电容(C3)的第一端连接于所述过压保护二极管(TVS1)和所述第一电感(L1)的第一端之间，所述去耦电容(C3)的第二端连接接地端(GND)。
  5. 根据权利要求1所述的射频功率放大器，其中，所述栅极直流通道模块(30)包括第二电阻(R2)、第一电感(L1)、去耦电容(C3)、和过压保护二极管(TVS1)；
    所述第一电感(L1)的第一端连接所述模拟开关(U1)的输出端，所述第一电感(L1)的第二端连接所述第二电阻(R2)的第一端，所述第二电阻(R2)的第二端连接所述偏置输入端；
    所述过压保护二极管(TVS1)连接所述第一电感(L1)的第一端；
    所述去耦电容(C3)连接于所述接地端(GND)和所述第二电阻(R2)的第一端之间，所述第二电阻(R2)的第二端连接所述偏置输入端。
  6. 根据权利要求4或5所述的射频功率放大器，其中，所述去耦电容(C3)的电容值小于所述第一滤波电容(C1)的电容值。
  7. 根据权利要求1-5任一项所述的射频功率放大器，其中，所述直流电源模块(10)包括线性稳压芯片(U2)、输入滤波电容(C4)、第三电阻(R3)、第四电阻(R4)、第五电阻(R5)和第六电阻(R6)；
    所述线性稳压芯片(U2)的输入端(IN)连接第一电源(VCC)、所述线性稳压芯片(U2)的输出端(OUT)连接所述第一滤波电容(C1)；
    所述输入滤波电容(C4)连接所述线性稳压芯片(U2)的输入端；
    所述第三电阻(R3)、第四电阻(R4)、第五电阻(R5)和第六电阻(R6)串联于所述线性稳压芯片(U2)的输出端(OUT)和所述接地端(GND)之间，所述第五电阻(R5)和所述第六电阻(R6)的公共连接端连接所述线性稳压芯片(U2)的反馈输入端(FB)，所述第四电阻(R4)为可调电阻，所述第五电阻(R5)为热敏电阻。
  8. 根据权利要求1-5任一项所述的射频功率放大器，其中，所述直流电源模块(10)包括运算放大器(U3)、第一输入滤波电容(C6)、三极管(Q2)、第七电阻(R7)、第八电阻(R8)、第九电阻(R9)、第十电阻(R10)和第十一电阻(R11)；
    所述第一输入滤波电容(C6)连接所述运算放大器(U3)的同相输入端(IN+)；
    所述第七电阻(R7)、第八电阻(R8)和第九电阻(R9)串联于第一电源(VCC)和接地端(GND)之间，所述第八电阻(R8)和所述第九电阻(R9)的公共端与所述运算放大器(U3)的同相输入端(IN+)电连接，以为所述运算放大器(U3)的同相输入端(IN+)提供参考电压；所述第八电阻(R8)为可 调电阻，所述第九电阻(R9)为热敏电阻；
    所述第十电阻(R10)连接于所述运算放大器(U3)的反相输入端(IN-)和接地端(GND)之间；
    所述第十一电阻(R11)连接于所述三极管(Q2)的集电极和第十电阻(R10)的第一端之间；
    所述三极管(Q2)的基极连接所述运算放大器(U3)的输出端，所述三极管(Q2)的集电极连接第一电源(VCC)，所述三极管(Q2)的发射极连接所述第一滤波电容(C1)。
  9. 根据权利要求1-5任一项所述的射频功率放大器，其中，所述射频功率放大模块(40)包括输入匹配及直流隔离电路(M1)、输出匹配及直流隔离电路(M2)、场效应管(Q1)、和第二电感(L2)；
    所述输入匹配及直流隔离电路(M1)连接于射频脉冲信号输入端(RF_in)和所述场效应管(Q1)的栅极之间，所述第二电感(L2)连接于所述场效应管(Q1)的第一极和第二电源(VDD)之间，所述输出匹配及直流隔离电路(M2)连接于射频脉冲信号输出端(RF_out)和所述场效应管(Q1)的第一极之间，所述场效应管(Q1)的第二极连接接地端(GND)，所述场效应管(Q1)的栅极作为所述偏置输入端。
  10. 根据权利要求1所述的射频功率放大器，其中，所述射频脉冲信号包括固定序列的脉冲和可变序列的脉冲。
PCT/CN2022/135078 2022-11-18 2022-11-29 射频功率放大器 WO2024103442A1 (zh)

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Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20030179048A1 (en) * 2002-03-25 2003-09-25 Arild Kolsrud Amplifier bias system and method
CN112311334A (zh) * 2019-07-23 2021-02-02 杭州灵芯微电子有限公司 一种用于5g通讯中的功率放大器
CN112631358A (zh) * 2020-12-28 2021-04-09 陕西烽火电子股份有限公司 一种ldmos功放管的栅极稳压电路
CN112803900A (zh) * 2021-03-30 2021-05-14 广州慧智微电子有限公司 偏置电路及射频功率放大器

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20030179048A1 (en) * 2002-03-25 2003-09-25 Arild Kolsrud Amplifier bias system and method
CN112311334A (zh) * 2019-07-23 2021-02-02 杭州灵芯微电子有限公司 一种用于5g通讯中的功率放大器
CN112631358A (zh) * 2020-12-28 2021-04-09 陕西烽火电子股份有限公司 一种ldmos功放管的栅极稳压电路
CN112803900A (zh) * 2021-03-30 2021-05-14 广州慧智微电子有限公司 偏置电路及射频功率放大器

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