WO2024060855A1 - 无桥隔离型ac-dc单级pfc变换器及其控制方法 - Google Patents

无桥隔离型ac-dc单级pfc变换器及其控制方法 Download PDF

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Publication number
WO2024060855A1
WO2024060855A1 PCT/CN2023/110969 CN2023110969W WO2024060855A1 WO 2024060855 A1 WO2024060855 A1 WO 2024060855A1 CN 2023110969 W CN2023110969 W CN 2023110969W WO 2024060855 A1 WO2024060855 A1 WO 2024060855A1
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Prior art keywords
rectifier
filter
transformer
bridge
output
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PCT/CN2023/110969
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English (en)
French (fr)
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张朝辉
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张丽娜
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Publication of WO2024060855A1 publication Critical patent/WO2024060855A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/38—Means for preventing simultaneous conduction of switches
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration

Definitions

  • the invention relates to a bridgeless isolation AC-DC single-stage PFC converter and a control method thereof. It is a high-efficiency electric energy conversion/switching power supply technology and belongs to the field of power electronics/new energy technology.
  • the single-stage PFC (Power Factor Correction) converter is an AC input/DC output switching converter; its function is to stabilize the required DC output with only one power conversion, while achieving power factor correction on the AC side.
  • the so-called power factor correction (PFC) means that the input AC current tracks the AC voltage so that their waveforms are consistent and in the same phase, thereby achieving a high power factor.
  • PFC (Power Factor Correction) converters are generally non-isolated.
  • Commonly used ones include Boost topology, Buck topology, buck-boost topology and totem pole topology.
  • Most of the isolated single-stage PFC converters use flyback topology, and there are also topologies such as half-bridge type.
  • the flyback topology is a single-ended conversion, and the transmitted power cannot be too large, so it is generally suitable for low-power applications.
  • topologies such as half-bridge can be applied to medium and high power, due to the existence of the power frequency rectifier bridge, the rectification loss is large in high-power applications.
  • the purpose of the present invention is to overcome the above-mentioned shortcomings of the prior art and propose a bridgeless isolation AC-DC single-stage PFC converter and a control method thereof.
  • the converter is an isolated single-stage PFC converter with input and output isolation, stable DC output and high power factor.
  • the converter is an AC-DC circuit without a power frequency rectifier bridge. It adopts a full-bridge topology and is a double-ended conversion. It can transmit high power and improve efficiency and reduce costs.
  • the converter includes two technical solutions: single transformer and double transformer, and the corresponding control methods are also divided into two types: one is single-arm PWM control, corresponding to the single-transformer solution; the other is double-arm PWM control, corresponding to For the dual transformer solution.
  • a bridgeless isolated AC-DC single-stage PFC converter has two technical solutions: the first is a single transformer solution, and the second is a dual transformer solution.
  • the two technical solutions have the same input filter circuit and conversion bridge.
  • the input filter circuit contains common mode and differential mode inductors, X capacitors, and Y capacitors, and has two AC output terminals.
  • the conversion bridge contains switching tubes Q1, Q2, Q3, and Q4. It is a full-bridge topology with two bridge arms. Switching tubes Q1 and Q2 constitute the first bridge arm, and switching tubes Q3 and Q4 constitute the second bridge arm.
  • the drain of switch Q1 and the source of Q2 are connected as nodes V1, the drain of switch Q3 and the source of Q4 are connected as node V2; the drains of switch Q2 and Q4 are connected as the positive terminal Vd of the transformer bridge, and the sources of switch Q1 and Q3 are connected as the ground terminal GND of the transformer bridge.
  • the single transformer solution consists of input filter circuit, capacitor network, converter bridge, transformer, rectifier and filter, which are connected in sequence.
  • the AC power supply Ua is connected to the input filter circuit, and the two AC output terminals of the input filter circuit provide a stable AC voltage; the filter supplies the DC voltage with high-frequency ripple filtered out as the output of the converter to the load.
  • the capacitor network includes capacitors Cs, Cd, C1, and C2, which all use film capacitors instead of electrolytic capacitors.
  • the two ends of the capacitor Cs are respectively connected to the two AC output terminals of the input filter circuit.
  • the two ends of the capacitor Cd are respectively connected to the positive terminal Vd and the ground terminal GND of the conversion bridge.
  • One end of the capacitor C1 and C2 is connected to an AC output terminal of the input filter circuit as Node Vs
  • the other end of the capacitor C1 is connected to the ground terminal GND of the conversion bridge
  • the other end of the capacitor C2 is connected to the positive terminal Vd of the conversion bridge
  • the node V2 of the conversion bridge is connected to the other AC output end of the input filter circuit.
  • the transformer has a primary winding Np and at least one secondary winding Ns.
  • the primary winding Np has two ends, and the secondary winding Ns has both ends and a center tap, or the center tap is removed.
  • One end of the primary winding Np is connected to the node Vs of the capacitor network, and the other end is connected to the node V1 of the transformer bridge. Both ends of the secondary winding Ns are connected to the two AC input terminals of the rectifier respectively; when the rectifier adopts the full-wave rectification topology, the middle tap of Ns serves as the positive or negative output terminal of the rectifier; when the rectifier adopts the full-bridge rectification topology, then Remove the center tap of Ns.
  • the rectifier adopts full-wave rectification topology or full-bridge rectification topology, with a positive output terminal, a negative output terminal and two AC input terminals.
  • the full-wave rectification topology includes diodes D1 and D2, which are connected in two ways.
  • the first connection method is called common cathode connection, that is, the cathodes of diodes D1 and D2 are connected together as the positive output terminal of the rectifier, the anode of diode D1 and the anode of D2 are used as the two AC input terminals of the rectifier, and the secondary winding of the transformer The center tap of Ns serves as the negative output terminal of the rectifier.
  • the second connection method is called common anode connection, that is, the anodes of diodes D1 and D2 are connected together as the negative output terminal of the rectifier, and the cathode of diode D1 and D2 are used as the two AC input terminals of the rectifier.
  • the secondary winding of the transformer The center tap of Ns serves as the positive output terminal of the rectifier.
  • the full-bridge rectifier topology includes diodes D1, D2, D3, and D4.
  • the anode of diode D1 is connected to the cathode of D2 as an AC input terminal of the rectifier.
  • the anode of diode D3 is connected to the cathode of D4 as another AC input terminal of the rectifier.
  • the diode The cathodes of D1 and D3 are connected as the positive output terminal of the rectifier, and the anodes of diodes D2 and D4 are connected as the negative output terminal of the rectifier.
  • the filter includes filter inductors Lo1, Lo2 and filter capacitor Co, forming a four-terminal network with a positive input terminal, a negative input terminal, a positive output terminal, and a negative output terminal.
  • the two ends of the filter inductor Lo1 serve as the positive input end and the positive output end of the filter respectively
  • the two ends of the filter inductor Lo2 serve as the negative input end and the negative output end of the filter respectively
  • the positive and negative poles of the filter capacitor Co are connected to the positive end of the filter respectively. output and negative output.
  • Filter inductors Lo1 and Lo2 are independent or coupled, or removed Lo1 or remove Lo2 to simplify the circuit; when Lo1 is removed, the positive input terminal and the positive output terminal of the filter are directly connected, and when Lo2 is removed, the negative input terminal and the negative output terminal of the filter are directly connected. Or remove the filter capacitor Co to apply to current source mode output. When Co is retained, it is a voltage source mode output.
  • the positive input terminal and the negative input terminal of the filter are respectively connected to the positive output terminal and the negative output terminal of the rectifier.
  • the positive output terminal and the negative output terminal of the filter are respectively used as the positive output terminal +Vo and the negative output terminal -Vo of the converter.
  • the switching tubes Q3 and Q4 of the converting bridge can be replaced with diodes D01 and D02 respectively, or the switching tubes Q3 and Q4 of the converting bridge can be replaced with unidirectional thyristors S1 and S2 respectively;
  • the replacement rule is that the anode and cathode of the diode or unidirectional thyristor correspond to the source and drain of the switch tube respectively.
  • the capacitors C1 and C2 of the capacitor network are retained. One end of the capacitors C1 and C2 is commonly connected to the node Vs or the node V2 of the conversion bridge. The other ends of the capacitors C1 and C2 are respectively connected to the ground terminal GND and the positive terminal Vd of the conversion bridge.
  • the dual transformer solution includes an input filter circuit, a capacitor network, a conversion bridge, a first transformer, a second transformer, a rectifier X, a rectifier Y, a filter or two filters, namely, filter X and filter Y.
  • the AC power source Ua is connected to the input filter circuit, and the two AC output terminals of the input filter circuit provide a stable AC voltage.
  • Rectifier X and rectifier Y are respectively connected to filter X and filter Y to form two groups of outputs; or, rectifier X and rectifier Y are connected to a filter in combination to form a group of outputs.
  • the capacitor network includes capacitors Cs, Cd, C1, C2, C3, and C4, which all use film capacitors instead of electrolytic capacitors.
  • the two ends of the capacitor Cs are connected to the two AC output terminals of the input filter circuit, and the two ends of the capacitor Cd are connected to the positive terminal Vd and the ground terminal GND of the transformer bridge respectively.
  • One end of the capacitor C1 and C3 is connected to an AC output terminal of the input filter circuit.
  • node Vs1 one end of capacitors C2 and C4 is connected to the other AC output end of the input filter circuit.
  • node Vs2 the other ends of capacitors C1 and C2 are connected to the ground terminal GND of the transformer bridge, and the other ends of capacitors C3 and C4 are connected to the transformer.
  • the first transformer has a primary winding Np1 and at least one secondary winding Ns1; the primary winding Np1 has two ends, and the secondary winding Ns1 has two ends and a center tap, or the center tap is removed.
  • the second transformer has a primary winding Np2 and at least one secondary winding Ns2; the primary winding Np2 has two ends, and the secondary winding Ns2 has two ends and a center tap, or the center tap is removed.
  • the two ends of the primary winding Np1 are respectively connected to the node Vs1 of the capacitor network and the node V1 of the transformer bridge.
  • the two ends of the primary winding Np2 are respectively connected to the node Vs2 of the capacitor network and the node V2 of the transformer bridge.
  • Both ends of the secondary winding Ns1 are connected to the two AC input terminals of the rectifier X respectively; when the rectifier X adopts the full-wave rectification topology, the middle tap of Ns1 serves as the positive or negative output terminal of the rectifier topology, remove the middle tap of Ns1.
  • Both ends of the secondary winding Ns2 are connected to the two AC input terminals of the rectifier Y respectively; when the rectifier Y adopts the full-wave rectification topology, the middle tap of Ns2 serves as the positive or negative output terminal of the rectifier Y; when the rectifier Y adopts the full-bridge rectification topology, remove the middle tap of Ns2.
  • Rectifier X and rectifier Y adopt full-wave rectification topology or full-bridge rectification topology, with a positive output terminal, a negative output terminal and two AC input terminals. Its internal diodes are connected in the same way as the rectifier described in the single transformer scheme, two copies of it.
  • a single filter, filter X, and filter Y are all four-terminal networks with positive input terminals, negative input terminals, positive output terminals, and negative output terminals.
  • the internal network structure is the same as the filter described in the single-transformer solution.
  • Rectifier X is connected to filter X and rectifier Y is connected to filter Y to form two sets of outputs, namely a first output and a second output.
  • the specific connection method is: the positive output terminal and the negative output terminal of the rectifier X are respectively connected to the positive input terminal and the negative input terminal of the filter X.
  • the positive output terminal and the negative output terminal of the rectifier Y are respectively connected to the positive input terminal and the negative input terminal of the filter Y.
  • the positive output terminal and the negative output terminal of the filter Y are respectively used as the positive terminal +Vo2 and the negative terminal of the second output of the converter. End-Vo2.
  • the first output and the second output can supply the load independently, or supply the load in parallel or series.
  • the rectifier X and the rectifier Y are jointly connected to a filter in a combined manner to form a set of outputs.
  • the positive output terminal and the negative output terminal of the filter are respectively used as the positive terminal +Vo and the negative terminal -Vo of the converter output.
  • the combination methods are divided into parallel combination and series combination.
  • the parallel combination is: the positive output terminals of rectifier X and rectifier Y are jointly connected to the positive input terminal of the filter, the negative output terminals of rectifier X and rectifier Y are jointly connected to the negative input terminal of the filter; the series combination is: the negative output terminal of rectifier The positive output terminal of the rectifier Y, the positive output terminal of the rectifier X and the negative output terminal of the rectifier Y are respectively connected to the positive input terminal and the negative input terminal of the filter.
  • the rectifier and the diodes in rectifier X and rectifier Y are replaced with switching tubes; the replacement rule is that the source and drain of the switching tube correspond to the anode and cathode of the diode respectively.
  • the so-called bidirectional conversion refers to the switching conversion in which electric energy flows in both directions between the AC side and the DC side.
  • the first derivative circuit is that the transformer has a parallel inductance with the primary windings of the first transformer and the second transformer.
  • the self-inductance of the parallel inductance is much smaller than the self-inductance of the primary winding.
  • the second derivative circuit is to add an inductor to the position of the transformer and the primary windings of the first transformer and the second transformer, and the primary windings are connected in series with capacitors.
  • connection method of the second derivative circuit is: the two ends of the inductor Lr are connected to the node V1 of the transformer bridge and the node Vs of the capacitor network respectively; one end of the primary winding Np of the transformer is connected to the node V1 and the other end of the transformer bridge. Connect one end of the capacitor Cr, and the other end of the capacitor Cr is connected to the node V2 of the conversion bridge, the positive terminal Vd, the ground terminal GND, or the node Vs of the capacitor network.
  • the connection method of the second derivative circuit is: the two ends of the inductor Lr1 are connected to the node V1 of the transformer bridge and the node Vs1 of the capacitor network respectively, and one end of the primary winding Np1 of the first transformer is connected to the nodes V1 and Vs1 of the transformer bridge.
  • the other end is connected to one end of the capacitor Cr1, and the other end of the capacitor Cr1 is connected to the positive terminal Vd of the transformer bridge or the ground terminal GND, or Connect the node Vs1 or node Vs2 of the capacitor network; the two ends of the inductor Lr2 are respectively connected to the node V2 of the transformer bridge and the node Vs2 of the capacitor network.
  • One end of the primary winding Np2 of the second transformer is connected to the node V2 of the transformer bridge, and the other end is connected to the capacitor Cr2.
  • One end of the capacitor Cr2 and the other end of the capacitor Cr2 are connected to the positive terminal Vd or the ground terminal GND of the conversion bridge, or to the node Vs1 or node Vs2 of the capacitor network.
  • the dual-bridge arm PWM control is further divided into two modes: frequency multiplication modulation and bipolar modulation.
  • the second bridge arm composed of switching tubes Q3 and Q4 switches at a low frequency at the frequency of the AC power supply Ua.
  • the positive half cycle of the AC power supply Ua that is, the voltage of the node Vs is higher than the voltage of the node V2
  • the upper switch Q4 is turned off and the lower switch Q3 is turned on
  • the negative half cycle of the AC power supply Ua that is, the node Vs The voltage is lower than the voltage of node V2, then the lower switch Q3 is turned off and the upper switch Q4 is turned on.
  • the first bridge arm composed of switching tubes Q1 and Q2 transforms at high frequency under complementary PWM control.
  • complementary PWM means that the sum of the duty ratios of switching tubes Q1 and Q2 is equal to 1 when the dead time is ignored.
  • Control rule During the positive half cycle of AC power supply Ua, the output voltage of the converter is proportional to the average value of D 1 ; during the negative half cycle of AC power supply Ua, the output voltage of the converter is proportional to the average value of D 2 .
  • the instantaneous values of D 1 and D 2 change according to the requirements of power factor correction, so that the input current tracks the sinusoidal AC voltage of the power supply Ua, so that their waveforms are consistent and have the same phase.
  • the control process is as follows:
  • the output variable On the DC side, the output variable is sampled, and the output variable is output voltage, output current or output power;
  • the AC voltage sampling signal controls the low-frequency switching of the second bridge arm and serves as the waveform phase reference of the input current
  • the control quantity is compared with the triangular wave to generate a pulse width modulation signal
  • the two complementary PWM signals drive the switching tubes Q1 and Q2 of the first bridge arm for high-frequency transformation.
  • the first bridge arm and the second bridge arm of the conversion bridge convert at high frequency under complementary PWM control at the same switching frequency.
  • the switching tubes Q3 and Q4 are synchronized with the conduction driving pulses of the switching tubes Q2 and Q1 respectively (i.e. in the same phase); in the frequency doubling modulation mode, the switching tubes Q3 and Q4 are respectively in synchronization with the conduction driving pulses of the switching tubes Q1 and Q2. Pass drive pulse synchronization.
  • Control law The output voltage of the converter is proportional to (D 1 ⁇ D 2 ) min /
  • the instantaneous values of D 1 and D 2 change according to the requirements of power factor correction.
  • the control process is as follows:
  • the output variable On the DC side, the output variable is sampled, and the output variable is output voltage, output current or output power;
  • the feedback variable is isolated and negatively fed back to the AC side.
  • the reference value is compared with the input current sampling signal to obtain an input current error
  • the four PWM signals drive the high-frequency transformation of the switch tubes of the first bridge arm and the second bridge arm respectively.
  • the present invention has the following advantages.
  • the present invention is an isolated single-stage PFC conversion, using a full-bridge topology, which is a double-ended conversion and can transmit high power.
  • the present invention is divided into two solutions: single transformer and double transformer. The latter can output greater power in various combinations.
  • the present invention simplifies the topology and eliminates the need for a power frequency rectifier bridge, thereby reducing costs, improving efficiency, and enhancing reliability.
  • the present invention has a wide output voltage range and is suitable for both voltage source mode and current source mode.
  • the present invention can adopt a variety of control methods such as single-bridge arm PWM, dual-bridge arm frequency doubling or bipolar PWM.
  • Figure 1 is the schematic diagram of the single transformer solution of the bridgeless isolated AC-DC single-stage PFC converter.
  • Figure 2 is a schematic diagram of a rectifier using a full-wave rectification topology with common cathode connection.
  • Figure 3 is a schematic diagram of a rectifier using a full-wave rectification topology with common anode connection.
  • Figure 4 is a schematic diagram of a rectifier using a full-bridge rectification topology.
  • Figure 5 is a schematic diagram of the network topology of the filter.
  • Figure 6 is the schematic diagram of the two outputs of the double transformer solution of the bridgeless isolated AC-DC single-stage PFC converter.
  • Figure 7 is a schematic diagram of the parallel output of the dual transformer solution of the bridgeless isolated AC-DC single-stage PFC converter.
  • Figure 8 is a schematic diagram of the series output of the dual transformer solution of the bridgeless isolated AC-DC single-stage PFC converter.
  • Figure 9 is a schematic diagram of a single transformer solution in which the switch tube of the converter bridge is replaced by a diode.
  • Figure 10 is a schematic diagram of a single transformer solution in which the diodes of the rectifier are replaced by switching tubes.
  • Figure 11 is the schematic diagram of the first derivative circuit of the single transformer solution.
  • Figure 12 is the second derivative circuit schematic diagram of the single transformer solution.
  • a bridgeless isolated AC-DC single-stage PFC converter has two technical solutions.
  • the first is a single transformer solution, see Figure 1; the second is a double transformer solution, see Figures 6, 7, and 8.
  • the two technical solutions have the same input filter circuit (1) and conversion bridge (3).
  • the input filter circuit (1) contains common mode and differential mode inductors, X capacitors, and Y capacitors, and has two AC output terminals.
  • the conversion bridge (3) contains switching tubes Q1, Q2, Q3, and Q4. It is a full-bridge topology with two bridge arms. Switching tubes Q1 and Q2 constitute the first bridge arm, and switching tubes Q3 and Q4 constitute the second bridge arm.
  • the drain of switch Q1 and the source of Q2 are connected as node V1, the drain of switch Q3 and the source of Q4 are connected as node V2; the drains of switch Q2 and Q4 are connected as the positive terminal Vd of the conversion bridge (3), and switch Q1 Connected to the source of Q3 as the ground terminal GND of the conversion bridge (3).
  • the switching tube uses but is not limited to MOSFET or IGBT.
  • the single transformer solution includes an input filter circuit (1), a capacitor network (2), a conversion bridge (3), a transformer (4), a rectifier (5) and a filter (6), which are connected in sequence.
  • the AC power source Ua is connected to the input filter circuit (1), and the two AC output terminals of the input filter circuit (1) provide a stable AC voltage; the filter (6) supplies the DC voltage with high-frequency ripples filtered out as the output of the converter to the load.
  • the capacitor network (2) includes capacitors Cs, Cd, C1, and C2, which all use small-capacity film capacitors instead of large-capacity electrolytic capacitors, especially Cd.
  • the two ends of the capacitor Cs are connected to the two AC output terminals of the input filter circuit (1) respectively, the two ends of the capacitor Cd are connected to the positive terminal Vd and the ground terminal GND of the transformer bridge (3) respectively; one end of the capacitors C1 and C2 are connected to the input filter
  • An AC output end of the circuit (1) serves as the node Vs.
  • the other end of the capacitor C1 is connected to the ground terminal GND of the conversion bridge (3).
  • the other end of the capacitor C2 is connected to the positive terminal Vd of the conversion bridge (3).
  • Node V2 is connected to the other AC output terminal of the input filter circuit (1). Either remove capacitors C1 and C2, or remove capacitor Cs, or remove capacitor Cd to simplify the circuit and reduce cost, but the ripple current of each capacitor will change.
  • the transformer (4) has a primary winding Np and at least one secondary winding Ns.
  • the primary winding Np has two ends, and the secondary winding Ns has both ends and a center tap, or the center tap is removed.
  • One end of the primary winding Np is connected to the node Vs of the capacitor network (2), and the other end is connected to the node V1 of the transformer bridge (3). Both ends of the secondary winding Ns are connected to the two AC input terminals of the rectifier (5) respectively; when the rectifier (5) adopts the full-wave rectification topology, the middle tap of Ns serves as the positive or negative output terminal of the rectifier (5); when When the rectifier (5) adopts the full-bridge rectification topology, the center tap of Ns is removed.
  • the rectifier (5) adopts a full-wave rectification topology or a full-bridge rectification topology, and has a positive output terminal, a negative output terminal and two AC input terminals. As shown in Figure 2, Figure 3, and Figure 4.
  • the full-wave rectification topology includes diodes D1 and D2, which are connected in two ways.
  • the first is called common cathode connection (see Figure 2), that is, the cathodes of diodes D1 and D2 are connected together as the positive output terminal of the rectifier (5), and the anode of diode D1 and the anode of D2 serve as the two terminals of the rectifier (5).
  • An AC input terminal, the center tap of the secondary winding Ns of the transformer (4) serves as the negative output terminal of the rectifier (5).
  • the second type is called common anode connection (see Figure 3), that is, the anodes of diodes D1 and D2 are connected together as the negative output terminal of the rectifier (5), and the cathode of diode D1 and the cathode of D2 serve as the two terminals of the rectifier (5).
  • An AC input terminal, the center tap of the secondary winding Ns of the transformer (4) serves as the positive output terminal of the rectifier (5).
  • the full-bridge rectifier topology includes diodes D1, D2, D3, and D4.
  • the anode of diode D1 is connected to the cathode of D2 as an AC input terminal of the rectifier (5).
  • the anode of diode D3 is connected to the cathode of D4 as the rectifier.
  • the cathodes of diodes D1 and D3 are connected as the positive output terminal of the rectifier (5)
  • the anodes of diodes D2 and D4 are connected as the negative output terminal of the rectifier (5).
  • the filter (6) includes filter inductors Lo1, Lo2 and filter capacitor Co, forming a four-terminal network with a positive input terminal, a negative input terminal, a positive output terminal, and a negative output terminal, as shown in Figure 5.
  • the two ends of the filter inductor Lo1 serve as the positive input end and the positive output end of the filter (6) respectively
  • the two ends of the filter inductor Lo2 serve as the negative input end and the negative output end of the filter (6) respectively
  • the positive and negative poles of the filter capacitor Co Connect the positive and negative output terminals of the filter (6) respectively.
  • the filter inductors Lo1 and Lo2 are independent or coupled, or Lo1 or Lo2 is removed to simplify the circuit; when Lo1 is removed, the positive input terminal and the positive output terminal of the filter (6) are directly connected, and when Lo2 is removed, the filter (6) The negative input terminal and the negative output terminal are directly connected. Or remove the filter capacitor Co to suit the current source mode output. When Co is retained, it is a voltage source mode output.
  • the positive input terminal and the negative input terminal of the filter (6) are respectively connected to the positive output terminal and the negative output terminal of the rectifier (5), and the positive output terminal and the negative output terminal of the filter (6) are respectively used as the positive output terminal of the converter. +Vo and output negative terminal -Vo.
  • Diodes are uncontrolled components
  • unidirectional thyristors are semi-controlled and low-frequency components
  • switching tubes are fully controlled and high-frequency components.
  • the double-transformer solution includes an input filter circuit (1), a capacitor network (2), a conversion bridge (3), a first transformer (41), a second transformer (42), and a rectifier.
  • the AC power supply Ua is connected to the input filter circuit (1), and the two AC output terminals of the input filter circuit (1) provide stable AC voltage.
  • Rectifier X (51) and rectifier Y (52) are connected to filter X (61) and filter Y (62) respectively to form two sets of outputs; alternatively, rectifier X (51) and rectifier Y (52) are jointly connected in a combined manner Filter (6), forming a set of outputs.
  • the capacitor network (2) includes capacitors Cs, Cd, C1, C2, C3, and C4, which all use small-capacity film capacitors instead of large-capacity electrolytic capacitors, especially Cd.
  • the two ends of the capacitor Cs are respectively connected to the two AC output terminals of the input filter circuit (1), the two ends of the capacitor Cd are connected to the positive terminal Vd and the ground terminal GND of the transformer bridge (3) respectively; one end of the capacitors C1 and C3 are connected to Connect one AC output end of the input filter circuit (1) as the node Vs1, one end of the capacitor C2 and C4 is connected to the other AC output end of the input filter circuit (1) as the node Vs2, and the other end of the capacitor C1 and C2 are connected to the transformer The ground terminal GND of the bridge (3), and the other ends of the capacitors C3 and C4 are connected to the positive terminal Vd of the conversion bridge (3). Or remove any two of C1, C2, C3, C4, or remove Cs or Cd, or remove Cs and Cd to simplify the circuit
  • the first transformer (41) has a primary winding Np1 and at least one secondary winding Ns1; the primary winding Np1 has two ends, and the secondary winding Ns1 has two ends and a center tap, or the center tap is removed.
  • the second transformer (42) has a primary winding Np2 and at least one secondary winding Ns2; the primary winding Np2 has two ends, and the secondary winding Ns2 has two ends and a center tap, or the center tap is removed.
  • the two ends of the primary winding Np1 are respectively connected to the node Vs1 of the capacitor network (2) and the node V1 of the transformer bridge (3).
  • the two ends of the primary winding Np2 are respectively connected to the node Vs2 of the capacitor network (2) and the node of the transformer bridge (3).
  • Both ends of the secondary winding Ns1 are connected to the two AC input terminals of the rectifier end; when the rectifier X(51) adopts the full-bridge rectification topology, remove the center tap of Ns1.
  • Both ends of the secondary winding Ns2 are connected to the two AC input terminals of the rectifier Y (52) respectively; when the rectifier Y (52) adopts the full-wave rectification topology, the center tap of Ns2 serves as the positive or negative output of the rectifier Y (52) end; when the rectifier Y (52) adopts the full-bridge rectification topology, remove the center tap of Ns2.
  • the rectifier X (51) and the rectifier Y (52) adopt a full-wave rectification topology or a full-bridge rectification topology, and have a positive output terminal, a negative output terminal and two AC input terminals.
  • the full-wave rectification topology includes diodes D1 and D2, and there are two connection methods;
  • the first connection method is called common cathode connection, that is, the cathodes of diodes D1 and D2 are connected together as the positive output terminal of the rectifier, and the anode of diode D1
  • the anodes of D2 and D2 serve as the two AC input terminals of the rectifier, the middle tap of the secondary winding Ns1 of the first transformer (41) serves as the negative output terminal of the rectifier X (51), and the middle tap of the secondary winding Ns2 of the second transformer (42)
  • the second connection method is called common anode connection, that is, the anodes of diodes D1 and D2 are connected together as the negative output terminal of the rectifier, and the cathode of diode D1 and the cathode of D2 serve as the rectifier.
  • the center tap of the secondary winding Ns1 of the first transformer (41) serves as the positive output terminal of the rectifier X (51), and the center tap of the secondary winding Ns2 of the second transformer (42) serves as the rectifier Y (52) the positive output terminal.
  • the full-bridge rectifier topology includes diodes D1, D2, D3, and D4.
  • the anode of diode D1 is connected to the cathode of D2 as an AC input terminal of the rectifier.
  • the anode of diode D3 is connected to the cathode of D4 as another AC input terminal of the rectifier.
  • the diode The cathodes of D1 and D3 are connected as the positive output terminal of the rectifier, and the anodes of diodes D2 and D4 are connected as the negative output terminal of the rectifier.
  • Filter Filter capacitor Co The two ends of the filter inductor Lo1 serve as the positive input end and the positive output end of the filter respectively, the two ends of the filter inductor Lo2 serve as the negative input end and the negative output end of the filter respectively; the positive and negative poles of the filter capacitor Co are connected to the positive end of the filter respectively. output and negative output.
  • filter The inductors Lo1 and Lo2 are independent or coupled, or Lo1 or Lo2 is removed to simplify the circuit; when Lo1 is removed, the positive input terminal and the positive output terminal of the filter are directly connected, and when Lo2 is removed, the negative input terminal and the negative output terminal of the filter are directly connected. directly connected. Or remove the filter capacitor Co to apply to current source mode output.
  • Rectifier X (51) and rectifier Y (52) are connected to filter X (61) and filter Y (62) respectively to form two sets of outputs, that is, rectifier X (51) is connected to filter The first output, rectifier Y (52) is connected to filter Y (62) to form the second output of the converter, see Figure 6.
  • the positive output terminal and the negative output terminal of the rectifier X (51) are respectively connected to the positive input terminal and the negative input terminal of the filter X (61).
  • the positive output terminal and the negative output terminal of the filter One output has the positive terminal +Vo1 and the negative terminal -Vo1.
  • the positive output terminal and the negative output terminal of the rectifier Y (52) are respectively connected to the positive input terminal and the negative input terminal of the filter Y (62).
  • the positive output terminal and the negative output terminal of the filter Y (62) are respectively used as the second output.
  • the first output and the second output can supply the load independently or in parallel or series.
  • rectifier terminal +Vo and negative terminal -Vo The combination methods are divided into parallel combination and series combination.
  • the parallel combination (see Figure 7) is: the positive output terminals of rectifier X (51) and rectifier Y (52) are jointly connected to the positive input terminal of filter (6), and the negative outputs of rectifier X (51) and rectifier Y (52) terminals are commonly connected to the negative input terminal of the filter (6);
  • the series combination (see Figure 8) is: the negative output terminal of the rectifier X (51) is connected to the positive output terminal of the rectifier Y (52), and the positive output terminal of the rectifier X (51) The terminal is connected to the positive input terminal of the filter (6), and the negative output terminal of the rectifier Y (52) is connected to the negative input terminal of the filter (6).
  • the diodes in the rectifier (5), rectifier X (51), and rectifier Y (52) are replaced with switching tubes.
  • the switching tubes are but not limited to MOSFET or IGBT; the replacement rules are,
  • the source and drain of the switch tube correspond to the anode and cathode of the diode respectively.
  • An example of a single transformer solution is shown in Figure 10.
  • the first derivative circuit is a parallel inductance of the primary windings of the transformer (4) and the first transformer (41) and the second transformer (42).
  • the self-inductance of the parallel inductance is much smaller than the self-inductance of the primary winding.
  • the primary winding Np of the transformer (4) is connected in parallel with the inductor Lr, so that the self-inductance of the inductor Lr is much smaller than the self-inductance of the primary winding Np. See Figure 11.
  • the primary winding Np1 of the first transformer (41) is connected in parallel with the inductor Lr1, so that the self-inductance of the inductor Lr1 is much smaller than the self-inductance of the primary winding Np1; in the primary winding of the second transformer (42) Np2 is connected in parallel with inductor Lr2, so that the self-inductance of inductor Lr2 is much smaller than the self-inductance of primary winding Np2.
  • the second derivative circuit is to add an inductor to the position of the primary winding of the transformer (4) and the first transformer (41) and the second transformer (42), and the primary winding is connected in series with a capacitor.
  • connection method of the second derivative circuit is: the two ends of the inductor Lr are respectively connected to the node V1 of the transformer bridge (3) and the node Vs of the capacitor network (2); one end of the primary winding Np of the transformer (4) Node connecting transform bridge (3) The other end of V1 is connected to one end of the capacitor Cr, and the other end of the capacitor Cr is connected to the node V2 of the conversion bridge (3) or the positive terminal Vd or the ground terminal GND, or the node Vs of the capacitor network (2). See Figure 12.
  • connection method of the second derivative circuit is: the two ends of the inductor Lr1 are respectively connected to the node V1 of the transformer bridge (3) and the node Vs1 of the capacitor network (2), and the primary winding of the first transformer (41)
  • One end of Np1 is connected to the node V1 of the conversion bridge (3), and the other end is connected to one end of the capacitor Cr1.
  • the other end of the capacitor Cr1 is connected to the positive terminal Vd or the ground terminal GND of the conversion bridge (3), or to the node Vs1 or the node of the capacitor network (2).
  • Node Vs2; both ends of the inductor Lr2 are respectively connected to the node V2 of the transformer bridge (3) and the node Vs2 of the capacitor network (2).
  • One end of the primary winding Np2 of the second transformer (42) is connected to the nodes V2 and V2 of the transformer bridge (3).
  • the other end is connected to one end of the capacitor Cr2, and the other end of the capacitor Cr2 is connected to the positive terminal Vd or the ground terminal GND of the conversion bridge (3), or to the node Vs1 or node Vs2 of the capacitor network (2).
  • the dual-bridge arm PWM control is further divided into two modes: frequency multiplication modulation and bipolar modulation.
  • the second bridge arm composed of switching tubes Q3 and Q4 switches at a low frequency at the frequency of the sinusoidal AC power supply Ua.
  • the upper switching tube Q4 is turned off and the lower switching tube Q3 is turned on; in the negative half cycle of the sinusoidal AC power supply Ua, that is, When the voltage of node Vs is lower than the voltage of node V2, the lower switch Q3 is turned off and the upper switch Q4 is turned on.
  • the first bridge arm composed of switching tubes Q1 and Q2 transforms at high frequency under complementary PWM control.
  • complementary PWM means that the sum of the duty ratios of switching tubes Q1 and Q2 is equal to 1 when the dead time is ignored.
  • Control law During the positive half cycle of the sinusoidal AC power supply Ua, the output voltage of the converter is proportional to the average value of D 1 ; during the negative half cycle of the sinusoidal AC power supply Ua, the output voltage of the converter is proportional to the average value of D 2 Directly proportional.
  • the instantaneous values of D 1 and D 2 change according to the requirements of power factor correction, so that the input current tracks the sinusoidal AC voltage of the power supply Ua, so that their waveforms are consistent and have the same phase.
  • the control process is as follows:
  • the output variable On the DC side, the output variable is sampled, and the output variable is output voltage, output current or output power;
  • the output variable is passed through a low-pass filter (bandwidth less than 20 Hz) to filter out the second harmonic and high-frequency ripple;
  • the AC voltage sampling signal controls the low-frequency switching of the second bridge arm and serves as the waveform phase reference of the input current
  • the two complementary PWM signals drive the switching tubes Q1 and Q2 of the first bridge arm for high-frequency transformation.
  • the first bridge arm and the second bridge arm of the conversion bridge (3) convert at high frequency under complementary PWM control at the same switching frequency.
  • bipolar modulation When the switching tubes Q3 and Q4 are synchronized with the conduction driving pulses of Q2 and Q1 respectively (that is, in the same phase), it is called bipolar modulation; when the switching tubes Q3 and Q4 are synchronized with the conduction driving pulses of Q1 and Q2 respectively, it is called bipolar modulation.
  • Do frequency doubling modulation There are three alignment methods for frequency-multiply modulated PWM pulses, namely center alignment, leading edge alignment, and trailing edge alignment.
  • Control law The output voltage of the converter is proportional to (D 1 ⁇ D 2 ) min /
  • the instantaneous values of D 1 and D 2 change according to the requirements of power factor correction.
  • the control process is as follows:
  • the output variable On the DC side, the output variable is sampled, and the output variable is output voltage, output current or output power;
  • the output variable is passed through a low-pass filter (bandwidth less than 20 Hz) to filter out the second harmonic and high-frequency ripple;
  • the AC voltage u a and the input current i a are sampled, and the former serves as the waveform phase reference of the input current;
  • the four PWM signals drive the high-frequency transformation of the switch tubes of the first bridge arm and the second bridge arm respectively.
  • the rectifier is synchronized with the driving pulse of the switching tube in the converter bridge, and the conduction duty cycle of the former is slightly smaller than that of the latter. That is to say, the rising edge of the driving pulse of the former is slightly behind the latter, and the falling edge of the driving pulse of the former is slightly ahead of the latter.
  • the lag/advance time must be accurately controlled and kept stable.
  • AC-DC bidirectional conversion For AC-DC bidirectional conversion, it is divided into two working conditions: rectification and inverter. If the electric energy flows from AC to DC, it is a rectifier working condition; if the electric energy flows from DC to AC, it is an inverter working condition.
  • the switching tubes of the rectifier work in the synchronous rectification state, and the requirements for the driving pulses of the rectifier and the converter bridge are the same as the aforementioned synchronous rectification.
  • the roles of the rectifier and the converter bridge are interchanged, that is, the function of the rectifier is “conversion” and the function of the converter bridge is "rectification". Since the rectifier is first connected to the inductor in the filter, the DC side input of the inverter condition is a current source, which requires an overlap time between the conduction drive pulses of the upper and lower tubes of the bridge arm. The rules of overlap time and dead time are opposite. During the dead time, both the upper and lower tubes of the bridge arm are turned off, and during the overlap time, both the upper and lower tubes of the bridge arm are turned on. Therefore, there are new requirements for the drive pulses of the rectifier and converter bridge.
  • the driving pulses of the rectifier and the switching tube in the converter bridge are synchronized, and the conduction duty cycle of the former is slightly larger than that of the latter. That is to say, the rising edge of the driving pulse of the former precedes the latter, and the falling edge of the driving pulse of the former lags behind the latter.
  • the lead/lag time must be accurately controlled and kept stable so that the overlap time of the rectifier arm is equal to (or slightly smaller than) the dead time of the converter bridge arm.
  • the single-transformer solution of this converter adopts single-bridge PWM control, and its working principle is detailed as follows.
  • the voltages between nodes Vs, V1, V2, Vd and GND are set to be u s , u 1 , u 2 , and u d , respectively.
  • u d is called the bus voltage;
  • the voltage between nodes Vs and V2 is recorded as u a , which is the sinusoidal AC voltage on the capacitor Cs after the AC power supply Ua passes through the input filter circuit;
  • the voltage between nodes Vs and V1 is recorded as up , which is the voltage of the primary winding Np of the transformer.
  • U R is the effective value of u a
  • ⁇ is the angular frequency of u a .
  • the second bridge arm composed of switching tubes Q3 and Q4 switches at low frequency at the frequency of sinusoidal AC power supply Ua; the first bridge arm composed of switching tubes Q1 and Q2 switches at high frequency under complementary PWM control.
  • the relationship between u p and u a is:
  • V R of its positive and negative output terminal voltages is:
  • V o can be decomposed into DC components (i.e. average value) and exchange components
  • D m is the minimum representative duty cycle corresponding to the effective value U R , which occurs at the peak moment of u a .
  • the converter must also implement PFC function (Power Factor Correction).
  • PFC Power Factor Correction
  • the so-called PFC means that the alternating current i a tracks the alternating voltage u a so that their waveforms are consistent and in the same phase, thereby achieving a high power factor.
  • the power factor PF PF ⁇ 1.
  • IR is the effective value of the alternating current i a .
  • ⁇ the efficiency of the converter
  • P a and P o are respectively:
  • the DC output voltage V o can be decomposed into the DC component plus communication component in the form of:
  • the double-transformer solution of this converter adopts double-arm frequency-doubled PWM control or double-arm bipolar PWM control. Its working principle is detailed as follows.
  • the voltages between nodes Vs1, Vs2, V1, V2, Vd and GND are recorded as u s1 , u s2 , u 1 , u 2 , and u d respectively.
  • u d is called the bus voltage; the voltage between nodes Vs1 and Vs2 is recorded as u a , u a is the filtered sinusoidal AC voltage of the AC power supply Ua.
  • u p1 The voltage between nodes Vs1 and V1 is denoted as u p1
  • u p1 is the voltage of the primary winding Np1 of the first transformer
  • u p2 the voltage between nodes Vs2 and V2 is denoted as u p2
  • u p2 is the voltage of the primary winding Np1 of the second transformer.
  • the first bridge arm and the second bridge arm convert at high frequency under complementary PWM control at the same switching frequency.
  • V R1 (u s1 ⁇ D 1 +(u d -u s1 ) ⁇ D 2 ) ⁇ n 1 (E-23)
  • rectifier Y also adopts full-bridge rectification topology, see Figure 4. Assume that the transformation ratio between the secondary winding Ns2 and the primary winding Np2 of the second transformer is n 2 .
  • V R2 (u s2 ⁇ D 3 +(u d -u s2 ) ⁇ D 4 ) ⁇ n 2 (E-26)
  • Rectifier X and rectifier Y can be connected to filter X and filter Y respectively to form two groups of outputs, as shown in Figure 6.
  • the two groups of DC outputs can supply the load independently, or be connected in parallel or in series to supply the load.
  • Rectifier X and rectifier Y can also be connected to a filter in combination to form a group of outputs; The formula is divided into parallel combination and series combination, see Figure 7 and Figure 8.
  • U RM is the maximum effective value of the AC voltage u a ;
  • max represent the minimum value of (D 1 ⁇ D 2 ) and the maximum value of
  • Formula (E-34) and Formula (E-35) are values for independent or parallel output. When output in series, the transformation ratio n of equation (E-34) is halved, and the transformation ratio n of equation (E-35) and double.
  • rectifier X and rectifier Y adopt full-wave rectification topology, it can be analyzed similarly.
  • the primary winding of the transformer also functions as a boost inductor.
  • the primary winding current i p has not only an AC component but also a DC component, and its DC component and AC component account for approximately half each.
  • two derivative circuits are proposed.
  • the first derivative circuit is that the primary winding of the transformer is connected in parallel with an inductance, and the self-inductance of the parallel inductance is much smaller than the self-inductance of the primary winding.
  • Figure 11 shows an example of parallel inductance for a single transformer solution. The quantitative relationship of self-inductance is:
  • L r is the self-inductance of the parallel inductor, is the required primary winding self-inductance after connecting the inductor in parallel, and L p is the required primary winding self-inductance when the inductor is not connected in parallel.
  • ⁇ 3 ⁇ 5 can be taken, then the DC component of the primary winding current is reduced to approximately 1/(1+ ⁇ ).
  • the second derivative circuit is to add an inductor to the primary winding of the transformer, and the primary winding is connected in series with the capacitor C r .
  • Figure 12 is an example of the second derivative circuit of the single-transformer solution.
  • the quantitative relationship of capacitance can refer to the following formula:
  • C r is the capacity of the series capacitor
  • Cs is the capacity of the capacitor Cs required when the capacitor is not connected in series.

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Abstract

本发明提出一种无桥隔离型AC-DC单级PFC变换器及其控制方法，属于电力电子/新能源领域。有单变压器和双变压器两种技术方案，前者包含输入滤波电路、电容网络、变换桥、变压器、整流器和滤波器，后者在前者基础上增加变压器和整流器各一个，且有多种输出方式。对应单桥臂PWM和双桥臂PWM两种控制方法，后者有倍频和双极性两种调制模式。为降低成本，单变压器方案的变换桥低频桥臂采用二极管。为双向变换或同步整流，则整流器采用开关管。为减小变压器原边绕组的直流电流，提出两种衍生电路。优越性：①隔离型单级变换，成本低；②省去工频整流桥，效率高；③变换功率大，输出电压宽；④适于电压源和电流源模式；⑤控制方法灵活可靠。

Description

无桥隔离型AC-DC单级PFC变换器及其控制方法 技术领域
本发明涉及一种无桥隔离型AC-DC单级PFC变换器及其控制方法，是一种电能高效变换/开关电源技术，属于电力电子/新能源技术领域。
背景技术
单级PFC(Power Factor Correction)变换器，是一种交流输入/直流输出的开关变换器；其功能是仅由一次功率变换即可稳定所需的直流输出量，同时实现交流侧的功率因数校正。所谓功率因数校正(PFC)，即是输入的交流电流跟踪交流电压，使它们波形一致相位相同，从而达到高功率因数。
目前，PFC(Power Factor Correction)变换器一般为非隔离型，常用的有Boost(升压)拓扑、Buck(降压)拓扑、升降压拓扑以及图腾柱拓扑。而隔离型的单级PFC变换器，大都采用反激式拓扑，也有半桥式等拓扑。反激式拓扑属于单端变换，所传输的功率不能太大，一般适合于小功率应用。半桥式等拓扑虽然能够应用于中大功率，但是，由于工频整流桥的存在，导致大功率应用时整流损耗较大。
上述内容仅用于辅助理解本发明的技术方案，并不表示上述都是现有技术。
发明内容
本发明的目的是，克服上述现有技术的不足，提出一种无桥隔离型AC-DC单级PFC变换器及其控制方法。该变换器是一种隔离型单级PFC变换器，输入与输出隔离，稳定直流输出且实现高功率因数。该变换器是一种无工频整流桥的AC-DC电路，采用全桥拓扑，属于双端变换，能传输大功率，并且提高效率降低成本。该变换器包括单变压器和双变压器两种技术方案，相应的控制方法也分为两种：一种是单桥臂PWM控制，对应于单变压器方案；另一种是双桥臂PWM控制，对应于双变压器方案。
本发明的技术方案如下。
一种无桥隔离型AC-DC单级PFC变换器，有两种技术方案；第一种是单变压器方案，第二种是双变压器方案。两种技术方案具有相同的输入滤波电路和变换桥。
输入滤波电路含有共模与差模电感和X电容、Y电容，具有两个交流输出端。
变换桥含有开关管Q1、Q2、Q3、Q4，为全桥拓扑具有两个桥臂，开关管Q1和Q2构成第一桥臂，开关管Q3和Q4构成第二桥臂。开关管Q1漏极和Q2源极相连作为节点 V1，开关管Q3漏极和Q4源极相连作为节点V2；开关管Q2和Q4漏极相连作为变换桥的正端Vd，开关管Q1和Q3源极相连作为变换桥的地端GND。
单变压器方案包含输入滤波电路、电容网络、变换桥、变压器、整流器和滤波器，它们依次连接。交流电源Ua连接输入滤波电路，输入滤波电路的两个交流输出端提供稳定的交流电压；滤波器将滤除高频纹波的直流电压作为变换器的输出而供给负载。
电容网络包括电容Cs、Cd、C1、C2，它们都采用薄膜电容而非电解电容。电容Cs两端分别连接输入滤波电路的两个交流输出端，电容Cd两端分别连接变换桥的正端Vd和地端GND；电容C1和C2的一端连接输入滤波电路的一个交流输出端而作为节点Vs，电容C1另一端连接变换桥的地端GND，电容C2另一端连接变换桥的正端Vd，变换桥的节点V2连接输入滤波电路的另一个交流输出端。或者去掉电容C1和C2，或者去掉电容Cs，或者去掉电容Cd，以简化电路降低成本，但各个电容的纹波电流会发生变化。
变压器具有原边绕组Np和至少一个副边绕组Ns。原边绕组Np具有两端，副边绕组Ns具有两端和中间抽头、或者去掉中间抽头。
原边绕组Np一端连接电容网络的节点Vs、另一端连接变换桥的节点V1。副边绕组Ns两端分别连接整流器的两个交流输入端；当整流器采用全波整流拓扑时，Ns的中间抽头作为整流器的正输出端或者负输出端；当整流器采用全桥整流拓扑时，则去掉Ns的中间抽头。
整流器采用全波整流拓扑或者全桥整流拓扑，具有正输出端、负输出端和两个交流输入端。
全波整流拓扑包括二极管D1、D2，其连接方式有两种。第一种连接方式称作共阴极连接，即是二极管D1、D2的阴极连接在一起作为整流器的正输出端，二极管D1的阳极和D2的阳极作为整流器的两个交流输入端，变压器副边绕组Ns的中间抽头作为整流器的负输出端。第二种连接方式称作共阳极连接，即是二极管D1、D2的阳极连接在一起作为整流器的负输出端，二极管D1的阴极和D2的阴极作为整流器的两个交流输入端，变压器副边绕组Ns的中间抽头作为整流器的正输出端。
全桥整流拓扑包括二极管D1、D2、D3、D4，二极管D1的阳极与D2的阴极相连作为整流器的一个交流输入端，二极管D3的阳极与D4的阴极相连作为整流器的另一个交流输入端，二极管D1、D3的阴极相连作为整流器的正输出端，二极管D2、D4的阳极相连作为整流器的负输出端。
滤波器包括滤波电感Lo1、Lo2和滤波电容Co，构成四端网络，具有正输入端、负输入端和正输出端、负输出端。滤波电感Lo1的两端分别作为滤波器的正输入端和正输出端，滤波电感Lo2的两端分别作为滤波器的负输入端和负输出端；滤波电容Co的正极和负极分别连接滤波器的正输出端和负输出端。滤波电感Lo1和Lo2独立或者耦合，或者去掉 Lo1或者去掉Lo2，以简化电路；当去掉Lo1时则滤波器的正输入端和正输出端直接相连，当去掉Lo2时则滤波器的负输入端和负输出端直接相连。或者去掉滤波电容Co，以适用于电流源模式输出。当保留Co时，则为电压源模式输出。
滤波器的正输入端和负输入端分别连接整流器的正输出端和负输出端，滤波器的正输出端和负输出端分别作为该变换器的输出正端+Vo和输出负端-Vo。
对于单变压器方案，为了降低成本(但损耗增加)，能够用二极管D01、D02分别替换变换桥的开关管Q3、Q4，或者用单向晶闸管S1、S2分别替换变换桥的开关管Q3、Q4；替换规则为，二极管或者单向晶闸管的阳极和阴极分别对应开关管的源极和漏极。同时保留电容网络的电容C1和C2，电容C1和C2的一端共同连接节点Vs或者变换桥的节点V2，电容C1和C2的另一端分别连接变换桥的地端GND和正端Vd。
双变压器方案包含输入滤波电路、电容网络、变换桥、第一变压器、第二变压器、整流器X、整流器Y、一个滤波器或者两个滤波器，即滤波器X和滤波器Y。交流电源Ua连接输入滤波电路，输入滤波电路的两个交流输出端提供稳定的交流电压。整流器X和整流器Y分别连接滤波器X和滤波器Y，形成两组输出；或者，整流器X与整流器Y以组合方式共同连接一个滤波器，形成一组输出。
电容网络包含电容Cs、Cd、C1、C2、C3、C4，它们都采用薄膜电容而非电解电容。电容Cs的两端分别连接输入滤波电路的两个交流输出端，电容Cd的两端分别连接变换桥的正端Vd和地端GND；电容C1和C3的一端连接输入滤波电路的一个交流输出端而作为节点Vs1，电容C2和C4的一端连接输入滤波电路的另一个交流输出端而作为节点Vs2，电容C1和C2的另一端连接变换桥的地端GND，电容C3和C4的另一端连接变换桥的正端Vd。或者去掉C1、C2、C3、C4之中的任意两个，或者去掉Cs或者去掉Cd，或者去掉Cs和Cd，以简化电路降低成本，但是电路对称性降低，并且各个电容的纹波电流将发生变化。
第一变压器具有原边绕组Np1和至少一个副边绕组Ns1；原边绕组Np1具有两端，副边绕组Ns1具有两端和中间抽头、或者去掉中间抽头。第二变压器具有原边绕组Np2和至少一个副边绕组Ns2；原边绕组Np2具有两端，副边绕组Ns2具有两端和中间抽头、或者去掉中间抽头。
原边绕组Np1两端分别连接电容网络的节点Vs1和变换桥的节点V1，原边绕组Np2两端分别连接电容网络的节点Vs2和变换桥的节点V2。副边绕组Ns1两端分别连接整流器X的两个交流输入端；当整流器X采用全波整流拓扑时，Ns1的中间抽头作为整流器X的正输出端或者负输出端；当整流器X采用全桥整流拓扑时，则去掉Ns1的中间抽头。副边绕组Ns2两端分别连接整流器Y的两个交流输入端；当整流器Y采用全波整流拓扑时，Ns2的中间抽头作为整流器Y的正输出端或者负输出端；当整流器Y采用全桥整流拓扑时，则去掉Ns2的中间抽头。
整流器X和整流器Y采用全波整流拓扑或者全桥整流拓扑，具有正输出端、负输出端和两个交流输入端。其内部二极管的连接方式与单变压器方案中所述的整流器一样，是它的两个复制。
单个滤波器和滤波器X、滤波器Y均为四端网络，具有正输入端、负输入端和正输出端、负输出端。其内部网络结构与单变压器方案中所述的滤波器一样。
整流器X连接滤波器X而整流器Y连接滤波器Y，构成两组输出，即第一输出和第二输出。具体连接方式为：整流器X的正输出端和负输出端分别连接滤波器X的正输入端和负输入端，滤波器X的正输出端和负输出端分别作为该变换器第一输出的正端+Vo1和负端-Vo1。整流器Y的正输出端和负输出端分别连接滤波器Y的正输入端和负输入端，滤波器Y的正输出端和负输出端分别作为该变换器第二输出的正端+Vo2和负端-Vo2。第一输出和第二输出能够独立供给负载，或者并联或者串联供给负载。
或者，整流器X与整流器Y以组合方式共同连接一个滤波器，构成一组输出，滤波器的正输出端和负输出端分别作为该变换器输出的正端+Vo和负端-Vo。组合方式分为并联组合与串联组合。并联组合为：整流器X和整流器Y的正输出端共同连接滤波器的正输入端，整流器X和整流器Y的负输出端共同连接滤波器的负输入端；串联组合为：整流器X负输出端连接整流器Y正输出端，整流器X正输出端和整流器Y负输出端分别连接滤波器的正输入端和负输入端。
为了实现AC-DC双向变换或者同步整流，则用开关管替换整流器和整流器X、整流器Y中的二极管；替换规则为，开关管的源极和漏极分别对应二极管的阳极和阴极。所谓双向变换，是指电能在交流侧与直流侧之间双向流动的开关变换。
为了减小单变压器方案中的变压器和双变压器方案中的第一变压器、第二变压器之原边绕组的直流电流，提出了两种衍生电路。
第一种衍生电路是，变压器和第一变压器、第二变压器的原边绕组并联电感，所并联电感的自感量远小于原边绕组的自感量。
第二种衍生电路是，在变压器和第一变压器、第二变压器之原边绕组的位置加入电感，而原边绕组串联电容。
基于单变压器方案，第二种衍生电路的连接方式为：电感Lr的两端分别连接变换桥的节点V1和电容网络的节点Vs；变压器的原边绕组Np一端连接变换桥的节点V1、另一端连接电容Cr的一端，电容Cr另一端连接变换桥的节点V2或正端Vd或地端GND、或者连接电容网络的节点Vs。
基于双变压器方案，第二种衍生电路的连接方式为：电感Lr1的两端分别连接变换桥的节点V1和电容网络的节点Vs1，第一变压器的原边绕组Np1一端连接变换桥的节点V1、另一端连接电容Cr1的一端，电容Cr1另一端连接变换桥的正端Vd或地端GND、或者连 接电容网络的节点Vs1或节点Vs2；电感Lr2的两端分别连接变换桥的节点V2和电容网络的节点Vs2，第二变压器的原边绕组Np2一端连接变换桥的节点V2、另一端连接电容Cr2的一端，电容Cr2另一端连接变换桥的正端Vd或地端GND、或者连接电容网络的节点Vs1或节点Vs2。
无桥隔离型AC-DC单级PFC变换器的控制方法，有两种。第一种是单桥臂PWM控制，适用于所述的单变压器方案；第二种是双桥臂PWM控制，适用于所述的双变压器方案，双桥臂PWM控制又细分为倍频调制和双极调制两种模式。
单桥臂PWM控制方法
由开关管Q3和Q4构成的第二桥臂，以交流电源Ua的频率低频切换。在交流电源Ua的正半周期，即节点Vs的电压高于节点V2的电压，则上部的开关管Q4关断而下部的开关管Q3导通；在交流电源Ua的负半周期，即节点Vs的电压低于节点V2的电压，则下部的开关管Q3关断而上部的开关管Q4导通。
由开关管Q1和Q2构成的第一桥臂，在互补PWM控制下高频变换。所谓互补PWM是指，在忽略死区时间的情况下开关管Q1和Q2的占空比之和等于1。设定开关管Q1和Q2的导通占空比分别为D1和D2，则D1+D2＝1。
控制规律：在交流电源Ua的正半周期，该变换器的输出电压与D1的平均值成正比；在交流电源Ua的负半周期，该变换器的输出电压与D2的平均值成正比。而D1和D2的瞬时值则按照功率因数校正的要求而变化，实现输入电流跟踪电源Ua的正弦交流电压，使它们波形一致相位相同。
控制流程如下：
在直流侧，取样输出变量，输出变量为输出电压、输出电流或者输出功率；
将输出变量经过(带宽小于20Hz)低通滤波器，滤除二次谐波和高频纹波；
将滤波后的输出变量与给定值进行比较，得到输出变量误差；
将输出变量误差进行PID或PI调整，得到反馈变量；
将反馈变量隔离后负反馈到交流侧。
在交流侧，取样交流电压和输入电流；
交流电压取样信号控制第二桥臂低频切换，并且作为输入电流的波形相位基准；
将交流电压取样信号与来自输出侧的反馈变量进行运算，得到输入电流的基准值；
将该基准值与输入电流取样信号进行比较，得到输入电流误差；
将输入电流误差进行PID或PI调整，得到控制量；
将该控制量与三角波进行比较，产生脉宽调制信号；
再插入死区时间，形成两路互补的PWM信号；
该两路互补PWM信号驱动第一桥臂的开关管Q1和Q2高频变换。
至此，完成双闭环反馈控制，从而稳定调节输出变量并且实现功率因数校正。
双桥臂PWM控制方法
变换桥的第一桥臂和第二桥臂，以相同的开关频率在互补PWM控制下高频变换。设定开关管Q1、Q2和Q3、Q4的导通占空比分别为D1、D2和D3、D4。忽略开关变换的死区时间，则D1+D2＝1＝D3+D4；并且要求D4＝D1和D3＝D2。
双极调制模式为，开关管Q3、Q4分别与开关管Q2、Q1的导通驱动脉冲同步(即同相位)；倍频调制模式为，开关管Q3、Q4分别与开关管Q1、Q2的导通驱动脉冲同步。倍频调制模式的PWM脉冲有三种对齐方式：中心对齐、前沿对齐、后沿对齐。
控制规律：该变换器的输出电压与(D1·D2)min/|D1-D2|max成正比，(D1·D2)min表示(D1·D2)的最小值，|D1-D2|max表示|D1-D2|的最大值，它们发生在交流电压ua的正负峰值时刻。而D1和D2的瞬时值则按照功率因数校正的要求而变化。
控制流程如下：
在直流侧，取样输出变量，输出变量为输出电压、输出电流或者输出功率；
将输出变量经过(带宽小于20Hz)低通滤波器，滤除二次谐波和高频纹波；
将滤波后的输出变量与给定值进行比较，得到输出变量误差；
将输出变量误差进行PID或PI调整，得到反馈变量；
将反馈变量隔离后负反馈到交流侧。
在交流侧，取样交流电压和输入电流，前者作为输入电流的波形相位基准；
将交流电压取样信号与来自输出侧的反馈变量进行运算，得到输入电流的基准值；
将该基准值与输入电流取样信号进行比较，得到输入电流误差；
将输入电流误差进行PID或PI调整，得到控制量；
将该控制量与三角波进行比较，产生脉宽调制信号；
再插入死区时间，扩充移相而形成两两互补的四路PWM信号；
该四路PWM信号，分别驱动第一桥臂和第二桥臂的开关管高频变换。
至此，完成双闭环反馈控制，从而稳定调节输出变量并且实现功率因数校正。
本发明与现有技术相比具有如下优越性。
1)本发明为隔离型单级PFC变换，采用全桥拓扑，属于双端变换，能够传输大功率。
2)本发明分为单变压器和双变压器两种方案，后者可以多种组合方式输出更大功率。
3)本发明拓扑精简，省去工频整流桥，降低成本、提高效率、增强可靠性。
4)本发明有很宽的输出电压范围，既适于电压源模式，又适于电流源模式。
5)本发明可以采用单桥臂PWM、双桥臂倍频或双极PWM等多种控制方法。
附图说明
图1为无桥隔离型AC-DC单级PFC变换器之单变压器方案的原理图。
图2为整流器采用全波整流拓扑共阴极连接的原理图。
图3为整流器采用全波整流拓扑共阳极连接的原理图。
图4为整流器采用全桥整流拓扑的原理图。
图5为滤波器的网络拓扑原理图。
图6为无桥隔离型AC-DC单级PFC变换器之双变压器方案两组输出的原理图。
图7为无桥隔离型AC-DC单级PFC变换器之双变压器方案并联输出的原理图。
图8为无桥隔离型AC-DC单级PFC变换器之双变压器方案串联输出的原理图。
图9为单变压器方案之变换桥的开关管被二极管替换的原理图。
图10为单变压器方案之整流器的二极管被开关管替换的原理图。
图11为单变压器方案的第一种衍生电路原理图。
图12为单变压器方案的第二种衍生电路原理图。
附图标号说明：
具体实施方式
下面将结合附图，以优选实施例，对本发明进行详细地描述与分析。所描述的实施例仅仅是本发明的一部分实施例而非全部。
1、本发明的优选实施例
一种无桥隔离型AC-DC单级PFC变换器，有两种技术方案。第一种是单变压器方案，见图1；第二种是双变压器方案，见图6、图7、图8。两种技术方案具有相同的输入滤波电路(1)和变换桥(3)。
输入滤波电路(1)含有共模与差模电感和X电容、Y电容，具有两个交流输出端。
变换桥(3)含有开关管Q1、Q2、Q3、Q4，为全桥拓扑具有两个桥臂，开关管Q1和Q2构成第一桥臂，开关管Q3和Q4构成第二桥臂。开关管Q1漏极和Q2源极相连作为节点V1，开关管Q3漏极和Q4源极相连作为节点V2；开关管Q2和Q4漏极相连作为变换桥(3)的正端Vd，开关管Q1和Q3源极相连作为变换桥(3)的地端GND。开关管采用但不限于MOSFET或者IGBT。
如图1所示，单变压器方案包含输入滤波电路(1)、电容网络(2)、变换桥(3)、变压器(4)、整流器(5)和滤波器(6)，它们依次连接。交流电源Ua连接输入滤波电路(1)，输入滤波电路(1)的两个交流输出端提供稳定的交流电压；滤波器(6)将滤除高频纹波的直流电压作为变换器的输出而供给负载。
电容网络(2)包括电容Cs、Cd、C1、C2，它们都采用容量较小的薄膜电容而非大容量的电解电容，尤其是Cd。电容Cs的两端分别连接输入滤波电路(1)的两个交流输出端，电容Cd的两端分别连接变换桥(3)的正端Vd和地端GND；电容C1和C2的一端连接输入滤波电路(1)的一个交流输出端而作为节点Vs，电容C1另一端连接变换桥(3)的地端GND，电容C2另一端连接变换桥(3)的正端Vd，变换桥(3)的节点V2连接输入滤波电路(1)的另一个交流输出端。或者去掉电容C1和C2，或者去掉电容Cs，或者去掉电容Cd，以简化电路降低成本，但各个电容的纹波电流会发生变化。
变压器(4)具有原边绕组Np和至少一个副边绕组Ns。原边绕组Np具有两端，副边绕组Ns具有两端和中间抽头、或者去掉中间抽头。
原边绕组Np一端连接电容网络(2)的节点Vs、另一端连接变换桥(3)的节点V1。副边绕组Ns两端分别连接整流器(5)的两个交流输入端；当整流器(5)采用全波整流拓扑时，Ns的中间抽头作为整流器(5)的正输出端或者负输出端；当整流器(5)采用全桥整流拓扑时，则去掉Ns的中间抽头。
整流器(5)采用全波整流拓扑或者全桥整流拓扑，具有正输出端、负输出端和两个交流输入端。如图2、图3、图4所示。
全波整流拓扑包括二极管D1、D2，其连接方式有两种。第一种称作共阴极连接(见图2)，即是二极管D1、D2的阴极连接在一起作为整流器(5)的正输出端，二极管D1的阳极和D2的阳极作为整流器(5)的两个交流输入端，变压器(4)副边绕组Ns的中间抽头作为整流器(5)的负输出端。第二种称作共阳极连接(见图3)，即是二极管D1、D2的阳极连接在一起作为整流器(5)的负输出端，二极管D1的阴极和D2的阴极作为整流器(5)的两个交流输入端，变压器(4)副边绕组Ns的中间抽头作为整流器(5)的正输出端。
全桥整流拓扑(见图4)包括二极管D1、D2、D3、D4，二极管D1的阳极与D2的阴极相连作为整流器(5)的一个交流输入端，二极管D3的阳极与D4的阴极相连作为整流器(5)的另一个交流输入端，二极管D1、D3的阴极相连作为整流器(5)的正输出端，二极管D2、D4的阳极相连作为整流器(5)的负输出端。
滤波器(6)包括滤波电感Lo1、Lo2和滤波电容Co，构成四端网络，具有正输入端、负输入端和正输出端、负输出端，如图5所示。滤波电感Lo1的两端分别作为滤波器(6)的正输入端和正输出端，滤波电感Lo2的两端分别作为滤波器(6)的负输入端和负输出端；滤波电容Co的正极和负极分别连接滤波器(6)的正输出端和负输出端。滤波电感Lo1和Lo2独立或者耦合，或者去掉Lo1或者去掉Lo2，以简化电路；当去掉Lo1时则滤波器(6)的正输入端和正输出端直接相连，当去掉Lo2时则滤波器(6)的负输入端和负输出端直接相连。或者去掉滤波电容Co，以适于电流源模式输出。当保留Co时，则为电压源模式输出。
滤波器(6)的正输入端和负输入端分别连接整流器(5)的正输出端和负输出端，滤波器(6)的正输出端和负输出端分别作为该变换器的输出正端+Vo和输出负端-Vo。
对于单变压器方案，为了降低成本(但损耗增加)，则用二极管D01、D02分别替换变换桥(3)的开关管Q3、Q4(见图9)，或者用单向晶闸管S1、S2分别替换变换桥(3)的开关管Q3、Q4；替换规则为，二极管或者单向晶闸管的阳极和阴极分别对应开关管的源极和漏极。同时保留电容网络(2)的电容C1和C2，电容C1和C2的一端共同连接节点Vs或者节点V2，电容C1和C2的另一端分别连接地端GND和正端Vd。补充说明：二极管为不控元件，单向晶闸管为半控且低频元件，而开关管为全控且高频元件。
如图6、图7、图8所示，双变压器方案包含输入滤波电路(1)、电容网络(2)、变换桥(3)、第一变压器(41)、第二变压器(42)、整流器X(51)、整流器Y(52)、滤波器(6)或者两个滤波器，即滤波器X(61)和滤波器Y(62)。交流电源Ua连接输入滤波电路(1)，输入滤波电路(1)的两个交流输出端提供稳定的交流电压。整流器X(51)和整流器Y(52)分别连接滤波器X(61)和滤波器Y(62)，形成两组输出；或者，整流器X(51)与整流器Y(52)以组合方式共同连接滤波器(6)，形成一组输出。
电容网络(2)包含电容Cs、Cd、C1、C2、C3、C4，它们都采用容量较小的薄膜电容而非大容量的电解电容，尤其是Cd。电容Cs的两端分别连接输入滤波电路(1)的两个交流输出端，电容Cd的两端分别连接变换桥(3)的正端Vd和地端GND；电容C1和C3的一端连 接输入滤波电路(1)的一个交流输出端而作为节点Vs1，电容C2和C4的一端连接输入滤波电路(1)的另一个交流输出端而作为节点Vs2，电容C1和C2的另一端连接变换桥(3)的地端GND，电容C3和C4的另一端连接变换桥(3)的正端Vd。或者去掉C1、C2、C3、C4之中的任意两个，或者去掉Cs或者去掉Cd，或者去掉Cs和Cd，以简化电路降低成本，但电路对称性降低，且各个电容的纹波电流将发生变化。
第一变压器(41)具有原边绕组Np1和至少一个副边绕组Ns1；原边绕组Np1具有两端，副边绕组Ns1具有两端和中间抽头、或者去掉中间抽头。第二变压器(42)具有原边绕组Np2和至少一个副边绕组Ns2；原边绕组Np2具有两端，副边绕组Ns2具有两端和中间抽头、或者去掉中间抽头。
原边绕组Np1两端分别连接电容网络(2)的节点Vs1和变换桥(3)的节点V1，原边绕组Np2两端分别连接电容网络(2)的节点Vs2和变换桥(3)的节点V2。副边绕组Ns1两端分别连接整流器X(51)的两个交流输入端；当整流器X(51)采用全波整流拓扑时，Ns1的中间抽头作为整流器X(51)的正输出端或者负输出端；当整流器X(51)采用全桥整流拓扑时，则去掉Ns1的中间抽头。副边绕组Ns2两端分别连接整流器Y(52)的两个交流输入端；当整流器Y(52)采用全波整流拓扑时，Ns2的中间抽头作为整流器Y(52)的正输出端或者负输出端；当整流器Y(52)采用全桥整流拓扑时，则去掉Ns2的中间抽头。
整流器X(51)和整流器Y(52)采用全波整流拓扑或者全桥整流拓扑，具有正输出端、负输出端和两个交流输入端。
全波整流拓扑包括二极管D1、D2，其连接方式有两种；第一种连接方式称作共阴极连接，即是二极管D1、D2的阴极连接在一起作为整流器的正输出端，二极管D1的阳极和D2的阳极作为整流器的两个交流输入端，第一变压器(41)副边绕组Ns1的中间抽头作为整流器X(51)的负输出端，第二变压器(42)副边绕组Ns2的中间抽头作为整流器Y(52)的负输出端；第二种连接方式称作共阳极连接，即是二极管D1、D2的阳极连接在一起作为整流器的负输出端，二极管D1的阴极和D2的阴极作为整流器的两个交流输入端，第一变压器(41)副边绕组Ns1的中间抽头作为整流器X(51)的正输出端，第二变压器(42)副边绕组Ns2的中间抽头作为整流器Y(52)的正输出端。
全桥整流拓扑包括二极管D1、D2、D3、D4，二极管D1的阳极与D2的阴极相连作为整流器的一个交流输入端，二极管D3的阳极与D4的阴极相连作为整流器的另一个交流输入端，二极管D1、D3的阴极相连作为整流器的正输出端，二极管D2、D4的阳极相连作为整流器的负输出端。
滤波器X(61)、滤波器Y(62)和滤波器(6)一样，均为四端网络，具有正输入端、负输入端和正输出端、负输出端，包括滤波电感Lo1、Lo2和滤波电容Co。滤波电感Lo1的两端分别作为滤波器的正输入端和正输出端，滤波电感Lo2的两端分别作为滤波器的负输入端和负输出端；滤波电容Co的正极和负极分别连接滤波器的正输出端和负输出端。滤波 电感Lo1和Lo2独立或者耦合，或者去掉Lo1或者去掉Lo2，以简化电路；当去掉Lo1时则滤波器的正输入端和正输出端直接相连，当去掉Lo2时则滤波器的负输入端和负输出端直接相连。或者去掉滤波电容Co，以适用于电流源模式输出。
整流器X(51)和整流器Y(52)分别连接滤波器X(61)和滤波器Y(62)，构成两组输出，即整流器X(51)连接滤波器X(61)构成该变换器的第一输出，整流器Y(52)连接滤波器Y(62)构成该变换器的第二输出，见图6。具体说明：整流器X(51)的正输出端和负输出端分别连接滤波器X(61)的正输入端和负输入端，滤波器X(61)的正输出端和负输出端分别作为第一输出的正端+Vo1和负端-Vo1。整流器Y(52)的正输出端和负输出端分别连接滤波器Y(62)的正输入端和负输入端，滤波器Y(62)的正输出端和负输出端分别作为第二输出的正端+Vo2和负端-Vo2。第一输出和第二输出能够独立或者并联或者串联供给负载。
或者，整流器X(51)与整流器Y(52)以组合方式共同连接滤波器(6)，构成一组输出，滤波器(6)的正输出端和负输出端分别作为该变换器输出的正端+Vo和负端-Vo。组合方式分为并联组合与串联组合。并联组合(见图7)为：整流器X(51)和整流器Y(52)的正输出端共同连接滤波器(6)的正输入端，整流器X(51)和整流器Y(52)的负输出端共同连接滤波器(6)的负输入端；串联组合(见图8)为：整流器X(51)的负输出端连接整流器Y(52)的正输出端，整流器X(51)的正输出端连接滤波器(6)的正输入端，整流器Y(52)的负输出端连接滤波器(6)的负输入端。
为了实现AC-DC双向变换或者同步整流，则用开关管替换整流器(5)和整流器X(51)、整流器Y(52)中的二极管，开关管采用但不限于MOSFET或者IGBT；替换规则为，开关管的源极和漏极分别对应二极管的阳极和阴极。单变压器方案的实施例见图10。
为了减小变压器(4)和第一变压器(41)、第二变压器(42)之原边绕组的直流电流，提出了两种衍生电路。
第一种衍生电路是，变压器(4)和第一变压器(41)、第二变压器(42)的原边绕组并联电感，所并联电感的自感量远小于原边绕组的自感量。
基于单变压器方案，在变压器(4)的原边绕组Np并联电感Lr，使电感Lr的自感量远小于原边绕组Np的自感量。见图11。
基于双变压器方案，在第一变压器(41)的原边绕组Np1并联电感Lr1，使电感Lr1的自感量远小于原边绕组Np1的自感量；在第二变压器(42)的原边绕组Np2并联电感Lr2，使电感Lr2的自感量远小于原边绕组Np2的自感量。
第二种衍生电路是，在变压器(4)和第一变压器(41)、第二变压器(42)之原边绕组的位置加入电感，而原边绕组串联电容。
基于单变压器方案，第二种衍生电路的连接方式为：电感Lr的两端分别连接变换桥(3)的节点V1和电容网络(2)的节点Vs；变压器(4)的原边绕组Np一端连接变换桥(3)的节点 V1、另一端连接电容Cr的一端，电容Cr另一端连接变换桥(3)的节点V2或正端Vd或地端GND、或者连接电容网络(2)的节点Vs。见图12。
基于双变压器方案，第二种衍生电路的连接方式为：电感Lr1的两端分别连接变换桥(3)的节点V1和电容网络(2)的节点Vs1，第一变压器(41)的原边绕组Np1一端连接变换桥(3)的节点V1、另一端连接电容Cr1的一端，电容Cr1另一端连接变换桥(3)的正端Vd或地端GND、或者连接电容网络(2)的节点Vs1或节点Vs2；电感Lr2的两端分别连接变换桥(3)的节点V2和电容网络(2)的节点Vs2，第二变压器(42)的原边绕组Np2一端连接变换桥(3)的节点V2、另一端连接电容Cr2的一端，电容Cr2另一端连接变换桥(3)的正端Vd或地端GND、或者连接电容网络(2)的节点Vs1或节点Vs2。
2、本发明的控制方法
无桥隔离型AC-DC单级PFC变换器的控制方法，有两种。第一种是单桥臂PWM控制，适用于所述的单变压器方案；第二种是双桥臂PWM控制，适用于所述的双变压器方案，双桥臂PWM控制又细分为倍频调制和双极调制两种模式。
单桥臂PWM控制方法
由开关管Q3和Q4构成的第二桥臂，以正弦交流电源Ua的频率低频切换。在正弦交流电源Ua的正半周期，即节点Vs的电压高于节点V2的电压，则上部的开关管Q4关断而下部的开关管Q3导通；在正弦交流电源Ua的负半周期，即节点Vs的电压低于节点V2的电压，则下部的开关管Q3关断而上部的开关管Q4导通。
由开关管Q1和Q2构成的第一桥臂，在互补PWM控制下高频变换。所谓互补PWM是指，在忽略死区时间的情况下开关管Q1和Q2的占空比之和等于1。设定开关管Q1和Q2的导通占空比分别为D1和D2，则D1+D2＝1。
控制规律：在正弦交流电源Ua的正半周期，该变换器的输出电压与D1的平均值成正比；在正弦交流电源Ua的负半周期，该变换器的输出电压与D2的平均值成正比。而D1和D2的瞬时值则按照功率因数校正的要求而变化，实现输入电流跟踪电源Ua的正弦交流电压，使它们波形一致相位相同。
控制流程如下：
在直流侧，取样输出变量，输出变量为输出电压、输出电流或者输出功率；
将输出变量经过(带宽小于20Hz)低通滤波器，滤除二次谐波和高频纹波；
将滤波后的输出变量与给定值进行比较，得到输出变量误差；
将输出变量误差进行PID或PI调整，得到反馈变量；
将反馈变量隔离后负反馈到交流侧。
在交流侧，取样交流电压ua和输入电流ia；
交流电压取样信号控制第二桥臂低频切换，并且作为输入电流的波形相位基准；
将交流电压取样信号与来自输出侧的反馈变量进行运算，得到输入电流的基准值；
将该基准值与输入电流取样信号进行比较，得到输入电流误差；
将输入电流误差进行PID或PI调整，得到控制量；
将该控制量与三角波进行比较，产生脉宽调制信号；
再插入死区时间，形成两路互补的PWM信号；
该两路互补PWM信号驱动第一桥臂的开关管Q1和Q2高频变换。
至此，完成双闭环反馈控制，从而稳定调节输出变量并且实现功率因数校正。
双桥臂PWM控制方法
变换桥(3)的第一桥臂和第二桥臂，以相同的开关频率在互补PWM控制下高频变换。设定开关管Q1、Q2和Q3、Q4的导通占空比分别为D1、D2和D3、D4。忽略开关变换的死区时间，则D1+D2＝1＝D3+D4；并且要求D4＝D1和D3＝D2。
当开关管Q3、Q4分别与Q2、Q1的导通驱动脉冲同步(即同相位)时，称作双极调制；当开关管Q3、Q4分别与Q1、Q2的导通驱动脉冲同步时，称作倍频调制。倍频调制的PWM脉冲有三种对齐方式，分别是中心对齐、前沿对齐、后沿对齐。
控制规律：该变换器的输出电压与(D1·D2)min/|D1-D2|max成正比，(D1·D2)min表示(D1·D2)的最小值，|D1-D2|max表示|D1-D2|的最大值，它们发生在交流电压ua的正负峰值时刻。而D1和D2的瞬时值则按照功率因数校正的要求而变化。
控制流程如下：
在直流侧，取样输出变量，输出变量为输出电压、输出电流或者输出功率；
将输出变量经过(带宽小于20Hz)低通滤波器，滤除二次谐波和高频纹波；
将滤波后的输出变量与给定值进行比较，得到输出变量误差；
将输出变量误差进行PID或PI调整，得到反馈变量；
将反馈变量隔离后负反馈到交流侧。
在交流侧，取样交流电压ua和输入电流ia，前者作为输入电流的波形相位基准；
将交流电压取样信号与来自输出侧的反馈变量进行运算，得到输入电流的基准值；
将该基准值与输入电流取样信号进行比较，得到输入电流误差；
将输入电流误差进行PID或PI调整，得到控制量；
将该控制量与三角波进行比较，产生脉宽调制信号；
再插入死区时间，扩充移相而形成两两互补的四路PWM信号；
该四路PWM信号，分别驱动第一桥臂和第二桥臂的开关管高频变换。
至此，完成双闭环反馈控制，从而稳定调节输出变量并且实现功率因数校正。
对于同步整流，要求整流器与变换桥中开关管的驱动脉冲同步，并且前者的导通占空比略小于后者。也就是说，前者之驱动脉冲的上升沿稍滞后于后者，前者之驱动脉冲的下降沿略超前于后者，滞后/超前的时间要精确控制并保持稳定。
对于AC-DC双向变换，分为整流和逆变两种工况。若电能从AC流向DC，则为整流工况；若电能从DC流向AC，则为逆变工况。
在整流工况下，整流器的开关管工作于同步整流状态，整流器与变换桥之驱动脉冲的要求与前述同步整流一样。
在逆变工况下，整流器与变换桥的角色产生了互换，即整流器的功能是“变换”，变换桥的功能为“整流”。由于整流器首先连接的是滤波器中的电感，因此逆变工况的直流侧输入为电流源，需要桥臂上下管的导通驱动脉冲有重叠时间。重叠时间与死区时间的规则相反，死区时间内桥臂上下管都关断，重叠时间内桥臂上下管均导通。所以，整流器和变换桥的驱动脉冲有新的要求。
对于逆变工况，要求整流器与变换桥中开关管的驱动脉冲同步，并且前者的导通占空比略大于后者。也就是说，前者之驱动脉冲的上升沿超前于后者，前者之驱动脉冲的下降沿滞后于后者。超前/滞后的时间要精确控制并保持稳定，使整流器之桥臂的重叠时间等于(或者稍微小于)变换桥之桥臂的死区时间。
3、单变压器方案的工作原理
该变换器的单变压器方案，采用单桥臂PWM控制，其工作原理详述如下。
3.1变压器的变比与输出电压的平均值
如图1所示，设定节点Vs、V1、V2、Vd与GND之间的电压，分别为us、u1、u2、ud。ud称作母线电压；节点Vs与V2之间的电压记作ua，ua即为交流电源Ua经过输入滤波电路后在电容Cs上的正弦交流电压；节点Vs与V1之间的电压记作up，up即为变压器原边绕组Np的电压。
电源Ua经过输入滤波电路后，在电容Cs上形成稳定的正弦交流电压ua。
式中，UR为ua的有效值，ω为ua的角频率。
由开关管Q3和Q4构成的第二桥臂，以正弦交流电源Ua的频率低频切换；由开关管Q1和Q2构成的第一桥臂，在互补PWM控制下高频变换。up与ua的关系为：
up＝ua+(u2-u1)             (E-2)
在正弦交流电源Ua的正半周期，即0＜ωt≤π,ua＞0，开关管Q4关断而Q3导通，则u2＝0。当开关管Q2关断而Q1导通时u1＝0，则up＝ua；当开关管Q1关断而Q2导通时u1＝ud，则up＝ua-ud。
在正弦交流电源Ua的负半周期，即π＜ωt≤2π,ua＜0，开关管Q3关断而Q4导通，则u2＝ud。当开关管Q2关断而Q1导通时u1＝0，则up＝ua+ud；当开关管Q1关断而Q2导通时u1＝ud，则up＝ua。
为了简化分析，忽略开关管Q1、Q2、Q3、Q4的导通压降，忽略Q1、Q2通断切换的死区时间。设定开关管Q1的导通占空比为D1，开关管Q2的导通占空比为D2，则D1+D2＝1。根据开关变换的伏秒值平衡原理，得出电压ua与ud的关系式。
下面以整流器采用全桥整流拓扑为例(见图4)，分析直流输出电压Vo与正弦交流电压ua的关系。Vo为输出+Vo与-Vo之间的电压，即是滤波器的滤波电容Co上的电压。设变压器的副边绕组Ns与原边绕组Np之变比为n。
忽略整流器的压降，则其正负输出端电压(即整流电压)之平均值VR为：
联立式(E-3)、式(E-4)和式(E-1)，得到VR和ud的统一解析式。

整流器的输出电压VR通过滤波器之后，得到直流输出电压Vo。Vo可以分解成直流分量(即平均值)和交流分量
由式(E-5)可见，当变比n确定之后，电压VR与ua、D两个变量有关。若要使VR为纯直流，则需要D与|sin(ωt)|成反比；而D的变化范围是0＜D＜1，VR不能够为纯直流，其含有一定的交流分量。因此该变换器需要调节D的平均值以稳定Vo的直流分量，而D的瞬时值则按照功率因数校正(PFC)的要求而控制。
那么，n如何确定？这需要一个约束条件——限定母线电压ud的最大峰值由于电容网络采用小容量电容，因此出现在交流电压ua的峰值时刻。设开关管Q1～Q4的耐压值为VDS，令A称作耐压利用率，一般取A＝0.85～0.8。
由式(E-5)和式(E-1)推导出：
理论分析和仿真实验证明：当以电压源输出(保留Co)时，交流电压ua峰值时刻对应的VR瞬时值等于输出电压的平均值当以电流源输出(去掉Co)时，交流电压ua峰值时刻对应的VR瞬时值等于输出电压的峰值
设定交流电压ua的范围是[UR-m,UR-M]，UR-m和UR-M分别为ua的最小有效值和最大有效值。根据式(E-7)和式(E-5)得出变比n和输出电压的统一关系式：

式中，和分别为输出电压的平均值和峰值，和分别为输出电压的最大平均值和最大峰值；Dm为有效值UR对应的最小表征占空比，它发生在ua峰值时刻。
说明：考虑二极管压降、死区时间、效率等因素，则变比n的实际值应该略大于式(E-8)给出的理论值。
当整流器采用全波整流拓扑时，可以得出类似的分析结论。
3.3功率因数校正与输出电压的交流分量
同时，该变换器还要实现PFC功能(Power Factor Correction—功率因数校正)。所谓PFC，即是交流电流ia跟踪交流电压ua，使它们波形一致相位相同，从而达到高功率因数。理论上，功率因数PF≤1。当PF＝1时即有：
式(E-13)中，IR为交流电流ia的有效值。设该变换器的效率为η，则其交流输入功 率Pa和直流输出功率Po分别为：
将输出功率Po分解成直流分量加上交流分量的形式：
下面以电压源输出、电阻负载为例，分析输出电压Vo中的交流分量。因为含有电抗分量(感抗或容抗)的负载与滤波电容并联，可以等效成电容与电阻的并联模型。
直流输出电压Vo可以分解成，直流分量加上交流分量的形式，即：
根据能量守恒定律、线性叠加定理和电路理论，得出如下微分方程：
其中，Co为滤波电容，Ro为负载电阻。考虑到式(E-17)简化为：
求解式(E-18)的微分方程，可以得出交流分量的表达式如下：
由式(E-19)可见，单级PFC变换器输出电压的交流分量的角频率为输入交流电压角频率的2倍，所以称为二次谐波。增大滤波电容Co可以减小二次谐波，但是不能完全消除。若要完全消除该二次谐波，则需采用另外的技术手段。
4、双变压器方案的工作原理
该变换器的双变压器方案，采用双桥臂倍频PWM控制或者双桥臂双极PWM控制，其工作原理详述如下。
4.1母线电压、整流电压与占空比的关系
如图6～图8所示，节点Vs1、Vs2、V1、V2、Vd与GND之间的电压，分别记作us1、us2、u1、u2、ud。ud称作母线电压；节点Vs1与Vs2之间的电压记作ua，ua即为交流电源Ua经过滤波后的正弦交流电压。节点Vs1与V1之间的电压记作up1，up1即为第一变压器之原边绕组Np1的电压；节点Vs2与V2之间的电压记作up2，up2即为第二变压器之原边绕组Np2的电压。则ua、up1、up2与us1、us2的关系为：
ua＝us1-us2,up1＝us1-u1,up2＝us2-u2    (E-20)
引入虚拟中点电压um，给定下列关系：
第一桥臂和第二桥臂，以相同的开关频率在互补PWM控制下高频变换。首先分析第一桥臂和第一变压器的工作情况。当开关管Q2关断而Q1导通时u1＝0，则up1＝us1；当开关管Q1关断而Q2导通时u1＝ud，则up1＝us1-ud。
为了简化分析，忽略开关管Q1、Q2的导通压降，忽略桥臂变换的死区时间。设定开关管Q1的导通占空比为D1，开关管Q2的导通占空比为D2，则D1+D2＝1。根据开关变换的伏秒值平衡原理，第一变压器的原边绕组Np1满足如下关系式。
us1·D1+(us1-ud)·D2＝0          (E-22)
下面以整流器X采用全桥整流拓扑(见图4)为例进行分析。设第一变压器的副边绕组Ns1与原边绕组Np1之变比为n1。
忽略整流器X的压降，则其整流输出电压之平均值VR1为：
VR1＝(us1·D1+(ud-us1)·D2)·n1         (E-23)
联立式(E-23)、式(E-22)和式(E-21)，得到：
然后分析第二桥臂和第二变压器的工作情况。当开关管Q4关断而Q3导通时u2＝0，则up2＝us2；当开关管Q3关断而Q4导通时u2＝ud，则up2＝us2-ud。
同样，忽略开关管Q3、Q4的导通压降和桥臂变换的死区时间。设定开关管Q3的导通占空比为D3，开关管Q4的导通占空比为D4，则D3+D4＝1。根据开关变换的伏秒值平衡原理，第二变压器的原边绕组Np2满足如下关系式。
us2·D3+(us2-ud)·D4＝0             (E-25)
同样，整流器Y也采用全桥整流拓扑，见图4。设第二变压器的副边绕组Ns2与原边绕组Np2之变比为n2。
同样，忽略整流器Y的压降，则其整流输出电压之平均值VR2为：
VR2＝(us2·D3+(ud-us2)·D4)·n2      (E-26)
联立式(E-26)、式(E-25)和式(E-21)，得到：
设定相同的整流电压平均值VR和变比n。
VR2＝VR1＝VR,n1＝n2＝n           (E-28)
联立式(E-28)和式(E-27)、式(E-24)，推导出：
解式(E-29)的二次方程，唯有D3＝1-D1满足实际工程要求，即：
将式(E-30)代入式(E-27)和式(E-24)，推导出：
um＝ud/2            (E-31)
4.2输出的组合方式与变压器的变比n
整流器X和整流器Y可以分别连接滤波器X和滤波器Y，形成两组输出，见图6。两组直流输出能够独立供给负载，或者并联或者串联供给负载。
整流器X与整流器Y还能够以组合方式共同连接一个滤波器，形成一组输出；组合方 式分为并联组合与串联组合，见图7和图8。
整流电压VR或其组合经过滤波器之后，得到直流输出电压Vo；Vo可以分解成直流分量(即平均值)和交流分量
由式(E-32)可见，该变换器需要调节D1的平均值以稳定输出电压Vo的直流分量，而D1的瞬时值则按照功率因数校正(PFC)的要求而控制。
与单变压器方案同样的思路，基于“限定母线电压最大峰值”的约束条件，确定变压器X和变压器Y的变比n。由于电容网络采用小容量电容，因此出现在交流电压ua的峰值时刻。在此重申，设开关管Q1～Q4的耐压值为VDS，令A称作耐压利用率，一般取A＝0.85～0.8。
由式(E-32)和式(E-1)得出：
仿真分析和理论推导证明：当以电压源输出(保留Co)时，交流电压ua峰值时刻对应的VR瞬时值等于输出电压的平均值当以电流源输出(去掉Co)时，交流电压ua峰值时刻对应的VR瞬时值等于输出电压的峰值
由式(E-33)和(E-32)得出，独立或并联输出时变比n和输出电压的统一公式。

式中，和分别为输出电压的平均值和峰值，和分别为输出电压的最大平均值和最大峰值；UR-M为交流电压ua的最大有效值；(D1·D2)min和|D1-D2|max分别表示(D1·D2)的最小值和|D1-D2|的最大值，它发生在ua的正负峰值时刻。
说明1：考虑死区时间、二极管压降和效率等因素，则变比n的实际值应该略大于式(E-34)给出的理论值。
说明2：式(E-34)和式(E-35)为独立或并联输出时的值。当串联输出时，式(E-34)的变比n减半，式(E-35)的和加倍。
当整流器X和整流器Y采用全波整流拓扑时，可以类似地分析。
双变压器方案的功率因数校正与输出电压的交流分量，可以得到与单变压器方案类似的结论。
5、两种衍生电路的说明
变压器的原边绕组兼有升压电感的功能，原边绕组电流ip中不但有交流分量而且有直流分量，其直流分量和交流分量大约各占一半。为了减小变压器原边绕组电流的直流分量，提出了两种衍生电路。
第一种衍生电路是，变压器的原边绕组并联电感，所并联电感的自感量远小于原边绕组的自感量。图11为单变压器方案并联电感的示例。自感量的量化关系为：
式中，Lr为并联电感的自感量，为并联电感后所需的原边绕组自感量，Lp是未并联电感时所需的原边绕组自感量。一般可取β＝3～5，则原边绕组电流的直流分量大约减小到原来的1/(1+β)。
第二种衍生电路是，在变压器之原边绕组的位置加入电感，而原边绕组串联电容Cr。图12为单变压器方案第二种衍生电路的示例。电容的量化关系可参考下面公式：
式中，Cr是串联电容的容量，是串联电容后所需的电容Cs的容量，Cs是未串联电容时所需的电容Cs的容量。
以上所述仅为本发明的优选实施例，并非因此限制本发明的专利范围，凡是在本发明的创新构思下，利用本发明说明书及附图内容所作的等效拓扑变换，或直接或间接运用在其他相关的技术领域，均包括在本发明的专利保护范围内。

Claims (11)

  1. 一种无桥隔离型AC-DC单级PFC变换器，为单变压器方案，包含输入滤波电路(1)、电容网络(2)、变换桥(3)、变压器(4)、整流器(5)和滤波器(6)，它们依次连接；输入滤波电路(1)含有共模与差模电感和X电容、Y电容，具有两个交流输出端；交流电源Ua连接输入滤波电路(1)，输入滤波电路(1)的两个交流输出端提供稳定的交流电压；滤波器(6)将滤除高频纹波的直流电压作为变换器的输出而供给负载；其特征在于：
    变换桥(3)含有开关管Q1、Q2、Q3、Q4，为全桥拓扑具有两个桥臂，开关管Q1和Q2构成第一桥臂，开关管Q3和Q4构成第二桥臂；开关管Q1漏极和Q2源极相连作为节点V1，开关管Q3漏极和Q4源极相连作为节点V2；开关管Q2和Q4漏极相连作为变换桥(3)的正端Vd，开关管Q1和Q3源极相连作为变换桥(3)的地端GND；
    电容网络(2)包括电容Cs、Cd、C1、C2，它们都采用薄膜电容而非电解电容；电容Cs的两端分别连接输入滤波电路(1)的两个交流输出端电容，Cd的两端分别连接变换桥(3)的正端Vd和地端GND；电容C1和C2的一端连接输入滤波电路(1)的一个交流输出端而作为节点Vs，电容C1另一端连接变换桥(3)的地端GND，电容C2另一端连接变换桥(3)的正端Vd，变换桥(3)的节点V2连接输入滤波电路(1)的另一个交流输出端；或者去掉电容C1和C2，或者去掉电容Cs，或者去掉电容Cd；
    变压器(4)具有原边绕组Np和至少一个副边绕组Ns；原边绕组Np具有两端，副边绕组Ns具有两端和中间抽头、或者去掉中间抽头；原边绕组Np一端连接电容网络(2)的节点Vs、另一端连接变换桥(3)的节点V1；副边绕组Ns两端分别连接整流器(5)的两个交流输入端；当整流器(5)采用全波整流拓扑时，Ns的中间抽头作为整流器(5)的正输出端或者负输出端；当整流器(5)采用全桥整流拓扑时，则去掉Ns的中间抽头；
    整流器(5)采用全桥整流拓扑或者全波整流拓扑，具有正输出端、负输出端和两个交流输入端；整流器(5)采用的全桥整流拓扑包括二极管D1、D2、D3、D4，二极管D1的阳极与D2的阴极相连作为整流器(5)的一个交流输入端，二极管D3的阳极与D4的阴极相连作为整流器(5)的另一个交流输入端，二极管D1、D3的阴极相连作为整流器(5)的正输出端，二极管D2、D4的阳极相连作为整流器(5)的负输出端；
    整流器(5)采用的全波整流拓扑包括二极管D1、D2，其连接方式有两种；第一种连接方式称作共阴极连接，即是二极管D1、D2的阴极连接在一起作为整流器(5)的正输出端，二极管D1的阳极和D2的阳极作为整流器(5)的两个交流输入端，变压器(4)副边绕组Ns的中间抽头作为整流器(5)的负输出端；第二种连接方式称作共阳极连接，即是二极管D1、D2的阳极连接在一起作为整流器(5)的负输出端，二极管D1的阴极和D2的阴极作为整流器(5)的两个交流输入端，变压器(4)副边绕组Ns的中间抽头作为整流器(5)的正输出端；
    滤波器(6)包括滤波电感Lo1、Lo2和滤波电容Co，构成四端网络，具有正输入端、负输入端和正输出端、负输出端；滤波电感Lo1的两端分别作为滤波器(6)的正输入端和正输 出端，滤波电感Lo2的两端分别作为滤波器(6)的负输入端和负输出端，滤波电容Co的正极和负极分别连接滤波器(6)的正输出端和负输出端；滤波电感Lo1和Lo2独立或者耦合，或者去掉Lo1或者去掉Lo2；当去掉Lo1时则滤波器(6)的正输入端和正输出端直接相连，当去掉Lo2时则滤波器(6)的负输入端和负输出端直接相连；或者去掉滤波电容Co，以适用于电流源模式输出；
    滤波器(6)的正输入端和负输入端分别连接整流器(5)的正输出端和负输出端，滤波器(6)的正输出端和负输出端分别作为该变换器的输出正端+Vo和输出负端-Vo。
  2. 根据权利要求1所述的一种无桥隔离型AC-DC单级PFC变换器，其特征在于：在单变压器方案中，用二极管D01、D02分别替换变换桥(3)的开关管Q3、Q4，或者用单向晶闸管S1、S2分别替换变换桥(3)的开关管Q3、Q4，替换规则为，二极管或者单向晶闸管的阳极和阴极分别对应开关管的源极和漏极；同时保留电容网络(2)的电容C1和C2，电容C1和C2的一端共同连接电容网络(2)的节点Vs或者连接变换桥(3)的节点V2，电容C1和C2的另一端分别连接变换桥(3)的地端GND和正端Vd。
  3. 根据权利要求1所述的一种无桥隔离型AC-DC单级PFC变换器，其特征在于：在单变压器方案中，用开关管替换整流器(5)中的二极管，替换规则为，开关管的源极和漏极分别对应二极管的阳极和阴极。
  4. 根据权利要求1所述的一种无桥隔离型AC-DC单级PFC变换器，其特征在于：基于单变压器方案有两种衍生电路；第一种衍生电路是，变压器(4)的原边绕组Np并联电感Lr，电感Lr的自感量远小于原边绕组Np的自感量；第二种衍生电路是，在变压器(4)之原边绕组Np的位置加入电感Lr，而原边绕组Np串联电容Cr，具体连接方式为：电感Lr的两端分别连接变换桥(3)的节点V1和电容网络(2)的节点Vs，原边绕组Np一端连接变换桥(3)的节点V1、另一端连接电容Cr的一端，电容Cr另一端连接变换桥(3)的节点V2或正端Vd或地端GND、或者连接电容网络(2)的节点Vs。
  5. 一种无桥隔离型AC-DC单级PFC变换器，为双变压器方案，包含输入滤波电路(1)、电容网络(2)、变换桥(3)、第一变压器(41)、第二变压器(42)、整流器X(51)、整流器Y(52)、一个滤波器(6)或者两个滤波器，即滤波器X(61)和滤波器Y(62)；输入滤波电路(1)含有共模与差模电感和X电容、Y电容，具有两个交流输出端；交流电源Ua连接输入滤波电路(1)，输入滤波电路(1)的两个交流输出端提供稳定的交流电压；其特征在于：
    变换桥(3)含有开关管Q1、Q2、Q3、Q4，为全桥拓扑具有两个桥臂，开关管Q1和Q2构成第一桥臂，开关管Q3和Q4构成第二桥臂；开关管Q1漏极和Q2源极相连作为节点V1，开关管Q3漏极和Q4源极相连作为节点V2；开关管Q2和Q4漏极相连作为变换桥(3)的正端Vd，开关管Q1和Q3源极相连作为变换桥(3)的地端GND；
    电容网络(2)包含电容Cs、Cd、C1、C2、C3、C4，它们都采用薄膜电容而非电解电容； 电容Cs的两端分别连接输入滤波电路(1)的两个交流输出端，电容Cd的两端分别连接变换桥(3)的正端Vd和地端GND；电容C1和C3的一端连接输入滤波电路(1)的一个交流输出端而作为节点Vs1，电容C2和C4的一端连接输入滤波电路(1)的另一个交流输出端而作为节点Vs2，电容C1和C2的另一端连接变换桥(3)的地端GND，电容C3和C4的另一端连接变换桥(3)的正端Vd；或者去掉电容C1、C2、C3、C4之中的任意两个，或者去掉电容Cs或者去掉电容Cd，或者去掉电容Cs和Cd；
    第一变压器(41)具有原边绕组Np1和至少一个副边绕组Ns1；原边绕组Np1具有两端，副边绕组Ns1具有两端和中间抽头、或者去掉中间抽头；第二变压器(42)具有原边绕组Np2和至少一个副边绕组Ns2；原边绕组Np2具有两端，副边绕组Ns2具有两端和中间抽头、或者去掉中间抽头；
    原边绕组Np1两端分别连接电容网络(2)的节点Vs1和变换桥(3)的节点V1，原边绕组Np2两端分别连接电容网络(2)的节点Vs2和变换桥(3)的节点V2；副边绕组Ns1两端分别连接整流器X(51)的两个交流输入端；当整流器X(51)采用全波整流拓扑时，Ns1的中间抽头作为整流器X(51)的正输出端或者负输出端；当整流器X(51)采用全桥整流拓扑时，则去掉Ns1的中间抽头；副边绕组Ns2两端分别连接整流器Y(52)的两个交流输入端；当整流器Y(52)采用全波整流拓扑时，Ns2的中间抽头作为整流器Y(52)的正输出端或者负输出端；当整流器Y(52)采用全桥整流拓扑时，则去掉Ns2的中间抽头；
    整流器X(51)和整流器Y(52)采用全波整流拓扑或者全桥整流拓扑，具有正输出端、负输出端和两个交流输入端；整流器X(51)和整流器Y(52)采用的全桥整流拓扑包括二极管D1、D2、D3、D4，二极管D1的阳极与D2的阴极相连作为整流器的一个交流输入端，二极管D3的阳极与D4的阴极相连作为整流器的另一个交流输入端，二极管D1、D3的阴极相连作为整流器的正输出端，二极管D2、D4的阳极相连作为整流器的负输出端；
    整流器X(51)和整流器Y(52)采用的全波整流拓扑包括二极管D1、D2，其连接方式有两种；第一种连接方式称作共阴极连接，即是二极管D1、D2的阴极连接在一起作为整流器的正输出端，二极管D1的阳极和D2的阳极作为整流器的两个交流输入端，第一变压器(41)副边绕组Ns1的中间抽头作为整流器X(51)的负输出端，第二变压器(42)副边绕组Ns2的中间抽头作为整流器Y(52)的负输出端；第二种连接方式称作共阳极连接，即是二极管D1、D2的阳极连接在一起作为整流器的负输出端，二极管D1的阴极和D2的阴极作为整流器的两个交流输入端，第一变压器(41)副边绕组Ns1的中间抽头作为整流器X(51)的正输出端，第二变压器(42)副边绕组Ns2的中间抽头作为整流器Y(52)的正输出端；
    滤波器X(61)、滤波器Y(62)和滤波器(6)一样，均为四端网络，具有正输入端、负输入端和正输出端、负输出端；均包括滤波电感Lo1、Lo2和滤波电容Co；滤波电感Lo1的两端分别作为滤波器的正输入端和正输出端，滤波电感Lo2的两端分别作为滤波器的负输入端和负输出端；滤波电容Co的正极和负极分别连接滤波器的正输出端和负输出端；滤波电感Lo1和Lo2独立或者耦合，或者去掉Lo1或者去掉Lo2，以简化电路；当去掉Lo1 时则滤波器的正输入端和正输出端直接相连，当去掉Lo2时则滤波器的负输入端和负输出端直接相连；或者去掉滤波电容Co，以适用于电流源模式输出；
    整流器X(51)和整流器Y(52)分别连接滤波器X(61)和滤波器Y(62)，形成两组输出；或者整流器X(51)与整流器Y(52)以组合方式共同连接滤波器(6)，形成一组输出。
  6. 根据权利要求5所述的一种无桥隔离型AC-DC单级PFC变换器，其特征在于：整流器X(51)连接滤波器X(61)构成该变换器的第一输出，整流器Y(52)连接滤波器Y(62)构成该变换器的第二输出；具体说明：整流器X(51)的正输出端和负输出端分别连接滤波器X(61)的正输入端和负输入端，滤波器X(61)的正输出端和负输出端分别作为第一输出的正端+Vo1和负端-Vo1；整流器Y(52)的正输出端和负输出端分别连接滤波器Y(62)的正输入端和负输入端，滤波器Y(62)的正输出端和负输出端分别作为第二输出的正端+Vo2和负端-Vo2；第一输出和第二输出能够独立或者并联或者串联供给负载。
  7. 根据权利要求5所述的一种无桥隔离型AC-DC单级PFC变换器，其特征在于：整流器X(51)与整流器Y(52)以组合方式共同连接滤波器(6)而构成一组输出，滤波器(6)的正输出端和负输出端分别作为该变换器输出的正端+Vo和负端-Vo；组合方式分为并联组合与串联组合；并联组合为，整流器X(51)和整流器Y(52)的正输出端共同连接滤波器(6)的正输入端，整流器X(51)和整流器Y(52)的负输出端共同连接滤波器(6)的负输入端；串联组合为，整流器X(51)的负输出端连接整流器Y(52)的正输出端，整流器X(51)的正输出端和整流器Y(52)的负输出端分别连接滤波器(6)的正输入端和负输入端。
  8. 根据权利要求5所述的一种无桥隔离型AC-DC单级PFC变换器，其特征在于：在双变压器方案中，用开关管替换整流器X(51)和整流器Y(52)中的二极管，替换规则为，开关管的源极和漏极分别对应二极管的阳极和阴极。
  9. 根据权利要求5所述的一种无桥隔离型AC-DC单级PFC变换器，其特征在于：基于双变压器方案有两种衍生电路；第一种衍生电路是：第一变压器(41)的原边绕组Np1并联电感Lr1，电感Lr1的自感量远小于原边绕组Np1的自感量；第二变压器(42)的原边绕组Np2并联电感Lr2，电感Lr2的自感量远小于原边绕组Np2的自感量；第二种衍生电路是：在第一变压器(41)之原边绕组Np1的位置加入电感Lr1，而原边绕组Np1串联电容Cr1，在第二变压器(42)之原边绕组Np2的位置加入电感Lr2，而原边绕组Np2串联电容Cr2；具体连接方式为：电感Lr1的两端分别连接变换桥(3)的节点V1和电容网络(2)的节点Vs1，原边绕组Np1一端连接变换桥(3)的节点V1、另一端连接电容Cr1的一端，电容Cr1另一端连接变换桥(3)的正端Vd或地端GND、或者连接电容网络(2)的节点Vs1或节点Vs2；电感Lr2的两端分别连接变换桥(3)的节点V2和电容网络(2)的节点Vs2，原边绕组Np2一端连接变换桥(3)的节点V2、另一端连接电容Cr2的一端，电容Cr2另一端连接变换桥(3)的正端Vd或地端GND、或者连接电容网络(2)的节点Vs1或节点Vs2。
  10. 一种无桥隔离型AC-DC单级PFC变换器的控制方法，采用单桥臂PWM控制，适用于所述变换器的单变压器方案；其特征是：
    由开关管Q3和Q4构成的第二桥臂，以交流电源Ua的频率低频切换；在交流电源Ua的正半周期，即节点Vs的电压高于节点V2的电压，则上部的开关管Q4关断而下部的开关管Q3导通；在交流电源Ua的负半周期，即节点Vs的电压低于节点V2的电压，则下部的开关管Q3关断而上部的开关管Q4导通；
    由开关管Q1和Q2构成的第一桥臂，在互补PWM控制下高频变换；所谓互补PWM是指，在忽略死区时间的情况下开关管Q1和Q2的占空比之和等于1；设定开关管Q1和Q2的导通占空比分别为D1和D2，则D1+D2＝1；在交流电源Ua的正半周期，该变换器的输出电压与D1的平均值成正比；在交流电源Ua的负半周期，该变换器的输出电压与D2的平均值成正比；而D1和D2的瞬时值则按照功率因数校正的要求而变化；
    控制流程如下：
    在直流侧，取样输出变量，输出变量为输出电压、输出电流或者输出功率；
    将输出变量经过(带宽小于20Hz)低通滤波器，滤除二次谐波和高频纹波；
    将滤波后的输出变量与给定值进行比较，得到输出变量误差；
    将输出变量误差进行PID或PI调整，得到反馈变量；
    将反馈变量隔离后负反馈到交流侧；
    在交流侧，取样交流电压和输入电流；
    交流电压取样信号控制第二桥臂低频切换，并且作为输入电流的波形相位基准；
    将交流电压取样信号与来自输出侧的反馈变量进行运算，得到输入电流的基准值；
    将该基准值与输入电流取样信号进行比较，得到输入电流误差；
    将输入电流误差进行PID或PI调整，得到控制量；
    将该控制量与三角波进行比较，产生脉宽调制信号；
    再插入死区时间，形成两路互补的PWM信号；
    该两路互补PWM信号驱动第一桥臂的开关管Q1和Q2高频变换。
  11. 一种无桥隔离型AC-DC单级PFC变换器的控制方法，采用双桥臂PWM控制，分为倍频调制和双极调制两种模式，适用于所述变换器的双变压器方案；其特征是：
    变换桥(3)的第一桥臂和第二桥臂，以相同的开关频率在互补PWM控制下高频变换；设定开关管Q1、Q2和Q3、Q4的导通占空比分别为D1、D2和D3、D4；忽略开关变换的死区时间，则D1+D2＝1＝D3+D4；并且要求D4＝D1和D3＝D2；
    双极调制模式为，开关管Q3、Q4分别与开关管Q2、Q1的导通驱动脉冲同步(即同相位)；倍频调制模式为，开关管Q3、Q4分别与开关管Q1、Q2的导通驱动脉冲同步；倍频调制模式的PWM脉冲有三种对齐方式：中心对齐、前沿对齐、后沿对齐；
    控制规律：该变换器的输出电压与(D1·D2)min/|D1-D2|max成正比，(D1·D2)min表示(D1·D2)的最小值，|D1-D2|max表示|D1-D2|的最大值，它们发生在交流电压ua的正负峰值时刻；而D1和D2的瞬时值则按照功率因数校正的要求而变化；
    控制流程如下：
    在直流侧，取样输出变量，输出变量为输出电压、输出电流或者输出功率；
    将输出变量经过(带宽小于20Hz)低通滤波器，滤除二次谐波和高频纹波；
    将滤波后的输出变量与给定值进行比较，得到输出变量误差；
    将输出变量误差进行PID或PI调整，得到反馈变量；
    将反馈变量隔离后负反馈到交流侧；
    在交流侧，取样交流电压ua和输入电流ia，前者作为输入电流的波形相位基准；
    将交流电压取样信号与来自输出侧的反馈变量进行运算，得到输入电流的基准值；
    将该基准值与输入电流取样信号进行比较，得到输入电流误差；
    将输入电流误差进行PID或PI调整，得到控制量；
    将该控制量与三角波进行比较，产生脉宽调制信号；
    再插入死区时间，扩充移相而形成两两互补的四路PWM信号；
    该四路PWM信号，分别驱动第一桥臂和第二桥臂的开关管高频变换。
PCT/CN2023/110969 2022-09-25 2023-08-03 无桥隔离型ac-dc单级pfc变换器及其控制方法 WO2024060855A1 (zh)

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