WO2023240449A1 - 不连续脉宽调制方法及三相逆变器调制电路 - Google Patents

不连续脉宽调制方法及三相逆变器调制电路 Download PDF

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WO2023240449A1
WO2023240449A1 PCT/CN2022/098699 CN2022098699W WO2023240449A1 WO 2023240449 A1 WO2023240449 A1 WO 2023240449A1 CN 2022098699 W CN2022098699 W CN 2022098699W WO 2023240449 A1 WO2023240449 A1 WO 2023240449A1
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igbt device
phase
sequence component
wave
angle range
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PCT/CN2022/098699
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English (en)
French (fr)
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魏琪康
刘超厚
施鑫淼
张祥平
王建星
魏斯旺
孙宇
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浙江艾罗网络能源技术股份有限公司
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Priority to PCT/CN2022/098699 priority Critical patent/WO2023240449A1/zh
Priority to CN202280006943.8A priority patent/CN116686201A/zh
Publication of WO2023240449A1 publication Critical patent/WO2023240449A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/538—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a push-pull configuration
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present application relates to the technical field of three-phase inverter modulation, and in particular to a discontinuous pulse width modulation method and a three-phase inverter modulation circuit.
  • Discontinuous pulse width modulation also called bus clamp modulation
  • DPWM Discontinuous pulse width modulation
  • the use of discontinuous pulse width modulation method can reduce the number of switching actions, which is beneficial to reducing the switching losses of semiconductor devices.
  • DPWM0 is one of the most widely used discontinuous modulation methods.
  • This modulation method has the advantage of lower switching loss than space vector modulation (SVPWM) when the modulation ratio is high.
  • SVPWM space vector modulation
  • Discontinuous modulation is very suitable for inverters with stable DC input and AC voltages.
  • the input voltage range of PV can change from 100V to 1000V, and the AC voltage may also fluctuate from 180V to 260V.
  • the modulation ratio mainly depends on the ratio of AC voltage and DC bus voltage. Therefore, the modulation ratio will vary widely. This results in the application of conventional discontinuous modulation methods being limited.
  • inverters In order to avoid problems such as harmonics and stability caused by discontinuous modulation, some inverters only use SVPWM modulation. There are also some inverters that use multiple modulation switching methods. For example, DPWM0 is used when the modulation is relatively high, and SVPWM modulation is used when the modulation is relatively low. This modulation mode switching method will bring stability risks. If you switch to SVPWM modulation, compared with discontinuous modulation, there will also be the disadvantage of increased loss and reduced efficiency.
  • This application provides a discontinuous pulse width modulation method and a three-phase inverter modulation circuit.
  • this application provides a discontinuous pulse width modulation method, which is applied to a three-phase inverter circuit.
  • the discontinuous pulse width modulation method includes:
  • the signal wave with added zero sequence component is used as a modulated wave and compared with the high-frequency triangular carrier wave to generate the switching tube driving signal of each IGBT device;
  • each IGBT device is controlled based on the switching tube drive signal of each IGBT device.
  • the present application provides a three-phase inverter modulation circuit.
  • the three-phase inverter modulation circuit includes:
  • a sampling circuit electrically connected to the three-phase inverter circuit
  • the modulator is electrically connected to the three-phase inverter circuit, the modulator is also electrically connected to the sampling circuit, the modulator is used to perform the discontinuous pulse width modulation method as mentioned in the foregoing content .
  • the present application relates to a discontinuous pulse width modulation method and a three-phase inverter modulation circuit.
  • the discontinuous pulse width modulation method reduces the number of switching actions and reduces the power consumption compared to the traditional SVPWM modulation method. consumption, which improves the working efficiency of the inverter.
  • the stability is greatly improved. It can still work stably when the modulation ratio is constantly changing, taking into account the two advantages of low power consumption and high stability.
  • the discontinuous pulse width modulation method provided by this application does not require switching between different modulation modes.
  • Figure 1 is a method flow chart of a discontinuous pulse width modulation method provided by an embodiment of the present application.
  • FIG. 2 is a circuit diagram of a three-phase inverter modulation circuit provided by an embodiment of the present application.
  • Figure 3 is a circuit diagram of a three-phase inverter modulation circuit provided by another embodiment of the present application.
  • FIG. 4 is a schematic diagram comparing the waveforms of the A-phase modulation wave and the high-frequency triangular carrier wave when the modulation ratio is 1 in the discontinuous pulse width modulation method provided by an embodiment of the present application.
  • Figure 5 shows the switching tube driving signal of the first IGBT device, the switching tube driving signal of the second IGBT device, and the switching tube driving signal of the third IGBT device in a discontinuous pulse width modulation method provided by this application when the modulation ratio is 1.
  • Signal and waveform diagram of the switch tube driving signal of the fourth IGBT device are shown.
  • Figure 6 is a composite waveform diagram of the original signal wave of phase A of the traditional SVPWM modulation method and the switching tube drive signal of the first IGBT device when the modulation ratio is 1;
  • Figure 7 shows the composite waveform diagram of the A-phase modulation wave and zero-sequence component of the traditional SVPWM modulation method when the modulation ratio is 1;
  • Figure 8 is a composite waveform diagram of the original signal wave of phase A and the switch tube drive signal of the first IGBT device using the traditional DPWM0 modulation method when the modulation ratio is 1;
  • Figure 9 shows the composite waveform diagram of the modulation wave and zero-sequence component of phase A of the traditional DPWM0 modulation method when the modulation ratio is 1;
  • Figure 10 is a composite waveform diagram of the original signal wave of phase A and the switch tube driving signal of the first IGBT device using a discontinuous pulse width modulation method provided by this application when the modulation ratio is 1;
  • Figure 11 is a composite waveform diagram of the A-phase modulation wave and zero-sequence component using a discontinuous pulse width modulation method provided by this application when the modulation ratio is 1;
  • Figure 12 is a composite waveform diagram of the original signal wave of phase A and the switch tube drive signal of the first IGBT device using the traditional SVPWM modulation method when the modulation ratio is 0.75;
  • Figure 13 shows the composite waveform diagram of the A-phase modulation wave and zero-sequence component of the traditional SVPWM modulation method when the modulation ratio is 0.75;
  • Figure 14 is a composite waveform diagram of the original signal wave of phase A and the switch tube drive signal of the first IGBT device using the traditional DPWM0 modulation method when the modulation ratio is 0.75;
  • Figure 15 is a composite waveform diagram of the modulation wave and zero sequence component of phase A of the traditional DPWM0 modulation method when the modulation ratio is 0.75;
  • Figure 16 is a composite waveform diagram of the A-phase original signal wave and the switch tube driving signal of the first IGBT device using a discontinuous pulse width modulation method provided by this application when the modulation ratio is 0.75;
  • Figure 17 is a composite waveform diagram of the A-phase modulation wave and zero-sequence component of a discontinuous pulse width modulation method provided by this application when the modulation ratio is 0.75;
  • Figure 18 is a composite waveform diagram of the original signal fundamental wave of phase A and the switch tube drive signal of the first IGBT device using the traditional SVPWM modulation method when the modulation ratio is 0.5;
  • Figure 19 is a composite waveform diagram of the modulation wave and zero sequence component of phase A of the traditional SVPWM modulation method when the modulation ratio is 0.5;
  • Figure 20 is a composite waveform diagram of the original signal wave of phase A and the switch tube drive signal of the first IGBT device using the traditional DPWM0 modulation method when the modulation ratio is 0.5;
  • Figure 21 is a composite waveform diagram of the modulation wave and zero sequence component of phase A of the traditional DPWM0 modulation method when the modulation ratio is 0.5;
  • Figure 22 is a composite waveform diagram of the original signal wave of phase A and the switch tube driving signal of the first IGBT device using a discontinuous pulse width modulation method provided by this application when the modulation ratio is 0.5;
  • Figure 23 is a composite waveform diagram of the A-phase modulation wave and zero-sequence component of a discontinuous pulse width modulation method provided by this application when the modulation ratio is 0.5.
  • 10-three-phase inverter circuit 110-DC bus; 111-positive bus capacitor; 112-negative bus capacitor;
  • FIG. 3 is a circuit diagram of a three-phase inverter modulation circuit provided by an embodiment of the present application.
  • the discontinuous pulse width modulation method provided by the present application is applied to the three-phase inverter modulation circuit shown in FIG. 3 .
  • the three-phase inverter modulation circuit shown in Figure 3 is a T-shaped three-phase three-level circuit.
  • the discontinuous pulse width modulation method provided by this application does not limit its execution subject.
  • the execution subject of the discontinuous pulse width modulation method provided by this application may be a modulator in a three-phase inverter modulation circuit.
  • the modulator may include one or more controllers.
  • the discontinuous pulse width modulation method includes the following S100 to S400:
  • the original signal wave of the three-phase voltage is a sine wave.
  • the dotted lines in Figure 6, Figure 8, Figure 10, Figure 12, Figure 14, Figure 16, Figure 18, Figure 20 and Figure 22 are all three-phase voltages.
  • the dotted line part in Figure 10, Figure 16 and Figure 22 is the waveform of phase A in the three-phase voltage original signal wave of the discontinuous pulse width modulation method provided by this application.
  • the abscissa is time, and the ordinate is the amplitude generated after normalizing the voltage value.
  • the modulator controls the sampling circuit to obtain the DC bus voltage, three-phase voltage and three-phase current. After the sampling is completed, the sampling circuit inputs the DC bus voltage, three-phase voltage and three-phase current to the modulator. Further, the modulator performs calculations based on the DC bus voltage, three-phase voltage and three-phase current to generate the three-phase voltage original signal wave.
  • the original signal wave of the three-phase voltage is composed of the superposition of the fundamental wave and some harmonics.
  • the signal wave to which the zero-sequence component is added is subsequently used as a modulated wave.
  • the solid line part in Figure 11, Figure 17 and Figure 23 is the modulation waveform of phase A in the three-phase voltage original signal wave of the discontinuous pulse width modulation method provided by this application, and the dotted line part is the waveform of the zero sequence component.
  • the abscissa is time, and the ordinate is the amplitude generated after normalizing the voltage value.
  • the modulation wave is compared with two high-frequency triangular carrier waves.
  • One of the two high-frequency triangular carrier waves has an amplitude of 0. to a high frequency triangular carrier of amplitude -1 to 0, and the other is a high frequency triangular carrier of amplitude -1 to 0.
  • the amplitudes mentioned here are the normalized amplitudes of the voltage values.
  • the frequency range of high frequency refers to the frequency range with a frequency greater than or equal to 3kHz and less than or equal to 50kHz.
  • S400 controls the operation of each IGBT device based on the switch tube drive signal of each IGBT device.
  • This application relates to a discontinuous pulse width modulation method.
  • the traditional SVPWM modulation method it reduces the number of switching actions, reduces power consumption, and improves the working efficiency of the switching tube.
  • the traditional DPWM0 modulation method The stability is greatly improved, and it can still work stably when the modulation ratio is constantly changing, taking into account the two advantages of low power consumption and high stability, and the discontinuous pulse width modulation method provided by this application does not require switching between different modulation methods. switch.
  • the S100 includes the following S110 to S120:
  • Equation 1 the voltage value of each phase in the three-phase voltage original signal wave is as shown in Equation 1.
  • m is the modulation ratio.
  • ⁇ is the vector angle.
  • Ua is the voltage value of phase A.
  • Ub is the B-phase voltage value.
  • Uc is the C-phase voltage value.
  • Uah is the harmonic component of phase A.
  • Ubh is the harmonic component of phase B.
  • Uch is the harmonic component of phase C.
  • this step normalizes the voltage values Ua, Ub, and Uc of each phase. Specifically, the voltage value of each phase is divided by one-half of the DC bus voltage value, and the voltage value of each phase is The value is converted into a value between -1 and 1. This value is called the amplitude in this application to facilitate subsequent comparison with the high-frequency triangular carrier.
  • the carrier wave is a continuous triangular wave with an amplitude of 1, so m is the amplitude of the three-phase voltage and is also the modulation ratio.
  • Equation 2 The maximum value of the three-phase voltage and the minimum value of the three-phase voltage are defined as shown in Equation 2.
  • Umax is the maximum value of the three-phase voltage.
  • Umin is the minimum value of the three-phase voltage.
  • Ua is the voltage value of phase A.
  • Ub is the B-phase voltage value.
  • Uc is the C-phase voltage value.
  • this step is to respectively obtain the maximum value of the three-phase voltage and the minimum value of the three-phase voltage.
  • the S200 includes the following S210:
  • Uz is the zero sequence component.
  • Umax is the maximum value of the three-phase voltage.
  • Umin is the minimum value of the three-phase voltage.
  • k is the zero sequence component calculation coefficient.
  • k can be 0 or 1, and k can also alternately change between 0 and 1.
  • the S200 further includes:
  • S220 Add the zero-sequence component to the voltage value of each phase in the three-phase voltage sinusoidal original signal wave to obtain a signal wave to which the zero-sequence component will be added.
  • the voltage value of each phase of the signal wave with the zero sequence component added is shown in Equation 4.
  • Uas is the phase A voltage value after adding the zero sequence component.
  • Ubs is the B-phase voltage value after adding the zero sequence component.
  • Ucs is the C-phase voltage value after adding the zero sequence component.
  • this application generates a modulated wave by superimposing a zero-sequence component on the basis of the original signal wave.
  • the S300 includes the following S310 to S360:
  • the first angle range is an angle range greater than or equal to 0 plus ⁇ and less than or equal to one-third ⁇ minus ⁇ .
  • the second angle range is an angle range that is greater than or equal to two-thirds of ⁇ plus ⁇ and less than or equal to ⁇ minus ⁇ .
  • the third angle range is an angle range that is greater than or equal to four-thirds ⁇ plus ⁇ and less than or equal to five-thirds ⁇ minus ⁇ .
  • the fourth angle range is an angle range that is greater than or equal to one-third ⁇ plus ⁇ and less than or equal to two-thirds ⁇ minus ⁇ .
  • the fifth angle range is an angle range that is greater than or equal to ⁇ plus ⁇ and less than or equal to four-thirds of ⁇ minus ⁇ .
  • the sixth angle range is an angle range that is greater than or equal to five-thirds ⁇ plus ⁇ and less than or equal to 0 minus ⁇ . ⁇ is the preset angle value.
  • Figure 10 is a composite waveform diagram of the original signal wave of phase A and the switch tube driving signal of the first IGBT device using a discontinuous pulse width modulation method provided by this application when the modulation ratio is 1.
  • Figure 10 contains two waveforms. One waveform is the original signal wave of phase A, that is, the waveform between point X and the zero line in Figure 3 (the dotted sinusoidal part in Figure 10). Another waveform is the waveform of the switching tube driving signal of the first IGBT device (solid line square wave part in Figure 10).
  • the amplitude of the original signal wave is 1 from 60 degrees to 120 degrees, corresponding to the driving constant conduction of the first IGBT device. That is to say, within 180 degrees, there is a total of 60 degrees of time when the switch is inactive, so the loss can be reduced.
  • phase A takes phase A as an example, because Figures 6 to 23 shown in this application are all waveform diagrams of phase A.
  • the first IGBT device, the second IGBT device, the third IGBT device and the fourth IGBT device form the bridge arm of phase A.
  • the first IGBT device serves as the upper bridge arm switch tube of phase A
  • the second IGBT device As the A-phase lower arm switch tube
  • the third IGBT device and the fourth IGBT device serve as the two middle bridge arm switch tubes of the A-phase.
  • the modulated wave in S300 of this application is compared with the high-frequency triangular carrier wave, and finally the switching tube driving signal of each IGBT device is generated.
  • the switching tube driving signal of the first IGBT device and the switching tube driving signal of the second IGBT device are generated.
  • the switching tube driving signal of the third IGBT device and the switching tube driving signal of the fourth IGBT device that is, a total of four switching tube driving signals are finally generated.
  • the dotted line is the A-phase modulated wave. Since this embodiment is applied to a three-phase three-level circuit, the modulated wave is combined with a high-frequency triangle wave with an amplitude of 0 to 1 and a high-frequency triangle wave with an amplitude of -1 to 0. High frequency triangle wave for comparison.
  • the dotted sine wave in Figure 4 is the waveform of the modulation wave
  • the solid line triangle wave is a high-frequency triangle wave with an amplitude ranging from 0 to 1
  • the dotted line triangle wave is a high-frequency triangle wave with an amplitude ranging from -1 to 0.
  • the method of obtaining the switching tube driving signal of the first IGBT device and the switching tube driving signal of the third IGBT device is to compare the waveform of the modulation wave with a high-frequency triangular wave whose amplitude ranges from 0 to 1. .
  • the switching tube driving signal of the first IGBT device outputs a high level at this time.
  • the switching tube driving signal of the first IGBT device outputs a low level at this time.
  • the third IGBT device is complementary to the first IGBT device, that is, at the same time node, when the amplitude of the modulation wave is greater than the amplitude of the high-frequency triangular wave between 0 and 1, the switching tube driving signal of the third IGBT device outputs low at this time. level.
  • the switching tube driving signal of the third IGBT device outputs a high level.
  • the method of obtaining the switching tube driving signal of the second IGBT device and the switching tube driving signal of the fourth IGBT device is to combine the waveform of the modulation wave and a high-frequency triangular wave with an amplitude ranging from -1 to 0. Compare. At the same time node, when the amplitude of the modulation wave is greater than the amplitude of the high-frequency triangular wave in the range of -1 to 0, the switch tube driving signal of the fourth IGBT device outputs a high level at this time. When the amplitude of the modulation wave is smaller than the amplitude of the high-frequency triangular wave in the range of -1 to 0, at this time, the switching tube driving signal of the fourth IGBT device outputs a low level.
  • the second IGBT device and the fourth IGBT device are complementary, that is, at the same time node, when the amplitude of the modulation wave is greater than the amplitude of the high-frequency triangular wave between -1 and 0, the switch tube driving signal of the second IGBT device is output at this time. low level.
  • the switching tube driving signal of the second IGBT device outputs a high level.
  • FIG. 5 The four waveform diagrams in Figure 5 respectively show the waveform diagram of the switching tube driving signal of the first IGBT device, the waveform diagram of the switching tube driving signal of the third IGBT device, and the waveform diagram of the switching tube driving signal of the fourth IGBT device.
  • the waveform diagram of the switching tube driving signal of the first IGBT device in Figure 5 is actually part of the square wave waveform in Figure 10. Therefore, it can be known that the square wave waveform in Figure 10 is obtained through Figure 5, that is, through modulation It is obtained by comparing the wave and the high-frequency triangle wave. By analogy, the square wave waveforms in other figures are also obtained using the above method, which will not be described again later.
  • the DPWM0 modulation method which is fixed within a certain 60-degree range and keeps the switching tube inactive, can reduce losses, but it will also cause its zero-sequence component to be biased towards the positive or negative side within this 60-degree range, as shown in Figure 9 shown. Especially when the modulation ratio becomes smaller, the positive and negative bus deviation will become larger, which is also the reason why its output characteristics will deteriorate.
  • ⁇ can be set to a fixed value or a changing value.
  • ⁇ is set to change with the modulation ratio m.
  • the switch will not perform switching near the zero crossing and the peak, that is, the first IGBT device remains open at the zero crossing and the first IGBT device remains closed at the peak.
  • the modulation ratio m is 1, 0.75 and 0.5 respectively. It can be understood that in the discontinuous pulse width modulation method provided by this application, the greater the modulation ratio m, the first IGBT device at the peak The longer it remains closed, the shorter it remains off near the zero crossing. The smaller the modulation ratio m, the shorter the time the first IGBT device remains closed at the peak value, and the longer the time it remains off near the zero-crossing point.
  • the discontinuous pulse width modulation method provided by this application improves the maximum value of the zero sequence component, and the change of the zero sequence component is also relatively gentle.
  • FIG 6, Figure 12, and Figure 18 are the composite waveform diagrams of the original signal wave of the traditional SVPWM modulation method and the switching tube drive signal of the first IGBT device.
  • the first IGBT device does not have any continuous conduction period in the positive half wave, so This will lead to frequent switching of switch states, low work efficiency, and high energy consumption.
  • the discontinuous pulse width modulation method of this application reduces the number of switching states by three thirds of the number of switching states of the traditional SVPWM modulation method.
  • the S300 further includes:
  • the zero sequence is controlled
  • the components satisfy formula 5.
  • Uz is the zero sequence component.
  • Umax is the maximum value of the three-phase voltage.
  • Umin is the minimum value of the three-phase voltage.
  • the zero sequence is controlled
  • the component is equal to the sum of Umax and Umin. Because when the zero sequence component is this value, the amplitude of the modulation wave near the zero-crossing point can be exactly 0, that is, the first IGBT device can be kept in an off state.
  • ⁇ is a preset fixed value.
  • ⁇ there are many ways to select ⁇ , and it can be selected as a fixed value.
  • the opening and closing range of the traditional DPWM0 IGBT device is one-third ⁇ to one-third ⁇ . That is, the length of the non-action angle interval of the two switches is the same, which is one-third ⁇ .
  • ⁇ can be set between 0 and one-sixth of ⁇ . Set to 0 degrees, it becomes the traditional DPWM0 modulation scheme. Setting it to one-sixth ⁇ means that it is fixed between 0 and one-sixth ⁇ and five-sixths ⁇ and ⁇ .
  • the first IGBT device remains turned off, and at the same time, the first IGBT device loses maintenance conduction near 90 degrees. range of angles.
  • ⁇ is a variable, and when the modulation ratio is larger, the angle value of controlling ⁇ is smaller, and when the modulation ratio is smaller, the angle value of controlling ⁇ is larger.
  • ⁇ there are many ways to select ⁇ , which can change with the change of the k modulation ratio.
  • This embodiment changes with the change of the k modulation ratio.
  • the ⁇ value changes with the modulation ratio is that, on the one hand, when the modulation ratio is small, the ⁇ value should be larger to obtain a smaller zero-sequence component. On the other hand, when the modulation is relatively large, the ⁇ value should be smaller, so that the time the switch tube remains on near 90 degrees can be increased as much as possible. Because in general, the greater the current, the greater the loss of one switching action. The current is generally larger near 90 degrees, and the current is generally smaller near zero. Although the sum of the angles near 90 degrees plus the inaction of the switch near zero is 60 degrees, the proportion of inaction near 90 degrees is larger, which is more conducive to reducing losses. .
  • This application also provides a three-phase inverter modulation circuit.
  • the three-phase inverter modulation circuit includes a three-phase inverter circuit 10 , a sampling circuit 20 and a modulator 30 .
  • the sampling circuit 20 is electrically connected to the three-phase inverter circuit 10 .
  • the modulator 30 is electrically connected to the three-phase inverter circuit 10 , and the modulator 30 is also electrically connected to the sampling circuit 20 .
  • the modulator 30 is used to perform the discontinuous pulse width modulation method mentioned in any of the previous embodiments.
  • this embodiment introduces a T-shaped three-phase three-level circuit.
  • discontinuous pulse width modulation method can also be applied to a three-phase two-level circuit, or a midpoint clamped three-level inverter circuit.
  • This application only introduces the T-type three-phase three-level circuit. Circuit structure, the structure of other circuits will not be described in detail.
  • the three-phase inverter circuit 10 includes a DC bus 110, a neutral line 190, a first IGBT device 121, a second IGBT device 122, and a third IGBT device 123 , the fourth IGBT device 124, the fifth IGBT device 131, the sixth IGBT device 132, the seventh IGBT device 133, the eighth IGBT device 134, the ninth IGBT device 141, the tenth IGBT device 142, the eleventh IGBT device 143, The twelfth IGBT device 144 , the first filter inductor 151 , the second filter inductor 152 , the third filter inductor 153 , the first filter capacitor 160 , the second filter capacitor 170 and the third filter capacitor 180 .
  • the DC bus 110 is provided with a positive bus capacitor 111 and a negative bus capacitor 112 .
  • Point W is provided on the connection link between the positive bus capacitor 111 and the negative bus capacitor 112 .
  • Point W is connected to the zero line.
  • One end of the first IGBT device 121 is connected to the positive bus capacitor 111 .
  • the other end of the first IGBT device 121 is connected to the second IGBT device 122 .
  • One end of the second IGBT device 122 is connected to the first IGBT device 121 , and the other end of the second IGBT device 122 is connected to the negative bus capacitor 112 .
  • One end of the third IGBT device 123 is connected to point W, and the other end of the third IGBT device 123 is connected to the fourth IGBT device 124.
  • One end of the fourth IGBT device 124 is connected to the third IGBT device 123 , and the other end of the fourth IGBT device 124 is connected to the first filter inductor 151 .
  • One end of the first filter inductor 151 is connected to the fourth IGBT device 124, and the other end of the first filter inductor 151 is connected to the A-phase voltage output point.
  • One end of the fifth IGBT device 131 is connected to the positive bus capacitor 111 , and the other end of the fifth IGBT device 131 is connected to the sixth IGBT device 132 .
  • One end of the sixth IGBT device 132 is connected to the fifth IGBT device 131 , and the other end of the sixth IGBT device 132 is connected to the negative bus capacitor 112 .
  • One end of the seventh IGBT device 133 is connected to point W, and the other end of the seventh IGBT device 133 is connected to the eighth IGBT device 134 .
  • One end of the eighth IGBT device 134 is connected to the seventh IGBT device 133 , and the other end of the eighth IGBT device 134 is connected to the second filter inductor 152 .
  • One end of the second filter inductor 152 is connected to the eighth IGBT device 134, and the other end of the second filter inductor 152 is connected to the B-phase voltage output point.
  • One end of the ninth IGBT device 141 is connected to the positive bus capacitor 111 , and the other end of the ninth IGBT device 141 is connected to the tenth IGBT device 142 .
  • One end of the tenth IGBT device 142 is connected to the ninth IGBT device 141 , and the other end of the tenth IGBT device 142 is connected to the negative bus capacitor 112 .
  • One end of the eleventh IGBT device 143 is connected to point W, and the other end of the eleventh IGBT device 143 is connected to the twelfth IGBT device 144 .
  • One end of the twelfth IGBT device 144 is connected to the eleventh IGBT device 143 , and the other end of the twelfth IGBT device 144 is connected to the third filter inductor 153 .
  • One end of the third filter inductor 153 is connected to the twelfth IGBT device 144, and the other end of the third filter inductor 153 is connected to the C-phase voltage output point.
  • the first IGBT device 121, the second IGBT device 122, the fourth IGBT device 124 and the first filter inductor 151 are connected to the same point X.
  • the fifth IGBT device 131, the sixth IGBT device 132, the eighth IGBT device 134 and the second filter inductor 152 are connected to the same point Y.
  • the ninth IGBT device 141, the tenth IGBT device 142, the twelfth IGBT device 144 and the third filter inductor 153 are connected to the same point Z.
  • the three-phase inverter circuit 10 further includes a first filter capacitor 160 , a second filter capacitor 170 and a third filter capacitor 180 .
  • the first filter capacitor 160 has an upper plate 161 connected to the connection link between the first filter inductor 151 and the A-phase voltage output point.
  • the lower plate 162 of the first filter capacitor 160 is connected to the neutral line 190 .
  • the second filter capacitor 170 has an upper plate 171 connected to the connection link between the second filter inductor 152 and the B-phase voltage output point.
  • the lower plate 172 of the second filter capacitor 170 is connected to the neutral line 190 .
  • the third filter capacitor 180 has an upper plate 181 connected to the connection link between the third filter inductor 153 and the C-phase voltage output point, and a lower plate 182 of the third filter capacitor 180 connected to zero. Line 190.
  • the discontinuous pulse width modulation method mentioned in any of the previous embodiments is used to generate a modulation wave between the The switching tube driving signal, the switching tube driving signal of the third IGBT device 123 and the switching tube driving signal of the fourth IGBT device 124.
  • the discontinuous pulse width modulation method mentioned in any of the previous embodiments is also used to generate a modulation wave between the Y point and the zero line, and modulate and generate the switching tube driving signal of the fifth IGBT device 131 and the sixth IGBT device 132 The switching tube driving signal of the seventh IGBT device 133 and the switching tube driving signal of the eighth IGBT device.
  • the discontinuous pulse width modulation method mentioned in any of the previous embodiments is also used to generate a modulation wave between the Z point and the zero line, and modulate and generate the switching tube driving signal of the ninth IGBT device 141 and the tenth IGBT device 142
  • the switching tube of the three-phase inverter circuit 10 can be an IGBT device or a MOSFET device.
  • the antiparallel diode drawn in Figure 3 is the body diode of the IGBT device and is a part of the IGBT device. If you choose a MOSFET device, you can add an additional diode in parallel. The use of any type of switching device will not affect the protection scope of this application.
  • the positive bus capacitor 111 has a positive bus voltage Vbus at both ends, and the negative bus capacitor 112 has a negative bus voltage -Vbus at both ends.
  • the W point between the positive bus capacitor 111 and the negative bus capacitor 112 is the zero point.
  • the first IGBT device 121 , the second IGBT device 122 , the third IGBT device 123 and the fourth IGBT device 124 form the bridge arm of phase A.
  • the first IGBT device 121 serves as the A-phase upper arm switch tube
  • the second IGBT device 122 serves as the A-phase lower arm switch tube.
  • the third IGBT device 123 and the fourth IGBT device 124 serve as the two middle bridge arm switching tubes of the A phase.
  • the fifth IGBT device 131, the sixth IGBT device 132, the seventh IGBT device 133 and the eighth IGBT device 134 form the bridge arm of the B phase.
  • the fifth IGBT device 131 serves as the B-phase upper arm switch tube
  • the sixth IGBT device 132 serves as the B-phase lower arm switch tube.
  • the seventh IGBT device 133 and the eighth IGBT device 134 serve as the two middle bridge arm switching tubes of the B phase.
  • the ninth IGBT device 141 , the tenth IGBT device 142 , the eleventh IGBT device 143 and the twelfth IGBT device 144 form a C-phase bridge arm.
  • the ninth IGBT device 141 serves as the C-phase upper arm switch tube
  • the tenth IGBT device 142 serves as the C-phase lower arm switch tube.
  • the eleventh IGBT device 143 and the twelfth IGBT device 144 serve as two intermediate bridge arm switching tubes of the C phase.
  • the upper bridge arm switch tube and the lower bridge arm switch tube in each phase bridge arm are in a complementary conduction relationship.
  • the voltage at point X is positive Vbus.
  • the voltage at point X is 0.
  • the voltage at point X is negative Vbus.
  • the square wave waveform of the switch tube drive signal shown in Figures 10, 16 and 22 of this application shows that the positive half wave only has positive Vbus and 0, and the negative half wave only has negative Vbus and 0.
  • a period can be set from 20 microseconds to 1 millisecond.

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Abstract

本申请涉及一种不连续脉宽调制方法及三相逆变器调制电路，其中，所述不连续脉宽调制方法，一方面，相对于传统SVPWM调制方法降低了开关动作次数，减小了功耗，提高了逆变器工作效率，另一方面，相对于传统DPWM调制方法大大提升了稳定性，在调制比不断变化时仍然能够稳定工作，兼顾了功耗低和稳定性高两个优点，且本申请提供的不连续脉宽调制方法并不需要进行不同调制方式之间的切换。

Description

不连续脉宽调制方法及三相逆变器调制电路 技术领域
本申请涉及三相逆变器调制技术领域，特别是涉及一种不连续脉宽调制方法及三相逆变器调制电路。
背景技术
不连续脉宽调制(DPWM)也叫母线钳位调制，是三相逆变器常用的调制方式之一。DPWM调制方式主要有DPWM0，DPWM1，DPWM2，DPWM3，DPWMMIN，DPWMMAX这6种方式。采用不连续脉宽调制方法可以使开关动作次数降低，有利于降低半导体器件的开关损耗。
DPWM0是应用最多的不连续调制方式之一。这种调制方式在调制比较高时，相比空间矢量调制(SVPWM)有开关损耗低的优势。但是当调制比变低时，采用DPWM调制方式的逆变器输出电流谐波会变差，运行稳定性也变差，逆变器甚至无法正常工作。对于直流输入和交流电压都比较稳定的逆变器而言，不连续调制是非常合适的。但是实际工作环境中，比如光伏逆变器，PV的输入电压范围可以从100V变到1000V，交流电压也可能从180V波动到260V。而调制比主要取决交流电压和直流母线电压的比值。因此调制比会有较大范围的变化。这就导致常规的不连续调制方法的应用受到了限制。
为了避免不连续调制导致的诸如谐波、稳定性等问题，有些逆变器只采用SVPWM调制。也有些逆变器采用多种调制方式切换的方式，例如调制比较高时采用DPWM0，调制比较低时改用SVPWM调制。这种调制方式切换的方法会带来稳定性的隐患，如果切换到SVPWM调制，相比不连续调制也会有损耗增加效率降低的不足。
因此，传统的DPWM0调制方式无法同时兼顾低损耗和高稳定性。
发明内容
基于此，有必要针对传统三相逆变器调制方法无法同时兼顾低损耗和高稳定性的问题，提供一种不连续脉宽调制方法及三相逆变器调制电路。
本申请提供一种不连续脉宽调制方法及三相逆变器调制电路。
一方面，本申请提供了一种不连续脉宽调制方法，应用于三相逆变电路。
所述不连续脉宽调制方法包括：
生成三相电压原始信号波；
将三相电压原始信号波加入一个零序分量，得到加入零序分量的信号波；
将加入零序分量的信号波作为调制波与高频的三角载波作比较，生成每一个IGBT器件的开关管驱动信号；
依据每一个IGBT器件的开关管驱动信号控制每一个IGBT器件工作。
另一方面，本申请提供了一种三相逆变器调制电路。
所述三相逆变器调制电路包括：
三相逆变器电路；
采样电路，与所述三相逆变器电路电连接；
调制器，所述调制器与所述三相逆变器电路电连接，所述调制器还与所述采样电路电连接，所述调制器用于执行如前述内容提及的不连续脉宽调制方法。
本申请涉及一种不连续脉宽调制方法及三相逆变器调制电路，其中，所述不连续脉宽调制方法，一方面，相对于传统SVPWM调制方法降低了开关动作次数，减小了功耗，提高了逆变器工作效率，另一方面，相对于传统DPWM0调制方法大大提升了稳定性，在调制比不断变化时仍然能够稳定工作，兼顾了功耗低和稳定性高两个优点，且本申请提供的不连续脉宽调制方法并不需要进行不同调制方式之间的切换。
附图说明
图1为本申请一实施例提供的不连续脉宽调制方法的方法流程图。
图2为本申请一实施例提供的三相逆变器调制电路的电路图。
图3为本申请另一实施例提供的三相逆变器调制电路的电路图。
图4为本申请一实施例提供的不连续脉宽调制方法中，调制比为1时，A相的调制波与高频的三角载波的波形比较示意图。
图5为调制比为1时，本申请提供的一种不连续脉宽调制方法中第一IGBT器件的开关管驱动信号，第二IGBT器件的开关管驱动信号，第三IGBT器件的开关管驱动信号和第四IGBT器件的开关管驱动信号的波形示意图。
图6为调制比为1时，传统SVPWM调制方法的A相的原始信号波和第一IGBT器件的开关管驱动信号的复合波形图；
图7为调制比为1时，传统SVPWM调制方法的A相的调制波和零序分量的复合波形图；
图8为调制比为1时，传统DPWM0调制方法的A相的原始信号波和第一IGBT器件的开关管驱动信号的复合波形图；
图9为调制比为1时，传统DPWM0调制方法的A相的调制波和零序分量复合波形图；
图10为调制比为1时，本申请提供的一种不连续脉宽调制方法的A相的原始信号波和第一IGBT器件的开关管驱动信号的复合波形图；
图11为调制比为1时，传本申请提供的一种不连续脉宽调制方法的A相的调制波和零序分量的复合波形图；
图12为调制比为0.75时，传统SVPWM调制方法的A相的原始信号波和第一IGBT器件的开关管驱动信号的复合波形图；
图13为调制比为0.75时，传统SVPWM调制方法的A相的调制波和零序分 量的复合波形图；
图14为调制比为0.75时，传统DPWM0调制方法的A相的原始信号波和第一IGBT器件的开关管驱动信号的复合波形图；
图15为调制比为0.75时，传统DPWM0调制方法的A相的调制波和零序分量的复合波形图；
图16为调制比为0.75时，本申请提供的一种不连续脉宽调制方法的A相的原始信号波和第一IGBT器件的开关管驱动信号的复合波形图；
图17为调制比为0.75时，本申请提供的一种不连续脉宽调制方法的A相的调制波和零序分量的复合波形图；
图18为调制比为0.5时，传统SVPWM调制方法的A相的原始信号基波和第一IGBT器件的开关管驱动信号的复合波形图；
图19为调制比为0.5时，传统SVPWM调制方法的A相的调制波和零序分量的复合波形图；
图20为调制比为0.5时，传统DPWM0调制方法的A相的原始信号波和第一IGBT器件的开关管驱动信号的复合波形图；
图21为调制比为0.5时，传统DPWM0调制方法的A相的调制波和零序分量的复合波形图；
图22为调制比为0.5时，本申请提供的一种不连续脉宽调制方法的A相的原始信号波和第一IGBT器件的开关管驱动信号的复合波形图；
图23为调制比为0.5时，本申请提供的一种不连续脉宽调制方法的A相的调制波和零序分量的复合波形图。
附图标记：
10-三相逆变器电路；110-直流母线；111-正母线电容；112-负母线电容；
121-第一IGBT器件；122-第二IGBT器件；123-第三IGBT器件；
124-第四IGBT器件；131-第五IGBT器件；132-第六IGBT器件；
133-第七IGBT器件；134-第八IGBT器件；1141-第九IGBT器件；
142-第十IGBT器件；143-第十一IGBT器件；144-第十二IGBT器件；
151-第一滤波电感；152-第二滤波电感；153-第三滤波电感；
160-第一滤波电容；161-第一滤波电容的上极板；
162-第一滤波电容的下极板；170-第二滤波电容；
161-第二滤波电容的上极板；162-第二滤波电容的下极板；
180-第三滤波电容；181-第三滤波电容的上极板；
182-第三滤波电容的下极板；190-零线；20-采样电路；30-调制器。
具体实施方式
为了使本申请的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本申请进行进一步详细说明。应当理解，此处所描述的具体实施例仅 仅用以解释本申请，并不用于限定本申请。
本申请提供一种不连续脉宽调制方法。需要说明的是，本申请提供的不连续脉宽调制方法应用于三相逆变电路。具体地，本申请提供的不连续脉宽调制方法可以应用于三相两电平的电路，也可以应用于三相三电平的电路。图3为本申请一实施例提供的三相逆变器调制电路的电路图，本申请提供的不连续脉宽调制方法应用于图3所示的三相逆变器调制电路。图3所示的三相逆变器调制电路是T型三相三电平的电路。
此外，本申请提供的不连续脉宽调制方法不限制其执行主体。可选地，本申请提供的不连续脉宽调制方法的执行主体可以为一种三相逆变器调制电路中的调制器。所述调制器可以包括一个或多个控制器。
如图1所示，在本申请的一实施例中，所述不连续脉宽调制方法包括如下S100至S400：
S100，生成三相电压原始信号波。
具体地，三相电压原始信号波是一种正弦波，图6，图8，图10，图12，图14，图16，图18，图20和图22中的虚线部分均是三相电压原始信号波中A相的波形。
图10，图16和图22中的虚线部分是本申请提供的不连续脉宽调制方法的三相电压原始信号波中A相的波形。横坐标是时间，纵坐标是电压值进行过归一化后生成的幅值。
首先调制器控制采样电路获取直流母线电压，三相电压和三相电流，采样电路在采样结束后，将直流母线电压，三相电压和三相电流输入至调制器。进一步地，调制器依据直流母线电压，三相电压和三相电流，进行计算，生成三相电压原始信号波。
三相电压原始信号波由基波和一部分谐波叠加而成。
S200，将三相电压原始信号波加入一个零序分量，得到加入零序分量的信号波。
具体地，加入零序分量的信号波后续作为调制波。图11，图17和图23中的实线部分是本申请提供的不连续脉宽调制方法的三相电压原始信号波中A相的调制波波形，虚线部分是零序分量的波形。横坐标是时间，纵坐标是电压值进行过归一化后生成的幅值。
S300，将加入零序分量的信号波作为调制波与高频的三角载波作比较，生成每一个IGBT器件的开关管驱动信号。
具体地，本申请提供的不连续脉宽调制方法应用到三相三电平电路时，调制波是与两个高频的三角载波分别做比较，两个高频的三角载波一个是幅值0到1的高频三角载波，另一个是幅值-1到0的高频三角载波。这里提到的幅值都是电压值经过归一化处理的幅值。高频的频率范围指的是频率大于等于3kHz 且小于等于50kHz的频率范围。
S400，依据每一个IGBT器件的开关管驱动信号控制每一个IGBT器件工作。
本申请涉及一种不连续脉宽调制方法，一方面，相对于传统SVPWM调制方法降低了开关动作次数，减小了功耗，提高了开关管工作效率，另一方面，相对于传统DPWM0调制方法大大提升了稳定性，在调制比不断变化时仍然能够稳定工作，兼顾了功耗低和稳定性高两个优点，且本申请提供的不连续脉宽调制方法并不需要进行不同调制方式之间的切换。
在本申请的一实施例中，所述S100包括如下S110至S120：
S110，生成三相电压原始信号波，三相电压原始信号波中每一相的电压值如公式1所示。
Figure PCTCN2022098699-appb-000001
其中，m为调制比。θ为矢量角。Ua为A相电压值。Ub为B相电压值。Uc为C相电压值。Uah为A相的谐波分量。，Ubh为B相的谐波分量。Uch为C相的谐波分量。
具体地，本步骤对每一相的电压值Ua，Ub，Uc进行了归一化处理，具体是每一相的电压值除以二分之一直流母线电压值，将每一相的电压值转化成-1到1之间的一个数值，这个数值在本申请中称为幅值，便于后续与高频的三角载波作比较。归一化处理后，载波是幅值为1的连续三角波，因此m是三相电压的幅值，也是调制比。
S120，获取三相电压最大值和三相电压最小值，三相电压最大值和三相电压最小值的定义方式如公式2所示。
Figure PCTCN2022098699-appb-000002
其中，Umax为三相电压最大值。Umin为三相电压最小值。Ua为A相电压值。Ub为B相电压值。Uc为C相电压值。
具体地，本步骤是分别取三相电压最大值和三相电压最小值。
在本申请的一实施例中，所述S200包括如下S210：
S210，定义零序分量。所述零序分量的表达式如公式3所示。
U z＝-kU max-(1-k)U min+(2k-1)  公式3
其中，Uz为零序分量。Umax为三相电压最大值。Umin为三相电压最小值。k为零序分量计算系数。
具体地，本申请提供的不连续脉宽调制方法，k可以取0，也可以取1，k也可以在0到1之间交替变化。
在本申请的一实施例中，所述S200还包括：
S220，向三相电压正弦原始信号波中每一相的电压值均加入所述零序分量，得到将加入零序分量的信号波。加入零序分量的信号波的每一相的电压值如公式4所示。
Figure PCTCN2022098699-appb-000003
其中，Uas为加入零序分量后的A相电压值。Ubs为加入零序分量后的B相电压值。Ucs为加入零序分量后的C相电压值。
具体地，本申请通过在原始信号波的基础上叠加零序分量来生成调制波。
在本申请的一实施例中，所述S300包括如下S310至S360：
S310，当矢量角位于第一角度范围时，控制零序分量计算系数取0。所述第一角度范围为大于等于0加β且小于等于三分之一π减β的角度范围。
S320，当矢量角位于第二角度范围时，控制零序分量计算系数取0。所述第二角度范围为大于等于三分之二π加β且小于等于π减β的角度范围。
S330，当矢量角位于第三角度范围时，控制零序分量计算系数取0。所述第三角度范围为大于等于三分之四π加β且小于等于三分之五π减β的角度范围。
S340，当矢量角位于第四角度范围时，控制零序分量计算系数取1。所述第四角度范围为大于等于三分之一π加β且小于等于三分之二π减β的角度范围。
S350，当矢量角位于第五角度范围时，控制零序分量计算系数取1。所述第五角度范围为大于等于π加β且小于等于三分之四π减β的角度范围。
S360，当矢量角位于第六角度范围时，控制零序分量计算系数取1。所述第六角度范围为大于等于三分之五π加β且小于等于0减β的角度范围。β为预设角度值。
具体地，图10为调制比为1时，本申请提供的一种不连续脉宽调制方法的 A相的原始信号波和第一IGBT器件的开关管驱动信号的复合波形图。图10包含了两种波形，一种波形是A相的原始信号波，即图3中X点和零线之间的波形(图10中虚线正弦曲线部分)。另一种波形是第一IGBT器件的开关管驱动信号的波形(图10中实线方波部分)。
在图10中，原始信号波的幅值在60度到120度为1，对应第一IGBT器件的驱动恒导通。也即在180度内累计有60度的时间开关是不动作的，因此能够减小损耗。
下面解释下调制波与高频的三角载波作比较的过程。
下述内容都是以A相为例，因为本申请中展示的图6至图23都是关于A相的波形图。
如图3所示，第一IGBT器件，第二IGBT器件，第三IGBT器件和第四IGBT器件组成了A相的桥臂，第一IGBT器件作为A相上桥臂开关管，第二IGBT器件作为A相下桥臂开关管，第三IGBT器件和第四IGBT器件作为A相的两个中间桥臂开关管。本申请S300中调制波与高频的三角载波作比较，最终生成每一个IGBT器件的开关管驱动信号，实际上就是生成第一IGBT器件的开关管驱动信号，第二IGBT器件的开关管驱动信号，第三IGBT器件的开关管驱动信号和第四IGBT器件的开关管驱动信号，即最终生成一共四个开关管驱动信号。
那么下面详细阐述如何通过调制波与高频的三角载波作比较得到这四个开关管驱动信号的过程。
如图4所示，虚线为A相调制波，由于本实施例应用于三相三电平电路，那么是将调制波分别与幅值0到1的高频三角波和幅值-1到0的高频三角波做对比。
图4中的虚线正弦波为调制波的波形，实线三角波为幅值在0至1区间的高频三角波，虚线三角波为幅值在-1至0区间的高频三角波。
如图4所示，求取第一IGBT器件的开关管驱动信号和第三IGBT器件的开关管驱动信号的方法，是将调制波的波形和幅值在0至1区间的高频三角波做比较。在同一时间节点，当调制波的幅值大于0至1区间的高频三角波的幅值时，此时第一IGBT器件的开关管驱动信号输出高电平。当调制波的幅值小于0至1区间的高频三角波的幅值时，此时第一IGBT器件的开关管驱动信号输出低电平。
第三IGBT器件与第一IGBT器件互补，即在同一时间节点，当调制波的幅值大于0至1区间的高频三角波的幅值时，此时第三IGBT器件的开关管驱动信号输出低电平。当调制波的幅值小于0至1区间的高频三角波的幅值时，此时第三IGBT器件的开关管驱动信号输出高电平。
如图4所示，求取第二IGBT器件的开关管驱动信号和第四IGBT器件的开关管驱动信号的方法，是将调制波的波形和幅值在-1至0区间的高频三角波做 比较。在同一时间节点，当调制波的幅值大于-1至0区间的高频三角波的幅值时，此时第四IGBT器件的开关管驱动信号输出高电平。当调制波的幅值小于-1至0区间的高频三角波的幅值时，此时第四IGBT器件的开关管驱动信号输出低电平。
第二IGBT器件与第四IGBT器件互补，即在同一时间节点，当调制波的幅值大于-1至0区间的高频三角波的幅值时，此时第二IGBT器件的开关管驱动信号输出低电平。当调制波的幅值小于-1至0区间的高频三角波的幅值时，此时第二IGBT器件的开关管驱动信号输出高电平。
通过是上述方法，最终得到图5，图5中的四张波形图分别展示了第一IGBT器件的开关管驱动信号的波形图，第三IGBT器件的开关管驱动信号的波形图，第四IGBT器件的开关管驱动信号的波形图，以及第二IGBT器件的开关管驱动信号的波形图。他们都是呈现方波状。
图5中的第一IGBT器件的开关管驱动信号的波形图，其实就是图10中的方波波形的部分，因此可以知晓图10中的方波波形是通过图5得到的，也即通过调制波和高频三角波比较得到的，以此类推，其他图中的方波波形也是使用上述方法得到，后文不再赘述。
DPWM0调制方法这种固定在某个60度范围内保持开关管不动作的方式可以减小损耗，但是也会导致其在这60度范围内零序分量往正或是往负偏，如图9所示。特别是调制比变小的时候，会导致正负母线偏差变大，也是其输出特性变差的原因。
而本申请提供的不连续脉宽调制方法，β可以设置为固定值，也可以是变化值。本实施例β设置成随调制比m而变化，调制比越大，β越小。如图10所示，在过零处附近和峰值处附近会使开关不做开关切换，即在过零处第一IGBT器件保持断开，在峰值处第一IGBT器件保持闭合。且对比图10，图16，图22，调制比m分别为1，0.75和0.5，可以理解，本申请提供的不连续脉宽调制方法中，调制比m越大，第一IGBT器件在峰值处保持闭合的时间就越长，过零点附近保持关断的时间就越短。调制比m越小，第一IGBT器件在峰值处保持闭合的时间就越短，过零点附近保持关断的时间就越长。
而图8，图14，图20的传统DPWM0调制方法在不同调制比的条件下，在正半波第一IGBT器件维持导通的时间几乎维持不变，其代价就是低调制比下迅速增大的零序分量。而零序分量过大会导致正负母线偏差增大，使得逆变器输出特性变差，稳定性降低。
如图9，图15，图21，传统DPWM0调制方案的零序分量的最大值随着调制比的减小，迅速增加，并且会直接从正的最大值切换到负的最大值，变化剧烈。
在调制比m为1的时候，尚能接受，调制比一旦下降，锯齿状越来越明显，在调制比m到0.5的时候，如图21，稳定性已经近似崩溃。
本申请提供的不连续脉宽调制方法则改善了零序分量的最大值，并且零序分量的变化也比较缓和。
参见图6，图12，图18是传统SVPWM调制方法的原始信号波和第一IGBT器件的开关管驱动信号复合波形图，第一IGBT器件在正半波没有任何持续导通的时间段，就会导致开关状态频繁切换，工作效率低，能耗高。而通过本申请图10，图16，图22可以看出，明显本申请的不连续脉宽调制方法相对于传统SVPWM调制方法，开关状态切换次数减少了传统SVPWM调制方法开关状态切换次数的三分之一，大大降低了功耗，而对比传统SVPWM调制方法的图7，图11和图17和本申请图11，图17，图23，本申请的不连续脉宽调制方法稳定性相比传统SVPWM调制方法，也没有落后，本申请的稳定性也很高。
因此传统技术必须在调制比高的时候使用DPWM0调制方法，在调制比低的时候使用SVPWM方法，但是频繁的切换调制方法会导致三相逆变器电路产生振荡。而且需要提前设计2套不同的调制方案，极为麻烦。
而本申请只需要应用一套调制方案，从头调制到尾，不需要切换调制方案，兼有功耗低和稳定性高两个优点。
在本申请的一实施例中，所述S300还包括：
当矢量角位于360度中除所述第一角度范围、第二角度范围、第三角度范围、第四角度范围、第五角度范围和第六角度范围之外的其他角度范围时，控制零序分量满足公式5。
Uz＝Umax+Umin    公式5
其中，Uz为零序分量。Umax为三相电压最大值。Umin为三相电压最小值。
当矢量角位于360度中除所述第一角度范围、第二角度范围、第三角度范围、第四角度范围、第五角度范围和第六角度范围之外的其他角度范围时，控制零序分量等于Umax和Umin的和。因为零序分量为该值时，可以恰好使得过零点附近的调制波幅值大小为0，即让第一IGBT器件保持断开状态。
在本申请的一实施例中，β为预设固定值。
具体地，β的选取有多种实施方式，可以选取为一个固定数值。β值越大，代表过零点附近(即正半波的0度和180度)第一IGBT器件维持断开的角度范围越大，同时90度附近维持导通的角度范围越小。也即开关管状态不切换的区间有三个：0度到β，三分之一π+β到三分之一π-β，π-β到π。三个区间总和的IGBT器件开关不动作角度区间长度是三分之一π。传统DPWM0的IGBT器件开不动作区间为三分之一π到三分之一π。即两者的开关不动作角度区间长度是一样的都是三分之一π。
表现在零序分量上，即β值越大，零序分量的最大值会变小。β可以在0到六分之一π之间设置。设置成0度，则变成传统的DPWM0调制方案。设置成六分之一π，则意味着固定在0到六分之一π和六分之五π到π之间，第一IGBT 器件维持关断，同时90度附近第一IGBT器件失去维持导通的角度区间。
在本申请的一实施例中，β为变量，且当调制比越大时控制β的角度数值越小，当调制比越小时控制β的角度数值越大。
具体地，β的选取有多种实施方式，可以随着k调制比的变化而变化，本实施例就是随着k调制比的变化而变化。
β值之所以要随调制比变化，一方面在调制比较小时，β值应当要大一些，以获得较小的零序分量。另一方面在调制比较大时，β值应当要小一些，这样可以让90度附近开关管维持导通的时间能够尽量增加。因为一般情况下电流越大，开关动作切换一次的损耗越大。90度附近一般电流较大，零点附近一般电流较小，虽然90度附近加零点附近开关不动作的角度总和是60度不变，90度附近不动作占比更大，更有利于减小损耗。
本申请还提供一种三相逆变器调制电路。
为了行文简洁，本申请只在三相逆变器调制电路的各个实施例中对各个元器件进行标号，而不对不连续脉宽调制方法的各个实施例中出现的同名称元器件进行标号。
如图2所示，在本申请的一实施例中，所述三相逆变器调制电路包括三相逆变电路10、采样电路20和调制器30。
采样电路20与所述三相逆变器电路10电连接。所述调制器30与所述三相逆变器电路10电连接，所述调制器30还与所述采样电路20电连接。所述调制器30用于执行前述任意一个实施例中提及的不连续脉宽调制方法。
具体地，本实施例介绍的是T型三相三电平的电路。
当然，本申请提供的不连续脉宽调制方法也可以应用于三相两电平的电路，或中点钳位三电平逆变器电路，本申请只介绍T型三相三电平电路的电路结构，对其他电路的结构不进行赘述。
如图3所示，在本申请的一实施例中，所述三相逆变器电路10包括直流母线110、零线190、第一IGBT器件121、第二IGBT器件122、第三IGBT器件123、第四IGBT器件124、第五IGBT器件131、第六IGBT器件132、第七IGBT器件133、第八IGBT器件134、第九IGBT器件141、第十IGBT器件142、第十一IGBT器件143、第十二IGBT器件144、第一滤波电感151、第二滤波电感152、第三滤波电感153、第一滤波电容160、第二滤波电容170和第三滤波电容180。
所述直流母线110上设置有正母线电容111和负母线电容112。正母线电容111和负母线电容112之间的连接链路上设置有W点。W点与零线连接。
第一IGBT器件121的一端与正母线电容111连接。第一IGBT器件121的另一端与第二IGBT器件122连接。第二IGBT器件122的一端与第一IGBT器件121连接，第二IGBT器件122的另一端与负母线电容112连接。第三IGBT器件 123的一端连接于W点，第三IGBT器件123的另一端与第四IGBT器件124连接。第四IGBT器件124的一端与第三IGBT器件123连接，第四IGBT器件124的另一端与第一滤波电感151连接。
第一滤波电感151的一端与第四IGBT器件124连接，第一滤波电感151的另一端连接A相电压输出点。
第五IGBT器件131的一端与正母线电容111连接，第五IGBT器件131的另一端与第六IGBT器件132连接。第六IGBT器件132的一端与第五IGBT器件131连接，第六IGBT器件132的另一端与负母线电容112连接。第七IGBT器件133的一端连接于W点，第七IGBT器件133的另一端与第八IGBT器件134连接。第八IGBT器件134的一端与第七IGBT器件133连接，第八IGBT器件134的另一端与第二滤波电感152连接。
第二滤波电感152的一端与第八IGBT器件134连接，第二滤波电感152的另一端连接B相电压输出点。
第九IGBT器件141的一端与正母线电容111连接，第九IGBT器件141的另一端与第十IGBT器件142连接。第十IGBT器件142的一端与第九IGBT器件141连接，第十IGBT器件142的另一端与负母线电容112连接。第十一IGBT器件143的一端连接于W点，第十一IGBT器件143的另一端与第十二IGBT器件144连接。第十二IGBT器件144的一端与第十一IGBT器件143连接，第十二IGBT器件144的另一端与第三滤波电感153连接。
第三滤波电感153的一端与第十二IGBT器件144连接，第三滤波电感153的另一端连接C相电压输出点。
第一IGBT器件121，第二IGBT器件122，第四IGBT器件124和第一滤波电感151连接于同一点X点。第五IGBT器件131，第六IGBT器件132，第八IGBT器件134和第二滤波电感152连接于同一点Y点。第九IGBT器件141，第十IGBT器件142，第十二IGBT器件144和第三滤波电感153连接于同一点Z点。
所述三相逆变器电路10还包括第一滤波电容160、第二滤波电容170和第三滤波电容180。
第一滤波电容160，第一滤波电容160的上极板161连接于第一滤波电感151与A相电压输出点之间的连接链路上。
所述第一滤波电容160的下级板162连接于零线190。第二滤波电容170，第二滤波电容170的上极板171连接于第二滤波电感152与B相电压输出点之间的连接链路上。第二滤波电容170的下极板172连接于零线190。
第三滤波电容180，第三滤波电容180的上极板181连接于第三滤波电感153与C相电压输出点之间的连接链路上，第三滤波电容180的下极板182连接于零线190。
前述任意一个实施例中所提及的不连续脉宽调制方法用于生成X点至零线 之间的调制波，并调制生成第一IGBT器件121的开关管驱动信号、第二IGBT器件122的开关管驱动信号、第三IGBT器件123的开关管驱动信号和第四IGBT器件124的开关管驱动信号。
前述任意一个实施例中所提及的不连续脉宽调制方法还用于生成Y点至零线之间的调制波，并调制生成第五IGBT器件131的开关管驱动信号、第六IGBT器件132的开关管驱动信号、第七IGBT器件133的开关管驱动信号和第八IGBT器件的开关管驱动信号。
前述任意一个实施例中所提及的不连续脉宽调制方法还用于生成Z点至零线之间的调制波，并调制生成第九IGBT器件141的开关管驱动信号、第十IGBT器件142的开关管驱动信号、第十一IGBT器件143的开关管驱动信号和第十二IGBT器件144的开关管驱动信号。
具体地，三相逆变器电路10的开关管可以选择IGBT器件也可以选择MOSFET器件，图3中绘制的反并二极管是IGBT器件器件的体二极管，是IGBT器件的一部分。如果选MOSFET器件则可以额外再反并一个二极管。使用任何类型的开关管器件并不会影响本申请的保护范围。
正母线电容111两端具有正母线电压Vbus，负母线电容112两端具有负母线电压-Vbus，正母线电容111和负母线电容112之间的W点是零点。
如图3所示，第一IGBT器件121，第二IGBT器件122，第三IGBT器件123和第四IGBT器件124组成了A相的桥臂。第一IGBT器件121作为A相上桥臂开关管，第二IGBT器件122作为A相下桥臂开关管。第三IGBT器件123和第四IGBT器件124作为A相的两个中间桥臂开关管。
第五IGBT器件131，第六IGBT器件132，第七IGBT器件133和第八IGBT器件134组成了B相的桥臂。第五IGBT器件131作为B相上桥臂开关管，第六IGBT器件132作为B相下桥臂开关管。第七IGBT器件133和第八IGBT器件134作为B相的两个中间桥臂开关管。
第九IGBT器件141，第十IGBT器件142，第十一IGBT器件143和第十二IGBT器件144组成了C相的桥臂。第九IGBT器件141作为C相上桥臂开关管，第十IGBT器件142作为C相下桥臂开关管。第十一IGBT器件143和第十二IGBT器件144作为C相的两个中间桥臂开关管。
每一相桥臂中的上桥臂开关管和下桥臂开关管为互补导通的关系。
以A相桥臂举例说明。可以理解，IGBT器件的开关管驱动信号输出高电平就代表IGBT器件导通，IGBT器件的开关管驱动信号输出低电平代表IGBT器件导通。那么X点的电压情况有三种：
当第一IGBT器件121导通，第二IGBT器件122断开，第三IGBT器件123和第四IGBT器件124均断开时，X点的电压为正的Vbus。
当第一IGBT器件121断开，第二IGBT器件122断开，第三IGBT器件123 和第四IGBT器件124均导通时，X点的电压为0。
当第一IGBT器件121断开，第二IGBT器件122导通，第三IGBT器件123和第四IGBT器件124均断开时，X点的电压为负的Vbus。
本申请图10，图16，图22示出的的开关管驱动信号的方波波形显示出的正半波只有正的Vbus和0，负半波只有负的Vbus和0。
关于本申请提及到的“持续”，需要说明的是，只要一个周期内出现了一次开关状态的切换，即可以认为没有持续的关断，也没有持续的导通。一个周期可以设定为20微秒到1毫秒。
在图6至图23中，横坐标都是时间，纵坐标均为归一化处理之后的电压幅值。
以上所述实施例的各技术特征可以进行任意的组合，各方法步骤也并不做执行顺序的限制，为使描述简洁，未对上述实施例中的各个技术特征所有可能的组合都进行描述，然而，只要这些技术特征的组合不存在矛盾，都应当认为是本说明书记载的范围。
以上所述实施例仅表达了本申请的几种实施方式，其描述较为具体和详细，但并不能因此而理解为对本申请专利范围的限制。应当指出的是，对于本领域的普通技术人员来说，在不脱离本申请构思的前提下，还可以做出若干变形和改进，这些都属于本申请的保护范围。因此，本申请的保护范围应以所附权利要求为准。

Claims (10)

  1. 一种不连续脉宽调制方法，其特征在于，应用于三相逆变电路，所述方法包括：
    生成三相电压原始信号波；
    将三相电压原始信号波加入一个零序分量，得到加入零序分量的信号波；
    将加入零序分量的信号波作为调制波与高频的三角载波作比较，生成每一个IGBT器件的开关管驱动信号；
    依据每一个IGBT器件的开关管驱动信号控制每一个IGBT器件工作。
  2. 根据权利要求1所述的不连续脉宽调制方法，所述生成三相电压原始信号波，包括：
    生成三相电压原始信号波，所述三相电压原始信号波中每一相的电压值如公式1所示；
    Figure PCTCN2022098699-appb-100001
    其中，m为调制比，θ为矢量角，Ua为A相电压值，Ub为B相电压值，Uc为C相电压值，Uah为A相的谐波分量，Ubh为B相的谐波分量，Uch为C相的谐波分量；
    获取三相电压最大值和三相电压最小值，三相电压最大值和三相电压最小值的定义方式如公式2所示；
    Figure PCTCN2022098699-appb-100002
    其中，Umax为三相电压最大值，Umin为三相电压最小值，Ua为A相电压值，Ub为B相电压值，Uc为C相电压值。
  3. 根据权利要求2所述的不连续脉宽调制方法，所述将三相电压原始信号波加入一个零序分量，得到加入零序分量的信号波，包括：
    定义零序分量，所述零序分量的表达式如公式3所示；
    U z＝-kU max-(1-k)U min+(2k-1)  公式3；
    其中，Uz为零序分量，Umax为三相电压最大值，Umin为三相电压最小值，k为零序分量计算系数。
  4. 根据权利要求3所述的不连续脉宽调制方法，所述将三相电压原始信号 波加入一个零序分量，得到加入零序分量的信号波，还包括：
    向三相电压原始信号波中每一相的电压值均加入所述零序分量，得到加入零序分量的信号波；加入零序分量的信号波的每一相的电压值如公式4所示；
    Figure PCTCN2022098699-appb-100003
    其中，Uas为加入零序分量后的A相电压值，Ubs为加入零序分量后的B相电压值，Ucs为加入零序分量后的C相电压值。
  5. 根据权利要求4所述的不连续脉宽调制方法，其特征在于，所述将加入零序分量的信号波作为调制波与高频的三角载波作比较，生成每一个IGBT器件的开关管驱动信号，包括：
    当矢量角位于第一角度范围时，控制零序分量计算系数取0；所述第一角度范围为大于等于0加β且小于等于三分之一π减β的角度范围；
    当矢量角位于第二角度范围时，控制零序分量计算系数取0；所述第二角度范围为大于等于三分之二π加β且小于等于π减β的角度范围；
    当矢量角位于第三角度范围时，控制零序分量计算系数取0；所述第三角度范围为大于等于三分之四π加β且小于等于三分之五π减β的角度范围；
    当矢量角位于第四角度范围时，控制零序分量计算系数取1；所述第四角度范围为大于等于三分之一π加β且小于等于三分之二π减β的角度范围；
    当矢量角位于第五角度范围时，控制零序分量计算系数取1；所述第五角度范围为大于等于π加β且小于等于三分之四π减β的角度范围；
    当矢量角位于第六角度范围时，控制零序分量计算系数取1；所述第六角度范围为大于等于三分之五π加β且小于等于0减β的角度范围；β为预设角度值。
  6. 根据权利要求5所述的不连续脉宽调制方法，其特征在于，所述将加入零序分量的信号波作为调制波与高频的三角载波作比较，生成每一个IGBT器件的开关管驱动信号，还包括：
    当矢量角位于360度中除所述第一角度范围、第二角度范围、第三角度范围、第四角度范围、第五角度范围和第六角度范围之外的其他角度范围时，控制零序分量满足公式5；
    Uz＝Umax+Umin  公式5；
    其中，Uz为零序分量，Umax为三相电压最大值，Umin为三相电压最小值。
  7. 根据权利要求6所述的不连续脉宽调制方法，其特征在于，β为预设固定值。
  8. 根据权利要求6所述的不连续脉宽调制方法，其特征在于，β为变量，且当调制比越大时控制β的角度数值越小，当调制比越小时控制β的角度数值越大。
  9. 一种三相逆变器调制电路，其特征在于，包括：
    三相逆变器电路；
    采样电路，与所述三相逆变器电路电连接；
    调制器，所述调制器与所述三相逆变器电路电连接，所述调制器还与所述采样电路电连接，所述调制器用于执行如权利要求1-8中任意一项所述的不连续脉宽调制方法。
  10. 根据权利要求9所述的不连续脉宽调制方法，其特征在于，所述三相逆变器电路包括直流母线、零线、第一IGBT器件、第二IGBT器件、第三IGBT器件、第四IGBT器件、第五IGBT器件、第六IGBT器件、第七IGBT器件、第八IGBT器件、第九IGBT器件、第十IGBT器件、第十一IGBT器件、第十二IGBT器件、第一滤波电感、第二滤波电感、第三滤波电感、第一滤波电容、第二滤波电容和第三滤波电容；
    直流母线，所述直流母线上设置有正母线电容和负母线电容；正母线电容和负母线电容之间的连接链路上设置有W点；
    零线，连接于W点；
    第一IGBT器件，一端与正母线电容连接，另一端与第二IGBT器件连接；
    第二IGBT器件，一端与第一IGBT器件连接，另一端与负母线电容连接；
    第三IGBT器件，一端连接于W点，另一端与第四IGBT器件连接；
    第四IGBT器件，一端与第三IGBT器件连接，另一端与第一滤波电感连接；
    第一滤波电感，一端与第四IGBT器件连接，另一端连接A相电压输出点；
    第五IGBT器件，一端与正母线电容连接，另一端与第六IGBT器件连接；
    第六IGBT器件，一端与第五IGBT器件连接，另一端与负母线电容连接；
    第七IGBT器件，一端连接于W点，另一端与第八IGBT器件连接；
    第八IGBT器件，一端与第七IGBT器件连接，另一端与第二滤波电感连接；
    第二滤波电感，一端与第八IGBT器件连接，另一端连接B相电压输出点；
    第九IGBT器件，一端与正母线电容连接，另一端与第十IGBT器件连接；
    第十IGBT器件，一端与第九IGBT器件连接，另一端与负母线电容连接；
    第十一IGBT器件，一端连接于W点，另一端与第十二IGBT器件连接；
    第十二IGBT器件，一端与第十一IGBT器件连接，另一端与第三滤波电感连接；
    第三滤波电感，一端与第十二IGBT器件连接，另一端连接C相电压输出点；
    第一IGBT器件，第二IGBT器件，第四IGBT器件和第一滤波电感连接于同一点X点；
    第五IGBT器件，第六IGBT器件，第八IGBT器件和第二滤波电感连接于同一点Y点；
    第九IGBT器件，第十IGBT器件，第十二IGBT器件和第三滤波电感连接于同一点Z点；
    所述三相逆变器电路还包括：
    第一滤波电容，第一滤波电容的上极板连接于第一滤波电感与A相电压输出点之间的连接链路上；所述第一滤波电容的下级板连接于零线；
    第二滤波电容，第二滤波电容的上极板连接于第二电滤波感与B相电压输出点之间的连接链路上；第二滤波电容的下极板连接于零线；
    第三滤波电容，第三滤波电容的上极板连接于第三滤波电感与C相电压输出点之间的连接链路上，第三滤波电容的下极板连接于零线；
    如权利要求1-8中任意一项所述的不连续脉宽调制方法用于生成X点至零线之间的调制波，并调制生成第一IGBT器件的开关管驱动信号、第二IGBT器件的开关管驱动信号、第三IGBT器件的开关管驱动信号和对第四IGBT器件的开关管驱动信号；
    如权利要求1-8中任意一项所述的不连续脉宽调制方法还用于生成Y点至零线之间的调制波，并调制生成第五IGBT器件的开关管驱动信号、第六IGBT器件的开关管驱动信号、第七IGBT器件的开关管驱动信号和第八IGBT器件的开关管驱动信号；
    如权利要求1-8中任意一项所述的不连续脉宽调制方法还用于生成Z点至零线之间的调制波，并调制生成第九IGBT器件的开关管驱动信号、第十IGBT器件的开关管驱动信号、第十一IGBT器件的开关管驱动信号和第十二IGBT器件的开关管驱动信号。
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