WO2023157185A1 - Circuit d'attaque de grille et dispositif de conversion de puissance - Google Patents

Circuit d'attaque de grille et dispositif de conversion de puissance Download PDF

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Publication number
WO2023157185A1
WO2023157185A1 PCT/JP2022/006413 JP2022006413W WO2023157185A1 WO 2023157185 A1 WO2023157185 A1 WO 2023157185A1 JP 2022006413 W JP2022006413 W JP 2022006413W WO 2023157185 A1 WO2023157185 A1 WO 2023157185A1
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WIPO (PCT)
Prior art keywords
gate
semiconductor switching
switching element
drive circuit
source
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PCT/JP2022/006413
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English (en)
Japanese (ja)
Inventor
翔太 森崎
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三菱電機株式会社
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Priority to PCT/JP2022/006413 priority Critical patent/WO2023157185A1/fr
Publication of WO2023157185A1 publication Critical patent/WO2023157185A1/fr

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present disclosure relates to a gate drive circuit that drives semiconductor switching elements, and a power conversion device that includes the gate drive circuit.
  • a power conversion device such as an inverter device, a servo amplifier device, or a switching power supply device includes a power conversion main circuit having one or more semiconductor switching elements.
  • the semiconductor switching element In the semiconductor switching element, the conduction state between the drain main terminal and the source main terminal changes according to the electric signal applied between the gate terminal and the source main terminal.
  • the gate drive circuit receives a command signal from a higher-level controller and applies an electric signal between the gate terminal and the source main terminal of the semiconductor switching element to drive the semiconductor switching element.
  • Patent Document 1 describes detecting a voltage generated in an inductance that may exist between a source control terminal and a source main terminal of a semiconductor switching element, and varying a gate driving voltage or a gate driving resistance based on the detected value.
  • a gate drive circuit is disclosed that provides control to allow
  • Damage to the element can be dealt with by slowing the switching speed, but in that case the switching loss will increase. That is, there is a trade-off relationship between surge and noise and switching loss.
  • Patent Literature 1 By using the method of Patent Document 1, it is possible to drive semiconductor switching elements at high speed. However, the technique of Patent Literature 1 has a problem that the surge, noise, and switching loss, which are in a trade-off relationship, cannot be optimized.
  • the present disclosure has been made in view of the above, and aims to obtain a gate drive circuit capable of driving a semiconductor switching element by optimizing the surge and noise, which are in a trade-off relationship, and switching loss.
  • the gate drive circuit drives a semiconductor device module having a drain main terminal, a gate terminal, a source main terminal, and first and second source terminals. It is a gate drive circuit.
  • a semiconductor switching element has a gate electrode, a drain electrode and a source electrode.
  • the drain main terminal is connected to the drain electrode and the gate terminal is connected to the gate electrode.
  • the source main terminal and the first source terminal are connected to the source electrode, and the second source terminal is connected to the source main terminal.
  • the gate drive circuit includes a gate driver, a feedback section, and a feedback intensity adjustment section. The gate driver applies an electric signal between the gate terminal and the source main terminal to gate-drive the semiconductor switching element.
  • the feedback unit feeds back to the gate driver an electromotive force induced between the first source terminal and the second source terminal by a main circuit current flowing between the drain main terminal and the source main terminal.
  • the feedback strength adjustment section is configured to be able to individually adjust the feedback strength, which is the strength of the voltage fed back from the feedback section to the gate driver, when the semiconductor switching element is turned on and when it is turned off.
  • the semiconductor switching element can be driven by optimizing the surge, noise, and switching loss, which are in a trade-off relationship.
  • FIG. 1 is a diagram showing a configuration example of a power conversion device including a gate drive circuit according to Embodiment 1;
  • FIG. 2 shows a detailed configuration of the gate drive circuit of the first embodiment together with semiconductor switching elements to be driven;
  • FIG. 4 is a diagram for explaining the operation of the semiconductor switching element when the gate drive circuit according to the first embodiment is turned on;
  • FIG. 4 shows an example of operation waveforms when the semiconductor switching element is turned on by the gate drive circuit of the first embodiment;
  • FIG. 4 is a diagram for explaining the operation of the semiconductor switching element when the gate drive circuit according to the first embodiment is turned off;
  • FIG. 4 shows an example of operation waveforms when the semiconductor switching element is turned off by the gate drive circuit of the first embodiment;
  • FIG. 4 is a diagram for explaining the effect of using the feedback intensity adjustment unit of the first embodiment;
  • FIG. 10 is a diagram showing the detailed configuration of the gate drive circuit of the second embodiment together with semiconductor switching elements to be driven;
  • connection includes both cases in which components are directly connected to each other and cases in which components are indirectly connected to each other via other components.
  • FIG. 1 is a diagram showing a configuration example of a power conversion device 1 including a gate drive circuit 3 according to Embodiment 1.
  • FIG. 2 is a circuit diagram showing a configuration example of the inverter circuit 2 shown in FIG.
  • FIG. 3 is a diagram showing the detailed configuration of the gate drive circuit 3 of the first embodiment together with the semiconductor switching element 6a to be driven.
  • the power conversion device 1 includes an inverter circuit 2, a gate drive circuit 3, and a control section 4.
  • a DC power supply 50 is connected to an input terminal of the inverter circuit 2 .
  • the DC power supply 50 is a supply source of DC power for applying a DC voltage to the inverter circuit 2, and corresponds to a power supply device, a converter, a power capacitor, and the like.
  • the inverter circuit 2 is a power conversion circuit that converts the DC power supplied from the DC power supply 50 into AC power.
  • the inverter circuit 2 has at least one semiconductor switching element 6a.
  • An example of the semiconductor switching element 6a is a metal-oxide-semiconductor field effect transistor (Metal-Oxide-Semiconductor Field-Effect Transistor: MOSFET) as shown in FIG.
  • MOSFET Metal-Oxide-Semiconductor Field-Effect Transistor
  • anti-parallel is a connection form in which the anode of the diode is connected to the source of the MOSFET and the cathode of the diode is connected to the drain of the MOSFET.
  • FIG. 1 exemplifies a MOSFET as the semiconductor switching element 6a, but the present invention is not limited to this.
  • An insulated gate bipolar transistor (IGBT) may be used instead of the MOSFET.
  • a motor 52 as a load is connected to the output terminal of the inverter circuit 2 .
  • the control unit 4 generates a control signal CS for controlling the semiconductor switching element 6a and outputs it to the gate drive circuit 3.
  • the gate drive circuit 3 generates a drive signal GS for driving the semiconductor switching element 6a based on the control signal CS and outputs the drive signal GS to the inverter circuit 2 .
  • the semiconductor switching element 6a performs a switching operation according to the drive signal GS, converts the DC power supplied from the DC power supply 50 into AC power, and drives the motor 52.
  • the inverter circuit 2 has legs 6A, 6B and 6C, as shown in FIG. Leg 6A, leg 6B, and leg 6C are connected in parallel with each other between DC bus 16 and DC bus 17 .
  • the leg 6A is a series circuit portion in which a U-phase upper arm semiconductor switching element 6UP and a U-phase lower arm semiconductor switching element 6UN are connected in series.
  • the leg 6B is a series circuit portion in which a V-phase upper arm semiconductor switching element 6VP and a V-phase lower arm semiconductor switching element 6VN are connected in series.
  • the leg 6C is a series circuit portion in which the W-phase upper arm semiconductor switching element 6WP and the W-phase lower arm semiconductor switching element 6WN are electrically connected in series. That is, the inverter circuit 2 is a bridge circuit including three legs that are series circuit portions.
  • motor 52 which is the load, is a three-phase motor in FIGS. 1 and 2, it is not limited to this.
  • Motor 52 may be a single-phase motor. If the motor 52 is a single phase motor, a single phase inverter circuit is used.
  • the single-phase inverter circuit has a configuration including a single-phase bridge circuit including two legs that are series circuit portions.
  • the load is a motor, but it is not limited to this.
  • the load may be a rechargeable battery.
  • a DCDC (Direct Current to Direct Current) converter is used instead of the inverter circuit 2 .
  • the minimum configuration of a DCDC converter is a half bridge circuit with one leg.
  • the semiconductor switching element 6a which is a MOSFET, has a drain electrode 61, a gate electrode 62 and a source electrode 63.
  • the semiconductor switching element 6a is housed in an electrically insulating case to form a semiconductor element module 6.
  • the semiconductor device module 6 has a drain main terminal 64 , a source main terminal 65 , a gate terminal 66 , a first source terminal 67 and a second source terminal 68 .
  • the drain main terminal 64 is connected to the drain electrode 61 of the semiconductor switching element 6a.
  • the gate terminal 66 is connected to the gate electrode 62 of the semiconductor switching element 6a.
  • the first source terminal 67 is connected to the source electrode 63 of the semiconductor switching element 6a.
  • a second source terminal 68 is connected to the source main terminal 65 of the semiconductor element module 6 .
  • An inductance 69 is shown between the source electrode 63 and the source main terminal 65 .
  • An inductance 69 is an inductance parasitic between the source electrode 63 and the source main terminal 65 .
  • the first source terminal 67 is connected to the source electrode 63 and the second source terminal 68 is connected to the source main terminal 65 . Therefore, the inductance 69 may be rephrased as "a parasitic inductance between the first source terminal 67 and the second source terminal 68".
  • the semiconductor switching element 6a may be an IGBT.
  • the semiconductor switching element 6a has a gate electrode, a collector electrode instead of a drain electrode, and an emitter electrode instead of a source electrode. If the type of the semiconductor switching element 6a is different, the names of some electrodes in the semiconductor switching element 6a and some terminals in the semiconductor element module 6 also change, but the effect obtained in the power converter 1 of the present disclosure does not change. . Therefore, in the following description, the description “drain electrode” may include the meaning of “collector electrode”, and the description “source electrode” may include the meaning of "emitter electrode”. Similarly, the description “drain main terminal” may include the meaning of "collector main terminal”, and the descriptions “source main terminal” and “source terminal” may include “emitter main terminal” and “emitter terminal”. ” may be included.
  • the drain main terminal 64 and the source main terminal 65 of the semiconductor element module 6 are connected to a load, a power capacitor, a reactor, or another module.
  • the semiconductor switching element 6a switches between an ON state and an OFF state according to the electric signal to perform switching operation.
  • the power converter 1 performs power conversion processing.
  • the gate drive circuit 3 includes a gate driver 32, a feedback section 34, and a feedback intensity adjustment section 36, as shown in FIG.
  • One gate drive circuit 3 is provided for one semiconductor element module 6 .
  • the gate driver 32 is a drive unit that applies an electric signal between the gate terminal 66 and the source main terminal 65 to gate-drive the semiconductor switching element 6a.
  • the feedback section 34 is a circuit section that feeds back the voltage induced in the inductance 69 to the gate driver 32 .
  • the voltage induced in inductance 69 is the induced electromotive force generated between first source terminal 67 and second source terminal 68 .
  • the feedback intensity adjustment unit 36 is a circuit unit that individually adjusts the feedback intensity when the semiconductor switching element 6a is turned on and when it is turned off.
  • the feedback strength is the strength of the voltage that the feedback section 34 feeds back to the gate driver 32 . Detailed functions and operations of the feedback section 34 and the feedback intensity adjustment section 36 will be described later.
  • the gate driver 32 has a positive bias power supply 321 and a negative bias power supply 322 .
  • the positive bias power supply 321 and the negative bias power supply 322 are connected in series, and a midpoint 329 that is a connection point of the series connection is connected to one end of the feedback section 34 and one end of the feedback strength adjustment section 36 .
  • the other end of the feedback section 34 is connected to the first source terminal 67 and the other end of the feedback strength adjustment section 36 is connected to the second source terminal 68 .
  • an on-drive switch 323 is connected to the positive terminal of a positive bias power supply 321
  • an on-drive gate resistor 325 is connected to the tip of the on-drive switch 323
  • An off-drive switch 324 is connected to the negative electrode of the negative bias power supply 322
  • an off-drive gate resistor 326 is connected to the tip of the off-drive switch 324 .
  • a connection point between the on-drive gate resistor 325 and the off-drive gate resistor 326 is connected to the gate terminal 66 of the semiconductor element module 6 .
  • the connection order of the on-drive switch 323 and the on-drive gate resistor 325 may be reversed, and the connection order of the off-drive switch 324 and the off-drive gate resistor 326 may be reversed. It may be.
  • the feedback section 34 is composed of a resistive element 341 .
  • One end of resistive element 341 is connected to midpoint 329 and the other end of resistive element 341 is connected to first source terminal 67 .
  • the feedback strength adjustment section 36 has a diode element 361 and a resistance element 363 .
  • the diode element 361 and the resistance element 363 are connected in series to form a series circuit.
  • this series circuit is appropriately called a "first series circuit”.
  • the feedback strength adjustment section 36 has a diode element 362 and a resistance element 364 .
  • the diode element 362 and the resistance element 364 are connected in series to form a series circuit.
  • this series circuit is appropriately called a "second series circuit”. These first and second series circuits are connected in parallel with each other.
  • the diode element 361 should be connected in a direction in which current flows through the diode element 361 when the potential of the middle point 329 is higher than the potential of the second source terminal 68 . Therefore, the connection order of the diode element 361 and the resistance element 363 may be reversed.
  • the diode element 362 may be connected in a direction in which current flows through the diode element 362 when the potential of the middle point 329 is lower than the potential of the second source terminal 68 . Therefore, the connection order of the diode element 362 and the resistance element 364 may be reversed.
  • the gate driver 32 has a processor 328 .
  • a processor 328 As the processor 328, a CPLD (Complex Programmable Logic Device), an ASIC (Application Specific Integrated Circuit), or a logic IC can be used.
  • the processor 328 controls the opening and closing of the ON drive switch 323 and the opening and closing of the OFF drive switch 324 .
  • the on-drive switch 323 When the on-drive switch 323 is closed and the off-drive switch 324 is open, the voltage from the positive bias power supply 321 is applied to the gate terminal 66 of the semiconductor element module 6 via the on-drive gate resistor 325 . As a result, the semiconductor switching element 6 a is turned on, and a drain current Id flows between the drain main terminal 64 and the source main terminal 65 . When the semiconductor switching element 6a is used in the power conversion main circuit, this drain current Id is also called main circuit current.
  • the negative bias power supply 322 is applied to the gate terminal 66 of the semiconductor element module 6 via the off-drive gate resistor 326 .
  • the semiconductor switching element 6a is turned off.
  • the feedback section 34 and the feedback intensity adjustment section 36 are components outside the gate driver 32, but the configuration is not limited to this.
  • the feedback section 34 and the feedback strength adjustment section 36 may be incorporated inside the gate driver 32 .
  • only the feedback section 34 may be incorporated inside the gate driver 32 .
  • FIG. 4 is a diagram for explaining the operation when the semiconductor switching element 6a is turned on by the gate drive circuit 3 of the first embodiment.
  • FIG. 5 is a diagram showing an example of operation waveforms when the semiconductor switching element 6a is turned on by the gate drive circuit 3 of the first embodiment.
  • a drain current Id flows through the semiconductor switching element 6a.
  • the drain current Id is indicated by a thick solid line.
  • an induced electromotive force ⁇ V is generated in the inductance 69 according to the current change rate (dId/dt) at turn-on.
  • the induced electromotive force ⁇ V can be expressed by the following equation (1).
  • Ls is the value of the inductance 69, that is, the inductance value of the inductance 69.
  • the induced electromotive force ⁇ V has a polarity that hinders the change in the drain current Id. Therefore, the potential of the first source terminal 67 is higher than the potential of the second source terminal 68 when turned on.
  • a gate current Ig flows immediately before the semiconductor switching element 6a turns on. Further, when the semiconductor switching element 6a is turned on, a return current Ia caused by the induced electromotive force ⁇ V flows.
  • the gate current Ig is indicated by a thick dashed line
  • the return current Ia is indicated by a thick dashed-dotted line. As shown, the gate current Ig starts from the positive bias power supply 321 and flows through the on-drive switch 323, the on-drive gate resistance 325, the semiconductor switching element 6a, and the resistance element 341.
  • the return current Ia starts from the inductance 69 and flows through the resistance element 341 of the feedback section 34 , the diode element 361 , and the resistance element 363 of the feedback strength adjustment section 36 . Since the polarity of the induced electromotive force ⁇ V at turn-on is positive, no current flows to the diode element 362 side.
  • V0 is the potential of the midpoint 329
  • Rfb is the resistance value of the resistance element 341
  • RaON is the resistance value of the resistance element 363.
  • the above formula (3) can also be expressed as the following formulas (4) and (5).
  • the voltage Vs of the first source terminal 67 at turn-on can be derived by transforming the equations in the following procedure.
  • FIG. 5 shows the waveforms of the gate-source voltage Vgs, the drain current Id, and the drain-source voltage Vds in order from the top.
  • the dashed line represents the operation waveform when only the feedback section 34 is provided and the feedback strength adjustment section 36 is not provided
  • the solid line indicates the operation when both the feedback section 34 and the feedback strength adjustment section 36 are provided.
  • the state of change in the portion surrounded by the dashed ellipse is schematically shown with respect to the operation waveform of the gate-source voltage Vgs.
  • the semiconductor switching element 6a when the semiconductor switching element 6a is turned on, an induced electromotive force ⁇ V is generated in the inductance 69 according to the current change rate (dId/dt) of the drain current Id. Due to this induced electromotive force ⁇ V, the source potential, which is the potential of the source main terminal 65, rises above the potential V0 of the midpoint 329, and as a result, the gate-source voltage Vgs decreases. That is, when the gate drive circuit 3 has the feedback strength adjustment section 36, the gate-source voltage Vgs is lower than when the feedback strength adjustment section 36 is not provided.
  • the current change rate (dId/dt) of the drain current Id immediately before recovery decreases, so that the ringing of the drain current Id decreases as shown in FIG.
  • "immediately before recovery” means immediately before the recovery current flows through the other arm of the same leg.
  • the arm to be driven is the semiconductor switching element 6UN of the lower arm
  • the other arm of the same leg is the semiconductor switching element 6UP of the upper arm.
  • FIG. 6 is a diagram for explaining the operation when the semiconductor switching element 6a is turned off by the gate drive circuit 3 of the first embodiment.
  • FIG. 7 is a diagram showing an example of operation waveforms when the semiconductor switching element 6a is turned off by the gate drive circuit 3 of the first embodiment.
  • the semiconductor switching element 6a When the semiconductor switching element 6a is turned off, the drain current Id flowing through the semiconductor switching element 6a stops flowing. At this time, the inductance 69 generates an induced electromotive force ⁇ V represented by the above equation (1). However, since the induced electromotive force ⁇ V has a polarity that hinders the change in the drain current Id, it is opposite to that at turn-on. Therefore, the potential of the first source terminal 67 at turn-off is lower than the potential of the second source terminal 68 .
  • the gate current Ig flows in the direction opposite to that in FIG. 4 in order to discharge the charge accumulated in the gate of the semiconductor switching element 6a.
  • the gate current Ig is indicated by a thick dashed line. As shown, the gate current Ig starts from the negative bias power supply 322 and flows along the path of the resistive element 341 , the semiconductor switching element 6 a , the off-drive gate resistor 326 , and the off-drive switch 324 .
  • a return current Ia caused by the induced electromotive force ⁇ V flows.
  • the return current Ia is indicated by a thick dashed-dotted line.
  • the direction in which return current Ia flows is opposite to that in FIG. Specifically, the return current Ia starts from the inductance 69 and flows along the path of the resistance element 364 of the feedback strength adjustment section 36 , the diode element 362 , and the resistance element 341 of the feedback section 34 . Since the polarity of the induced electromotive force ⁇ V at turn-off is negative, no current flows to the diode element 361 side.
  • the voltage Vs of the first source terminal 67 at the time of turn-off can be expressed by the following equation (7).
  • the procedure for formula derivation is the same as that at the time of turn-on, and detailed description thereof will be omitted.
  • Vs V0 ⁇ Ig ⁇ Rfb -( ⁇ V ⁇ Ig ⁇ Rfb) ⁇ Rfb/(RaOFF+Rfb) ⁇ (7)
  • FIG. 7 shows the waveforms of the gate-source voltage Vgs, the drain current Id, and the drain-source voltage Vds in order from the top.
  • the dashed line represents the operation waveform when only the feedback section 34 is provided and the feedback strength adjustment section 36 is not provided
  • the solid line indicates the operation when both the feedback section 34 and the feedback strength adjustment section 36 are provided.
  • the state of change in the portion surrounded by the dashed ellipse is schematically shown with respect to the operation waveform of the gate-source voltage Vgs.
  • the polarity of the induced electromotive force ⁇ V generated in the inductance 69 is negative when the semiconductor switching element 6a is turned off. Therefore, the source potential, which is the potential of the source main terminal 65, drops below the potential V0 of the midpoint 329, and as a result, the gate-source voltage Vgs rises. That is, when the gate drive circuit 3 has the feedback strength adjustment section 36, the gate-source voltage Vgs increases compared to the case where the feedback strength adjustment section 36 is not provided. As the gate-source voltage Vgs increases, the drain-source voltage Vds is clamped as shown in FIG. If the drain-source voltage Vds is clamped, the surge is also clamped. As a result, surge can be reduced as compared with the case where the feedback strength adjusting section 36 is not provided.
  • FIG. 8 is a diagram for explaining the effect of using the feedback strength adjusting section 36 of the first embodiment.
  • the upper part of FIG. 8 shows the relationship between the feedback strength and the degree of effect during the turn-on operation
  • the lower part of FIG. 8 shows the relationship between the feedback strength and the degree of effect during the turn-off operation.
  • the resistance value RaON of the resistance element 363 and the resistance value RaOFF of the resistance element 364 can be adjusted individually. As a result, it is possible to achieve a trade-off between the respective losses without being mutually dependent on the turn-on loss and the turn-off loss of the semiconductor switching element 6a.
  • a typical example of the semiconductor switching element 6a is a semiconductor switching element made of silicon (Si), but the present invention is not limited to this.
  • a wide bandgap element which is a semiconductor switching element formed of a wide bandgap semiconductor such as silicon carbide (SiC), gallium nitride (GaN), gallium oxide (Ga 2 O 3 ), or diamond, can be used.
  • the semiconductor switching element 6a is a wide bandgap element
  • high-speed switching control is possible.
  • High-speed switching using a wide bandgap element can reduce switching loss.
  • surge and noise increase, so it is effective when used in the gate drive circuit 3 of the first embodiment.
  • the voltage resistance is high and the allowable current density is also high, so that the semiconductor element module 6 can be miniaturized.
  • the value of the induced electromotive force ⁇ V increases. Therefore, even if the inductance value of the inductance 69 is small, the necessary induced electromotive force ⁇ V can be obtained. As a result, the distance between the first source terminal 67 and the second source terminal 68 can be reduced, so that the semiconductor element module 6 can be further miniaturized.
  • the feedback strength which is the strength of the voltage fed back from the feedback section to the gate driver, is set independently when the semiconductor switching element is turned on and when it is turned off.
  • a feedback intensity adjustment unit configured to be adjustable to .
  • the feedback section can be configured by a resistive element. By using a resistance element, the feedback section can be configured simply and at low cost. Further, the feedback strength adjusting section can be configured by a series circuit of a resistance element and a diode element. By using a resistance element and a diode element, the feedback strength adjustment section can be configured simply and at low cost.
  • the series circuit of the feedback strength adjustment unit includes a first series circuit that adjusts the first feedback strength when the semiconductor switching element is turned on, and a second feedback strength when the semiconductor switching element is turned on. and a second series circuit. If the first and second series circuits are configured to be connected in parallel with each other, the first and second feedback strengths can be individually and independently adjusted. In addition, since the first series circuit and the second series circuit can be configured by connecting the diode elements of both in opposite directions to each other, parts can be shared, and the configuration can be simple and low cost. .
  • the resistance element of the feedback intensity adjustment section may be a variable resistance element whose resistance value can be adjusted.
  • each semiconductor switching element can be individually adjusted according to the state of deterioration of the semiconductor switching element. As a result, it is possible to optimize the surge, noise, and switching loss even during the operation of the power converter.
  • FIG. 9 is a diagram showing the detailed configuration of the gate drive circuit 3A of the second embodiment together with the semiconductor switching element 6a to be driven. Comparing the gate drive circuit 3A with the gate drive circuit 3 shown in FIG. 3, in FIG. 9, the feedback strength adjustment section 36 is replaced with a feedback strength adjustment section 36A. In the feedback strength adjusting section 36A, the resistive element 363 is replaced with a capacitive element 365, and the resistive element 364 is replaced with a capacitive element 366.
  • FIG. Other configurations are the same as or equivalent to those of the gate drive circuit 3 of FIG. 3, and the same or equivalent components are denoted by the same reference numerals, and overlapping descriptions are omitted.
  • the gate drive circuit 3 of Embodiment 1 since the gate current Ig flows through the feedback strength adjustment section 36 even during the mirror period, it also affects the change in the drain-source voltage Vds, which is the main voltage.
  • the mirror period is a period during which the gate-source voltage Vgs becomes flat during charging and discharging of the gate capacitance of the semiconductor switching element 6a.
  • the element for adjusting the feedback intensity is changed from the resistance element to the capacitive element.
  • This configuration prevents the gate current Ig from flowing to the feedback intensity adjusting section 36A during the mirror period.
  • the change of the circuit constant in the feedback strength adjusting section 36A does not affect the change of the drain-source voltage Vds.
  • the gate drive circuit 3A of the second embodiment in addition to the effects of the first embodiment, it is possible to obtain the effect of facilitating the change of the circuit constant in the feedback strength adjusting section 36A.

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Abstract

L'invention concerne un circuit d'attaque de grille (3) comprenant : un circuit d'attaque de grille (32) qui applique un signal électrique entre une borne de grille (66) et une borne principale de source (65) et la grille pilote un élément de commutation semiconducteur (6a) ; une unité de rétroaction (34) qui renvoie, au circuit d'attaque de grille (32), une force électromotrice induite générée entre une première borne de source (67) et une deuxième borne de source (68) par un courant de circuit principal qui circule entre une borne principale de drain (64) et la borne principale de source (65) ; et une unité de réglage d'intensité de rétroaction (36) qui est configurée pour être apte à régler l'intensité de rétroaction, laquelle est l'intensité de la tension renvoyée au circuit d'attaque de grille (32) à partir de l'unité de rétroaction (34), séparément lorsque l'élément de commutation semiconducteur (6a) est mis en circuit et lorsque celui-ci est mis hors circuit.
PCT/JP2022/006413 2022-02-17 2022-02-17 Circuit d'attaque de grille et dispositif de conversion de puissance WO2023157185A1 (fr)

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Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2000089838A (ja) * 1998-08-12 2000-03-31 Abb Ind Oy 制御回路
JP2012039460A (ja) * 2010-08-09 2012-02-23 Honda Motor Co Ltd 半導体素子の駆動装置及び方法
US20160301308A1 (en) * 2013-11-14 2016-10-13 Tm4 Inc. Compensation circuit, commutation cell and power converter controlling turn-on and turn-off of a power electronic switch
WO2016189585A1 (fr) * 2015-05-22 2016-12-01 日産自動車株式会社 Dispositif de conversion de puissance
JP2021150977A (ja) * 2020-03-16 2021-09-27 株式会社豊田自動織機 ドライバ回路及び電力変換装置

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JP2000089838A (ja) * 1998-08-12 2000-03-31 Abb Ind Oy 制御回路
JP2012039460A (ja) * 2010-08-09 2012-02-23 Honda Motor Co Ltd 半導体素子の駆動装置及び方法
US20160301308A1 (en) * 2013-11-14 2016-10-13 Tm4 Inc. Compensation circuit, commutation cell and power converter controlling turn-on and turn-off of a power electronic switch
WO2016189585A1 (fr) * 2015-05-22 2016-12-01 日産自動車株式会社 Dispositif de conversion de puissance
JP2021150977A (ja) * 2020-03-16 2021-09-27 株式会社豊田自動織機 ドライバ回路及び電力変換装置

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