WO2023098127A1 - 一种dc-dc开关电源的电感电流预估方法 - Google Patents

一种dc-dc开关电源的电感电流预估方法 Download PDF

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WO2023098127A1
WO2023098127A1 PCT/CN2022/110105 CN2022110105W WO2023098127A1 WO 2023098127 A1 WO2023098127 A1 WO 2023098127A1 CN 2022110105 W CN2022110105 W CN 2022110105W WO 2023098127 A1 WO2023098127 A1 WO 2023098127A1
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voltage
output
digital
module
input voltage
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French (fr)
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徐申
杨晨曦
钱毅杰
刘玉洁
于利民
孙伟锋
时龙兴
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东南大学
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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
    • G01R19/0038—Circuits for comparing several input signals and for indicating the result of this comparison, e.g. equal, different, greater, smaller (comparing pulses or pulse trains according to amplitude)
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/157—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators with digital control
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R15/00—Details of measuring arrangements of the types provided for in groups G01R17/00 - G01R29/00, G01R33/00 - G01R33/26 or G01R35/00
    • G01R15/04—Voltage dividers
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
    • G01R19/25—Arrangements for measuring currents or voltages or for indicating presence or sign thereof using digital measurement techniques
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1588—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load comprising at least one synchronous rectifier element
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention belongs to the field of switching power supplies, in particular to a method for estimating the inductance current of a DC-DC switching power supply.
  • the inductor current is an important feedback signal of the control circuit, which is used for the loop control of the current mode controller, such as the average current mode controller, the peak current mode controller, the current hysteresis controller, etc. It is also used for the over-current protection of the converter, so in most DC-DC switching power supply systems, it is necessary to measure the inductor current information in real time to keep the system running safely.
  • current measurement methods for DC-DC switching power supplies can be divided into voltage drop measurement methods and observer-based measurement methods. In the voltage drop based approach, the current information is extracted from the voltage drop caused by the current flowing through the sense resistor or MOSFET. Observer-based systems typically estimate the current from the voltage across the power-stage inductor.
  • the signal output by the sensor needs to be transmitted to the digital controller through an analog-to-digital converter (Analog to Digital Converter, ADC).
  • ADC Analog to Digital Converter
  • the present invention proposes a method for estimating the inductance current of a DC-DC switching power supply, which can accurately estimate the real-time inductance under the condition of only sampling the input and output voltages Current information, the present invention does not need to add additional analog sampling circuits such as resistors, capacitors, and op amps, and does not need to use high-speed ADCs much higher than the switching frequency, which can reduce costs and circuit volumes, and has high versatility sex.
  • a method for estimating the inductor current of a DC-DC switching power supply including a voltage sampling module, a data conversion module, a switch signal counting module, an inductor voltage calculation module, and a digital filter module; the voltage sampling module inputs the input voltage of the DC-DC switching power supply and the output voltage to obtain the input voltage digital quantity and the output voltage digital quantity, and then perform calculation and digit conversion through the data conversion module to obtain the converted input voltage and the converted output voltage with the same digit, and output them to the inductance voltage calculation module ;
  • the node voltage is compared with the reference voltage by the comparator, the actual switching signal is output, and then the duty ratio is obtained by the switching signal counting module, and output to the inductor voltage calculation module; then the inductor voltage calculation module calculates the voltage at both ends of the inductor and the parasitic resistance The average value is then filtered by the digital filter module to finally obtain the estimated inductor current.
  • the voltage sampling module includes a sampling circuit and an analog-to-digital converter (Analog to Digital Converter, ADC).
  • ADC Analog to Digital Converter
  • the sampling circuit uses a voltage dividing resistor to sample the input voltage and the output voltage according to the scaled voltage. Output to the corresponding ADC, and finally get the input voltage digital quantity and the output voltage digital quantity.
  • the data conversion module receives the input voltage digital quantity and the output voltage digital quantity, and obtains the actual input voltage V in and output voltage V o according to the following formula:
  • D in [n] is the input voltage digital quantity
  • D o [n] is the output voltage digital quantity
  • n indicates that the corresponding digital quantity is in the nth cycle
  • the ADC digits of the input voltage and output voltage are N 1 and N respectively 2
  • the input ranges are ⁇ V 1 and ⁇ V 2 respectively
  • G 1 and G 2 are the input sampling gain coefficient and the output sampling gain coefficient respectively;
  • the input voltage V in and the output voltage V o obtained by the operation are converted into digital quantities with the same number of digits, that is, the converted input voltage and the converted output voltage.
  • the input sampling gain coefficient and the output sampling gain coefficient are respectively calculated according to the following formula:
  • R 1 and R 2 are the pair of voltage dividing resistors for the input voltage
  • R 3 and R 4 are the pair of voltage dividing resistors for the output voltage
  • a 1 and A 2 are the single-ended to differential operational amplifiers for sampling the input voltage and output voltage respectively magnification factor.
  • the switch signal counting module is used to detect and count the switch signal, compare the node voltage with the reference voltage through the comparator, and output the switch signal: if the upper switch tube is turned off, the lower switch tube is turned on, and the node voltage is ground Potential, the node voltage is less than the reference voltage, the comparator outputs a switch signal 0; if the upper switch is turned on, the lower switch is turned off, the node voltage is the input voltage at this time, the node voltage is greater than the reference voltage, the comparator outputs a switch signal 1; The number of times the switch signal is 1 in a fixed cycle T s is counted by a high-frequency counter, and the duty cycle is output once in each cycle.
  • the duty cycle is calculated as follows:
  • duty[n] is the duty cycle
  • m is the total number of allowed counting times in the fixed period T s
  • k n is the count value of the switch signal in the nth fixed period T s .
  • the average voltage across the inductance and parasitic resistance is calculated as follows:
  • V iL [n] duty[n] ⁇ V in [n]-V o [n]
  • V iL [n] is the average voltage across the inductor and parasitic resistance
  • V in [n] is the converted input voltage
  • V o [n] is the converted output voltage
  • the DC-DC switching power supply inductor current estimation method adopted in the present invention uses a BUCK switching power supply as a typical application. Compared with the traditional inductor current sampling scheme, it does not need to use an analog sampling circuit, which avoids complicated sampling circuit design , which greatly reduces the cost of current sampling;
  • the digital filter scheme used in the present invention to calculate the inductor current can effectively reduce the sampling frequency of the input voltage and the output voltage compared to the traditional filtering scheme, avoiding the use of expensive high-speed ADCs, and taking The method of average value only needs to complete one-step operation within multiple counting cycles, which greatly improves the cycle of single-step operation and can effectively reduce the algorithm's requirements for calculation speed;
  • the inductor current estimation method proposed by the present invention can not only estimate the average value of the inductor current, but also estimate the ripple value of the inductor current, and is suitable for current ripple-based Control circuit; in addition, the inductor current estimation method has a fast speed and low delay, and can quickly and accurately follow the actual value of the inductor current;
  • the present invention can be realized by a pure digital circuit, and the algorithm can be embedded in various application occasions without adding complex peripheral circuits to realize digital sampling of the inductor current, and has high flexibility and integration.
  • Fig. 1 is a block diagram of the system structure applied in the BUCK switching power supply by the inductance current estimation method of the present invention
  • Fig. 2 is the circuit diagram of voltage sampling module
  • Figure 3a is the switching characteristics of the BUCK switching power supply when it is turned on
  • Figure 3b is the switching characteristics of the BUCK switching power supply when it is turned off
  • Figure 4 is a block diagram of the inductor current estimation hardware
  • Figure 5 is a timing diagram of counting and sampling
  • Fig. 6 is the design block diagram of infinite impulse response (Infinite Impulse Response, IIR) filter
  • Fig. 7 is a simulation waveform diagram of simplis applied in the BUCK switching power supply of the present invention.
  • Fig. 1 is a block diagram of the system structure of the inductance current estimation method applied in the BUCK switching power supply in the present invention, including a voltage sampling module, a data conversion module, a switching signal counting module, an inductance voltage calculation module and a digital filter module; the synchronous rectification BUCK passes Control the turn-on and turn-off time of the upper switch MOS1 and the lower switch MOS2 to control the output voltage.
  • the voltage sampling module inputs the input voltage V in and output voltage V o of the DC-DC switching power supply, and obtains the input voltage digital quantity D in [n] and the output voltage digital quantity D o [n], and then performs calculation through the data conversion module and digital conversion, the output corresponding to the input voltage V in and output voltage V o analog digital quantity is the converted input voltage V in [n] and the converted output voltage V o [n]; the upper switch tube MOS1 and the lower
  • the node voltage V x of the middle node of the switching tube MOS2 is compared with the reference voltage V ref by the comparator, and the actual switching signal SW is output, and then the number of times the switching signal SW is 1 in one cycle is counted by the switching signal counting module, so as to obtain the duty cycle Ratio duty[n];
  • the above digital quantities output the average voltage V iL [n] of the inductance L and the parasitic resistance R L in a fixed period through the inductance voltage calculation module, and then pass through the infinite impulse response (
  • the digital filter can finally output the estimated inductor current I L [n] within a fixed period T s , if the digital filter samples The period T s is much shorter than the switching period T sw , so the final output inductor current I L [n] not only includes the average value I L_A of the actual inductor current IL but also reflects the ripple information of the actual inductor current.
  • FIG. 2 is a structural diagram of the voltage sampling module.
  • the voltage sampling module includes a sampling circuit and an analog-to-digital converter ADC.
  • the sampling circuit scales the input voltage V in and the output voltage V o through a voltage divider resistor, and then converts it through a single-ended differential operation amplifier.
  • the signal is amplified, and the corresponding input voltage digital quantity D in and output voltage digital quantity D o are output through the ADC.
  • the final output differential signal of the sampling circuit shown in Figure 2 can be expressed as equations (1) and (2):
  • R 1 and R 2 are the pair of voltage dividing resistors for the input voltage V in
  • R 3 and R 4 are the pair of voltage dividing resistors for the output voltage V o
  • a 1 and A 2 Respectively, the single-ended-to-differential operational amplifier amplification coefficients of input voltage V in and output voltage V o are sampled.
  • G 1 and G 2 are the input sampling gain coefficient of the input signal V in and the output sampling gain coefficient of the output signal V o respectively.
  • the input voltage V in and the output voltage V o need to be calculated under the same unit, and the voltage sampling module has different sampling amplification factors for the input voltage V in and the output voltage V o , and because the output voltage V o
  • the sampling accuracy requirements of the input voltage V in are often higher than that of the input voltage V in , and the number of ADC bits used to sample the input voltage V in and the output voltage V o is also different. Therefore, the data conversion module needs to convert the ADC output according to formulas (5) and (6).
  • the input voltage digital quantity D in [n] and the output voltage digital quantity D o [n] can be used for subsequent calculations after initial calculation:
  • the ADC digits of the input voltage V in and the output voltage V o are N 1 and N 2 respectively, and the input ranges are ⁇ V 1 and ⁇ V 2 .
  • the data conversion module converts the input voltage V in and output voltage V o obtained from the operation into digital quantities with the same number of digits through digit conversion, that is, the converted input voltage V in [n] and the converted output voltage V o [n ].
  • the average node voltage V x_ave and the average output voltage V o_ave of the middle node of the upper switch MOS1 and the lower switch MOS2 are equal.
  • the relationship between the output voltage average V o_ave and the node voltage average V x_ave and the inductor current average I L_A is given by (7):
  • T sw is the switching cycle
  • RL is the parasitic resistance value
  • the above formula assumes that the information of input voltage V in and output voltage V o is available, and they are constant within a switching cycle T sw , which is an important condition that the above formula must meet when it is used for current estimation.
  • the switching period T sw will change with the change of the working condition, but in the digital control system, it is difficult to use a digital scheme to average a certain value in an uncertain period.
  • one switching period T sw can be switched into multiple small fixed periods T s of equal duration, and the frequency is f s .
  • the switching period T sw is not fixed, but the average value of the inductor current in each small fixed period T s can be calculated, assuming that the upper switching tube MOS1 conducts for t on in the fixed period T s , and Regardless of the on-resistance of the switch tube, the average value of the inductor current I L_A in the fixed period T s is expressed as (8):
  • V in_ave is the average value of the input voltage.
  • the average value V x_ave of the node voltage in one cycle depends on the conduction time t on of the upper switching transistor MOS1 in the fixed cycle T s .
  • the PWM signal and the node voltage Vx have the same voltage timing distribution, and the change of the pulse width of the PWM signal can represent the change of the inductor current, so the conduction time of the PWM signal can be used to estimate the inductor current.
  • ⁇ i L_A is the inductor current ripple value
  • L is the inductance value
  • Figure 3 shows the switching characteristics of the MOSFET on and off when the BUCK switching power supply is turned on and off.
  • the gate-source voltage V gs , source-drain current I ds , source-drain voltage V ds and node voltage V of the upper switching tube MOS1 are drawn.
  • the turn-on delay t r and the turn-off delay t f are not the same, and have a nonlinear relationship with the inductor current, which eventually leads to a certain difference between the pulse width of the PWM signal and the actual node voltage.
  • the node voltage V x does not use the PWM signal to calculate, and a simple comparator can be used to detect the level change of the node voltage V x , compare the node voltage V x with the reference voltage V ref , and then use this information as a
  • the bit switch signal SW is provided to the digital controller, and since the delay of the comparator is fixed, the change of the node voltage Vx can be reproduced truly in the digital controller.
  • the switching signal counting module is used to detect and count the switching signals, compare the node voltage Vx with the reference voltage Vref through a comparator, and output the switching signal SW: if the upper switching tube MOS1 is turned off, the lower switching tube MOS2 is turned on, the node voltage V x is at the ground potential at this time, the node voltage V x is less than the reference voltage V ref , the comparator output switching signal SW is 0; if the upper switch MOS1 is turned on, the lower switch MOS2 is turned off, at this time the node The voltage V x is the input voltage V in , the node voltage V x is greater than the reference voltage V ref , and the comparator outputs the switch signal SW to be 1; then the number of times the switch signal SW is 1 in a fixed period T s is counted by a high-frequency counter, and each Periodically output duty cycle duty[n].
  • FIG. 4 is a hardware block diagram of inductor current estimation.
  • the switching signal SW output by the comparator is counted by the counter, and the counting clock period is 250M clocks, where clk_s is the clock after frequency division by the counter, and the frequency division clock period is T s , the counting result cnt_on[4:0] of the switch signal SW is output once at the end of each fixed cycle T s , and the count is set to zero at the beginning of the cycle.
  • the fixed period T s of the clock after frequency division is used as the sampling period of the IIR digital filter.
  • Fig. 5 is a timing diagram of switching signal SW counting and digital filter sampling in the inductor current prediction method, in which the switching signal SW is input to the digital controller through a comparator, and the switching signal SW is counted at the highest frequency clk_cnt, and the digital filter uses The lower frequency clk_s is used as the sampling frequency, and the counting frequency clk_cnt is m times of the sampling frequency clk_s.
  • Ts if the number of cycles of SW high level detected in m clk_cnt periods is k n times, then the average voltage V across the inductor L and the parasitic resistance RL in this sampling period can be calculated iL [n], as shown in formula (10).
  • the inductor current I L is expressed as an s-domain expression, as shown in (11). Discretize the expression in s domain, and through bilinear transformation, the form of difference equation shown in (12) can be finally obtained.
  • I L [n] c 1 ⁇ I L [n-1]+c 2 ⁇ (V iL [n]+V iL [n-1]) (12)
  • I L represents the instantaneous value of the inductor current in the continuous domain
  • I L [n] represents the inductor current in the nth sampling period in the discrete domain, that is, the inductor current estimated by the digital filter
  • V iL represents the inductor L
  • V iL [n] represents the average value of the voltage across the inductance L and the parasitic resistance RL of the nth sampling period in the discrete domain
  • c 1 and c 2 are digital filter coefficients.
  • Figure 6 is a digital design block diagram of a first-order IIR filter.
  • the above differential equation can be realized by a first-order IIR filter. Since the digital filter coefficients c 1 and c 2 are in decimal form, the data needs to be shifted to the left in the digital implementation process N bits, that is to say, become C 1 and C 2 after proportionally amplifying 2 N times.
  • the average voltage V iL [n] at both ends of the inductor L and the parasitic resistance R L is taken as input, and the delay unit outputs V iL [n-1] as the input of the previous sampling cycle, where the nth cycle represents the current cycle, Then the n-1th cycle represents the previous cycle; the inductor current I L [n] is used as the output, and the delay unit outputs I L [n-1] as the output of the previous sampling cycle.
  • C 1 and C 2 are fixed values under certain working conditions, so the realization of the multiplier can improve the operation speed through the method of shifting. Since the filter coefficients are enlarged in proportion during the calculation process, the final output needs to pass through a first-stage divider, and the division can be realized by shifting right by N bits.
  • Fig. 7 is a Simplis simulation waveform diagram applied in the BUCK switching power supply of the present invention, wherein PWM is the control signal waveform of the upper switching tube MOS1, I_buck is the actual value of the inductor current, and IL_est is the estimated value of the inductor current. It can be seen that the estimated value of the inductor current is The estimation can accurately restore the actual inductor current average value and ripple value.

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  • Physics & Mathematics (AREA)
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Abstract

一种DC-DC开关电源的电感电流预估方法，输入电压(V in)和输出电压(V o)经过电压采样模块和数据转换模块，得到位数相同的转换后的输入电压(D in[n])和转换后的输出电压(D o[n])；节点电压(V x)与参考电压(V ref)进行比较，再通过开关信号计数模块得到占空比(duty[n])；然后电感电压计算模块输出电感(L)和寄生电阻(R L)两端的电压平均值(V iL[n])，再经过数字滤波器模块最终得到预估的电感电流(I L[n])。

Description

一种DC-DC开关电源的电感电流预估方法 技术领域
本发明属于开关电源领域，尤其涉及一种DC-DC开关电源的电感电流预估方法。
技术背景
随着科技的不断发展，多种多样的电子设备出现在人们的生活中，小到各种便携穿戴电子设备，如手机、智能手表、平板电脑，大到汽车电子、医疗器械等，都影响和改善着人们的生活。消费类电子不断迭代更新，功能也是越来越强大，这也要求电子产品有一个稳定的供电系统来维持其正常工作。同时电子产品体积越来越小，在电池技术没有突破性的革新前，只能通过高性能的电源管理芯片来提高其供电效率。
目前，开关电源的发展趋势主要集中在高效率，小面积，快速瞬态响应，大驱动能力和数字化，其中数字化的研究是开关电源中发展最快的分支之一。预计到2022年数字管理集成电路(Integrated Circuit，IC)的全球市场将高达42亿美元，其中数字电源管理芯片在照明，消费电子，工业和汽车电子等市场的占用率也在不断增加。相对于模拟控制的电源芯片，数字控制DC-DC开关电源更容易嵌入到各种SoC芯片中，从而实现高集成度。另外，数字控制可以实现更复杂的控制算法，数字开关电源在调试和监测芯片各种参数时会更加灵活，数字控制能减小控制系统分立器件的数量，改善系统可靠性。
在DC-DC开关电源中，电感电流是控制电路的重要反馈信号，用于电流模式控制器的环路控制，如平均电流模式控制器、峰值电流模式控制器、电流滞后控制器等。它也用于转换器的过电流保护，因此在大多数DC-DC开关电源系统中，需要实时测量电感电流信息，来保持系统安全运行。一般来说，DC-DC开关电源的电流测量方法可分为电压降测量法和基于观测器的测量法。在基于电压降的方法中，通过电流流过感测电阻器或MOSFET引起的电压降中提取电流信息。基于观测器的系统通常通过功率级电感上的电压来估计电流。
在大多数情况下，现有的方法不太适合开关电源的数字控制器集成，其中总体尺寸、系统成本和总体效率是主要难点。基于电压降的方法要么降低转换器的效率，要么需要高带宽的放大器，这在最新的CMOS数字工艺中非常具有挑战性。这是由于标准数字电路的电源电压非常有限，传统的模拟结构无法使用。因此，这种架构需要体积更大、可靠性更低的多芯片解决方案，需要用不同的IC技术实现传感电路和控制器。另一方面，基于 观测器的方法准确性有限，电流估算依赖于电感值和等效串联电阻值的先验数据，这取决于工作条件和外部影响下的变化。此外，对于数字控制器的实现，传感器输出的信号还需要通过模数转换器(Analog to Digital Converter，ADC)传输到数字控制器中，如果要在控制器中使用电感电流的开关频率分量，即交流纹波量，则需要具有远高于开关频率的采样率的高带宽电流传感器和ADC。由于ADC的要求非常昂贵，这对于开关频率很高的DC-DC开关电源来说是一个明显的缺点。因此，期望在数字控制器中消除用于电感电流采样的高速ADC和高带宽传感器。
发明内容
发明目的：为克服现有技术的局限以及不足，本发明提出了一种DC-DC开关电源的电感电流预估方法，可以在只采样输入和输出电压的情况下，准确的估算出实时的电感电流信息，本发明不需要增加额外的电阻、电容和运放等模拟采样电路，另外不需要用到远高于开关频率的高速ADC，可以降低成本、减小电路体积，具有很高的泛用性。
技术方案:为实现上述目的，本发明采用如下的技术方案：
一种DC-DC开关电源的电感电流预估方法，包括电压采样模块、数据转换模块、开关信号计数模块、电感电压计算模块和数字滤波器模块；电压采样模块输入DC-DC开关电源的输入电压和输出电压，得到输入电压数字量和输出电压数字量，再经过数据转换模块进行运算和位数转换，得到位数相同的转换后的输入电压和转换后的输出电压，输出给电感电压计算模块；节点电压通过比较器和参考电压进行比较，输出实际的开关信号，再通过开关信号计数模块得到占空比，输出给电感电压计算模块；然后电感电压计算模块计算得到电感和寄生阻两端的电压平均值，再经过数字滤波器模块进行滤波，最终得到预估的电感电流。
其中，
所述电压采样模块包括采样电路和模数转换器(Analog to DigitalConverter，ADC)，采样电路利用分压电阻采样输入电压和输出电压按照比例缩放后的电压，再分别通过单端转差分运放后输出给相应的ADC，最终得到输入电压数字量和输出电压数字量。
所述数据转换模块接收输入电压数字量和输出电压数字量，按下式运算得到实际的输入电压V in和输出电压V o：
Figure PCTCN2022110105-appb-000001
Figure PCTCN2022110105-appb-000002
其中，D in[n]为输入电压数字量，D o[n]为输出电压数字量，n表示相应数字量处于第n个周期；输入电压和输出电压的ADC位数分别为N 1和N 2，输入范围分别为±V 1和±V 2；G 1和G 2分别为输入采样增益系数和输出采样增益系数；
最后通过位数转换将运算得到的输入电压V in和输出电压V o转换为相同位数的数字量即转换后的输入电压和转换后的输出电压。
所述输入采样增益系数和所述输出采样增益系数分别按照下式计算：
Figure PCTCN2022110105-appb-000003
Figure PCTCN2022110105-appb-000004
其中，R 1和R 2为输入电压的分压电阻对，R 3和R 4为输出电压的分压电阻对，A 1和A 2分别为输入电压和输出电压采样的单端转差分运放的放大系数。
所述开关信号计数模块用于检测开关信号并进行计数，将节点电压与参考电压通过比较器进行比较，输出开关信号：若上开关管关断，下开关管导通，此时节点电压为地电位，节点电压小于参考电压，比较器输出开关信号0；若上开关管导通，下开关管关断，此时节点电压为输入电压，节点电压大于参考电压，比较器输出开关信号1；再通过高频计数器计数一个固定周期T s内开关信号为1的次数，每个周期输出一次占空比。
所述占空比按下式计算：
duty[n]＝k n/m
其中，duty[n]为占空比，m为固定周期T s内允许计数总次数，k n为第n个固定周期T s内开关信号计数值。
所述电感和寄生阻两端的电压平均值按下式计算：
V iL[n]＝duty[n]·V in[n]-V o[n]
其中，V iL[n]为电感和寄生阻两端的电压平均值，V in[n]为转换后的输入电压，V o[n]为转换后的输出电压。
有益效果：与现有技术相比，本发明的优点及显著效果：
1.本发明所采用的DC-DC开关电源电感电流预估方法，以BUCK开关电源作为典型应用，相比于传统的电感电流采样方案，不需要使用模拟采样电路，避免了复杂的采样电路设计，大大减小了电流采样的成本；
2.本发明所采用的用于计算电感电流的数字滤波器方案，相对于传统的滤波方案，可 以有效减小输入电压和输出电压的采样频率，避免了使用昂贵的高速ADC，并且通过计数取平均值的方法，只需要在多个计数周期内完成一步运算，大大提高了单步运算的周期，能够有效降低算法对于计算速度的要求；
3.本发明所提出的电感电流预估方法，相对于传统的数字预估方案，不仅能预估电感电流的平均值，还能预估电感电流的纹波值，适用于基于电流纹波的控制电路；另外该电感电流预估方法速度快延迟低，能够快速并准确的跟随电感电流实际值变化；
4.本发明可以通过纯数字电路实现，该算法可以在不需要添加复杂外围电路的情况后嵌入到多种应用场合，实现电感电流的数字采样，具有高度的灵活性和集成度。
附图说明
图1是本发明电感电流预估方法在BUCK开关电源中应用的系统结构框图；
图2是电压采样模块的电路图；
图3a是BUCK开关电源在导通时的开关特性；
图3b是BUCK开关电源在关断时的开关特性；
图4是电感电流预估硬件框图；
图5是计数和采样的时序图；
图6是无限脉冲响应(Infinite Impulse Response，IIR)滤波器的设计框图；
图7是本发明在BUCK开关电源中应用的simplis仿真波形图。
图中有：输入电压V in，输出电压V o，输入电压数字量D in[n]，输出电压数字量D o[n]，转换后的输入电压V in[n]，转换后的输出电压V o[n]，上开关管MOS1，下开关管MOS2，节点电压V x，参考电压V ref，占空比duty[n]，电感L，寄生阻R L，电感L和寄生阻R L两端的电压平均值V iL[n]，电感电流I L[n]。
具体实施方式
为了更清楚地说明本发明，下面将结合附图对本发明的技术方案进行进一步解释。
图1为本发明中电感电流预估方法在BUCK开关电源中应用的系统结构框图，包括电压采样模块、数据转换模块、开关信号计数模块、电感电压计算模块和数字滤波器模块；同步整流BUCK通过控制上开关管MOS1和下开关管MOS2两个开关管的导通和关断时间来控制输出电压。其中，电压采样模块输入DC-DC开关电源的输入电压V in和输出电压V o，得到输入电压数字量D in[n]和输出电压数字量D o[n]，再经过数据转换模块进行运算和位数转换，输出对应输入电压V in和输出电压V o模拟量的数字量即转换后的输入电压V in[n]和转换后的输出电压V o[n]；上开关管MOS1和下开关管MOS2中间节点的节点 电压V x通过比较器和参考电压V ref进行比较，输出实际的开关信号SW，再通过开关信号计数模块计数一个周期内开关信号SW为1的次数，从而得到占空比duty[n]；以上数字量通过电感电压计算模块输出固定周期内电感L和寄生阻R L两端的电压平均值V iL[n]，再经过无限脉冲响应(Infinite Impulse Response，IIR)数字滤波器模块进行滤波，最终输出预估的电感电流I L[n]，其中n表示的是相应数字量处于第n个周期。
若数字滤波器的极点系数和电路中的电感L和寄生阻R L参数相匹配，数字滤波器最终可以输出一个固定周期T s内预估的电感电流I L[n]，若数字滤波器采样周期T s远小于开关周期T sw，则最终输出的电感电流I L[n]不仅包含实际电感电流I L的平均值I L_A还反映了实际电感电流的纹波信息。
图2是电压采样模块的结构图，电压采样模块包括采样电路和模数转换器ADC，采样电路通过分压电阻对输入电压V in和输出电压V o进行缩放，再经过单端转差分运放对信号进行放大，通过ADC输出对应的输入电压数字量D in和输出电压数字量D o。图2中所示的采样电路最终输出的差分信号可以表示为式(1)和(2)：
Figure PCTCN2022110105-appb-000005
Figure PCTCN2022110105-appb-000006
其中，
Figure PCTCN2022110105-appb-000007
和
Figure PCTCN2022110105-appb-000008
是输入电压V in经采样电路输出的差分信号，
Figure PCTCN2022110105-appb-000009
和
Figure PCTCN2022110105-appb-000010
是输出电压V o经采样电路输出的差分信号，R 1和R 2为输入电压V in的分压电阻对，R 3和R 4为输出电压V o的分压电阻对，A 1和A 2分别为输入电压V in和输出电压V o采样的单端转差分运放放大系数。
可以得到：
Figure PCTCN2022110105-appb-000011
Figure PCTCN2022110105-appb-000012
其中，G 1和G 2分别为输入信号V in的输入采样增益系数和输出信号V o的输出采样增益系数。
考虑到在后续计算中需要在同一单位下对输入电压V in和输出电压V o进行计算，而电压采样模块对于输入电压V in和输出电压V o的采样放大倍数不同，另外由于输出电压 V o的采样精度要求往往要高于输入电压V in，用于采样输入电压V in和输出电压V o的ADC位数也不同，因此数据转换模块需要按照式(5)和(6)对ADC输出的输入电压数字量D in[n]和输出电压数字量D o[n]进行初步计算后才能用于后面的运算：
Figure PCTCN2022110105-appb-000013
Figure PCTCN2022110105-appb-000014
其中，输入电压V in和输出电压V o的ADC位数分别为N 1和N 2，输入范围为±V 1和±V 2。
数据转换模块最后通过位数转换将运算得到的输入电压V in和输出电压V o转换为相同位数的数字量即转换后的输入电压V in[n]和转换后的输出电压V o[n]。
若BUCK变换器是理想且无损的，则上开关管MOS1和下开关管MOS2中间节点的节点电压平均值V x_ave和输出电压平均值V o_ave是相等的。但是，在实际情况下，考虑寄生阻R L时，输出电压平均值V o_ave和节点电压平均值V x_ave以及电感电流平均值I L_A之间的关系由(7)给出：
Figure PCTCN2022110105-appb-000015
其中，T sw是开关周期，R L为寄生阻阻值。
上式假设输入电压V in和输出电压V o信息可用，并且它们在一个开关周期T sw内是常数，这是上述公式用于电流预估时必须满足的一个重要条件。在不定频控制中，开关周期T sw会随着工况的变化发生改变，然而在数字控制系统中，很难用数字方案实现在一个不确定的周期内对某个值求平均。为了解决上述问题，可以将一个开关周期T sw切换成多个等长时间的小固定周期T s，频率为f s。尽管在不定频控制中，开关周期T sw不固定，但是可以只计算每个小固定周期T s内电感电流平均值，假设上开关管MOS1在固定周期T s内导通时间为t on，并且不考虑开关管的导通电阻，则固定周期T s内电感电流平均值I L_A表示为(8)：
Figure PCTCN2022110105-appb-000016
其中，V in_ave为输入电压平均值。
从(8)中可以看出，在一个周期内节点电压平均值V x_ave取决于上开关管MOS1在固定周期T s内的导通时间t on。对于理想的开关电源BUCK变换器，PWM信号和节点电压V x具有相同的电压时序分布，PWM信号脉冲宽度的变化可以表示电感电流的变化， 因此可以利用PWM信号的导通时间来估算电感电流。通过提供转换后的输入电压V in[n]和转换后的输出电压V o[n]给数字控制器，再依靠数字控制器内部生成的脉宽调制信号，很容易在数字控制器内部估算出电感电流平均值。对于电感电流纹波值的计算，通过公式(9)在控制器内部实现。
Figure PCTCN2022110105-appb-000017
其中，Δi L_A为电感电流纹波值，L为电感值。
上述方案在忽略BUCK开关电源开关延时的情况下实行的，在实际应用中，随着负载电流和开关频率的增加，PWM信号和节点电压V x之间的开关延时t d非线性增加，利用PWM信号进行电流预估会产生很大的估算误差。
图3给出了BUCK开关电源的MOSFET导通和关断时的开关特性，图中绘制了上开关管MOS1的栅源电压V gs、源漏电流I ds、源漏电压V ds和节点电压V x在开关时的变化曲线，其中V th为开关管阈值电压，V PWM为PWM驱动信号电压。参考图3(a)，当PWM信号由0V变为12V后，经过导通延时t r时间延时后节点电压V x才上升到输入电压12V；参考图3(b)，当PWM信号有12V变为0V后，经过关断延时t f时间延时节点电压V x才下降到0。在上开关管MOS1导通期间，开关延时t d的一部分取决于电感电流(t 2～t 3)，电感电流越大，这部分延时越大；在上开关管MOS1关断期间，延时对电感电流的依赖性较小(t 3～t 4)。导通延时t r和关断延时t f并不相同，且和电感电流呈非线性关系，最终导致PWM信号的脉冲宽度和实际节点电压之间存在一定差异。在实际应用中，节点电压V x不使用PWM信号计算，可以使用简单的比较器来检测节点电压V x电平变化，将节点电压V x同参考电压V ref进行比较，再将该信息作为一位开关信号SW提供给数字控制器，由于比较器的延迟是固定的，因此节点电压V x的变化可以在数字控制器中真实再现。
本实施例中开关信号计数模块用于检测开关信号并进行计数，将节点电压V x与参考电压V ref通过比较器进行比较，输出开关信号SW：若上开关管MOS1关断，下开关管MOS2导通，此时节点电压V x为地电位，节点电压V x小于参考电压V ref，比较器输出开关信号SW为0；若上开关管MOS1导通，下开关管MOS2关断，此时节点电压V x为输入电压V in，节点电压V x大于参考电压V ref，比较器输出开关信号SW为1；再通过高频计数器计数一个固定周期T s内开关信号SW为1的次数，每个周期输出一次占空比duty[n]。
图4是电感电流预估的硬件框图，比较器输出的开关信号SW通过计数器进行计数，计数时钟周期为250M时钟，其中clk_s为通过计数器分频后的时钟，分频的时钟周期为 周期T s，每个固定周期T s结束输出一次开关信号SW计数结果cnt_on[4:0]，周期开始时计数置零。分频后的时钟固定周期T s作为IIR数字滤波器的采样周期。
图5是电感电流预估算法中开关信号SW计数和数字滤波器采样的时序图，其中开关信号SW通过比较器输入给数字控制器，开关信号SW以最高的频率clk_cnt进行计数，数字滤波器采用较低频率clk_s作为采样频率，计数频率clk_cnt是采样频率clk_s的m倍。在第n个采样周期Ts内，如果m个clk_cnt周期中检测到SW高电平的周期数为k n次，则可以计算出该采样周期内电感L和寄生阻R L两端的电压平均值V iL[n]，如式(10)所示。
Figure PCTCN2022110105-appb-000018
其中，
Figure PCTCN2022110105-appb-000019
在连续域中，电感电流I L表示成s域表达式，如(11)所示。将s域表达式离散化，通过双线性变换，最终可以得到(12)所示的差分方程形式。
Figure PCTCN2022110105-appb-000020
I L[n]＝c 1·I L[n-1]+c 2·(V iL[n]+V iL[n-1])      (12)
Figure PCTCN2022110105-appb-000021
Figure PCTCN2022110105-appb-000022
其中，I L表示连续域中电感电流瞬时值，I L[n]表示离散域中第n个采样周期的电感电流，也就是数字滤波器估计的电感电流，V iL表示连续域中电感L和寄生阻R L两端的电压瞬时值，V iL[n]表示离散域中第n个采样周期的电感L和寄生阻R L两端的电压平均值；c 1和c 2为数字滤波器系数。
图6是一阶IIR滤波器的数字设计框图，上述差分方程可以通过一阶的IIR滤波器实现，由于数字滤波器系数c 1和c 2为小数形式，在数字实现过程中需要对数据左移N位，即等比例放大2 N倍后变为C 1和C 2。电感L和寄生阻R L两端的电压平均值V iL[n]作为输入，经过延时单元输出V iL[n-1]作为上一个采样周期的输入，其中，第n个周期表示当前周期，则第n-1个周期表示上一个周期；电感电流I L[n]作为输出，经过延时单元输出I L[n-1] 作为上一个采样周期的输出。其中，C 1和C 2在一定工况下为固定的值，因此乘法器的实现均可通过移位的方法，提高运算速度。由于滤波器系数在计算过程中按照比例放大了，因此最终输出需要经过一级除法器，可以采用右移N位的方式实现除法。
图7是本发明在BUCK开关电源中应用的Simplis仿真波形图，其中PWM为上开关管MOS1控制信号波形，I_buck为电感电流实际值，IL_est为电感电流预估值，可以看出通过电感电流预估值能够准确还原实际电感电流平均值和纹波值。
以上内容是结合图示对本发明所做的进一步详细说明，不能认定本发明的具体实施只局限于这些说明，以上所述仅是本发明的优选实施方式。对于本技术领域的技术人员来说，在不脱离本发明原理的前提下，所做的任何修改、等同替换、改进等，均应包含在本发明的保护范围之内。

Claims (7)

  1. 一种DC-DC开关电源的电感电流预估方法，其特征在于，包括电压采样模块、数据转换模块、开关信号计数模块、电感电压计算模块和数字滤波器模块；电压采样模块输入DC-DC开关电源的输入电压(V in)和输出电压(V o)，得到输入电压数字量(D in[n])和输出电压数字量(D o[n])，再经过数据转换模块进行运算和位数转换，得到位数相同的转换后的输入电压(V in[n])和转换后的输出电压(V o[n])，输出给电感电压计算模块；节点电压(V x)通过比较器和参考电压(V ref)进行比较，输出实际的开关信号，再通过开关信号计数模块得到占空比(duty[n])，输出给电感电压计算模块；然后电感电压计算模块计算得到电感(L)和寄生阻(R L)两端的电压平均值(V iL[n])，再经过数字滤波器模块进行滤波，最终得到预估的电感电流(I L[n])。
  2. 根据权利要求1所述的一种DC-DC开关电源的电感电流预估方法，其特征在于，所述电压采样模块包括采样电路和模数转换器(Analog to DigitalConverter，ADC)，采样电路利用分压电阻采样输入电压(V in)和输出电压(V o)按照比例缩放后的电压，再分别通过单端转差分运放后输出给相应的ADC，最终得到输入电压数字量(D in[n])和输出电压数字量(D o[n])。
  3. 根据权利要求1所述的一种DC-DC开关电源的电感电流预估方法，其特征在于，所述数据转换模块接收输入电压数字量(D in[n])和输出电压数字量(D o[n])，按下式运算得到实际的输入电压V in和输出电压V o：
    Figure PCTCN2022110105-appb-100001
    Figure PCTCN2022110105-appb-100002
    其中，D in[n]为输入电压数字量，D o[n]为输出电压数字量，n表示相应数字量处于第n个周期；输入电压(V in)和输出电压(V o)的ADC位数分别为N 1和N 2，输入范围分别为±V 1和±V 2；G 1和G 2分别为输入采样增益系数和输出采样增益系数；
    最后通过位数转换将运算得到的输入电压V in和输出电压V o转换为相同位数的数字量即转换后的输入电压(V in[n])和转换后的输出电压(V o[n])。
  4. 根据权利要求3所述的一种DC-DC开关电源的电感电流预估方法，其特征在于，所述输入采样增益系数和所述输出采样增益系数分别按照下式计算：
    Figure PCTCN2022110105-appb-100003
    Figure PCTCN2022110105-appb-100004
    其中，R 1和R 2为输入电压(V in)的分压电阻对，R 3和R 4为输出电压(V o)的分压电阻对，A 1和A 2分别为输入电压(V in)和输出电压(V o)采样的单端转差分运放的放大系数。
  5. 根据权利要求1所述的一种DC-DC开关电源的电感电流预估方法，其特征在于，所述开关信号计数模块用于检测开关信号并进行计数，将节点电压(V x)与参考电压(V ref)通过比较器进行比较，输出开关信号：若上开关管(MOS1)关断，下开关管(MOS2)导通，此时节点电压(V x)为地电位，节点电压(V x)小于参考电压(V ref)，比较器输出开关信号0；若上开关管(MOS1)导通，下开关管(MOS2)关断，此时节点电压(V x)为输入电压(V in)，节点电压(V x)大于参考电压(V ref)，比较器输出开关信号1；再通过高频计数器计数一个固定周期T s内开关信号为1的次数，每个周期输出一次占空比(duty[n])。
  6. 根据权利要求5所述的一种DC-DC开关电源的电感电流预估方法，其特征在于，所述占空比按下式计算：
    duty[n]＝k n/m
    其中，duty[n]为占空比，m为固定周期T s内允许计数总次数，k n为第n个固定周期T s内开关信号计数值。
  7. 根据权利要求1所述的一种DC-DC开关电源的电感电流预估方法，其特征在于，所述电感(L)和寄生阻(R L)两端的电压平均值(V iL[n])按下式计算：
    V iL[n]＝duty[n]·V in[n]-V o[n]
    其中，V iL[n]为电感(L)和寄生阻(R L)两端的电压平均值，V in[n]为转换后的输入电压，V o[n]为转换后的输出电压。
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