WO2023041023A1 - 一种谐振变换器的控制方法 - Google Patents

一种谐振变换器的控制方法 Download PDF

Info

Publication number
WO2023041023A1
WO2023041023A1 PCT/CN2022/119255 CN2022119255W WO2023041023A1 WO 2023041023 A1 WO2023041023 A1 WO 2023041023A1 CN 2022119255 W CN2022119255 W CN 2022119255W WO 2023041023 A1 WO2023041023 A1 WO 2023041023A1
Authority
WO
WIPO (PCT)
Prior art keywords
voltage
midpoint
vab
switching tube
resonant converter
Prior art date
Application number
PCT/CN2022/119255
Other languages
English (en)
French (fr)
Inventor
王志燊
李永昌
王晨阳
Original Assignee
广州金升阳科技有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 广州金升阳科技有限公司 filed Critical 广州金升阳科技有限公司
Publication of WO2023041023A1 publication Critical patent/WO2023041023A1/zh

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates to the control of switching converters, and in particular to the closed-loop control of resonant converters.
  • the lowest input voltage is 4:1, but between the lowest input voltage of the half bridge and the highest input voltage of the full bridge, due to the different operating frequency and the working state of one of the bridge arms Different, therefore, a transient transition process needs to be introduced to switch between half-bridge and full-bridge, the solution in the paper increases the complexity of the control, and the frequency range is large, the lowest operating frequency is about the highest operating frequency 1/4 of that, which is not conducive to miniaturization design.
  • FIG. 1 shows a schematic diagram of a typical clamped resonant converter, including an inverter circuit, a resonant circuit, a clamping branch, a transformer, and a secondary rectification and filtering circuit;
  • the inverter circuit includes a switch tube Q1, a switch tube Q2, and a switch tube Q3 and the switching tube Q4, the drain of the switching tube Q1 is connected to the drain of the switching tube Q3 and the positive terminal of the input voltage at the same time, the source of the switching tube Q1 is connected to the drain of the switching tube Q2, and the connection point is the left bridge
  • the midpoint of the arm, denoted as a the source of the switching tube Q3 and the drain of the switching tube Q4 are connected together, this connection point is the midpoint of the right bridge arm, denoted as b, the source of the switching tube Q2 is connected to the switch at the same time
  • the resonant circuit includes a reson
  • the clamped resonant converter shown in Figure 1 has a narrower operating frequency range than conventional resonant within the same input voltage range.
  • the following working modes are proposed for the clamped resonant converter to improve efficiency:
  • Figure 2 can widen the input voltage range or output voltage range of the clamped resonant converter within a relatively narrow frequency range, but it still needs to introduce a transient transition process between the full-bridge PWM mode and the half-bridge PWM mode, the control The complexity is still relatively high.
  • the technical problem to be solved by the present invention is to propose a control method for a resonant converter, which can control the steady-state control between each input voltage point while ensuring a wide range of output voltage gain of the clamped resonant converter.
  • a sudden change in the control signal and there is no need to introduce a dynamic transition process between the modes, reducing the complexity of the control.
  • a method for controlling a resonant converter wherein the resonant converter includes an inverter circuit, a resonant circuit, a clamping branch, a transformer, and a secondary side rectification and filtering circuit, characterized in that:
  • the input voltage is divided into three intervals by the first set voltage and the second set voltage, the first set voltage ⁇ the second set voltage;
  • the resonant converter works in asymmetric PWM mode: the midpoint voltage Vab of the left and right bridge arms of the inverter circuit is one of positive voltage and negative voltage The ratio is 50%, and the ratio of the other is less than 50%.
  • the gain of the converter is controlled by adjusting the ratio of the midpoint voltage Vab to the positive voltage and the negative voltage.
  • the proportion of the midpoint voltage Vab being a positive voltage is 50%
  • the method of adjusting the proportion of the midpoint voltage Vab being a positive voltage and a negative voltage is as follows: the proportion of the midpoint voltage Vab being a positive voltage remains unchanged, and the midpoint voltage is changed Vab is the ratio of negative voltage.
  • the midpoint voltage Vab is a positive voltage at the beginning of each cycle, then becomes a negative voltage, and finally becomes a zero voltage, by changing the midpoint voltage Vab from a negative voltage The position of the rising edge becomes zero voltage, thereby changing the proportion of the midpoint voltage Vab being a negative voltage.
  • the second method of changing the ratio of the midpoint voltage Vab to a negative voltage is: the midpoint voltage Vab is a positive voltage at the beginning of each cycle, then becomes a zero voltage, then becomes a negative voltage, and finally becomes a zero voltage, keeping the midpoint
  • the central position of the voltage Vab being a negative voltage remains unchanged, by changing the position of the falling edge of the midpoint voltage Vab from zero voltage to negative voltage and the rising edge of the midpoint voltage Vab from negative voltage to zero voltage at the same time, thereby changing the midpoint
  • the voltage Vab is a proportion of the negative voltage.
  • the third method of changing the ratio of the midpoint voltage Vab to a negative voltage is: the midpoint voltage Vab is a positive voltage at the beginning of each cycle, then becomes a zero voltage, and finally becomes a negative voltage, by changing the midpoint voltage Vab to a zero voltage Change to the position of the falling edge of the negative voltage, thereby changing the ratio of zero to negative voltage.
  • the proportion of the midpoint voltage Vab being a negative voltage is 50%
  • the method for adjusting the proportion of the midpoint voltage Vab being a positive voltage and a negative voltage is as follows: the proportion of the midpoint voltage Vab being a negative voltage remains unchanged, and the midpoint voltage Vab is changed Vab is the ratio of positive voltage.
  • the first set voltage is NVo
  • the second set voltage is 2NVo
  • N is the turn ratio of the primary winding to the secondary winding of the transformer
  • Vo is the output voltage of the resonant converter.
  • the resonant converter works in the full-bridge PFM mode: the positive voltage and the negative voltage of the midpoint voltage Vab of the left and right bridge arms of the inverter circuit account for 50% respectively , by adjusting the switching frequency to control the gain of the converter.
  • the clamped resonant converter works in the half-bridge PWM mode: one switch tube in one of the left and right bridge arms of the inverter circuit is always on, and the other switch The tube is always turned off; the other bridge arm is the working bridge arm, and the driving phases of the two switching tubes differ by 180°, the duty cycle is the same, and both are less than 50%; the working timing of the clamping branch is when the working bridge arm When any one of the switching tubes of the working bridge arm is turned on, the clamping branch is disconnected. When both switching tubes of the working bridge arm are turned off, the clamping branch is turned on; by adjusting the duty of the two switching tubes of the working bridge arm ratio controls the gain of the converter.
  • the inverter circuit includes a switching tube Q1, a switching tube Q2, a switching tube Q3 and a switching tube Q4, the drain of the switching tube Q1 is connected to the drain of the switching tube Q3 and the positive terminal of the input voltage, and the switching tube Q1
  • the source of the switching tube Q2 is connected together with the drain of the switching tube Q2. This connection point is the midpoint of the left bridge arm, which is denoted as a.
  • the source of the switching tube Q3 and the drain of the switching tube Q4 are connected together.
  • the connecting point is The midpoint of the right bridge arm is denoted as b, the source of the switching tube Q2 is connected to the source of the switching tube Q4 and the negative terminal of the input voltage at the same time;
  • the resonant circuit includes a resonant inductance Lr, an excitation inductance Lm and a capacitor Cr, the capacitor One end of Cr is connected to point a, the other end of capacitor Cr is connected to one end of resonant inductance Lr, the other end of resonant inductance Lr is connected to one end of excitation inductance Lm and one end of transformer primary winding, the other end of excitation inductance Lm is connected to transformer primary side The other end of the winding is connected to point b at the same time;
  • the clamping branch includes a switch tube Q5 and a switch tube Q6, the drain of the switch tube Q5 is connected to the other end of the capacitor Cr, and the source of the switch tube Q5 is connected to the source of the switch tube Q6 pole, and
  • Figure 1 is a schematic diagram of a typical clamped resonant converter
  • Figure 2 shows the driving curves of each switching tube of the clamped resonant converter in the full-bridge PFM mode
  • Figure 3 shows the midpoint voltage curves of the two bridge arms of the three implementation methods of the clamped resonant converter in the asymmetrical PWM mode
  • Fig. 4 is the driving curve of each switching tube in the third implementation method of the clamped resonant converter in the asymmetrical PWM mode
  • Fig. 5 is the driving curve of each switching tube of the clamped resonant converter in the half-bridge PWM mode
  • Figure 6 is the output voltage gain curve of the clamped resonant converter.
  • the schematic diagram of the clamped resonant converter involved in the present invention is the prior art shown in Figure 1.
  • the innovation of the present invention is to propose the following control method for the steady-state control of the clamped resonant converter:
  • the input voltage is divided into three intervals by the first set voltage and the second set voltage, the first set voltage ⁇ the second set voltage;
  • the present invention is preferably here
  • the voltage ratios are 50% each, and the gain of the converter is controlled by adjusting the switching frequency.
  • the clamping branch composed of the switching tubes Q5 and Q6 is always in the off state.
  • the clamped resonant converter works in the full-bridge PFM mode.
  • the present invention requires the clamped resonant converter to work in an asymmetrical PWM mode: the midpoint voltage Vab of the two bridge arms on the primary side is positive and negative asymmetrical, including The following two situations:
  • the proportion of positive voltage is 50%, the proportion of negative voltage is less than 50%, the proportion of positive voltage remains unchanged, and the gain of the converter is changed by changing the proportion of negative voltage.
  • Figure 3 shows three implementation methods of maintaining the positive voltage ratio of 50% and changing the negative voltage ratio: Method 1, the midpoint voltage Vab of the bridge arm is a positive voltage at the beginning of each cycle, then becomes a negative voltage, and finally becomes zero voltage , by changing the position of the rising edge where the midpoint voltage Vab is a negative voltage and becomes zero voltage, thereby changing the proportion of the midpoint voltage Vab being a negative voltage, and changing the gain of the converter; The point voltage Vab is a positive voltage, then becomes zero voltage, then becomes a negative voltage, and finally becomes a zero voltage, keeping the center position of the midpoint voltage Vab as a negative voltage, and changing the midpoint voltage Vab to zero voltage at the same time becomes The falling edge of the negative voltage and the position of the rising edge where the midpoint voltage Vab becomes negative voltage becomes zero voltage, thereby changing the proportion of the mid
  • FIG. 4 shows the driving curve of the switching tube in the third implementation method, wherein, the driving of the switching tube Q1 and the switching tube Q2 are complementary, and the duty cycle is 50% respectively, and the switching tube Q4
  • the rising edge is consistent with the switching tube Q1
  • the duty cycle of the switching tube Q4 is greater than 50%
  • the driving of the switching tube Q3 and the switching tube Q4 are complementary.
  • the present invention preferably operates in This voltage range works in the half-bridge PWM mode: one switch tube in one of the two bridge arms on the primary side is always on, the other switch tube is always off, and the other bridge arm is the working bridge arm, and the driving phase difference between the upper and lower two tubes 180°, the duty cycle is the same, both less than 50%, the clamping branch and the two tubes of the working bridge arm work complementary, that is, when any switch tube of the working bridge arm is turned on, the clamping branch is disconnected, when the working bridge arm When both tubes of the bridge arm are turned off, the clamping branch is turned on.
  • the gain of the converter is adjusted by adjusting the duty cycle of the two switching tubes of the working bridge arm.
  • the driving of each switching tube is shown in Figure 5, wherein, the driving duty ratio of switching tube Q1 and switching tube Q2 is the same, the phase difference is 180°, switching tube Q3 is normally off, switching tube Q4 is normally on, switching tube Q5 is driven and Q1 is driven Complementary, the switching tube Q6 and Q1 drive complementary.
  • the gain of the converter is adjusted by adjusting the duty ratio of the switch tube Q1.
  • Figure 6 is the gain curve of the above control method.
  • the axis of abscissa represents the switching frequency fs
  • the axis of ordinate represents the gain G.
  • the conversion The converter works in the low-voltage full-bridge PFM mode, and the gain is adjusted by adjusting the operating frequency; when the switching frequency fs is equal to the resonant frequency fr, and the duty cycle of the switching tube Q1 is 50%, the converter works in the asymmetrical PWM working mode, and the switching tube is adjusted by adjusting
  • the duty cycle D4 of Q4 adjusts the gain; when the switching frequency fs is equal to the resonant frequency fr and the duty cycle of the switch tube Q4 is 100%, the converter works in the half-bridge PWM mode, and the gain is adjusted by adjusting the duty cycle D1 of the switch tube Q1 .
  • the duty cycle D4 of the switch tube Q1 is adjusted by adjusting the duty cycle D1 of the switch tube Q1 .
  • clamped resonant converter shown in Figure 1 should not be regarded as a limitation to the specific circuit to which the present invention is applicable, and other types of resonant converters with clamping branches also have the problems described in the background , the present invention is also applicable.
  • the resonant inductance Lr in Fig. 1 is moved between the center tap of the secondary winding and the connection point between the output filter capacitor Co and the output negative terminal Vo-. Structure.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

本发明公开了一种谐振变换器的控制方法，所述的谐振变换器包括逆变电路、谐振谐振腔、钳位支路、变压器以及副边整流滤波电路，通过第一设定电压和第二设定电压将输入电压分为三个区间，第一设定电压＜第二设定电压；当"第一设定电压＜输入电压≤第二设定电压"时，谐振变换器工作于不对称PWM模式：逆变电路左右两个桥臂中点电压Vab为正电压和负电压之一的比例为50％，另外一个的比例小于50％，通过调节中点电压Vab为正电压和负电压的比例控制变换器的增益。本发明可以使谐振变换器在整个输入电压范围内电压增益连续，不存在电压跳变，不需要增加过渡过程，并且频率变化范围较窄。

Description

一种谐振变换器的控制方法 技术领域
本发明涉及开关变换器控制，特别涉及谐振变换器的闭环控制。
背景技术
由于谐振可以实现ZVS(零电压开关)以及准ZCS(零电流开关)，因此常常用于高频高功率密度变换器中。当输入电压或输出电压范围较宽时，若采用传统的PFM方式对谐振进行闭环控制，则需要进行较宽范围的变频，不利于滤波器的设计。
Milan M.
Figure PCTCN2022119255-appb-000001
在论文《On-the-Fly Topology-Morphing Control Efficiency Optimization Method for谐振Resonant Converters Operating in Wide Input-and/or Output-Voltage Range》中，对传统谐振拓扑，采用半桥——全桥切换的方案，使得谐振可工作于最高输入电压：最低输入电压为4:1的场合，但在半桥的最低输入电压和全桥的最高输入电压之间，由于工作频率不同，并且其中一桥臂的工作状态不同，因此，需要引入一个瞬态过渡过程，在半桥和全桥之间进行切换，论文中的解决方案增加了控制的复杂度，而且频率变化范围较大，最低工作频率约为最高工作频率的1/4，不利于小型化设计。
图1所示为典型的箝位谐振变换器原理图，包括逆变电路、谐振电路、钳位支路、变压器以及副边整流滤波电路；逆变电路包括开关管Q1、开关管Q2、开关管Q3以及开关管Q4，开关管Q1的漏极同时连接开关管Q3的漏极和输入电压的正端，开关管Q1的源极和开关管Q2的漏极连接在一起，该连接点为左桥臂的中点，记为a，开关管Q3的源极和开关管Q4的漏极连接在一起，该连接点为右桥臂的中点，记为b，开关管Q2的源极同时连接开关管Q4的源极和输入电压的负端；谐振电路包括谐振电感Lr、励磁电感Lm以及电容Cr，电容Cr的一端连接a点，电容Cr的另一端连接谐振电感Lr的一端，谐振电感Lr的另一端同时连接励磁电感Lm的一端和变压器原边绕组的一端，励磁电感Lm的另一端和变压器原边绕组的另一端同时连接b点；钳位支路包括开关管Q5以及开关管Q6，开关管Q5的漏极连接电容Cr的另一端，开关管Q5的源极连接 开关管Q6的源极，开关管Q6的漏极连接b点，钳位支路的作用为对谐振电感Lr中的电流进行钳位，从而实现软启动过程中逆变电路中的开关管零电压开通。
图1所示的箝位谐振变换器在相同的输入电压范围内，工作频率范围会比传统谐振窄。公开号为CN110768535A的专利文献《一种变拓扑谐振谐振变换器的宽增益控制方法》中，对箝位谐振变换器提出以下工作模式以提升效率：
低压输入时工作于全桥PFM模式；中压输入时工作于全桥PWM模式；高压输入时工作于半桥PWM模式。其控制方法示意图如图2所示，开关管Q1和开关管Q4的驱动一致，开关管Q2和开关管Q3的驱动一致，开关管Q1驱动与开关管Q2互补，各开关管的占空比为50％，开关管Q5和Q6关断。
图2可以在相对较窄的频率变化范围内拓宽箝位谐振变换器的输入电压范围或输出电压范围，但在全桥PWM模式和半桥PWM模式之间依然需要引入一个瞬态过渡过程，控制复杂度还是比较高。
综上，在箝位谐振变换器中，对同样的输入和输出电压，可以采用多种控制方法，目前在输入电压较宽的范围内，怎样设置控制方法使得输入电压或输出电压范围较宽有待研究创新。
发明内容
有鉴于此，本发明要解决的技术问题是：提出一种谐振变换器的控制方法，在保证箝位谐振变换器输出电压增益范围较宽的情况下，稳态控制各个输入电压点之间不存在控制信号突变，不需要在模态之间引入动态过渡过程，降低控制的复杂度。
解决上述技术问题，本发明所采用的技术方案如下：
一种谐振变换器的控制方法，所述的谐振变换器包括逆变电路、谐振电路、钳位支路、变压器以及副边整流滤波电路，其特征在于：
通过第一设定电压和第二设定电压将输入电压分为三个区间，第一设定电压＜第二设定电压；
当“第一设定电压＜输入电压≤第二设定电压”时，谐振变换器工作于不对称PWM模式：逆变电路左右两个桥臂中点电压Vab为正电压和负电压之一的比例为50％，另外一个的比例小于50％，通过调节中点电压Vab为正电压和负电压的比例控制变换器的增益。
进一步地，中点电压Vab为正电压的比例为50％，调节中点电压Vab为正电压和负电压的比例的方法为：中点电压Vab为正电压的比例保持不变，改变中点电压Vab为负电压的比例。
改变中点电压Vab为负电压的比例的方法之一为：每个周期开始时中点电压Vab为正电压，然后变为负电压，最后变成零电压，通过改变中点电压Vab由负电压变为零电压的上升沿的位置，从而改变中点电压Vab为负电压的占比。
改变中点电压Vab为负电压的比例的方法之二为：每个周期开始时中点电压Vab为正电压，然后变为零电压，接着变为负电压，最后变成零电压，保持中点电压Vab为负电压的中心位置不变，通过同时改变中点电压Vab由零电压变为负电压的下降沿以及中点电压Vab由负电压变为零电压的上升沿的位置，从而改变中点电压Vab为负电压的占比。
改变中点电压Vab为负电压的比例的方法之三为：每个周期开始时中点电压Vab为正电压，然后变为零电压，最后变成负电压，通过改变中点电压Vab为零电压变为负电压的下降沿的位置，从而改变零为负电压的占比。
进一步地，中点电压Vab为负电压的比例为50％，调节中点电压Vab为正电压和负电压的比例的方法为：中点电压Vab为负电压的比例保持不变，改变中点电压Vab为正电压的比例。
优选地，第一设定电压为NVo，第二设定电压为2NVo，N为变压器原边绕组与副边绕组的匝比，Vo为谐振变换器的输出电压。
进一步地，当“输入电压≤第一设定电压”时，谐振变换器工作于全桥PFM模式：逆变电路左右两个桥臂中点电压Vab的正电压与负电压占比各为50％，通过调节开关频率控制变换器的增益。
进一步地，当“输入电压＞第二设定电压”时，箝位谐振变换器工作于半桥PWM模式：逆变电路左右两个桥臂之一中的一个开关管恒导通，另外一个开关管恒关断；另一个桥臂为工作桥臂，其中的两个开关管的驱动相位相差180°，占空比相同，且都小于50％；箝位支路的工作时序为当工作桥臂的任意一个开关管导通时，箝位支路断开，当工作桥臂的两个开关管都关断时，箝位支路导通；通过调节工作桥臂的两个开关管的占空比控制变换器的增益。
优选地，所述的逆变电路包括开关管Q1、开关管Q2、开关管Q3以及开关 管Q4，关管Q1的漏极同时连接开关管Q3的漏极和输入电压的正端，开关管Q1的源极和开关管Q2的漏极连接在一起，该连接点为左桥臂的中点，记为a，开关管Q3的源极和开关管Q4的漏极连接在一起，该连接点为右桥臂的中点，记为b，开关管Q2的源极同时连接开关管Q4的源极和输入电压的负端；所述的谐振电路包括谐振电感Lr、励磁电感Lm以及电容Cr，电容Cr的一端连接a点，电容Cr的另一端连接谐振电感Lr的一端，谐振电感Lr的另一端同时连接励磁电感Lm的一端和变压器原边绕组的一端，励磁电感Lm的另一端和变压器原边绕组的另一端同时连接b点；所述的钳位支路包括开关管Q5以及开关管Q6，开关管Q5的漏极连接电容Cr的另一端，开关管Q5的源极连接开关管Q6的源极，开关管Q6的漏极连接b点。
基于以上技术方案，与现有技术相比，本发明有益效果如下：
(1)通过将输入电压划分为三个区间，设置输入电压为中压段时箝位谐振变换器工作于不对称PWM模式，通过调节桥臂中点电压为正负电压的比例控制变换器的增益，使得箝位谐振变换器可工作于输入电压范围较宽或者输出电压范围较宽的场合；
(2)箝位谐振变换器在三个输入电压区间的增益连续，工作模式切换不需过渡，控制复杂度低。
附图说明
利用附图对本发明作进一步说明，但附图中的实施例不构成对本发明的任何限制，对于本领域的普通技术人员，在不付出创造性劳动的前提下，还可以根据以下附图获得其它的附图。
图1为典型的箝位谐振变换器原理图；
图2为箝位谐振变换器在全桥PFM模式下各开关管驱动曲线；
图3为箝位谐振变换器在不对称PWM模式下三种实现方法两桥臂中点电压曲线；
图4为箝位谐振变换器在不对称PWM模式下第三种实现方法中各开关管驱动曲线；
图5为箝位谐振变换器在半桥PWM模式下各开关管驱动曲线；
图6为箝位谐振变换器的输出电压增益曲线。
具体实施方式
本发明涉及的箝位谐振变换器原理图为图1所示的现有技术，本发明的创新点在于针对箝位谐振变换器的稳态控制提出如下控制方法：
通过第一设定电压和第二设定电压将输入电压分为三个区间，第一设定电压＜第二设定电压；
当“输入电压≤第一设定电压”时，对箝位谐振变换器的工作模式不作要求，由于该区域内箝位谐振变换器工作于全桥PFM模式效率较高，故本发明优选在此电压区间工作于全桥PFM模式：当fs＝fr时，fs为开关频率，fr为谐振频率，各开关管的驱动与图2相同，原边两桥臂中点相对电压Vab为正电压与负电压占比各为50％，通过调节开关频率对变换器的增益进行控制，在该控制方式下，开关管Q5和Q6组成的箝位支路一直处于关断状态。另外，箝位谐振变换器工作于全桥PFM模式，当fs＝fr时，钳位谐振变换器的增益G＝Vo/Vin＝1/N，故Vin＝NVo，因此优选第一设定电压为NVo。
当“第一设定电压时＜输入电压≤第二设定电压”时，本发明要求箝位谐振变换器工作于不对称PWM模式：原边两桥臂中点电压Vab正负不对称，包括如下两种情形：
(1)正电压的比例为50％，负电压的比例小于50％，正电压的比例保持不变，通过改变负电压的比例改变变换器的增益。图3展示了正电压比例维持50％、改变负电压比例的三种实现方法：方法一，每个周期开始时桥臂中点电压Vab为正电压，然后变为负电压，最后变成零电压，通过改变中点电压Vab为负电压变为零电压的上升沿的位置，从而改变中点电压Vab为负电压的占比，改变变换器的增益；方法二，每个周期开始时桥臂中点电压Vab为正电压，然后变为零电压，接着变为负电压，最后变成零电压，保持中点电压Vab为负电压中心位置不变，通过同时改变中点电压Vab为零电压变为负电压的下降沿以及中点电压Vab为负电压变为零电压的上升沿的位置，从而改变中点电压Vab为负电压的占比，改变变换器的增益；方法三，每个周期开始时桥臂中点电压Vab为正电压，然后变为零电压，最后变成负电压，通过改变中点电压Vab为零电压变为负电压的下降沿的位置，从而改变中点电压Vab为负电压的占比，改变变换器的增益；图4给出了第三种实现方法中的开关管驱动曲线，其中，开关 管Q1和开关管Q2驱动互补，占空比各为50％，开关管Q4上升沿与开关管Q1一致，开关管Q4占空比大于50％，开关管Q3和开关管Q4驱动互补。通过调节开关管Q3和Q4的占空比，就可以调节臂中点电压Vab为负电压的占比，从而改变变换器的增益。
(2)正电压的比例小于50％，负电压的比例为50％，负电压的比例保持不变，通过改变正电压的比例改变变换器的增益。
当“输入电压＞第二设定电压”时，对箝位谐振变换器的工作模式也不作要求，由于该区域内箝位谐振变换器工作于半桥PWM模式效率较高，故本发明优选在此电压区间工作于半桥PWM模式：原边两桥臂之一中的一个开关管恒导通，另外一个开关管恒关断，另一桥臂为工作桥臂，上下两管的驱动相位相差180°，占空比相同，都小于50％，箝位支路与工作桥臂的两管互补工作，即当工作桥臂的任意一个开关管导通时，箝位支路断开，当工作桥臂的两管都关断时，箝位支路导通。通过调节工作桥臂的两开关管的占空比调整变换器增益。各开关管的驱动如图5，其中，开关管Q1和开关管Q2驱动占空比相同，相位相差180°，开关管Q3常关断，开关管Q4常导通，开关管Q5驱动与Q1驱动互补，开关管Q6与Q1驱动互补。通过调节开关管Q1的占空比调节变换器增益。另外，箝位谐振变换器工作于半桥PWM模式，当fs＝fr时，G＝Vo/Vin＝1/2N，故Vin＝NVo，因此优选第二设定电压为2NVo。
图6为上述控制方法的增益曲线，横坐标轴代表开关频率fs、纵坐标轴代表增益G，从图6可以看出，当开关频率fs小于谐振电感Lr和电容Cr的谐振频率fr时，变换器工作于低压全桥PFM模式，通过调节工作频率调节增益；当开关频率fs等于谐振频率fr，开关管Q1占空比为50％时，变换器工作于不对称PWM工作模式，通过调节开关管Q4的占空比D4调节增益；当开关频率fs等于谐振频率fr，开关管Q4占空比为100％时，变换器工作于半桥PWM模式，通过调节开关管Q1的占空比D1调节增益。从图6可知，三种控制方式的增益是连续的，不存在跳变，因此在全输入电压控制过程中，不需要设置全桥和半桥之间的切换过程，降低了控制复杂度。
需要说明的是，图1所示的箝位谐振变换器不应当视为对本发明所适用的具体电路的限制，其他类型的带有钳位支路的谐振变换器也存在背景技术所述 的问题，本发明同样适用，例如将图1中的谐振电感Lr移动到副边绕组中心抽头与输出滤波电容Co与输出负端Vo-的连接点之间，又如，变压器采用原边串联副边并联的结构。
以上实施例的说明只是用于帮助理解本申请的发明构思，并不用以限制本发明，对于本技术领域的普通技术人员来说，凡在不脱离本发明原理的前提下，所作的任何修改、等同替换、改进等，均应包含在本发明的保护范围之内。

Claims (10)

  1. 一种谐振变换器的控制方法，所述的谐振变换器包括逆变电路、谐振电路、钳位支路、变压器以及副边整流滤波电路，其特征在于：
    通过第一设定电压和第二设定电压将输入电压分为三个区间，第一设定电压＜第二设定电压；
    当“第一设定电压＜输入电压≤第二设定电压”时，谐振变换器工作于不对称PWM模式：逆变电路左右两个桥臂中点电压Vab为正电压和负电压之一的比例为50％，另外一个的比例小于50％，通过调节中点电压Vab为正电压和负电压的比例控制变换器的增益。
  2. 根据权利要求1所述的谐振变换器的控制方法，其特征在于，中点电压Vab为正电压的比例为50％，调节中点电压Vab为正电压和负电压的比例的方法为：中点电压Vab为正电压的比例保持不变，改变中点电压Vab为负电压的比例。
  3. 根据权利要求2所述的谐振变换器的控制方法，其特征在于，改变中点电压Vab为负电压的比例的方法为：每个周期开始时中点电压Vab为正电压，然后变为负电压，最后变成零电压，通过改变中点电压Vab由负电压变为零电压的上升沿的位置，从而改变中点电压Vab为负电压的占比。
  4. 根据权利要求2所述的谐振变换器的控制方法，其特征在于，改变中点电压Vab为负电压的比例的方法为：每个周期开始时中点电压Vab为正电压，然后变为零电压，接着变为负电压，最后变成零电压，保持中点电压Vab为负电压的中心位置不变，通过同时改变中点电压Vab由零电压变为负电压的下降沿以及中点电压Vab由负电压变为零电压的上升沿的位置，从而改变中点电压Vab为负电压的占比。
  5. 根据权利要求2所述的谐振变换器的控制方法，其特征在于，改变中点电压Vab为负电压的比例的方法为：每个周期开始时中点电压Vab为正电压，然后变为零电压，最后变成负电压，通过改变中点电压Vab为零电压变为负电压的下降沿的位置，从而改变零为负电压的占比。
  6. 根据权利要求1所述的谐振变换器的控制方法，其特征在于，中点电压Vab为负电压的比例为50％，调节中点电压Vab为正电压和负电压的比例的方法 为：中点电压Vab为负电压的比例保持不变，改变中点电压Vab为正电压的比例。
  7. 根据权利要求1所述的谐振变换器的控制方法，其特征在于：第一设定电压为NVo，第二设定电压为2NVo，N为变压器原边绕组与副边绕组的匝比，Vo为谐振变换器的输出电压。
  8. 根据权利要求1所述的谐振变换器的控制方法，其特征在于：当“输入电压≤第一设定电压”时，谐振变换器工作于全桥PFM模式：逆变电路左右两个桥臂中点电压Vab的正电压与负电压占比各为50％，通过调节开关频率控制变换器的增益。
  9. 根据权利要求1所述的谐振变换器的控制方法，其特征在于：当“输入电压＞第二设定电压”时，箝位谐振变换器工作于半桥PWM模式：逆变电路左右两个桥臂之一中的一个开关管恒导通，另外一个开关管恒关断；另一个桥臂为工作桥臂，其中的两个开关管的驱动相位相差180°，占空比相同，且都小于50％；箝位支路的工作时序为当工作桥臂的任意一个开关管导通时，箝位支路断开，当工作桥臂的两个开关管都关断时，箝位支路导通；通过调节工作桥臂的两个开关管的占空比控制变换器的增益。
  10. 根据权利要求1所述的谐振变换器的控制方法，其特征在于：
    所述的逆变电路包括开关管Q1、开关管Q2、开关管Q3以及开关管Q4，关管Q1的漏极同时连接开关管Q3的漏极和输入电压的正端，开关管Q1的源极和开关管Q2的漏极连接在一起，该连接点为左桥臂的中点，记为a，开关管Q3的源极和开关管Q4的漏极连接在一起，该连接点为右桥臂的中点，记为b，开关管Q2的源极同时连接开关管Q4的源极和输入电压的负端；
    所述的谐振电路包括谐振电感Lr、励磁电感Lm以及电容Cr，电容Cr的一端连接a点，电容Cr的另一端连接谐振电感Lr的一端，谐振电感Lr的另一端同时连接励磁电感Lm的一端和变压器原边绕组的一端，励磁电感Lm的另一端和变压器原边绕组的另一端同时连接b点；
    所述的钳位支路包括开关管Q5以及开关管Q6，开关管Q5的漏极连接电容Cr的另一端，开关管Q5的源极连接开关管Q6的源极，开关管Q6的漏极连接b点。
PCT/CN2022/119255 2021-09-18 2022-09-16 一种谐振变换器的控制方法 WO2023041023A1 (zh)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
CN202111104034.1 2021-09-18
CN202111104034.1A CN115833592A (zh) 2021-09-18 2021-09-18 一种谐振变换器的控制方法

Publications (1)

Publication Number Publication Date
WO2023041023A1 true WO2023041023A1 (zh) 2023-03-23

Family

ID=85516020

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/CN2022/119255 WO2023041023A1 (zh) 2021-09-18 2022-09-16 一种谐振变换器的控制方法

Country Status (2)

Country Link
CN (1) CN115833592A (zh)
WO (1) WO2023041023A1 (zh)

Citations (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN102723876A (zh) * 2012-06-08 2012-10-10 上海市电力公司 一种全桥串联谐振变换器拓扑结构
US20120300503A1 (en) * 2011-05-23 2012-11-29 System General Corp. Control circuit with zvs-lock and asymmetrical pwm for resonant power converter
US20140036545A1 (en) * 2012-07-31 2014-02-06 General Electric Corporation Llc converter with dynamic gain transformation for wide input and output range
CN105978356A (zh) * 2016-07-05 2016-09-28 陕西科技大学 一种有源钳位串并联全桥dc/dc变换器
CN110768535A (zh) * 2019-10-22 2020-02-07 广州金升阳科技有限公司 一种变拓扑llc谐振变换器的宽增益控制方法
CN112087147A (zh) * 2020-09-28 2020-12-15 广州金升阳科技有限公司 一种变换器宽增益控制方法及其应用
CN112713780A (zh) * 2020-12-30 2021-04-27 江苏南自通华智慧能源股份有限公司 一种双有源桥变换器不对称电压匹配移相控制方法

Patent Citations (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20120300503A1 (en) * 2011-05-23 2012-11-29 System General Corp. Control circuit with zvs-lock and asymmetrical pwm for resonant power converter
CN102723876A (zh) * 2012-06-08 2012-10-10 上海市电力公司 一种全桥串联谐振变换器拓扑结构
US20140036545A1 (en) * 2012-07-31 2014-02-06 General Electric Corporation Llc converter with dynamic gain transformation for wide input and output range
CN105978356A (zh) * 2016-07-05 2016-09-28 陕西科技大学 一种有源钳位串并联全桥dc/dc变换器
CN110768535A (zh) * 2019-10-22 2020-02-07 广州金升阳科技有限公司 一种变拓扑llc谐振变换器的宽增益控制方法
CN112087147A (zh) * 2020-09-28 2020-12-15 广州金升阳科技有限公司 一种变换器宽增益控制方法及其应用
CN112713780A (zh) * 2020-12-30 2021-04-27 江苏南自通华智慧能源股份有限公司 一种双有源桥变换器不对称电压匹配移相控制方法

Also Published As

Publication number Publication date
CN115833592A (zh) 2023-03-21

Similar Documents

Publication Publication Date Title
CN110768535B (zh) 一种变拓扑llc谐振变换器的宽增益控制方法
CN110649812B (zh) 一种宽增益范围llc谐振变换器及其控制方法
CN110601543B (zh) 一种llc谐振变换器的宽增益控制方法及其谐振变换器
CN108028605B (zh) 具有保持操作的转换器
WO2021042773A1 (zh) 一种llc谐振变换器及控制方法
CN112202336B (zh) 功率方向可自动切换的双向clllc型变换器控制方法
CN106685231B (zh) 一种原边钳位型软开关全桥变换器及其不对称控制方法
US6310785B1 (en) Zero voltage switching DC-DC converter
CN110165895B (zh) 一种实现宽增益fb-hb llc谐振变换器电路结构和控制方法
CN104578804B (zh) 一种llc谐振变换器
WO2018157796A1 (zh) 一种谐振变换器
US11848604B2 (en) Single-stage AC-DC converter circuit with power factor correction function
WO2022000217A1 (zh) 一种谐振变换器及电压转换方法
CN106130356A (zh) 基于变压器变比可调的宽输入llc谐振变换器及调控方法
WO2024160049A1 (zh) 一种谐振变换器的控制方法
EP3905502A1 (en) Full-bridge circuit and full-bridge converter
WO2023041023A1 (zh) 一种谐振变换器的控制方法
WO2018157797A1 (zh) 一种全桥谐振变换器
CN116191893A (zh) 一种llc谐振变换器
WO2023041021A1 (zh) Llc谐振电路的控制方法以及控制装置
CN110224605B (zh) 一种全桥变换电路
CN109194135B (zh) 一种谐振状态可调型功率变换器的自适应效率优化方法
CN116613993A (zh) 一种谐振变换器的控制方法、电路及谐振变换器
CN114649959A (zh) 一种基于双极性对称移相调制策略的Buck-Boost LLC变换器
CN209982343U (zh) 一种实现宽增益fb-hb llc谐振变换器电路结构

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 22869389

Country of ref document: EP

Kind code of ref document: A1

NENP Non-entry into the national phase

Ref country code: DE

122 Ep: pct application non-entry in european phase

Ref document number: 22869389

Country of ref document: EP

Kind code of ref document: A1