WO2023041021A1 - Llc谐振电路的控制方法以及控制装置 - Google Patents

Llc谐振电路的控制方法以及控制装置 Download PDF

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Publication number
WO2023041021A1
WO2023041021A1 PCT/CN2022/119253 CN2022119253W WO2023041021A1 WO 2023041021 A1 WO2023041021 A1 WO 2023041021A1 CN 2022119253 W CN2022119253 W CN 2022119253W WO 2023041021 A1 WO2023041021 A1 WO 2023041021A1
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Prior art keywords
bridge
mode
switching tube
llc resonant
time point
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PCT/CN2022/119253
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English (en)
French (fr)
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王晨阳
王志燊
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广州金升阳科技有限公司
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Publication of WO2023041021A1 publication Critical patent/WO2023041021A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present application relates to the field of switching converters, and in particular, relates to a control method, a control device, a computer-readable storage medium, a processor, and an LLC resonant system for an LLC resonant circuit.
  • LLC converters Due to its unique topological characteristics, LLC converters are widely used in switching power supplies. Compared with traditional power frequency transformers, LLC converters are light in weight, small in size, low in cost, and have high power quality. Resonant technology, as an optimization method to realize soft switching, has attracted widespread attention.
  • the resonant network is its basic conversion unit. It is turned on or off under certain conditions, reducing switching loss and achieving the purpose of soft switching.
  • the LLC converter can realize the ZVS (Zero Voltage Switch) of the primary side switch tube and the quasi-ZCS (Zero Current Switch, zero current switch) of the secondary side rectifier tube, and is often used in high frequency and high power density power supply products middle.
  • the turn-on will generate a large turn-on current, which will resonate with the inductance of the PCB trace, which will cause the voltage stress of the primary side switch tube to rise; at the same time, due to the hard turn-on of the switch tube of the inverter circuit, a large power loop will also be generated.
  • Current and voltage spikes lead to poor EMI (Electromagnetic Interference) performance of the product and cause reliability risks.
  • EMI Electromagnetic Interference
  • the main purpose of the present application is to provide a control method, a control device, a computer-readable storage medium, a processor and an LLC resonant system for an LLC resonant circuit, so as to solve the problems in the converter between the full-bridge mode and the half-bridge mode in the prior art.
  • ZVS cannot be realized, and the reliability is poor.
  • a method for controlling an LLC resonant circuit includes a full-bridge inverter circuit, the full-bridge inverter circuit includes four three-terminal switch tubes, the The method includes: when the input voltage of the LLC resonant circuit is greater than a set threshold and the LLC resonant circuit is in a full-bridge mode, the time point corresponding to the rising edge of the driving voltage of the switching tube in the control part does not change, and the time point corresponding to the falling edge changes according to the first rule, so that the LLC resonant circuit is converted from the full-bridge mode to the half-bridge mode; when the input voltage is less than or equal to the set threshold, And when the LLC resonant circuit is in the half-bridge mode, the time point corresponding to the rising edge of the driving voltage of the switching tube in the control part remains unchanged, and the time point corresponding to the falling edge changes according to the second rule, so that the LLC resonant circuit is converted from the half
  • the four switch tubes are respectively the first switch tube, the second switch tube, the third switch tube and the fourth switch tube, wherein the first end of the first switch tube and the first end of the third switch tube end connection, the second end of the first switch tube is connected to the first end of the second switch tube, the second end of the third switch tube is connected to the first end of the fourth switch tube, and the second switch tube
  • the second end of the tube is connected to the second end of the fourth switch tube
  • the third ends of the first switch tube, the second switch tube, the third switch tube, and the fourth switch tube are respectively
  • the time point corresponding to the rising edge of the driving voltage of the switching tube in the control part is unchanged, and the time point corresponding to the falling edge changes according to the first rule, including: controlling the time point corresponding to the third rising edge And the time point corresponding to the fourth rising edge remains unchanged, the time point corresponding to the third falling edge and the time point corresponding to the fourth falling edge change according to the first rule, wherein the third rising edge is the third The rising edge
  • the time point corresponding to the third rising edge and the time point corresponding to the fourth rising edge are controlled to remain unchanged, and the time point corresponding to the third falling edge and the time point corresponding to the fourth falling edge are changed according to the first rule, Including: controlling the time point corresponding to the third rising edge to remain unchanged, and controlling the time point corresponding to the third falling edge to move, so that the duty cycle of the third switching tube gradually decreases to 0;
  • the duty cycle of the third switching tube is 0, the time point corresponding to the fourth rising edge is controlled to remain unchanged, and the time point corresponding to the fourth falling edge is controlled to move, so that the time point corresponding to the fourth switching tube The duty cycle gradually increases to 1.
  • the full-bridge mode includes a full-bridge PFM (Pulse Frequency Modulation, pulse frequency modulation) mode and a full-bridge PWM (Pulse Width Modulation, pulse width modulation) mode
  • the half-bridge mode includes a half-bridge PFM mode and half-bridge PWM mode, when the LLC resonant circuit is in a steady state, the full-bridge PFM mode, the full-bridge PWM mode, the half-bridge PFM mode and the half-bridge PWM mode
  • the input voltages corresponding to the modes increase sequentially.
  • the LLC resonant circuit further includes an LLC resonant cavity
  • the LLC resonant cavity includes a fifth switch transistor with three terminals and a sixth switch transistor with three terminals, the first end of the fifth switch transistor is connected to the The second end of the first switching tube is connected, the second end of the fifth switching tube is connected to the first end of the sixth switching tube, the second end of the sixth switching tube is connected to the fourth switching tube connected to the first end of the LLC resonant circuit, when the input voltage of the LLC resonant circuit is greater than the set threshold, and the LLC resonant circuit is in the full-bridge mode
  • the method further includes: when the LLC resonant circuit is in the In the case of a full-bridge PWM mode, obtain the real-time frequency and the real-time duty cycle of the first switch tube or the second switch tube; equal to the resonant frequency at the real-time frequency, and the real-time duty cycle is equal to the first When three predetermined values are reached, control the duty cycle of the fifth switching tube and the sixth
  • the method further includes: when the LLC resonant circuit is in the In the case of a half-bridge PWM mode, obtain the real-time frequency and the real-time duty ratio; when the real-time frequency is equal to the resonant frequency, and the real-time duty ratio is equal to the third predetermined value, control the The duty cycle of the fifth switch tube and the sixth switch tube becomes the third predetermined value, so that the LLC resonant circuit is converted from the half-bridge PWM mode to the full-bridge PFM mode.
  • a control device for an LLC resonant circuit includes a full-bridge inverter circuit, and the full-bridge inverter circuit includes four three-terminal switch tubes,
  • the device includes a first control unit and a second control unit, wherein the first control unit is configured to operate when the input voltage of the LLC resonant circuit is greater than a set threshold, and the LLC resonant circuit is in a full-bridge mode.
  • the time point corresponding to the rising edge of the driving voltage of the switching tube in the control part remains unchanged, and the time point corresponding to the falling edge changes according to the first rule, so that the LLC resonant circuit is changed from the full bridge mode switch to the half-bridge mode;
  • the second control unit is used to control all the The time point corresponding to the rising edge of the driving voltage of the switch tube remains unchanged, and the time point corresponding to the falling edge changes according to the second rule, so that the LLC resonant circuit is converted from the half-bridge mode to the full-bridge mode state.
  • processor configured to run a program, wherein, when the program is running, any one of the methods described above is executed.
  • an LLC resonant system including an LLC resonant circuit and a control device for the LLC resonant circuit, wherein the LLC resonant circuit includes a full-bridge inverter circuit, and the full-bridge
  • the bridge inverter circuit includes four three-terminal switching tubes; the control device is used to implement any one of the methods described above.
  • the full-bridge inverter circuit in the resonant circuit is controlled
  • the time corresponding to the rising edge of the driving voltage of the part of the switch tube remains unchanged, and the time corresponding to the falling edge of the switch is controlled to change according to the first rule, so that the LLC resonant circuit is switched from the full-bridge mode to the half-bridge mode; in the LLC resonant
  • the input voltage of the circuit is less than or equal to the set threshold and in the case of half-bridge mode, the time corresponding to the rising edge of the driving voltage of the part of the switching tube of the control full-bridge inverter circuit remains unchanged, and the time corresponding to its falling edge is controlled Change according to the second rule, so that the LLC resonant circuit is switched from the half-bridge mode to the full-bridge mode.
  • the method uses the input voltage as the judgment condition for switching between the full-bridge mode and the half-bridge mode, and when the mode switching is required, the corresponding time of the rising edge of the driving voltage of the switching tube in the control part remains unchanged, only Change the corresponding moment of the falling edge to realize the gradual change of the duty cycle, so that during the mode switching process, with the gradual change of the duty cycle of the switching tube, the corresponding time of the rising edge remains unchanged, that is, the position of the rising edge does not change.
  • This ensures that the resonant current always meets the ZVS condition during the switching process of the half-bridge mode and the full-bridge mode, and ensures that the stress of the switch tube is low, thus ensuring the reliability of the circuit and ensuring the circuit's ability to resist electromagnetic interference. powerful.
  • FIG. 1 shows a schematic flow chart of a control method of an LLC resonant circuit according to an embodiment of the present application
  • FIG. 2 shows a schematic diagram of an LLC resonant circuit according to an embodiment of the present application
  • FIG. 3 and FIG. 4 show a waveform sequence diagram of switching from a full-bridge mode to a half-bridge mode according to an embodiment of the present application
  • FIG. 5 and FIG. 6 show a waveform sequence diagram of switching from a half-bridge mode to a full-bridge mode according to an embodiment of the present application
  • Fig. 7 shows the schematic diagram of the steady-state control scheme of existing scheme
  • Fig. 8 shows a schematic diagram of a steady-state control scheme according to an embodiment of the present application
  • Fig. 9 shows the efficiency comparison chart corresponding to the solutions in Fig. 7 and Fig. 8;
  • Figure 10 shows the gain curves of the LLC resonant circuit at different frequencies when the duty cycle is 0.5;
  • FIG. 11 shows a schematic diagram of a control device of an LLC resonant circuit according to an embodiment of the present application
  • a typical In an embodiment a method for controlling an LLC resonant circuit, a control device, a computer-readable storage medium, a processor, and an LLC resonant system are provided.
  • a method for controlling an LLC resonant circuit is provided.
  • FIG. 1 is a flowchart of a method for controlling an LLC resonant circuit according to an embodiment of the present application.
  • the above-mentioned LLC resonant circuit includes a full-bridge inverter circuit 101, and the above-mentioned full-bridge inverter circuit 101 includes four three-terminal switching tubes.
  • the method includes the following steps:
  • Step S101 when the input voltage of the LLC resonant circuit is greater than the set threshold and the LLC resonant circuit is in the full-bridge mode, the time point corresponding to the rising edge of the driving voltage of the switching tube in the control part remains unchanged, and The time point corresponding to the falling edge is changed according to the first rule, so that the above-mentioned LLC resonant circuit is converted from the above-mentioned full-bridge mode to the half-bridge mode;
  • Step S102 when the input voltage is less than or equal to the set threshold and the LLC resonant circuit is in the half-bridge mode, the time point corresponding to the rising edge of the driving voltage of the switching tube in the control part remains unchanged, and The time point corresponding to the falling edge is changed according to the second rule, so that the above-mentioned LLC resonant circuit is converted from the above-mentioned half-bridge mode to the above-mentioned full-bridge mode.
  • the above method uses the input voltage as the judgment condition for switching between the full-bridge mode and the half-bridge mode, and when the mode switching is required, the corresponding time of the rising edge of the driving voltage of the control part of the above-mentioned switching tube remains unchanged, and only the falling edge is changed. The corresponding moment of the rising edge is used to realize the gradual change of the duty cycle.
  • the resonant current always meets the ZVS condition, which ensures that the stress of the switch tube is low, thereby ensuring the reliability of the circuit and ensuring the circuit's strong anti-electromagnetic interference ability.
  • the above-mentioned LLC resonant circuit may be any suitable LLC circuit in the prior art.
  • the above-mentioned LLC resonant circuit is a clamped LLC circuit.
  • the above-mentioned set threshold is the input voltage point at which mode switching occurs.
  • the set threshold in practical applications, in order to prevent the above-mentioned LLC resonant circuit from switching back and forth between two modes, the set threshold usually has a hysteresis property. Mode switching is not performed when the above-mentioned input voltage is greater than the above-mentioned threshold value and the above-mentioned LLC resonant circuit is working in the half-bridge state, and the above-mentioned input voltage is less than or equal to the above-mentioned threshold value, and the above-mentioned LLC resonant circuit is not performing mode switching when working in the full-bridge state .
  • the four above-mentioned switch tubes are respectively the first switch tube S1, the second switch tube S2, the third switch tube S3 and the fourth switch tube S4, wherein the first switch tube S1 of the first switch tube S1 One end is connected to the first end of the third switching tube S3, the second end of the first switching tube S1 is connected to the first end of the second switching tube S2, and the second end of the third switching tube S3 is connected to the fourth switching tube S3.
  • the first end of the tube S4 is connected, the second end of the second switching tube S2 is connected to the second end of the fourth switching tube S4, the first switching tube S1, the second switching tube S2, the third switching tube S3 and the third end of the fourth switching tube S4 are respectively used to connect to the drive circuit, the time point corresponding to the rising edge of the driving voltage of the above switching tube in the control part remains unchanged, and the time point corresponding to the falling edge changes according to the first rule , including: controlling the time point corresponding to the third rising edge and the time point corresponding to the fourth rising edge to remain unchanged, and the time point corresponding to the third falling edge and the time point corresponding to the fourth falling edge to change according to the above-mentioned first rule, wherein, The third rising edge is the rising edge of the driving voltage of the third switching transistor S3, the fourth rising edge is the rising edge of the driving voltage of the fourth switching transistor S4, and the third falling edge is the rising edge of the driving voltage of the fourth switching transistor S4.
  • the corresponding falling edge, the above-mentioned fourth falling edge is the falling edge corresponding to the above-mentioned fourth rising edge
  • the time point corresponding to the rising edge of the driving voltage of the above-mentioned switching tube in the control part remains unchanged, and the time point corresponding to the falling edge is according to the first Two rule changes, including: controlling the time point corresponding to the above-mentioned third rising edge and the time point corresponding to the above-mentioned fourth rising edge unchanged, and the time point corresponding to the above-mentioned third falling edge and the time point corresponding to the above-mentioned fourth falling edge according to the above-mentioned Second rule changes.
  • the above method further ensures that the resonant current always satisfies the ZVS condition during the circuit mode switching process by controlling the rising edge corresponding time of the driving voltage of the third switching tube and the fourth switching tube to be constant, and controlling the falling edge to change, thereby It further ensures the reliability and performance of the circuit is better.
  • the driving of the third switch S3 is the same as that of the second switch S2;
  • the driving of the tube S4 is the same as that of the above-mentioned first switching tube S1.
  • the rising edge position of the driving voltage of the above-mentioned third switching tube is controlled to remain unchanged and the falling edge position is gradually moved to the left.
  • the duty cycle of the third switching tube is gradually reduced to 0, that is, the third switching tube is always turned off, and then the rising edge position of the driving voltage of the fourth switching tube is controlled to remain unchanged and the falling edge position is gradually right shift, so that the duty cycle of the fourth switching tube is gradually increased to 1, that is, the fourth switching tube is always turned on, which further ensures that the above-mentioned first switching tube is switched from the full-bridge mode to the half-bridge mode.
  • the three switch tubes and the above-mentioned fourth switch tube are always ZVS.
  • the driving of the first switching tube S1 maintains closed-loop control; the driving of the second switching tube S2 occupies The space ratio is the same as S1, and the phase difference is 180°.
  • the control of the third switching transistor and the fourth switching transistor is an open-loop control.
  • the third switch S3 when the LLC resonant circuit shown in FIG. 2 is in the half-bridge mode, the third switch S3 is always turned off; the fourth switch S4 is always turned on.
  • the abscissa is time and the ordinate is voltage
  • the time point corresponding to the third rising edge and the time point corresponding to the fourth rising edge are controlled to remain unchanged, and the third falling edge corresponds to The time point corresponding to the above-mentioned fourth falling edge and the time point corresponding to the above-mentioned fourth falling edge change according to the above-mentioned second rule, including: controlling the time point corresponding to the above-mentioned fourth rising edge to remain unchanged, and controlling the time point corresponding to the above-mentioned fourth falling edge to move, so that the above-mentioned
  • the duty cycle of the fourth switching tube S4 gradually decreases to a first predetermined value, and the first predetermined value is the duty cycle of the first switching tube; when the duty cycle of the fourth switching
  • the mode switching time can be adjusted by adjusting the speed of gradual change of the duty cycle of the third switching transistor and the fourth switching transistor. This ensures that the length of time is controllable, thereby further ensuring that the control strategy is simple.
  • the above-mentioned full-bridge mode includes a full-bridge PFM mode and a full-bridge PWM mode
  • the above-mentioned half-bridge mode includes half-bridge PFM mode and half-bridge PWM mode.
  • the above-mentioned input voltage corresponding to the PWM mode increases sequentially, that is, this application adopts the steady-state control scheme of low-voltage full-bridge PFM, medium-low voltage full-bridge PWM, medium-high voltage half-bridge PFM, and high-voltage PWM.
  • the present application adopts the half-bridge PFM mode control when the voltage is high, so that the shutdown current is lower than that of the full-bridge PWM control, thereby improving the efficiency.
  • the efficiency curves of the steady-state control scheme in the prior art and the steady-state control scheme of the present application are shown in Figure 9, wherein the abscissa is the voltage value, the ordinate is the efficiency, and the dotted line is the steady-state control in the prior art
  • the efficiency curve corresponding to the scheme is implemented as the efficiency curve corresponding to the steady-state control scheme of the present application. It can be seen that the efficiency of the steady-state control method of the present application is significantly higher than that of the steady-state control scheme in the prior art.
  • the adoption of the above-mentioned control scheme of the present application ensures that the overshoot and undershoot of the output voltage are relatively small during the switching process between the half-bridge mode and the full-bridge mode.
  • switching from the above-mentioned full-bridge PFM mode or the above-mentioned full-bridge PWM mode to the above-mentioned half-bridge PFM mode or the above-mentioned half-bridge PWM mode includes four situations.
  • the first one is switching from the above-mentioned full-bridge PFM mode To the above-mentioned half-bridge PFM mode; second, switch from the above-mentioned full-bridge PFM mode to the above-mentioned half-bridge PWM mode; third, switch from the above-mentioned full-bridge PWM mode to the above-mentioned half-bridge PFM mode; fourth One, switching from the above-mentioned full-bridge PWM mode to the above-mentioned half-bridge PWM mode.
  • Switching from the above-mentioned half-bridge PFM mode or the above-mentioned half-bridge PWM mode to the above-mentioned full-bridge PFM mode or the above-mentioned full-bridge PWM mode also includes four situations.
  • the first one is to switch from the above-mentioned half-bridge PFM mode to by The above-mentioned full-bridge PFM mode;
  • the second type switching from the above-mentioned half-bridge PFM mode to the above-mentioned full-bridge PWM mode;
  • the third type switching from the above-mentioned half-bridge PWM mode to the above-mentioned full-bridge PFM mode;
  • the fourth type switching from the above-mentioned half-bridge PWM mode to the above-mentioned full-bridge PWM mode.
  • There are modal jumps in the above eight situations and the above-mentioned method of the present application is used for control switching.
  • the LLC resonant circuit further includes an LLC resonant cavity 102, and the LLC resonant cavity 102 includes a fifth switch tube S5 with three terminals and a sixth switch tube S6 with three terminals.
  • the first end of the fifth switching tube S5 is connected to the second end of the first switching tube, the second end of the fifth switching tube S5 is connected to the first end of the sixth switching tube S6, and the sixth switching tube S6 The second end is connected to the first end of the fourth switching tube S4, as shown in Figure 3 and Figure 4, Vgs5 is the driving waveform of the fifth switching tube S5, Vgs6 is the driving waveform of the sixth switching tube S6, in the above LLC
  • the above method further includes: when the above-mentioned LLC resonant circuit is in the above-mentioned full-bridge PWM mode, obtaining the above-mentioned first switch The real-time frequency and real-time duty ratio of the tube S1 or the above-mentioned second switch tube S2; when the above-mentioned real-time frequency is equal to the resonant frequency, and the above-ment
  • the above-mentioned method it also includes: obtaining the above-mentioned real-time frequency and the above-mentioned real-time duty cycle when the above-mentioned LLC resonant circuit is in the above-mentioned half-bridge PWM mode; when the above-mentioned real-time frequency is equal to the resonant frequency, and the above-mentioned real-time duty cycle is equal to the above-mentioned third predetermined value , controlling the duty cycle of the fifth switching tube and the sixth switching tube to become the third predetermined value, so that the LLC resonant circuit is converted from the half-bridge PWM mode to the full-bridge PFM mode. In this way, the influence of the working states of the fifth
  • the above-mentioned third predetermined value is 0.5
  • the above-mentioned fourth predetermined value is 0.
  • the gain curves of the LLC circuit at different frequencies are shown in FIG. 10 .
  • the solid line in the figure represents that the clamping branch composed of the fifth switching tube and the sixth switching tube does not work, and the dotted line represents the working of the clamping branch. It can be seen from Figure 10 that when the frequency is the resonant frequency fr of the circuit, Whether the clamping branch works or not has no influence on the circuit gain. Therefore, in this mode, directly turning off the fifth switch tube and the sixth switch tube has no influence on the output of the circuit, and will not cause the output voltage to jump.
  • the frequencies of the first switching tube, the second switching tube, the third switching tube, the fourth switching tube, the fifth switching tube and the sixth switching tube are all controlled by a closed loop , affected by the closed-loop control, the duty cycle of the third switching tube and the fourth switching tube is controlled by the switch, and the driving duty cycle of the fifth switching tube and the sixth switching tube changes between 0 and 0.5 during the switching process It may occur at any stage, and it is also possible that the driving duty of the fifth switching tube and the sixth switching tube oscillates back and forth between 0 and 0.5 during the switching process. In practical applications, the fifth switching tube and the sixth switching tube The sudden change control of the switch tube often adds some hysteresis.
  • the above-mentioned LLC resonant circuit further includes a transformer 103 and a secondary rectification network 104
  • the above-mentioned LLC resonant cavity 102 also includes a resonant inductance Lr, an excitation inductance Lm and a resonant capacitor Cr
  • the transformer 103 includes a transformer 103 composed of a primary winding P1, a secondary winding Q1 and a secondary winding Q2
  • the secondary rectification network 104 includes a full-wave rectification circuit composed of synchronous rectifier tubes SR1 and SR2 and an output filter capacitor Cout.
  • the above-mentioned switch tube may be any feasible three-terminal switch tube in the prior art, such as a triode or a MOS tube.
  • each of the above-mentioned switching tubes is a MOS tube, and the drain of the first switching tube S1 and the drain of the third switching tube S3 are connected together as an LLC resonant circuit
  • the positive input terminal of the input power supply Vin is used to connect the positive terminal of the input power supply Vin.
  • the source of the first switching tube S1 is connected to the drain of the second switching tube S2 and one end of the resonant capacitor Cr, and the other end of the resonant capacitor Cr is connected to the resonant inductor.
  • the source of the fourth switching tube S4 and the source of the second switching tube S2 are connected together as the input negative terminal of the LLC resonant circuit, which is used to connect the negative pole of the input power supply Vin; one end of the secondary winding Q1 of the transformer 103 is connected to The drain of the secondary synchronous rectifier SR2, the source of the synchronous rectifier SR2, the source of the synchronous rectifier SR1 and one end of the output filter capacitor Cout are connected together
  • the above-mentioned LLC resonant circuit in the present application is not limited to the circuit structure shown in FIG. 2 , and it can be any suitable LLC resonant circuit in the prior art.
  • replacing the secondary side rectification network 104 with a bridge rectification structure composed of four switching tubes or diodes, or exchanging the positions of the resonant capacitor Cr and the resonant inductor Lr, LLC resonant circuits with different structures can be obtained.
  • Table 1 shows the state of each switching tube corresponding to FIG. 2 in each mode.
  • the driving duty cycle of the first switching tube S1 is 50%, and the frequency is determined by the closed loop; the driving frequency and duty cycle of the second switching tube S2 are the same as those of the first switching tube S1.
  • the phase difference is 180°; the driving of the third switching tube S3 is the same as that of the second switching tube S2; the driving of the fourth switching tube S4 is the same as that of the first switching tube S1; the fifth switching tube S5 is always off, and the sixth switching tube S5 is always off .
  • the driving frequency of the first switching tube S1 is the resonant frequency, and the duty cycle is determined by the closed loop; the driving frequency and duty cycle of the second switching tube S2 are the same as those of the first switching tube S1, and the phase difference is 180°;
  • the drive of the third switch S3 is the same as that of the second switch S2; the drive of the fourth switch S4 is the same as that of the first switch S1; the drive of the fifth switch S5 is complementary to that of the first switch S1; the drive of the sixth switch S6 is the same as that of the second switch Switch S2 is complementary.
  • the driving duty cycle of the first switching tube S1 is 50%, and the frequency is determined by the closed loop; the driving frequency and duty cycle of the second switching tube S2 are the same as those of the first switching tube S1, and the phase difference is 180°;
  • the third switch tube S3 is always off; the fourth switch tube S4 is always on; the fifth switch tube S5 is always off, and the sixth switch tube S6 is always off.
  • the driving frequency of the first switching tube S1 is the resonant frequency, and the duty cycle is determined by the closed loop; the driving frequency and duty cycle of the second switching tube S2 are the same as those of the first switching tube S1, and the phase difference is 180°;
  • the third switching tube S3 is always turned off; the fourth switching tube S4 is always on; the driving of the fifth switching tube S5 is complementary to that of the first switching tube S1; the driving of the sixth switching tube S6 is complementary to that of the second switching tube S2.
  • the above-mentioned control scheme of the present application can be realized through analog control or digital control.
  • the clamp LLC resonant circuit with output 24V/600W according to the above control scheme of this application, an experimental prototype was built for mode switching verification.
  • the mode switching time was set to 20mS, and the output voltage overshoot and undershoot caused by the switching process was within 3%. Within, and the switching process can realize the ZVS of the switching tube of the inverter circuit.
  • the embodiment of the present application also provides a control device for the LLC resonant circuit. It should be noted that the control device for the LLC resonant circuit in the embodiment of the present application can be used to implement the control for the LLC resonant circuit provided in the embodiment of the present application. method. The control device for the LLC resonant circuit provided in the embodiment of the present application is introduced below.
  • Fig. 11 is a schematic diagram of the control device of the LLC resonant circuit according to the embodiment of the present application.
  • the above-mentioned LLC resonant circuit includes a full-bridge inverter circuit 101, and the above-mentioned full-bridge inverter circuit 101 includes four three-terminal switches Tube, as shown in Figure 11, the device includes a first control unit 10 and a second control unit 20, wherein the first control unit 10 is used to make the input voltage of the LLC resonant circuit greater than the set threshold, and the LLC resonant
  • the circuit is in full-bridge mode, the time point corresponding to the rising edge of the driving voltage of the switch tube in the control part remains unchanged, and the time point corresponding to the falling edge changes according to the first rule, so that the above-mentioned LLC resonant circuit is formed by the above-mentioned
  • the full-bridge mode is converted to the half-bridge mode; the second control unit 20 is used to control the above-mentioned The time point corresponding to
  • the above-mentioned first control unit controls the part of the switching tubes of the full-bridge inverter circuit in the resonant circuit.
  • the time corresponding to the rising edge of the driving voltage remains unchanged, and the time corresponding to the falling edge is controlled to change according to the first rule, so that the LLC resonant circuit is switched from the full-bridge mode to the half-bridge mode; when the input voltage of the LLC resonant circuit is less than Or equal to the set threshold and in the case of the half-bridge mode, the second control unit controls the time corresponding to the rising edge of the driving voltage of the part of the switching tube of the full-bridge inverter circuit to remain unchanged, and controls the time corresponding to the falling edge Change according to the second rule, so that the LLC resonant circuit is switched from the half-bridge mode to the full-bridge mode.
  • the above-mentioned device uses the output voltage as the judgment condition for switching between the full-bridge mode and the half-bridge mode, and when the mode switching is required, the corresponding time of the rising edge of the driving voltage of the control part of the above-mentioned switching tube remains unchanged, and only the falling The corresponding moment of the rising edge is used to realize the gradual change of the duty cycle.
  • the resonant current always meets the ZVS condition, which ensures that the stress of the switch tube is low, thereby ensuring the reliability of the circuit and ensuring the circuit's strong anti-electromagnetic interference ability.
  • the above-mentioned LLC resonant circuit may be any suitable LLC circuit in the prior art.
  • the above-mentioned LLC resonant circuit is a clamped LLC circuit.
  • the above-mentioned set threshold is the input voltage point at which mode switching occurs.
  • the set threshold in practical applications, in order to prevent the above-mentioned LLC resonant circuit from switching back and forth between two modes, the set threshold usually has a hysteresis property. Mode switching is not performed when the above-mentioned input voltage is greater than the above-mentioned threshold value and the above-mentioned LLC resonant circuit is working in the half-bridge state, and the above-mentioned input voltage is less than or equal to the above-mentioned threshold value, and the above-mentioned LLC resonant circuit is not performing mode switching when working in the full-bridge state .
  • the four above-mentioned switch tubes are respectively the first switch tube S1, the second switch tube S2, the third switch tube S3 and the fourth switch tube S4, wherein the first switch tube S1 of the first switch tube S1 One end is connected to the first end of the third switching tube S3, the second end of the first switching tube S1 is connected to the first end of the second switching tube S2, and the second end of the third switching tube S3 is connected to the fourth switching tube S3.
  • the first end of the tube S4 is connected, the second end of the second switching tube S2 is connected to the second end of the fourth switching tube S4, the first switching tube S1, the second switching tube S2, the third switching tube S3 and the third end of the fourth switching tube S4 are respectively used to connect to the drive circuit,
  • the first control unit includes a first control module, and the first control module is used to control the time point corresponding to the third rising edge and the fourth rising edge
  • the time point corresponding to the edge remains unchanged, and the time point corresponding to the third falling edge and the time point corresponding to the fourth falling edge change according to the first rule, wherein the third rising edge is the driving voltage of the third switching tube S3 rising edge, the fourth rising edge is the rising edge of the driving voltage of the fourth switching tube S4, the third falling edge is the falling edge corresponding to the third rising edge, and the fourth falling edge is the rising edge corresponding to the fourth rising edge.
  • the second control unit includes a second control module, the second control module is used to control the time point corresponding to the third rising edge and the time point corresponding to the fourth rising edge unchanged, the third rising edge
  • the time point corresponding to the falling edge and the time point corresponding to the fourth falling edge change according to the above second rule.
  • the device further ensures that the resonant current always satisfies the ZVS condition during the circuit mode switching process by controlling the rising edge corresponding time of the driving voltage of the third switching tube and the fourth switching tube to be constant, and controlling the falling edge to change. It further ensures the reliability and performance of the circuit is better.
  • the driving of the third switch S3 is the same as that of the second switch S2;
  • the driving of the tube S4 is the same as that of the above-mentioned first switching tube S1.
  • Vgs1 is the driving waveform of the first switching tube S1
  • Vgs2 is the driving waveform of the second switching tube S2
  • Vgs3 is the driving waveform of the third switching tube S3
  • Vgs4 is the driving waveform of the fourth switching tube S4.
  • the first control module includes a first control submodule and a second control submodule, wherein the first control submodule is used to control the time point corresponding to the third rising edge to remain unchanged, and to control the third falling edge moving along the corresponding time point, so that the duty cycle of the third switching tube is gradually reduced to 0; the second control submodule is used to control the duty cycle of the third switching tube to 0
  • the time points corresponding to the four rising edges remain unchanged, and the time point corresponding to the fourth falling edge is controlled to move, so that the duty cycle of the fourth switching transistor S4 gradually increases to 1.
  • the rising edge position of the driving voltage of the above-mentioned third switching tube is controlled to remain unchanged and the falling edge position is gradually moved to the left.
  • the duty cycle of the third switching tube is gradually reduced to 0, that is, the third switching tube is always turned off, and then the rising edge position of the driving voltage of the fourth switching tube is controlled to remain unchanged and the falling edge position is gradually right shift, so that the duty cycle of the fourth switching tube is gradually increased to 1, that is, the fourth switching tube is always turned on, which further ensures that the above-mentioned first switching tube is switched from the full-bridge mode to the half-bridge mode.
  • the three switch tubes and the above-mentioned fourth switch tube are always ZVS.
  • the driving of the first switching tube S1 maintains closed-loop control; the driving of the second switching tube S2 occupies The space ratio is the same as S1, and the phase difference is 180°.
  • the control of the third switching transistor and the fourth switching transistor is an open-loop control.
  • the above-mentioned second control module includes a third control sub-module and a fourth control sub-module, wherein the above-mentioned third control sub-module is used to control the time point corresponding to the above-mentioned fourth rising edge to remain unchanged, And control the time point corresponding to the fourth falling edge to move, so that the duty cycle of the fourth switching tube S4 gradually decreases to a first predetermined value, and the first predetermined value is the duty cycle of the first switching tube;
  • the fourth control submodule is used to control the time point corresponding to the third rising edge to remain unchanged and control the time point corresponding to the third falling edge to move when the duty cycle of the fourth switching tube S4 is less than 0.5, The duty cycle of the third switching tube is gradually increased to a second predetermined value
  • the mode switching time can be adjusted by adjusting the speed of gradual change of the duty cycle of the third switching transistor and the fourth switching transistor. This ensures that the length of time is controllable, thereby further ensuring that the control strategy is simple.
  • the above-mentioned full-bridge mode includes a full-bridge PFM mode and a full-bridge PWM mode
  • the above-mentioned half-bridge mode includes half-bridge PFM mode and half-bridge PWM mode.
  • the above-mentioned input voltage corresponding to the PWM mode increases sequentially, that is, this application adopts the steady-state control scheme of low-voltage full-bridge PFM, medium-low voltage full-bridge PWM, medium-high voltage half-bridge PFM, and high-voltage PWM.
  • the present application adopts the half-bridge PFM mode control when the voltage is high, so that the shutdown current is lower than that of the full-bridge PWM control, thereby improving the efficiency.
  • the efficiency curves of the steady-state control scheme in the prior art and the steady-state control scheme of the present application are shown in FIG. From the efficiency curve corresponding to the steady-state control scheme, it can be seen that the efficiency of the steady-state control device of the present application is obviously higher than that of the steady-state control scheme in the prior art.
  • the adoption of the above-mentioned control scheme of the present application ensures that the overshoot and undershoot of the output voltage are relatively small during the switching process between the half-bridge mode and the full-bridge mode.
  • switching from the above-mentioned full-bridge PFM mode or the above-mentioned full-bridge PWM mode to the above-mentioned half-bridge PFM mode or the above-mentioned half-bridge PWM mode includes four situations.
  • the first one is switching from the above-mentioned full-bridge PFM mode To the above-mentioned half-bridge PFM mode; second, switch from the above-mentioned full-bridge PFM mode to the above-mentioned half-bridge PWM mode; third, switch from the above-mentioned full-bridge PWM mode to the above-mentioned half-bridge PFM mode; fourth One, switching from the above-mentioned full-bridge PWM mode to the above-mentioned half-bridge PWM mode.
  • Switching from the above-mentioned half-bridge PFM mode or the above-mentioned half-bridge PWM mode to the above-mentioned full-bridge PFM mode or the above-mentioned full-bridge PWM mode also includes four situations.
  • the first one is to switch from the above-mentioned half-bridge PFM mode to by The above-mentioned full-bridge PFM mode;
  • the second type switching from the above-mentioned half-bridge PFM mode to the above-mentioned full-bridge PWM mode;
  • the third type switching from the above-mentioned half-bridge PWM mode to the above-mentioned full-bridge PFM mode;
  • the fourth type switching from the above-mentioned half-bridge PWM mode to the above-mentioned full-bridge PWM mode.
  • There are modal jumps in the above eight situations and the above-mentioned method of the present application is used for control switching.
  • the LLC resonant circuit further includes an LLC resonant cavity 102, and the LLC resonant cavity 102 includes a fifth switch tube S5 with three terminals and a sixth switch tube S6 with three terminals.
  • the first end of the fifth switching tube S5 is connected to the second end of the first switching tube, the second end of the fifth switching tube S5 is connected to the first end of the sixth switching tube S6, and the sixth switching tube S6 The second end is connected to the first end of the fourth switching tube S4, as shown in Figure 3 and Figure 4, Vgs5 is the driving waveform of the fifth switching tube S5, Vgs6 is the driving waveform of the sixth switching tube S6, in the above LLC
  • the above-mentioned device further includes a first acquisition unit and a third control unit, wherein the above-mentioned first acquisition unit is used in the above-mentioned LLC When the resonant circuit is in the above-mentioned full-bridge PWM mode, obtain the real-time frequency and the real-time duty ratio of the above-mentioned first switch tube S1 or the above-mentioned second
  • the above-mentioned device when the above-mentioned input voltage is less than or equal to the above-mentioned set threshold, and the above-mentioned LLC resonant circuit is in the above-mentioned half-bridge mode, the above-mentioned device It also includes a second acquisition unit and a fourth control unit, wherein the second acquisition unit is used to acquire the real-time frequency and the real-time duty cycle when the LLC resonant circuit is in the half-bridge PWM mode; The fourth control unit is used to control the duty cycle of the fifth switching tube and the sixth switching tube to become the third predetermined value when the real-time frequency is equal to the resonance frequency and the real-time duty cycle is equal to the third predetermined value , so that the above-mentioned LLC resonant circuit is converted from the above-mentioned half-bridge PWM mode to the above-mentioned full-bridge PFM mode. In this way, the influence of the working states of the fifth light switching tube
  • the above-mentioned third predetermined value is 0.5
  • the above-mentioned fourth predetermined value is 0.
  • the gain curves of the LLC circuit at different frequencies are shown in FIG. 10 .
  • the solid line in the figure represents that the clamping branch composed of the fifth switching tube and the sixth switching tube does not work, and the dotted line represents the working of the clamping branch. It can be seen from Figure 10 that when the frequency is the resonant frequency fr of the circuit, Whether the clamping branch works or not has no influence on the circuit gain. Therefore, in this mode, directly turning off the fifth switch tube and the sixth switch tube has no influence on the output of the circuit, and will not cause the output voltage to jump.
  • the frequencies of the first switching tube, the second switching tube, the third switching tube, the fourth switching tube, the fifth switching tube and the sixth switching tube are all controlled by a closed loop , affected by the closed-loop control, the duty cycle of the third switching tube and the fourth switching tube is controlled by the switch, and the driving duty cycle of the fifth switching tube and the sixth switching tube changes between 0 and 0.5 during the switching process It may occur at any stage, and it is also possible that the driving duty of the fifth switching tube and the sixth switching tube oscillates back and forth between 0 and 0.5 during the switching process. In practical applications, the fifth switching tube and the sixth switching tube The sudden change control of the switch tube often adds some hysteresis.
  • the above-mentioned LLC resonant circuit further includes a transformer 103 and a secondary rectification network 104
  • the above-mentioned LLC resonant cavity 102 also includes a resonant inductance Lr, an excitation inductance Lm and a resonant capacitor Cr
  • the transformer 103 includes a transformer 103 composed of a primary winding P1, a secondary winding Q1 and a secondary winding Q2
  • the secondary rectification network 104 includes a full-wave rectification circuit composed of synchronous rectifier tubes SR1 and SR2 and an output filter capacitor Cout.
  • the above-mentioned switch tube may be any feasible three-terminal switch tube in the prior art, such as a triode or a MOS tube.
  • each of the above-mentioned switching tubes is a MOS tube, and the drain of the first switching tube S1 and the drain of the third switching tube S3 are connected together as an LLC resonant circuit
  • the positive input terminal of the input power supply Vin is used to connect the positive terminal of the input power supply Vin.
  • the source of the first switch tube S1 is connected to the drain of the second switch tube S2 and one end of the resonant capacitor Cr, and the other end of the resonant capacitor Cr is connected to the resonant inductor.
  • the source of the fourth switching tube S4 and the source of the second switching tube S2 are connected together as the input negative terminal of the LLC resonant circuit, which is used to connect the negative pole of the input power supply Vin; one end of the secondary winding Q1 of the transformer 103 is connected to The drain of the secondary synchronous rectifier SR2, the source of the synchronous rectifier SR2, the source of the synchronous rectifier SR1 and one end of the output filter capacitor Cout are connected together
  • the above-mentioned LLC resonant circuit in the present application is not limited to the circuit structure shown in FIG. 2 , and it can be any suitable LLC resonant circuit in the prior art.
  • replacing the secondary side rectification network 104 with a bridge rectification structure composed of four switching tubes or diodes, or exchanging the positions of the resonant capacitor Cr and the resonant inductor Lr, LLC resonant circuits with different structures can be obtained.
  • Table 1 shows the state of each switching tube corresponding to FIG. 2 in each mode.
  • the driving duty cycle of the first switching tube S1 is 50%, and the frequency is determined by the closed loop; the driving frequency and duty cycle of the second switching tube S2 are the same as those of the first switching tube S1.
  • the phase difference is 180°; the driving of the third switching tube S3 is the same as that of the second switching tube S2; the driving of the fourth switching tube S4 is the same as that of the first switching tube S1; the fifth switching tube S5 is always off, and the sixth switching tube S5 is always off .
  • the driving frequency of the first switching tube S1 is the resonant frequency, and the duty cycle is determined by the closed loop; the driving frequency and duty cycle of the second switching tube S2 are the same as those of the first switching tube S1, and the phase difference is 180°;
  • the drive of the third switch S3 is the same as that of the second switch S2; the drive of the fourth switch S4 is the same as that of the first switch S1; the drive of the fifth switch S5 is complementary to that of the first switch S1; the drive of the sixth switch S6 is the same as that of the second switch Switch S2 is complementary.
  • the driving duty cycle of the first switching tube S1 is 50%, and the frequency is determined by the closed loop; the driving frequency and duty cycle of the second switching tube S2 are the same as those of the first switching tube S1, and the phase difference is 180°;
  • the third switch tube S3 is always off; the fourth switch tube S4 is always on; the fifth switch tube S5 is always off, and the sixth switch tube S6 is always off.
  • the driving frequency of the first switching tube S1 is the resonant frequency, and the duty cycle is determined by the closed loop; the driving frequency and duty cycle of the second switching tube S2 are the same as those of the first switching tube S1, and the phase difference is 180°;
  • the third switching tube S3 is always turned off; the fourth switching tube S4 is always on; the driving of the fifth switching tube S5 is complementary to that of the first switching tube S1; the driving of the sixth switching tube S6 is complementary to that of the second switching tube S2.
  • the above-mentioned control scheme of the present application can be realized through analog control or digital control.
  • the clamp LLC resonant circuit with output 24V/600W according to the above control scheme of this application, an experimental prototype was built for mode switching verification.
  • the mode switching time was set to 20mS, and the output voltage overshoot and undershoot caused by the switching process was within 3%. Within, and the switching process can realize the ZVS of the switching tube of the inverter circuit.
  • An embodiment of the present invention provides a processor, the processor is used to run a program, wherein the above method for controlling the LLC resonant circuit is executed when the program is running.
  • an LLC resonant system including an LLC resonant circuit and a control device for the LLC resonant circuit, wherein the LLC resonant circuit includes a full-bridge inverter circuit, and the full-bridge
  • the inverter circuit includes four three-terminal switch tubes; the above control device is used to execute any one of the above methods.
  • the above-mentioned LLC resonant system includes an LLC resonant circuit and a control device for the LLC resonant circuit, and the control device is used to execute any one of the above-mentioned methods.
  • the above method uses the input voltage as the judgment condition for switching between the full-bridge mode and the half-bridge mode, and when the mode switching is required, the corresponding time of the rising edge of the driving voltage of the control part of the above-mentioned switching tube remains unchanged, and only the falling edge is changed. The corresponding moment of the rising edge is used to realize the gradual change of the duty cycle.
  • the disclosed technical content can be realized in other ways.
  • the device embodiments described above are only illustrative.
  • the division of the above-mentioned units can be a logical function division.
  • there may be another division method for example, multiple units or components can be combined or integrated. to another system, or some features may be ignored, or not implemented.
  • the mutual coupling or direct coupling or communication connection shown or discussed may be through some interfaces, and the indirect coupling or communication connection of units or modules may be in electrical or other forms.
  • the units described above as separate components may or may not be physically separated, and the components shown as units may or may not be physical units, that is, they may be located in one place, or may be distributed to multiple units. Part or all of the units can be selected according to actual needs to achieve the purpose of the solution of this embodiment.
  • each functional unit in each embodiment of the present invention may be integrated into one processing unit, each unit may exist separately physically, or two or more units may be integrated into one unit.
  • the above-mentioned integrated units can be implemented in the form of hardware or in the form of software functional units.
  • the above integrated units are realized in the form of software function units and sold or used as independent products, they can be stored in a computer-readable storage medium.
  • the essence of the technical solution of the present invention or the part that contributes to the prior art or all or part of the technical solution can be embodied in the form of a software product, and the computer software product is stored in a storage medium , including several instructions to make a computer device (which may be a personal computer, a server, or a network device, etc.) execute all or part of the steps of the above-mentioned methods in various embodiments of the present invention.
  • the aforementioned storage media include: U disk, read-only memory (ROM, Read-Only Memory), random access memory (RAM, Random Access Memory), mobile hard disk, magnetic disk or optical disc, etc., which can store program codes. .
  • the above method uses the input voltage as the judgment condition for switching between the full-bridge mode and the half-bridge mode, and when the mode switching is required, the corresponding time of the rising edge of the driving voltage of the control part of the above-mentioned switching tube remains unchanged, and only the falling edge is changed. The corresponding moment of the rising edge is used to realize the gradual change of the duty cycle.
  • the resonant current always meets the ZVS condition, which ensures that the stress of the switch tube is low, thereby ensuring the reliability of the circuit and ensuring the circuit's strong anti-electromagnetic interference ability.
  • the above-mentioned first control unit controls the full-bridge inverter circuit in the resonant circuit
  • the time corresponding to the rising edge of the driving voltage of the part of the switch tube remains unchanged, and the time corresponding to the falling edge of the switch is controlled to change according to the first rule, so that the LLC resonant circuit is switched from the full-bridge mode to the half-bridge mode;
  • the above-mentioned second control unit controls the time corresponding to the rising edge of the driving voltage of part of the switching tubes of the full-bridge inverter circuit to remain unchanged, and controls its The moment corresponding to the falling edge changes according to the second rule, so that the LLC resonant circuit switches from the half-bridge mode to
  • the above-mentioned device uses the output voltage as the judgment condition for switching between the full-bridge mode and the half-bridge mode, and when the mode switching is required, the corresponding time of the rising edge of the driving voltage of the control part of the above-mentioned switching tube remains unchanged, and only the falling The corresponding moment of the rising edge is used to realize the gradual change of the duty cycle.
  • the resonant current always meets the ZVS condition, which ensures that the stress of the switch tube is low, thereby ensuring the reliability of the circuit and ensuring the circuit's strong anti-electromagnetic interference ability.
  • the above-mentioned LLC resonant system of the present application includes an LLC resonant circuit and a control device for the above-mentioned LLC resonant circuit, and the control device is used to execute any one of the above-mentioned methods.
  • the above method uses the input voltage as the judgment condition for switching between the full-bridge mode and the half-bridge mode, and when the mode switching is required, the corresponding time of the rising edge of the driving voltage of the control part of the above-mentioned switching tube remains unchanged, and only the falling edge is changed. The corresponding moment of the rising edge is used to realize the gradual change of the duty cycle.

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Abstract

本申请提供了一种LLC谐振电路的控制方法以及控制装置，LLC谐振电路包括全桥逆变电路，该方法包括：在LLC谐振电路的输入电压大于设定阈值，且LLC谐振电路处于全桥模态的情况下，控制部分的开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第一规则变化，以使得LLC谐振电路由全桥模态转换至半桥模态；在输入电压小于或者等于设定阈值，且LLC谐振电路处于半桥模态的情况下，控制部分的开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第二规则变化，以使得LLC谐振电路由半桥模态转换至全桥模态。该方法保证了电路的可靠性较好。

Description

LLC谐振电路的控制方法以及控制装置 技术领域
本申请涉及开关变换器领域，具体而言，涉及一种LLC谐振电路的控制方法、控制装置、计算机可读存储介质、处理器以及LLC谐振系统。
背景技术
LLC变换器因其独有的拓扑特性，在开关电源中得到广泛应用，相对于传统的工频变压器，LLC变换器重量轻、体积小、成本底，同时电能质量较高。谐振技术作为实现软开关的优化方法，受到人们的普遍关注，谐振网络是其基本变换单元，发生谐振时，电路中的电流或电压周期性的降为零，从而开关管在零电流或零电压的情况下导通或关断，降低开关损耗，达到软开关的目的。LLC变换器可以实现原边开关管的ZVS(Zero Voltage Switch，零电压开关)以及副边整流管的准ZCS(Zero Current Switch，零电流开关)，常常被用于高频高功率密度的电源产品中。
然而，现有技术方案中针对LLC变换器全桥模态和半桥模态切换的过程中，随着半桥状态下常关断开关管占空比渐变，在其占空比较小时，均存在驱动信号上升沿出现在谐振腔电流过零点附近的情况，在半桥状态下常关断开关管开通前谐振电流已不满足其实现ZVS的条件，导致该开关管硬开通，实际产品应用时硬开通会产生很大的开通电流，该电流与PCB走线电感发生谐振会导致原边开关管的电压应力上升；同时由于逆变电路开关管的硬开通，还会在功率回路中产生较大的电流和电压尖峰，导致产品EMI(Electromagnetic Interference，电磁干扰)性能变差，引起可靠性风险。
在背景技术部分中公开的以上信息只是用来加强对本文所描述技术的背景技术的理解，因此，背景技术中可能包含某些信息，这些信息对于本领域技术人员来说并未形成在本国已知的现有技术。
发明内容
本申请的主要目的在于提供一种LLC谐振电路的控制方法、控制装置、计算机可读存储介质、处理器以及LLC谐振系统，以解决现有技术中变换器在全桥模式和半桥模式之间切换的过程中无法实现ZVS，可靠性较差的问题。
根据本发明实施例的一个方面，提供了一种LLC谐振电路的控制方法，所 述LLC谐振电路包括全桥逆变电路，所述全桥逆变电路包括四个三端子的开关管，所述方法包括：在所述LLC谐振电路的输入电压大于设定阈值，且所述LLC谐振电路处于全桥模态的情况下，控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第一规则变化，以使得所述LLC谐振电路由所述全桥模态转换至半桥模态；在所述输入电压小于或者等于所述设定阈值，且所述LLC谐振电路处于所述半桥模态的情况下，控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第二规则变化，以使得所述LLC谐振电路由所述半桥模态转换至所述全桥模态。
可选地，四个所述开关管分别为第一开关管、第二开关管、第三开关管以及第四开关管，其中，第一开关管的第一端和第三开关管的第一端连接，所述第一开关管的第二端和第二开关管的第一端连接，所述第三开关管的第二端与第四开关管的第一端连接，所述第二开关管的第二端与所述第四开关管的第二端连接，所述第一开关管、所述第二开关管、所述第三开关管以及所述第四开关管的第三端分别用于连接驱动电路，控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第一规则变化，包括：控制第三上升沿对应的时间点以及第四上升沿对应的时间点不变，第三下降沿对应的时间点以及第四下降沿对应的时间点按照所述第一规则变化，其中，所述第三上升沿为所述第三开关管的驱动电压的上升沿，所述第四上升沿为所述第四开关管的驱动电压的上升沿，所述第三下降沿为与所述第三上升沿对应的下降沿，所述第四下降沿为与所述第四上升沿对应的下降沿，控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第二规则变化，包括：控制所述第三上升沿对应的时间点以及所述第四上升沿对应的时间点不变，所述第三下降沿对应的时间点以及所述第四下降沿对应的时间点按照所述第二规则变化。
可选地，控制第三上升沿对应的时间点以及第四上升沿对应的时间点不变，第三下降沿对应的时间点以及第四下降沿对应的时间点按照所述第一规则变化，包括：控制所述第三上升沿对应的时间点不变，且控制所述第三下降沿对应的时间点移动，使得所述第三开关管的占空比逐渐减小至0；在所述第三开关管的占空比为0的情况下，控制所述第四上升沿对应的时间点不变，且控制所述第四下降沿对应的时间点移动，使得所述第四开关管的占空比逐渐增大至1。
可选地，控制所述第三上升沿对应的时间点以及所述第四上升沿对应的时间点不变，所述第三下降沿对应的时间点以及所述第四下降沿对应的时间点按照所述第二规则变化，包括：控制所述第四上升沿对应的时间点不变，且控制所述第四下降沿对应的时间点移动，使得所述第四开关管的占空比逐渐减小至第一预定值，所述第一预定值为所述第一开关管的占空比；在所述第四开关管的占空比小于0.5的情况下，控制所述第三上升沿对应的时间点不变，且控制所述第三下降沿对应的时间点移动，使得所述第三开关管的占空比逐渐增大至第二预定值，所述第二预定值为所述第二开关管的占空比。
可选地，所述全桥模态包括全桥PFM(Pulse Frequency Modulation，脉冲频率调制)模态以及全桥PWM(Pulse Width Modulation，脉冲宽度调制)模态，所述半桥模态包括半桥PFM模态以及半桥PWM模态，在所述LLC谐振电路处于稳态时，所述全桥PFM模态、所述全桥PWM模态、所述半桥PFM模态以及所述半桥PWM模态对应的所述输入电压依次增大。
可选地，所述LLC谐振电路还包括LLC谐振腔，所述LLC谐振腔包括三端子的第五开关管以及三端子的第六开关管，所述第五开关管的第一端与所述第一开关管的第二端连接，所述第五开关管的第二端与所述第六开关管的第一端连接，所述第六开关管的第二端与所述第四开关管的第一端连接，在所述LLC谐振电路的输入电压大于设定阈值，且所述LLC谐振电路处于全桥模态的情况下，所述方法还包括：在所述LLC谐振电路处于所述全桥PWM模态的情况下，获取所述第一开关管或者所述第二开关管的实时频率以及实时占空比；在所述实时频率等于谐振频率，且所述实时占空比等于第三预定值时，控制所述第五开关管以及所述第六开关管的占空比变为第四预定值，使得所述LLC谐振电路由所述全桥PWM模态转换至半桥PFM模态。
可选地，在所述输入电压小于或者等于所述设定阈值，且所述LLC谐振电路处于所述半桥模态的情况下，所述方法还包括：在所述LLC谐振电路处于所述半桥PWM模态的情况下，获取所述实时频率以及所述实时占空比；在所述实时频率等于谐振频率，且所述实时占空比等于所述第三预定值时，控制所述第五开关管以及所述第六开关管的占空比变为所述第三预定值，使得所述LLC谐振电路由所述半桥PWM模态转换至所述全桥PFM模态。
根据本发明实施例的另一方面，还提供了一种LLC谐振电路的控制装置， 所述LLC谐振电路包括全桥逆变电路，所述全桥逆变电路包括四个三端子的开关管，所述装置包括第一控制单元以及第二控制单元，其中，所述第一控制单元用于在所述LLC谐振电路的输入电压大于设定阈值，且所述LLC谐振电路处于全桥模态的情况下，控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第一规则变化，以使得所述LLC谐振电路由所述全桥模态转换至半桥模态；所述第二控制单元用于在所述输入电压小于或者等于所述设定阈值，且所述LLC谐振电路处于所述半桥模态的情况下，控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第二规则变化，以使得所述LLC谐振电路由所述半桥模态转换至所述全桥模态。
根据本发明实施例的又一方面，还提供了一种处理器，所述处理器用于运行程序，其中，所述程序运行时执行任意一种所述的方法。
根据本发明实施例的另一方面，还提供了一种LLC谐振系统，包括LLC谐振电路以及所述LLC谐振电路的控制装置，其中，所述LLC谐振电路包括全桥逆变电路，所述全桥逆变电路包括四个三端子的开关管；所述控制装置用于执行任一种所述的方法。
应用本发明所述的实施例，所述的LLC谐振电路的控制方法中，在LLC谐振电路的输入电压大于设定阈值且在全桥模态的情况下，控制谐振电路中全桥逆变电路的部分开关管的驱动电压的上升沿对应的时刻不变，且控制其下降沿对应的时刻按照第一规则变化，来使得LLC谐振电路由全桥模态切换至半桥模态；在LLC谐振电路的输入电压小于或者等于设定阈值且在半桥模态的情况下，控制全桥逆变电路的部分开关管的驱动电压的上升沿对应的时刻不变，且控制其下降沿对应的时刻按照第二规则变化，来使得LLC谐振电路由半桥模态切换至全桥模态。所述方法通过输入电压作为全桥模态和半桥模态之间切换的判断条件，且在需要进行模态切换时，通过控制部分所述开关管的驱动电压上升沿对应时刻不变，仅改变下降沿对应时刻，来实现占空比的渐变，这样在模态切换过程中，随着所述开关管的占空比渐变，其上升沿对应时刻不变，即上升沿位置不发生改变，这样保证了半桥模态以及全桥模态切换过程中谐振电流始终满足ZVS条件，保证了开关管应力较低，从而保证了电路的可靠性较好，同时保证了电路的抗电磁干扰能力较强。
附图说明
构成本申请的一部分的说明书附图用来提供对本申请的进一步理解，本申请的示意性实施例及其说明用于解释本申请，并不构成对本申请的不当限定。在附图中：
图1示出了根据本申请的实施例的LLC谐振电路的控制方法的流程示意图；
图2示出了根据本申请的实施例的LLC谐振电路原理图；
图3和图4示出了根据本申请的实施例的全桥模态切换为半桥模态的波形时序图；
图5和图6示出了根据本申请的实施例的半桥模态切换为全桥模态的波形时序图；
图7示出了现有方案的稳态控制方案示意图；
图8示出了根据本申请的实施例的稳态控制方案示意图；
图9示出了图7以及图8的方案对应的效率对比图；
图10示出了占空比为0.5时LLC谐振电路在不同频率下的增益曲线图；
图11示出了根据本申请的实施例的LLC谐振电路的控制装置的示意图；
其中，上述附图包括以下附图标记：
101、全桥逆变电路；102、LLC谐振腔；103、变压器；104、副边整流网络。
具体实施方式
需要说明的是，在不冲突的情况下，本申请中的实施例及实施例中的特征可以相互组合。下面将参考附图并结合实施例来详细说明本申请。
为了使本技术领域的人员更好地理解本申请方案，下面将结合本申请实施例中的附图，对本申请实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例仅仅是本申请一部分的实施例，而不是全部的实施例。基于本申请中的实施例，本领域普通技术人员在没有做出创造性劳动前提下所获得的所有其他实施例，都应当属于本申请保护的范围。
需要说明的是，本申请的说明书和权利要求书及上述附图中的术语“第一”、“第二”等是用于区别类似的对象，而不必用于描述特定的顺序或先后次序。应该理解这样使用的数据在适当情况下可以互换，以便这里描述的本申请的实施例。此外，术语“包括”和“具有”以及他们的任何变形，意图在于覆盖不 排他的包含，例如，包含了一系列步骤或单元的过程、方法、系统、产品或设备不必限于清楚地列出的那些步骤或单元，而是可包括没有清楚地列出的或对于这些过程、方法、产品或设备固有的其它步骤或单元。
应该理解的是，当元件(诸如层、膜、区域、或衬底)描述为在另一元件“上”时，该元件可直接在该另一元件上，或者也可存在中间元件。而且，在说明书以及权利要求书中，当描述有元件“连接”至另一元件时，该元件可“直接连接”至该另一元件，或者通过第三元件“连接”至该另一元件。
正如背景技术中所说的，现有技术中的变换器在全桥模式和半桥模式之间切换的过程中无法实现ZVS，可靠性较差，为了解决上述问题，本申请的一种典型的实施方式中，提供了一种LLC谐振电路的控制方法、控制装置、计算机可读存储介质、处理器以及LLC谐振系统。
根据本申请的实施例，提供了一种LLC谐振电路的控制方法。
图1是根据本申请实施例的LLC谐振电路的控制方法的流程图。如图2所示，上述LLC谐振电路包括全桥逆变电路101，上述全桥逆变电路101包括四个三端子的开关管，如图1所示，该方法包括以下步骤：
步骤S101，在上述LLC谐振电路的输入电压大于设定阈值，且上述LLC谐振电路处于全桥模态的情况下，控制部分的上述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第一规则变化，以使得上述LLC谐振电路由上述全桥模态转换至半桥模态；
步骤S102，在上述输入电压小于或者等于上述设定阈值，且上述LLC谐振电路处于上述半桥模态的情况下，控制部分的上述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第二规则变化，以使得上述LLC谐振电路由上述半桥模态转换至上述全桥模态。
上述的LLC谐振电路的控制方法中，在LLC谐振电路的输入电压大于设定阈值且在全桥模态的情况下，控制谐振电路中全桥逆变电路的部分开关管的驱动电压的上升沿对应的时刻不变，且控制其下降沿对应的时刻按照第一规则变化，来使得LLC谐振电路由全桥模态切换至半桥模态；在LLC谐振电路的输入电压小于或者等于设定阈值且在半桥模态的情况下，控制全桥逆变电路的部分开关管的驱动电压的上升沿对应的时刻不变，且控制其下降沿对应的时刻按照第二规则变化，来使得LLC谐振电路由半桥模态切换至全桥模态。上述方法通 过输入电压作为全桥模态和半桥模态之间切换的判断条件，且在需要进行模态切换时，通过控制部分上述开关管的驱动电压上升沿对应时刻不变，仅改变下降沿对应时刻，来实现占空比的渐变，这样在模态切换过程中，随着上述开关管的占空比渐变，其上升沿对应时刻不变，即上升沿位置不发生改变，这样保证了半桥模态以及全桥模态切换过程中谐振电流始终满足ZVS条件，保证了开关管应力较低，从而保证了电路的可靠性较好，同时保证了电路的抗电磁干扰能力较强。
需要说明的是，上述控制方法中，由上述半桥模态转换至上述全桥模态的过程中，以及由上述全桥模态转换至半桥模态的过程中，对应调整的开关管相同。
在实际的应用过程中，上述LLC谐振电路可以为现有技术中任意合适的LLC电路。一种具体的实施例中，上述LLC谐振电路为钳位LLC电路。采用本申请的上述方法控制上述钳位LLC电路，与现有技术相比，在相同输入电压范围内变频范围较窄，方便设计，保证了钳位LLC电路的实用性较强。
具体地，上述设定阈值即为发生模态切换的输入电压点，实际应用中为了避免上述LLC谐振电路在两个模态之间来回切换，该设定阈值通常带有滞回性质。在上述输入电压大于上述阈值，且上述LLC谐振电路工作在半桥状态时不进行模态切换，上述输入电压小于或者等于上述阈值，且上述LLC谐振电路工作在全桥状态时不进行模态切换。
为了进一步地避免LLC谐振电路的全桥逆变电路中开关管无法实现ZVS，进一步地避免LLC谐振电路的开关管硬开通导致其电压应力加大以及EMI性能变差，根据本申请的一种具体的实施例，如图2所示，四个上述开关管分别为第一开关管S1、第二开关管S2、第三开关管S3以及第四开关管S4，其中，第一开关管S1的第一端和第三开关管S3的第一端连接，上述第一开关管S1的第二端和第二开关管S2的第一端连接，上述第三开关管S3的第二端与第四开关管S4的第一端连接，上述第二开关管S2的第二端与上述第四开关管S4的第二端连接，上述第一开关管S1、上述第二开关管S2、上述第三开关管S3以及上述第四开关管S4的第三端分别用于连接驱动电路，控制部分的上述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第一规则变化，包括：控制第三上升沿对应的时间点以及第四上升沿对应的时间点不变，第三 下降沿对应的时间点以及第四下降沿对应的时间点按照上述第一规则变化，其中，上述第三上升沿为上述第三开关管S3的驱动电压的上升沿，上述第四上升沿为上述第四开关管S4的驱动电压的上升沿，上述第三下降沿为与上述第三上升沿对应的下降沿，上述第四下降沿为与上述第四上升沿对应的下降沿，控制部分的上述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第二规则变化，包括：控制上述第三上升沿对应的时间点以及上述第四上升沿对应的时间点不变，上述第三下降沿对应的时间点以及上述第四下降沿对应的时间点按照上述第二规则变化。上述方法通过控制上述第三开关管以及上述第四开关管的驱动电压的上升沿对应时刻不变，且控制下降沿变化，进一步地保证了电路模态切换过程中谐振电流始终满足ZVS条件，从而进一步地保证了电路的可靠性能较好。
根据本申请的另一种具体的实施例，如图2所示的上述LLC谐振电路处于全桥模态时，上述第三开关管S3的驱动与上述第二开关管S2相同；上述第四开关管S4的驱动与上述第一开关管S1相同。如图3和图4所示，其中，横坐标为时间，纵坐标为电压，Vgs1为第一开关管S1的驱动波形，Vgs2为第二开关管S2的驱动波形，Vgs3为第三开关管S3的驱动波形，Vgs4为第四开关管S4的驱动波形，控制第三上升沿对应的时间点以及第四上升沿对应的时间点不变，第三下降沿对应的时间点以及第四下降沿对应的时间点按照上述第一规则变化，包括：控制上述第三上升沿对应的时间点不变，且控制上述第三下降沿对应的时间点移动，使得上述第三开关管的占空比逐渐减小至0；在上述第三开关管的占空比为0的情况下，控制上述第四上升沿对应的时间点不变，且控制上述第四下降沿对应的时间点移动，使得上述第四开关管S4的占空比逐渐增大至1。在上述LLC谐振电路的输入电压大于设定阈值，且上述LLC谐振电路处于全桥模态的情况下，通过控制上述第三开关管的驱动电压的上升沿位置不变且下降沿位置逐渐左移，使得上述第三开关管的占空比逐渐减至0，即使得上述第三开关管恒关断，然后再控制上述第四开关管的驱动电压的上升沿位置不变且下降沿位置逐渐右移，使得上述第四开关管的占空比逐渐增至1，即使得上述第四开关管恒导通，这样进一步地保证电路由全桥模态切换为半桥模态的过程中，上述第三开关管以及上述第四开关管始终为ZVS。
如图3和图4所示，在上述第三开关管以及上述第四开关管按照上述第一 规则变化的过程中，上述第一开关管S1驱动保持闭环控制；上述第二开关管S2驱动占空比跟S1相同，相位差180°。并且，对上述第三开关管以及上述第四开关管的控制为开环控制。
本申请的再一种具体的实施例中，如图2所示的上述LLC谐振电路处于半桥模态时，上述第三开关管S3恒关断；上述第四开关管S4恒导通。如图5和图6所示，其中，横坐标为时间，纵坐标为电压，控制上述第三上升沿对应的时间点以及上述第四上升沿对应的时间点不变，上述第三下降沿对应的时间点以及上述第四下降沿对应的时间点按照上述第二规则变化，包括：控制上述第四上升沿对应的时间点不变，且控制上述第四下降沿对应的时间点移动，使得上述第四开关管S4的占空比逐渐减小至第一预定值，上述第一预定值为上述第一开关管的占空比；在上述第四开关管S4的占空比小于0.5的情况下，控制上述第三上升沿对应的时间点不变，且控制上述第三下降沿对应的时间点移动，使得上述第三开关管的占空比逐渐增大至第二预定值，上述第二预定值为上述第二开关管的占空比。在上述LLC谐振电路的输入电压小于或者等于设定阈值，且上述LLC谐振电路处于半桥模态的情况下，通过控制上述第四开关管的驱动电压的上升沿位置不变且下降沿位置逐渐左移，使得上述第四开关管的占空比逐渐减至上述第一开关管的占空比，即使得上述第四开关管的驱动与上述第一开关管相同，然后再控制上述第三开关管的驱动电压的上升沿位置不变且下降沿位置逐渐右移，使得上述第三开关管的占空比逐渐增至上述第二开关管的占空比，即使得上述第三开关管的驱动与上述第二开关管相同，这样进一步地保证电路由半桥模态切换为全桥模态的过程中，上述第三开关管的占空比渐变，其上升沿位置均不发生改变，即该开关管开通前写真电流始终满足ZVS条件，从而进一步地保证了模态切换过程中该开关管的应力较小，进一步地避免了其硬导通的问题。
如图5和图6所示，在上述第三开关管以及上述第四开关管按照上述第二规则变化的过程中上述第一开关管S1驱动保持闭环控制；上述第二开关管S2驱动占空比跟S1相同，相位差180°。
根据本申请的再一种具体的实施例，通过调整上述第三开关管以及第四开关管的占空比渐变的速度，即可调整模态切换时间。这样保证了时间长度可控，从而进一步地保证了控制策略简单。
现有技术中，对LLC谐振电路的稳态控制方案如图7所示，一共有三个工作区域，采用的控制方式分别为：低压输入时工作于全桥PFM模态、中压输入时为全桥PWM模态、高压输入时为半桥PWM模态。由于在输入电压较高时采用全桥PWM控制，各开关管的关断电流较大，导致效率较低。此种情况下，为了保证控制效率较高，根据本申请的又一种具体的实施例，如图8所示，上述全桥模态包括全桥PFM模态以及全桥PWM模态，上述半桥模态包括半桥PFM模态以及半桥PWM模态，在上述LLC谐振电路处于稳态时，上述全桥PFM模态、上述全桥PWM模态、上述半桥PFM模态以及上述半桥PWM模态对应的上述输入电压依次增大，即本申请采用低压全桥PFM、中低压全桥PWM、中高压半桥PFM以及高压PWM的稳态控制方案。本申请在电压较高时采用半桥PFM模态控制，这样关断电流比采用全桥PWM控制时的关断电流低，从而提升了效率。现有技术中的稳态控制方案与本申请的稳态控制方案的效率曲线图如图9所示，其中，横坐标为电压值，纵坐标为效率，虚线为现有技术中的稳态控制方案对应的效率曲线，实现为本申请的稳态控制方案对应的效率曲线，可以看出本申请的稳态控制方法的效率明显高于现有技术中的稳态控制方案。并且，采用本申请的上述控制方案，保证了在半桥模态和全桥模态切换过程中输出电压的过欠冲较小。
需要说明的是，上述全桥PFM模态与上述全桥PWM模态之间的切换不存在模态跳变，采用常规控制即可，上述半桥PFM模态以及上述半桥PWM模态之间的切换也不存在模态跳变，也采用常规控制即可。在由上述全桥PFM模态或者上述全桥PWM模态切换至上述半桥PFM模态或者上述半桥PWM模态，或者由上述半桥PFM模态或者上述半桥PWM模态切换至上述全桥PFM模态或者上述全桥PWM模态时，存在模态跳变，采用本申请的上述方法进行控制切换。
具体地，由上述全桥PFM模态或者上述全桥PWM模态切换至上述半桥PFM模态或者上述半桥PWM模态，包括四种情况，第一种，由上述全桥PFM模态切换至上述半桥PFM模态；第二种，由上述全桥PFM模态切换至上述半桥PWM模态；第三种，由上述全桥PWM模态切换至上述半桥PFM模态；第四种，由上述全桥PWM模态切换至上述半桥PWM模态。由上述半桥PFM模态或者上述半桥PWM模态切换至上述全桥PFM模态或者上述全桥PWM模态，也包括四种情况，第一种，由上述半桥PFM模态切换至由上述全桥PFM模态；第二种，由上述半桥PFM 模态切换至上述全桥PWM模态；第三种，由上述半桥PWM模态切换至上述全桥PFM模态；第四种，由上述半桥PWM模态切换至上述全桥PWM模态。上述八种情况均存在模态跳变，均采用本申请的上述方法进行控制切换。
在实际的应用过程中，如图2所示，上述LLC谐振电路还包括LLC谐振腔102，上述LLC谐振腔102包括三端子的第五开关管S5以及三端子的第六开关管S6，上述第五开关管S5的第一端与上述第一开关管的第二端连接，上述第五开关管S5的第二端与上述第六开关管S6的第一端连接，上述第六开关管S6的第二端与上述第四开关管S4的第一端连接，如图3和图4所示，Vgs5为第五开关管S5的驱动波形，Vgs6为第六开关管S6的驱动波形，在上述LLC谐振电路的输入电压大于设定阈值，且上述LLC谐振电路处于全桥模态的情况下，上述方法还包括：在上述LLC谐振电路处于上述全桥PWM模态的情况下，获取上述第一开关管S1或者上述第二开关管S2的实时频率以及实时占空比；在上述实时频率等于谐振频率，且上述实时占空比等于第三预定值时，控制上述第五开关管S5以及上述第六开关管S6的占空比变为第四预定值，使得上述LLC谐振电路由上述全桥PWM模态转换至半桥PFM模态。这样避免了上述第五开光管和上述第六开关管的工作状态对上述LLC谐振电路的增益的影响，避免了输出电压跳变。
需要说明的是，上述LLC谐振电路由上述全桥PWM模态转换至半桥PWM模态的过程中，上述第五开关管以及上述第六开关管的动作与上述过程类似，此处不再一一赘述。
本申请的另一种具体的实施例中，如图5和图6所示，在上述输入电压小于或者等于上述设定阈值，且上述LLC谐振电路处于上述半桥模态的情况下，上述方法还包括：在上述LLC谐振电路处于上述半桥PWM模态的情况下，获取上述实时频率以及上述实时占空比；在上述实时频率等于谐振频率，且上述实时占空比等于上述第三预定值时，控制上述第五开关管以及上述第六开关管的占空比变为上述第三预定值，使得上述LLC谐振电路由上述半桥PWM模态转换至上述全桥PFM模态。这样避免了上述第五开光管和上述第六开关管的工作状态对上述LLC谐振电路的增益的影响，避免了输出电压跳变。
需要说明的是，上述LLC谐振电路由上述半桥PWM模态转换至全桥PWM模态的过程中，上述第五开关管以及上述第六开关管的动作与上述过程类似，此 处不再一一赘述。
在实际的应用过程中，上述第三预定值为0.5，上述第四预定值为0。在上述占空比为0.5时，LLC电路在不同频率下的增益曲线图如图10所示。图中实线代表由第五开关管和第六开关管构成的钳位支路不工作，虚线代表钳位支路工作，从图10中可以看到，当频率为电路的谐振频率fr时，钳位支路工作与否对电路增益不影响。因此，在这种模态下，直接关断第五开关管和第六开关管对电路的输出没有影响，不会引起输出电压跳变。
上述模态切换过程所有情况下，上述第一开关管、上述第二开关管、上述第三开关管、上述第四开关管、上述第五开关管以及上述第六开关管的频率均由闭环控制，受闭环控制影响，上述第三开关管以及上述第四开关管的占空比受开关控制，切换过程中上述第五开关管和上述第六开关管驱动占空比在0和0.5之间突变可能发生在其中任一阶段，也有可能在上述切换过程中上述第五开关管和上述第六开关管驱动占空在0和0.5之间来回震荡，实际应用中上述第五开关管和上述第六开关管的突变控制往往会加入一些回差。
另外，除了上述第五开关管和上述第六开关管，其他开关管在切换过程中都不存在占空比跳变，因此，整个切换过程是平滑的，不会引起跳变。由于整个模态切换过程中开关管上升沿位置均未发生变化，谐振电流相位及大小仍然满足实现ZVS的条件，故该模态切换不会带来半桥模态下常关断的逆变电路开关管在切换过程中无法实现ZVS的问题，进一步地保证了开关管的应力较小，进一步地保证了电路的抗电磁干扰性能较高。
根据本申请的又一种具体的实施例，如图2所示，上述LLC谐振电路还包括变压器103和副边整流网络104，上述LLC谐振腔102还包括谐振电感Lr、励磁电感Lm和谐振电容Cr，变压器103包括由原边绕组P1、副边绕组Q1和副边绕组Q2组成的变压器103；副边整流网络104包括同步整流管SR1和SR2构成的全波整流电路以及输出滤波电容Cout。
在实际的应用过程中，上述开关管可以为现有技术中任意可行的三端子开关管，如三极管或者MOS管等。更为具体的一种实施例中，如图2所示，各上述开关管分别为MOS管，第一开关管S1的漏极和第三开关管S3的漏极连接在一起，作为LLC谐振电路的输入正端，用于连接输入电源Vin的正端，第一开关管S1的源极连于第二开关管S2的漏极和谐振电容Cr的一端，谐振电容Cr 的另一端连于谐振电感Lr的一端和第五开关管S5的漏极，谐振电感Lr的另一端连于励磁电感Lm的一端和变压器103原边绕组P1的一端，变压器103原边绕组P1的另一端连于励磁电感Lm的另一端、第三开关管S3的源极、第四开关管S4的漏极以及第六开关管S6的漏极，第五开关管S5的源极和第六开关管S6的源极相连，第四开关管S4的源极和第二开关管S2的源极连接在一起，作为LLC谐振电路的输入负端，用于连接输入电源Vin的负极；变压器103的副边绕组Q1的一端连于副边同步整流管SR2的漏极，同步整流管SR2的源极、同步整流管SR1的源极和输出滤波电容Cout的一端连接在一起，作为LLC谐振电路的输出负端，用于连接输出负载Ro的负极，变压器103的副边绕组Q1的另一端、变压器103的副边绕组Q2的一端和输出滤波电容Co的另一端连接在一起，作为LLC谐振电路的输出正端，用于连接输出负载Ro的正极，变压器103的副边绕组Q2的另一端连于副边同步整流管SR1的漏极。变压器103原边绕组P1与副边绕组Q1和副边绕组Q2的一端互为同名端，变压器原边绕组P1与副边绕组Q1和副边绕组Q2的另一端互为同名端。
当然，本申请上述的LLC谐振电路并不限于图2示出的电路结构，其可以为现有技术中任意合适的LLC谐振电路。如将副边整流网络104替换成四个开关管或二极管组成的桥式整流结构，或者交换谐振电容Cr和谐振电感Lr的位置，都可以得到不同结构的LLC谐振电路。
表1为图2对应的各开关管在各模态下的状态。如表1所示，全桥PFM模态下，第一开关管S1驱动占空比为50％，频率由闭环决定；第二开关管S2驱动频率与占空比跟第一开关管S1相同，相位差180°；第三开关管S3驱动与第二开关管S2相同；第四开关管S4驱动与第一开关管S1相同；第五开关管S5恒关断，第六开关管S5恒关断。全桥PWM模态下，第一开关管S1驱动工作频率为谐振频率，占空比由闭环决定；第二开关管S2驱动频率与占空比跟第一开关管S1相同，相位差180°；第三开关管S3驱动与第二开关S2相同；第四开关管S4驱动与第一开关管S1相同；第五开关管S5驱动与第一开关管S1互补；第六开关管S6驱动与第二开关S2互补。半桥PFM模态下，第一开关管S1驱动占空比为50％，频率由闭环决定；第二开关管S2驱动频率与占空比跟第一开关管S1相同，相位差180°；第三开关管S3恒关断；第四开关管S4恒导通；第五开关管S5恒关断，第六开关管S6恒关断。全桥PWM模态下，第一开关管 S1驱动工作频率为谐振频率，占空比由闭环决定；第二开关管S2驱动频率与占空比跟第一开关管S1相同，相位差180°；第三开关管S3恒关断；第四开关管S4恒导通；第五开关管S5驱动与第一开关管S1互补；第六开关管S6驱动与第二开关管S2互补。
表1
Figure PCTCN2022119253-appb-000001
本申请的上述控制方案，可以通过模拟控制实现，也可以通过数字控制实现。对输出24V/600W的钳位LLC谐振电路，根据本申请的上述控制方案，搭建了实验样机进行模态切换验证，模态切换时间设置为20mS，切换过程引起的输出电压过欠冲在3％以内，且切换过程均可实现逆变电路开关管的ZVS。
本申请实施例还提供了一种LLC谐振电路的控制装置，需要说明的是，本申请实施例的LLC谐振电路的控制装置可以用于执行本申请实施例所提供的用于LLC谐振电路的控制方法。以下对本申请实施例提供的LLC谐振电路的控制装置进行介绍。
图11是根据本申请实施例的LLC谐振电路的控制装置的示意图，如图2所示，上述LLC谐振电路包括全桥逆变电路101，上述全桥逆变电路101包括四个三端子的开关管，如图11所示，该装置包括第一控制单元10以及第二控制单元20，其中，上述第一控制单元10用于在上述LLC谐振电路的输入电压大于设定阈值，且上述LLC谐振电路处于全桥模态的情况下，控制部分的上述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第一规则变化，以使得上述LLC谐振电路由上述全桥模态转换至半桥模态；上述第二控制单元20用于在上述输入电压小于或者等于上述设定阈值，且上述LLC谐振 电路处于上述半桥模态的情况下，控制部分的上述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第二规则变化，以使得上述LLC谐振电路由上述半桥模态转换至上述全桥模态。
上述的LLC谐振电路的控制装置中，在LLC谐振电路的输入电压大于设定阈值且在全桥模态的情况下，上述第一控制单元控制谐振电路中全桥逆变电路的部分开关管的驱动电压的上升沿对应的时刻不变，且控制其下降沿对应的时刻按照第一规则变化，来使得LLC谐振电路由全桥模态切换至半桥模态；在LLC谐振电路的输入电压小于或者等于设定阈值且在半桥模态的情况下，上述第二控制单元控制全桥逆变电路的部分开关管的驱动电压的上升沿对应的时刻不变，且控制其下降沿对应的时刻按照第二规则变化，来使得LLC谐振电路由半桥模态切换至全桥模态。上述装置通过输出电压作为全桥模态和半桥模态之间切换的判断条件，且在需要进行模态切换时，通过控制部分上述开关管的驱动电压上升沿对应时刻不变，仅改变下降沿对应时刻，来实现占空比的渐变，这样在模态切换过程中，随着上述开关管的占空比渐变，其上升沿对应时刻不变，即上升沿位置不发生改变，这样保证了半桥模态以及全桥模态切换过程中谐振电流始终满足ZVS条件，保证了开关管应力较低，从而保证了电路的可靠性较好，同时保证了电路的抗电磁干扰能力较强。
需要说明的是，上述控制装置中，由上述半桥模态转换至上述全桥模态的过程中，以及由上述全桥模态转换至半桥模态的过程中，对应调整的开关管相同。
在实际的应用过程中，上述LLC谐振电路可以为现有技术中任意合适的LLC电路。一种具体的实施例中，上述LLC谐振电路为钳位LLC电路。采用本申请的上述装置控制上述钳位LLC电路，与现有技术相比，在相同输入电压范围内变频范围较窄，方便设计，保证了钳位LLC电路的实用性较强。
具体地，上述设定阈值即为发生模态切换的输入电压点，实际应用中为了避免上述LLC谐振电路在两个模态之间来回切换，该设定阈值通常带有滞回性质。在上述输入电压大于上述阈值，且上述LLC谐振电路工作在半桥状态时不进行模态切换，上述输入电压小于或者等于上述阈值，且上述LLC谐振电路工作在全桥状态时不进行模态切换。
为了进一步地避免LLC谐振电路的全桥逆变电路中开关管无法实现ZVS， 进一步地避免LLC谐振电路的开关管硬开通导致其电压应力加大以及EMI性能变差，根据本申请的一种具体的实施例，如图2所示，四个上述开关管分别为第一开关管S1、第二开关管S2、第三开关管S3以及第四开关管S4，其中，第一开关管S1的第一端和第三开关管S3的第一端连接，上述第一开关管S1的第二端和第二开关管S2的第一端连接，上述第三开关管S3的第二端与第四开关管S4的第一端连接，上述第二开关管S2的第二端与上述第四开关管S4的第二端连接，上述第一开关管S1、上述第二开关管S2、上述第三开关管S3以及上述第四开关管S4的第三端分别用于连接驱动电路，上述第一控制单元包括第一控制模块，上述第一控制模块用于控制第三上升沿对应的时间点以及第四上升沿对应的时间点不变，第三下降沿对应的时间点以及第四下降沿对应的时间点按照上述第一规则变化，其中，上述第三上升沿为上述第三开关管S3的驱动电压的上升沿，上述第四上升沿为上述第四开关管S4的驱动电压的上升沿，上述第三下降沿为与上述第三上升沿对应的下降沿，上述第四下降沿为与上述第四上升沿对应的下降沿，上述第二控制单元包括第二控制模块，上述第二控制模块用于控制上述第三上升沿对应的时间点以及上述第四上升沿对应的时间点不变，上述第三下降沿对应的时间点以及上述第四下降沿对应的时间点按照上述第二规则变化。上述装置通过控制上述第三开关管以及上述第四开关管的驱动电压的上升沿对应时刻不变，且控制下降沿变化，进一步地保证了电路模态切换过程中谐振电流始终满足ZVS条件，从而进一步地保证了电路的可靠性能较好。
根据本申请的另一种具体的实施例，如图2所示的上述LLC谐振电路处于全桥模态时，上述第三开关管S3的驱动与上述第二开关管S2相同；上述第四开关管S4的驱动与上述第一开关管S1相同。如图3和图4所示，Vgs1为第一开关管S1的驱动波形，Vgs2为第二开关管S2的驱动波形，Vgs3为第三开关管S3的驱动波形，Vgs4为第四开关管S4的驱动波形，上述第一控制模块包括第一控制子模块以及第二控制子模块，其中，上述第一控制子模块用于控制上述第三上升沿对应的时间点不变，且控制上述第三下降沿对应的时间点移动，使得上述第三开关管的占空比逐渐减小至0；上述第二控制子模块用于在上述第三开关管的占空比为0的情况下，控制上述第四上升沿对应的时间点不变，且控制上述第四下降沿对应的时间点移动，使得上述第四开关管S4的占空比逐渐 增大至1。在上述LLC谐振电路的输入电压大于设定阈值，且上述LLC谐振电路处于全桥模态的情况下，通过控制上述第三开关管的驱动电压的上升沿位置不变且下降沿位置逐渐左移，使得上述第三开关管的占空比逐渐减至0，即使得上述第三开关管恒关断，然后再控制上述第四开关管的驱动电压的上升沿位置不变且下降沿位置逐渐右移，使得上述第四开关管的占空比逐渐增至1，即使得上述第四开关管恒导通，这样进一步地保证电路由全桥模态切换为半桥模态的过程中，上述第三开关管以及上述第四开关管始终为ZVS。
如图3和图4所示，在上述第三开关管以及上述第四开关管按照上述第一规则变化的过程中，上述第一开关管S1驱动保持闭环控制；上述第二开关管S2驱动占空比跟S1相同，相位差180°。并且，对上述第三开关管以及上述第四开关管的控制为开环控制。
本申请的再一种具体的实施例中，如图2所示的上述LLC谐振电路处于半桥模态时，上述第三开关管S3恒关断；上述第四开关管S4恒导通。如图5和图6所示，上述第二控制模块包括第三控制子模块以及第四控制子模块，其中，上述第三控制子模块用于控制上述第四上升沿对应的时间点不变，且控制上述第四下降沿对应的时间点移动，使得上述第四开关管S4的占空比逐渐减小至第一预定值，上述第一预定值为上述第一开关管的占空比；上述第四控制子模块用于在上述第四开关管S4的占空比小于0.5的情况下，控制上述第三上升沿对应的时间点不变，且控制上述第三下降沿对应的时间点移动，使得上述第三开关管的占空比逐渐增大至第二预定值，上述第二预定值为上述第二开关管的占空比。在上述LLC谐振电路的输入电压小于或者等于设定阈值，且上述LLC谐振电路处于半桥模态的情况下，通过控制上述第四开关管的驱动电压的上升沿位置不变且下降沿位置逐渐左移，使得上述第四开关管的占空比逐渐减至上述第一开关管的占空比，即使得上述第四开关管的驱动与上述第一开关管相同，然后再控制上述第三开关管的驱动电压的上升沿位置不变且下降沿位置逐渐右移，使得上述第三开关管的占空比逐渐增至上述第二开关管的占空比，即使得上述第三开关管的驱动与上述第二开关管相同，这样进一步地保证电路由半桥模态切换为全桥模态的过程中，上述第三开关管的占空比渐变，其上升沿位置均不发生改变，即该开关管开通前写真电流始终满足ZVS条件，从而进一步地保证了模态切换过程中该开关管的应力较小，进一步地避免了其硬导通的问题。
如图5和图6所示，在上述第三开关管以及上述第四开关管按照上述第二规则变化的过程中上述第一开关管S1驱动保持闭环控制；上述第二开关管S2驱动占空比跟S1相同，相位差180°。
根据本申请的再一种具体的实施例，通过调整上述第三开关管以及第四开关管的占空比渐变的速度，即可调整模态切换时间。这样保证了时间长度可控，从而进一步地保证了控制策略简单。
现有技术中，对LLC谐振电路的稳态控制方案如图7所示，一共有三个工作区域，采用的控制方式分别为：低压输入时工作于全桥PFM模态、中压输入时为全桥PWM模态、高压输入时为半桥PWM模态。由于在输入电压较高时采用全桥PWM控制，各开关管的关断电流较大，导致效率较低。此种情况下，为了保证控制效率较高，根据本申请的又一种具体的实施例，如图8所示，上述全桥模态包括全桥PFM模态以及全桥PWM模态，上述半桥模态包括半桥PFM模态以及半桥PWM模态，在上述LLC谐振电路处于稳态时，上述全桥PFM模态、上述全桥PWM模态、上述半桥PFM模态以及上述半桥PWM模态对应的上述输入电压依次增大，即本申请采用低压全桥PFM、中低压全桥PWM、中高压半桥PFM以及高压PWM的稳态控制方案。本申请在电压较高时采用半桥PFM模态控制，这样关断电流比采用全桥PWM控制时的关断电流低，从而提升了效率。现有技术中的稳态控制方案与本申请的稳态控制方案的效率曲线图如图9所示，其中，虚线为现有技术中的稳态控制方案对应的效率曲线，实现为本申请的稳态控制方案对应的效率曲线，可以看出本申请的稳态控制装置的效率明显高于现有技术中的稳态控制方案。并且，采用本申请的上述控制方案，保证了在半桥模态和全桥模态切换过程中输出电压的过欠冲较小。
需要说明的是，上述全桥PFM模态与上述全桥PWM模态之间的切换不存在模态跳变，采用常规控制即可，上述半桥PFM模态以及上述半桥PWM模态之间的切换也不存在模态跳变，也采用常规控制即可。在由上述全桥PFM模态或者上述全桥PWM模态切换至上述半桥PFM模态或者上述半桥PWM模态，或者由上述半桥PFM模态或者上述半桥PWM模态切换至上述全桥PFM模态或者上述全桥PWM模态时，存在模态跳变，采用本申请的上述装置进行控制切换。
具体地，由上述全桥PFM模态或者上述全桥PWM模态切换至上述半桥PFM模态或者上述半桥PWM模态，包括四种情况，第一种，由上述全桥PFM模态切 换至上述半桥PFM模态；第二种，由上述全桥PFM模态切换至上述半桥PWM模态；第三种，由上述全桥PWM模态切换至上述半桥PFM模态；第四种，由上述全桥PWM模态切换至上述半桥PWM模态。由上述半桥PFM模态或者上述半桥PWM模态切换至上述全桥PFM模态或者上述全桥PWM模态，也包括四种情况，第一种，由上述半桥PFM模态切换至由上述全桥PFM模态；第二种，由上述半桥PFM模态切换至上述全桥PWM模态；第三种，由上述半桥PWM模态切换至上述全桥PFM模态；第四种，由上述半桥PWM模态切换至上述全桥PWM模态。上述八种情况均存在模态跳变，均采用本申请的上述方法进行控制切换。
在实际的应用过程中，如图2所示，上述LLC谐振电路还包括LLC谐振腔102，上述LLC谐振腔102包括三端子的第五开关管S5以及三端子的第六开关管S6，上述第五开关管S5的第一端与上述第一开关管的第二端连接，上述第五开关管S5的第二端与上述第六开关管S6的第一端连接，上述第六开关管S6的第二端与上述第四开关管S4的第一端连接，如图3和图4所示，Vgs5为第五开关管S5的驱动波形，Vgs6为第六开关管S6的驱动波形，在上述LLC谐振电路的输入电压大于设定阈值，且上述LLC谐振电路处于全桥模态的情况下，上述装置还包括第一获取单元以及第三控制单元，其中，上述第一获取单元用于在上述LLC谐振电路处于上述全桥PWM模态的情况下，获取上述第一开关管S1或者上述第二开关管S2的实时频率以及实时占空比；上述第三控制单元用于在上述实时频率等于谐振频率，且上述实时占空比等于第三预定值时，控制上述第五开关管S5以及上述第六开关管S6的占空比变为第四预定值，使得上述LLC谐振电路由上述全桥PWM模态转换至半桥PFM模态。这样避免了上述第五开光管和上述第六开关管的工作状态对上述LLC谐振电路的增益的影响，避免了输出电压跳变。
需要说明的是，上述LLC谐振电路由上述全桥PWM模态转换至半桥PWM模态的过程中，上述第五开关管以及上述第六开关管的动作与上述过程类似，此处不再一一赘述。
本申请的另一种具体的实施例中，如图5和图6所示，在上述输入电压小于或者等于上述设定阈值，且上述LLC谐振电路处于上述半桥模态的情况下，上述装置还包括第二获取单元以及第四控制单元，其中，上述第二获取单元用于在上述LLC谐振电路处于上述半桥PWM模态的情况下，获取上述实时频率以 及上述实时占空比；上述第四控制单元用于在上述实时频率等于谐振频率，且上述实时占空比等于上述第三预定值时，控制上述第五开关管以及上述第六开关管的占空比变为上述第三预定值，使得上述LLC谐振电路由上述半桥PWM模态转换至上述全桥PFM模态。这样避免了上述第五开光管和上述第六开关管的工作状态对上述LLC谐振电路的增益的影响，避免了输出电压跳变。
需要说明的是，上述LLC谐振电路由上述半桥PWM模态转换至全桥PWM模态的过程中，上述第五开关管以及上述第六开关管的动作与上述过程类似，此处不再一一赘述。
在实际的应用过程中，上述第三预定值为0.5，上述第四预定值为0。在上述占空比为0.5时，LLC电路在不同频率下的增益曲线图如图10所示。图中实线代表由第五开关管和第六开关管构成的钳位支路不工作，虚线代表钳位支路工作，从图10中可以看到，当频率为电路的谐振频率fr时，钳位支路工作与否对电路增益不影响。因此，在这种模态下，直接关断第五开关管和第六开关管对电路的输出没有影响，不会引起输出电压跳变。
上述模态切换过程所有情况下，上述第一开关管、上述第二开关管、上述第三开关管、上述第四开关管、上述第五开关管以及上述第六开关管的频率均由闭环控制，受闭环控制影响，上述第三开关管以及上述第四开关管的占空比受开关控制，切换过程中上述第五开关管和上述第六开关管驱动占空比在0和0.5之间突变可能发生在其中任一阶段，也有可能在上述切换过程中上述第五开关管和上述第六开关管驱动占空在0和0.5之间来回震荡，实际应用中上述第五开关管和上述第六开关管的突变控制往往会加入一些回差。
另外，除了上述第五开关管和上述第六开关管，其他开关管在切换过程中都不存在占空比跳变，因此，整个切换过程是平滑的，不会引起跳变。由于整个模态切换过程中开关管上升沿位置均未发生变化，谐振电流相位及大小仍然满足实现ZVS的条件，故该模态切换不会带来半桥模态下常关断的逆变电路开关管在切换过程中无法实现ZVS的问题，进一步地保证了开关管的应力较小，进一步地保证了电路的抗电磁干扰性能较高。
根据本申请的又一种具体的实施例，如图2所示，上述LLC谐振电路还包括变压器103和副边整流网络104，上述LLC谐振腔102还包括谐振电感Lr、励磁电感Lm和谐振电容Cr，变压器103包括由原边绕组P1、副边绕组Q1和副 边绕组Q2组成的变压器103；副边整流网络104包括同步整流管SR1和SR2构成的全波整流电路以及输出滤波电容Cout。
在实际的应用过程中，上述开关管可以为现有技术中任意可行的三端子开关管，如三极管或者MOS管等。更为具体的一种实施例中，如图2所示，各上述开关管分别为MOS管，第一开关管S1的漏极和第三开关管S3的漏极连接在一起，作为LLC谐振电路的输入正端，用于连接输入电源Vin的正端，第一开关管S1的源极连于第二开关管S2的漏极和谐振电容Cr的一端，谐振电容Cr的另一端连于谐振电感Lr的一端和第五开关管S5的漏极，谐振电感Lr的另一端连于励磁电感Lm的一端和变压器103原边绕组P1的一端，变压器103原边绕组P1的另一端连于励磁电感Lm的另一端、第三开关管S3的源极、第四开关管S4的漏极以及第六开关管S6的漏极，第五开关管S5的源极和第六开关管S6的源极相连，第四开关管S4的源极和第二开关管S2的源极连接在一起，作为LLC谐振电路的输入负端，用于连接输入电源Vin的负极；变压器103的副边绕组Q1的一端连于副边同步整流管SR2的漏极，同步整流管SR2的源极、同步整流管SR1的源极和输出滤波电容Cout的一端连接在一起，作为LLC谐振电路的输出负端，用于连接输出负载Ro的负极，变压器103的副边绕组Q1的另一端、变压器103的副边绕组Q2的一端和输出滤波电容Co的另一端连接在一起，作为LLC谐振电路的输出正端，用于连接输出负载Ro的正极，变压器103的副边绕组Q2的另一端连于副边同步整流管SR1的漏极。变压器103原边绕组P1与副边绕组Q1和副边绕组Q2的一端互为同名端，变压器原边绕组P1与副边绕组Q1和副边绕组Q2的另一端互为同名端。
当然，本申请上述的LLC谐振电路并不限于图2示出的电路结构，其可以为现有技术中任意合适的LLC谐振电路。如将副边整流网络104替换成四个开关管或二极管组成的桥式整流结构，或者交换谐振电容Cr和谐振电感Lr的位置，都可以得到不同结构的LLC谐振电路。
表1为图2对应的各开关管在各模态下的状态。如表1所示，全桥PFM模态下，第一开关管S1驱动占空比为50％，频率由闭环决定；第二开关管S2驱动频率与占空比跟第一开关管S1相同，相位差180°；第三开关管S3驱动与第二开关管S2相同；第四开关管S4驱动与第一开关管S1相同；第五开关管S5恒关断，第六开关管S5恒关断。全桥PWM模态下，第一开关管S1驱动工作 频率为谐振频率，占空比由闭环决定；第二开关管S2驱动频率与占空比跟第一开关管S1相同，相位差180°；第三开关管S3驱动与第二开关S2相同；第四开关管S4驱动与第一开关管S1相同；第五开关管S5驱动与第一开关管S1互补；第六开关管S6驱动与第二开关S2互补。半桥PFM模态下，第一开关管S1驱动占空比为50％，频率由闭环决定；第二开关管S2驱动频率与占空比跟第一开关管S1相同，相位差180°；第三开关管S3恒关断；第四开关管S4恒导通；第五开关管S5恒关断，第六开关管S6恒关断。全桥PWM模态下，第一开关管S1驱动工作频率为谐振频率，占空比由闭环决定；第二开关管S2驱动频率与占空比跟第一开关管S1相同，相位差180°；第三开关管S3恒关断；第四开关管S4恒导通；第五开关管S5驱动与第一开关管S1互补；第六开关管S6驱动与第二开关管S2互补。
本申请的上述控制方案，可以通过模拟控制实现，也可以通过数字控制实现。对输出24V/600W的钳位LLC谐振电路，根据本申请的上述控制方案，搭建了实验样机进行模态切换验证，模态切换时间设置为20mS，切换过程引起的输出电压过欠冲在3％以内，且切换过程均可实现逆变电路开关管的ZVS。
本发明实施例提供了一种处理器，上述处理器用于运行程序，其中，上述程序运行时执行上述LLC谐振电路的控制方法。
根据本申请的再一种典型的实施例，还提供了一种LLC谐振系统，包括LLC谐振电路以及上述LLC谐振电路的控制装置，其中，上述LLC谐振电路包括全桥逆变电路，上述全桥逆变电路包括四个三端子的开关管；上述控制装置用于执行任一种上述的方法。
上述的LLC谐振系统，包括LLC谐振电路以及上述LLC谐振电路的控制装置，该控制装置用于执行任一种上述的方法。上述方法通过输入电压作为全桥模态和半桥模态之间切换的判断条件，且在需要进行模态切换时，通过控制部分上述开关管的驱动电压上升沿对应时刻不变，仅改变下降沿对应时刻，来实现占空比的渐变，这样在模态切换过程中，随着上述开关管的占空比渐变，其上升沿对应时刻不变，即上升沿位置不发生改变，这样保证了半桥模态以及全桥模态切换过程中谐振电流始终满足ZVS条件，保证了开关管应力较低，从而保证了LLC谐振系统的可靠性较好，同时保证了LLC谐振系统的抗电磁干扰能力较强。
在本发明的上述实施例中，对各个实施例的描述都各有侧重，某个实施例中没有详述的部分，可以参见其他实施例的相关描述。
在本申请所提供的几个实施例中，应该理解到，所揭露的技术内容，可通过其它的方式实现。其中，以上所描述的装置实施例仅仅是示意性的，例如上述单元的划分，可以为一种逻辑功能划分，实际实现时可以有另外的划分方式，例如多个单元或组件可以结合或者可以集成到另一个系统，或一些特征可以忽略，或不执行。另一点，所显示或讨论的相互之间的耦合或直接耦合或通信连接可以是通过一些接口，单元或模块的间接耦合或通信连接，可以是电性或其它的形式。
上述作为分离部件说明的单元可以是或者也可以不是物理上分开的，作为单元显示的部件可以是或者也可以不是物理单元，即可以位于一个地方，或者也可以分布到多个单元上。可以根据实际的需要选择其中的部分或者全部单元来实现本实施例方案的目的。
另外，在本发明各个实施例中的各功能单元可以集成在一个处理单元中，也可以是各个单元单独物理存在，也可以两个或两个以上单元集成在一个单元中。上述集成的单元既可以采用硬件的形式实现，也可以采用软件功能单元的形式实现。
上述集成的单元如果以软件功能单元的形式实现并作为独立的产品销售或使用时，可以存储在一个计算机可读取存储介质中。基于这样的理解，本发明的技术方案本质上或者说对现有技术做出贡献的部分或者该技术方案的全部或部分可以以软件产品的形式体现出来，该计算机软件产品存储在一个存储介质中，包括若干指令用以使得一台计算机设备(可为个人计算机、服务器或者网络设备等)执行本发明各个实施例上述方法的全部或部分步骤。而前述的存储介质包括：U盘、只读存储器(ROM，Read-Only Memory)、随机存取存储器(RAM，Random Access Memory)、移动硬盘、磁碟或者光盘等各种可以存储程序代码的介质。
从以上的描述中，可以看出，本申请上述的实施例实现了如下技术效果：
1)、本申请上述的LLC谐振电路的控制方法中，在LLC谐振电路的输入电压大于设定阈值且在全桥模态的情况下，控制谐振电路中全桥逆变电路的部分开关管的驱动电压的上升沿对应的时刻不变，且控制其下降沿对应的时刻按照 第一规则变化，来使得LLC谐振电路由全桥模态切换至半桥模态；在LLC谐振电路的输入电压小于或者等于设定阈值且在半桥模态的情况下，控制全桥逆变电路的部分开关管的驱动电压的上升沿对应的时刻不变，且控制其下降沿对应的时刻按照第二规则变化，来使得LLC谐振电路由半桥模态切换至全桥模态。上述方法通过输入电压作为全桥模态和半桥模态之间切换的判断条件，且在需要进行模态切换时，通过控制部分上述开关管的驱动电压上升沿对应时刻不变，仅改变下降沿对应时刻，来实现占空比的渐变，这样在模态切换过程中，随着上述开关管的占空比渐变，其上升沿对应时刻不变，即上升沿位置不发生改变，这样保证了半桥模态以及全桥模态切换过程中谐振电流始终满足ZVS条件，保证了开关管应力较低，从而保证了电路的可靠性较好，同时保证了电路的抗电磁干扰能力较强。
2)、本申请上述的LLC谐振电路的控制装置中，在LLC谐振电路的输入电压大于设定阈值且在全桥模态的情况下，上述第一控制单元控制谐振电路中全桥逆变电路的部分开关管的驱动电压的上升沿对应的时刻不变，且控制其下降沿对应的时刻按照第一规则变化，来使得LLC谐振电路由全桥模态切换至半桥模态；在LLC谐振电路的输入电压小于或者等于设定阈值且在半桥模态的情况下，上述第二控制单元控制全桥逆变电路的部分开关管的驱动电压的上升沿对应的时刻不变，且控制其下降沿对应的时刻按照第二规则变化，来使得LLC谐振电路由半桥模态切换至全桥模态。上述装置通过输出电压作为全桥模态和半桥模态之间切换的判断条件，且在需要进行模态切换时，通过控制部分上述开关管的驱动电压上升沿对应时刻不变，仅改变下降沿对应时刻，来实现占空比的渐变，这样在模态切换过程中，随着上述开关管的占空比渐变，其上升沿对应时刻不变，即上升沿位置不发生改变，这样保证了半桥模态以及全桥模态切换过程中谐振电流始终满足ZVS条件，保证了开关管应力较低，从而保证了电路的可靠性较好，同时保证了电路的抗电磁干扰能力较强。
3)、本申请上述的LLC谐振系统，包括LLC谐振电路以及上述LLC谐振电路的控制装置，该控制装置用于执行任一种上述的方法。上述方法通过输入电压作为全桥模态和半桥模态之间切换的判断条件，且在需要进行模态切换时，通过控制部分上述开关管的驱动电压上升沿对应时刻不变，仅改变下降沿对应时刻，来实现占空比的渐变，这样在模态切换过程中，随着上述开关管的占空 比渐变，其上升沿对应时刻不变，即上升沿位置不发生改变，这样保证了半桥模态以及全桥模态切换过程中谐振电流始终满足ZVS条件，保证了开关管应力较低，从而保证了LLC谐振系统的可靠性较好，同时保证了LLC谐振系统的抗电磁干扰能力较强。
以上所述仅为本申请的优选实施例而已，并不用于限制本申请，对于本领域的技术人员来说，本申请可以有各种更改和变化。凡在本申请的精神和原则之内，所作的任何修改、等同替换、改进等，均应包含在本申请的保护范围之内。

Claims (10)

  1. 一种LLC谐振电路的控制方法，所述LLC谐振电路包括全桥逆变电路，所述全桥逆变电路包括四个三端子的开关管，其特征在于，所述方法包括：
    在所述LLC谐振电路的输入电压大于设定阈值，且所述LLC谐振电路处于全桥模态的情况下，控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第一规则变化，以使得所述LLC谐振电路由所述全桥模态转换至半桥模态；
    在所述输入电压小于或者等于所述设定阈值，且所述LLC谐振电路处于所述半桥模态的情况下，控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第二规则变化，以使得所述LLC谐振电路由所述半桥模态转换至所述全桥模态。
  2. 根据权利要求1所述的方法，其特征在于，四个所述开关管分别为第一开关管、第二开关管、第三开关管以及第四开关管，其中，第一开关管的第一端和第三开关管的第一端连接，所述第一开关管的第二端和第二开关管的第一端连接，所述第三开关管的第二端与第四开关管的第一端连接，所述第二开关管的第二端与所述第四开关管的第二端连接，所述第一开关管、所述第二开关管、所述第三开关管以及所述第四开关管的第三端分别用于连接驱动电路；
    控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第一规则变化，包括：
    控制第三上升沿对应的时间点以及第四上升沿对应的时间点不变，第三下降沿对应的时间点以及第四下降沿对应的时间点按照所述第一规则变化，其中，所述第三上升沿为所述第三开关管的驱动电压的上升沿，所述第四上升沿为所述第四开关管的驱动电压的上升沿，所述第三下降沿为与所述第三上升沿对应的下降沿，所述第四下降沿为与所述第四上升沿对应的下降沿，
    控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第二规则变化，包括：
    控制所述第三上升沿对应的时间点以及所述第四上升沿对应的时间点不变，所述第三下降沿对应的时间点以及所述第四下降沿对应的时间点按照所述第二规则变化。
  3. 根据权利要求2所述的方法，其特征在于，控制第三上升沿对应的时间点以及第四上升沿对应的时间点不变，第三下降沿对应的时间点以及第四下降沿对应的时间点按照所述第一规则变化，包括：
    控制所述第三上升沿对应的时间点不变，且控制所述第三下降沿对应的时间点移动，使得所述第三开关管的占空比逐渐减小至0；
    在所述第三开关管的占空比为0的情况下，控制所述第四上升沿对应的时间点不变，且控制所述第四下降沿对应的时间点移动，使得所述第四开关管的占空比逐渐增大至1。
  4. 根据权利要求2所述的方法，其特征在于，控制所述第三上升沿对应的时间点以及所述第四上升沿对应的时间点不变，所述第三下降沿对应的时间点以及所述第四下降沿对应的时间点按照所述第二规则变化，包括：
    控制所述第四上升沿对应的时间点不变，且控制所述第四下降沿对应的时间点移动，使得所述第四开关管的占空比逐渐减小至第一预定值，所述第一预定值为所述第一开关管的占空比；
    在所述第四开关管的占空比小于0.5的情况下，控制所述第三上升沿对应的时间点不变，且控制所述第三下降沿对应的时间点移动，使得所述第三开关管的占空比逐渐增大至第二预定值，所述第二预定值为所述第二开关管的占空比。
  5. 根据权利要求2至4中任一项所述的方法，其特征在于，所述全桥模态包括全桥PFM模态以及全桥PWM模态，所述半桥模态包括半桥PFM模态以及半桥PWM模态，在所述LLC谐振电路处于稳态时，所述全桥PFM模态、所述全桥PWM模态、所述半桥PFM模态以及所述半桥PWM模态对应的所述输入电压依次增大。
  6. 根据权利要求5所述的方法，其特征在于，所述LLC谐振电路还包括LLC谐振腔，所述LLC谐振腔包括三端子的第五开关管以及三端子的第六开关管，所述第五开关管的第一端与所述第一开关管的第二端连接，所述第五开关管的第二端与所述第六开关管的第一端连接，所述第六开关管的第二端与所述第四开关管的第一端连接，
    在所述LLC谐振电路的输入电压大于设定阈值，且所述LLC谐振电路处于全桥模态的情况下，所述方法还包括：
    在所述LLC谐振电路处于所述全桥PWM模态的情况下，获取所述第一开关管或 者所述第二开关管的实时频率以及实时占空比；
    在所述实时频率等于谐振频率，且所述实时占空比等于第三预定值时，控制所述第五开关管以及所述第六开关管的占空比变为第四预定值，使得所述LLC谐振电路由所述全桥PWM模态转换至半桥PFM模态。
  7. 根据权利要求6所述的方法，其特征在于，在所述输入电压小于或者等于所述设定阈值，且所述LLC谐振电路处于所述半桥模态的情况下，所述方法还包括：
    在所述LLC谐振电路处于所述半桥PWM模态的情况下，获取所述实时频率以及所述实时占空比；
    在所述实时频率等于谐振频率，且所述实时占空比等于所述第三预定值时，控制所述第五开关管以及所述第六开关管的占空比变为所述第三预定值，使得所述LLC谐振电路由所述半桥PWM模态转换至所述全桥PFM模态。
  8. 一种LLC谐振电路的控制装置，所述LLC谐振电路包括全桥逆变电路，所述全桥逆变电路包括四个三端子的开关管，其特征在于，所述装置包括：
    第一控制单元，用于在所述LLC谐振电路的输入电压大于设定阈值，且所述LLC谐振电路处于全桥模态的情况下，控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第一规则变化，以使得所述LLC谐振电路由所述全桥模态转换至半桥模态；
    第二控制单元，用于在所述输入电压小于或者等于所述设定阈值，且所述LLC谐振电路处于所述半桥模态的情况下，控制部分的所述开关管的驱动电压的上升沿对应的时间点不变，且下降沿对应的时间点按第二规则变化，以使得所述LLC谐振电路由所述半桥模态转换至所述全桥模态。
  9. 一种处理器，其特征在于，所述处理器用于运行程序，其中，所述程序运行时执行权利要求1至7中任意一项所述的方法。
  10. 一种LLC谐振系统，其特征在于，包括：
    LLC谐振电路，包括全桥逆变电路，所述全桥逆变电路包括四个三端子的开关管；
    所述LLC谐振电路的控制装置，所述控制装置用于执行权利要求1至7中任一项所述的方法。
PCT/CN2022/119253 2021-09-18 2022-09-16 Llc谐振电路的控制方法以及控制装置 WO2023041021A1 (zh)

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