WO2023001315A1 - 开关变换器 - Google Patents

开关变换器 Download PDF

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Publication number
WO2023001315A1
WO2023001315A1 PCT/CN2022/113213 CN2022113213W WO2023001315A1 WO 2023001315 A1 WO2023001315 A1 WO 2023001315A1 CN 2022113213 W CN2022113213 W CN 2022113213W WO 2023001315 A1 WO2023001315 A1 WO 2023001315A1
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Prior art keywords
voltage
current
switching converter
switching
reference voltage
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PCT/CN2022/113213
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English (en)
French (fr)
Inventor
张海波
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圣邦微电子（北京）股份有限公司
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Publication of WO2023001315A1 publication Critical patent/WO2023001315A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates to the technical field of electronic circuits, and more particularly relates to a switching converter.
  • Modern portable electronic devices are often provided with a power source, such as a battery, which acts as a direct current (DC) for the various electronic components within the device.
  • a power source such as a battery
  • DC direct current
  • these components will have different voltage requirements, so such devices typically employ one or more voltage converters that step down the nominal voltage associated with the power supply to a voltage suitable for the different electronic components.
  • a linear regulator the output voltage is regulated by adjusting a passive element such as a variable resistor to control the continuous flow of current from the voltage source to the load.
  • the switching converter controls the output voltage by connecting or disconnecting the current.
  • one or more switches and inductance and capacitance components are used to store and transfer energy to the load end.
  • the regulator controls the on and off of the switching element to Regulates the voltage delivered to the load, thereby controlling the amount of power delivered through the inductor in discrete current pulses.
  • the inductor and capacitor convert the delivered current pulses into a steady load current to regulate the load voltage.
  • regulation of the output voltage is achieved by adjusting the turn-on and turn-off times of the switching elements according to the feedback signals representing the output voltage and the load current.
  • Switching converters operating in current mode are widely used because they provide good linearity and load transient rejection, as well as good current limiting capability during fault conditions such as output short circuits.
  • Many current-mode DC-DC converters monitor the inductor current and compare it to the peak inductor current to determine when to turn off the main switching element to eliminate excess current delivery.
  • Many current-mode switching converter circuits consist of the following sections: a logic section; an output switch controlled by the logic section; an oscillator that provides a periodic timing signal to turn on the main switch; a current amplifier that relays the sense measuring voltage; an error amplifier, which adjusts its output voltage according to the load state; and a current comparator, which generates a signal for controlling when the sensed voltage is compared with the voltage from the error amplifier in a predetermined manner. Turn-off timing of the main switch.
  • Switching converters typically exhibit higher efficiencies (where efficiency is defined as the ratio of output voltage to input voltage) than linear regulators, resulting in significantly less unwanted heat dissipation.
  • efficiency is not always maximized and varies in proportion to the size of the load.
  • the efficiency of the switching converter is a function of the output current. When the load decreases, the switching loss and internal loss inside the switching converter will not decrease, so the efficiency of the switching converter will decrease at this time.
  • An effective alternative to forced continuous mode operation is to allow the switching converter to enter burst mode operation.
  • the regulator can ignore switching cycles at light loads, reducing transistor losses.
  • the active switching element and other unnecessary parts of the converter circuit can be kept off due to the load current dropping to a specified value, thereby reducing the internal loss and switching loss of the circuit, Improve circuit efficiency.
  • a switching converter capable of operating in burst mode basically uses the same circuit as the usual switching converter above, plus a burst comparator and circuitry to provide a burst threshold level, when the output load is below a certain value, the output The voltage will be charged by the set output current, and then the output feedback voltage is higher than the reference voltage, triggering the burst mode switching part of the circuit to reduce power consumption.
  • the defect of the existing switching converters working in burst mode is that the sampling current and the superimposed signal of the slope compensation are compared with the minimum clamping voltage to control the turn-off timing of the main switch in the circuit.
  • the slope voltage component is a function of the duty cycle. When the on-time is longer, the slope voltage component is larger and the peak value of the inductor current is smaller.
  • the switch duty cycle when the minimum clamping voltage is fixed, the switch duty cycle The smaller the ratio, the larger the current in burst mode and the larger the output ripple, which will generate a lot of noise in application and cannot be used.
  • the object of the present invention is to provide a switching converter, which can adaptively adjust the minimum peak inductor current of the switching converter according to the input voltage and output voltage when the switching converter works in burst mode, so that the burst mode The current under control remains relatively constant, effectively reducing output ripple and noise.
  • a switching converter comprising: an input terminal receiving an input voltage; an output terminal connected to a load to provide an output voltage; a power circuit connected to the input terminal and the output terminal, and the power circuit adopts an inductor And at least one main switch tube regulates the current provided to the load; a control circuit connected to the power circuit, the control circuit is used to control the turn-on and turn-off of the at least one main switch tube according to the received reference voltage , so as to control the minimum peak inductor current flowing through the inductor; and a regulating circuit connected to the control circuit, providing the reference voltage to the control circuit, when the switching converter operates in burst mode, the The regulating circuit is used for adaptively regulating the reference voltage according to the input voltage and the output voltage.
  • the adjustment circuit adjusts the reference voltage so that the reference voltage is positively correlated with the difference between the output voltage and the input voltage.
  • control circuit includes: a logic circuit, configured to control the turn-on and turn-off of the at least one main switch; an oscillator, configured to provide a clock signal to the logic circuit, and the clock signal is used for and controlling the conduction moment of the at least one main switch tube.
  • control circuit further includes: an error amplifier, configured to compare the feedback voltage of the output voltage with a reference voltage to obtain an error signal, and the error amplifier adjusts the error signal according to the load state; burst A clamping module, for comparing the error signal with a reference voltage, and setting a clamping voltage representing the minimum peak inductor current according to the comparison result; and a current comparator, for superimposing the current sampling signal and the slope compensation signal Compared with the clamping voltage, the turn-off time of the at least one switch tube is controlled according to the comparison result.
  • an error amplifier configured to compare the feedback voltage of the output voltage with a reference voltage to obtain an error signal, and the error amplifier adjusts the error signal according to the load state
  • burst A clamping module for comparing the error signal with a reference voltage, and setting a clamping voltage representing the minimum peak inductor current according to the comparison result
  • a current comparator for superimposing the current sampling signal and the slope compensation signal Compared with the clamping voltage, the turn-off time of the at least one switch tube is
  • control circuit further includes: a burst comparator, configured to compare the error signal with a set burst threshold voltage, and when the error signal is smaller than the burst threshold voltage, control The switching converter enters burst mode.
  • a burst comparator configured to compare the error signal with a set burst threshold voltage, and when the error signal is smaller than the burst threshold voltage, control The switching converter enters burst mode.
  • the burst clamping module is configured to set the clamping voltage equal to the reference voltage when the error signal is less than the reference voltage, and set the clamping voltage equal to the reference voltage when the error signal is greater than or equal to the reference voltage , setting the clamping voltage equal to the error signal.
  • the regulating circuit includes: a transconductance amplifier, the two input terminals of which respectively receive the input voltage and the output voltage, and are used to convert the voltage difference between the two into current information; the first resistor, the second One end is connected to the output end of the transconductance amplifier; and a voltage follower, one input end is used to receive the reference voltage, and the other input end and output end are connected to the second end of the first resistor, wherein A connection node between the transconductance amplifier and the first resistor is used to provide the compensated reference voltage to the burst clamping module.
  • control circuit further includes: a sampling tube and a sampling resistor for sampling the inductor current flowing through the at least one main switch tube; and a current amplifier for generating a sensing voltage across the sampling resistor , to obtain the current sampling signal.
  • the power circuit further includes at least one synchronous switch tube, and the turn-on and turn-off of the synchronous switch tube is opposite to that of the main switch tube.
  • control circuit further includes: an inverting current comparator.
  • the inverting current comparator monitors the current flowing through the synchronous switching tube to The synchronous switching tube is turned off when the current is reversed.
  • the switching converter is a buck converter, a boost converter or a buck-boost converter.
  • the switching converter is a synchronous switching converter or an asynchronous switching converter.
  • the switching converter in the embodiment of the present invention includes an adjustment circuit connected to the control circuit.
  • the adjustment circuit adaptively adjusts the minimum peak inductor current of the switching converter according to the input voltage and the output voltage, so that When the input voltage and output voltage fluctuate and the duty cycle changes, the influence of the duty cycle on the inductor current is offset, so that the controlled current in the burst mode remains relatively constant, effectively reducing the output ripple and noise.
  • FIG. 1 shows a schematic circuit diagram of a conventional buck-boost switching converter with burst mode
  • FIG. 2 shows a schematic diagram of the principle of a traditional buck-boost switching converter with burst mode
  • Fig. 3 shows the working principle diagram of the burst comparator in Fig. 1;
  • Fig. 4 shows the schematic diagram of the superposition of the current sampling signal and the slope compensation signal
  • Fig. 5 shows a schematic circuit diagram of a switching converter according to the first embodiment of the present invention
  • FIG. 6 shows a schematic circuit diagram of another switching converter according to the first embodiment of the present invention.
  • FIG. 7 shows a schematic circuit diagram of a switching converter according to a second embodiment of the present invention.
  • FIG. 8 shows a schematic circuit diagram of another switching converter according to the second embodiment of the present invention.
  • a “circuit” refers to a conductive loop formed by at least one element or sub-circuit through electrical or electromagnetic connections.
  • an element or circuit When an element or circuit is said to be “connected to” another element or an element/circuit is said to be “connected between” two nodes, it can be directly coupled or connected to the other element or there can be intervening elements and the connection between elements can be be physical, logical, or a combination thereof.
  • an element when an element is referred to as being “directly coupled to” or “directly connected to” another element, there are no intervening elements present.
  • the high-voltage transistor comprises an N-channel metal-oxide-semiconductor (NMOS) field-effect transistor (FET), wherein a high voltage is provided at a first terminal (ie, drain) and a second terminal ( ie source).
  • NMOS metal-oxide-semiconductor
  • FET field-effect transistor
  • an integrated controller circuit may be used to drive the power switch when regulating the energy provided to the load.
  • ground or ground potential in this application refers to a reference voltage or potential relative to which all electrical circuits or integrated circuits (ICs) are defined or measured. other voltages or potentials.
  • FIG. 1 shows a conventional buck-boost switching converter with burst mode.
  • inductor L and transistor M4 are sequentially connected between the input terminal and output terminal of the switching converter, transistor M2 is connected between the intermediate node of transistor M1 and inductor L and ground, and transistor M3 is connected between transistor M4 and inductor L between node and ground.
  • the transistors M1, M2 and the inductor L constitute the step-down circuit in the switching converter; the transistors M3, M4 and the inductor L constitute the boosting circuit in the switching converter.
  • the control circuit 110 is used to control the turn-on and turn-off of the transistors M1-M4 in the power circuit, so as to control the inductor to output energy in discontinuous pulses.
  • the control circuit 110 includes an oscillator 101 , a logic circuit 102 , a current amplifier 103 and a current inverse comparator 104 .
  • the oscillator 101 is used to provide an internal clock for switching timing for the circuit, and at the same time generate a sawtooth wave, which is provided to the PWM comparator and a compensation signal for the slope compensation circuit.
  • the logic circuit 102 is used to realize the logic control function of the system, process the logic signals of each module controlling the working state of the transistors M1-M4, and generate switch driving signals for the transistors M1-M4.
  • the logic circuit 102 may include a pulse width modulator (PWM) circuit or any other suitable circuit capable of controlling the duty cycle of the power switches M1-M4.
  • PWM pulse width modulator
  • the switching converter 100 works in buck mode, the transistor M3 is kept off, the transistor M4 is kept on, and the transistors M1 and M2 are turned on and off alternately.
  • the transistor M1 is also called the main switching tube, and the transistor M2 is also called the synchronous switching tube.
  • the current amplifier 103 indirectly samples the main switching tube M1 through the sampling tube M0, and the inductor current flows through sampling resistor Rs, and generate a sensing voltage at its two ends, the sensing voltage is approximately equal to the product of the inductor current and the sampling resistor value, and then the voltage is amplified by the current amplifier 103 to obtain a current sampling signal.
  • the current comparator 115 When the superposition signal Vsum of the current sampling signal and the slope compensation signal exceeds the voltage of the non-inverting input terminal of the current comparator 115 (that is, the clamping voltage Vc1 output by the first clamping module 112), the current comparator 115 provides a first reset signal RST1 is provided to the logic circuit 102 to control the main switch M1 to turn off and the synchronous switch M2 to turn on.
  • the switching converter 100 works in the boost mode, the transistor M1 is kept on, the transistor M2 is kept off, and the transistors M3 and M4 are turned on and off alternately.
  • the transistor M3 is also called the main switching tube, and the transistor M4 is also called the synchronous switching tube.
  • the current comparator 116 compares the superimposed signal Vsum of the current sampling signal and the slope compensation signal with the positive phase The voltages at the input terminals are compared (that is, the clamping voltage Vc2 output by the second clamping module 114), and when the superimposed signal Vsum is higher than the voltage at the non-inverting input terminal, the current comparator 116 provides a second reset signal RST2 to the logic circuit 102, To control the main switching tube M3 to turn off and the synchronous switching tube M4 to turn on.
  • clamping voltage Vc1 output by the first clamping module 112 is converted by the level conversion module 113, and the converted voltage signal is provided to the second clamping module 114, and the second clamping module 114 according to the received The voltage signal generates a clamping voltage Vc2.
  • the switching converter in Figure 1 has two modes of continuous operation and burst mode.
  • continuous operation mode the switching converter can reduce noise, RF interference and output voltage ripple.
  • burst mode the main switching tube and the synchronous switching tube in the circuit are intermittently turned on under light load. By reducing the switching loss and quiescent current of the circuit under light load, higher efficiency can be obtained.
  • the current inverting comparator 104 In Burst Mode, the current inverting comparator 104 is activated and the inductor current is not allowed to go negative.
  • the current inversion comparator 104 monitors the inductor current flowing through the synchronous switch M4 and provides a signal to the logic circuit 102 to turn off the synchronous switch M2 or M4 when the current experiences a current inversion state.
  • the first clamping module 112 and the second clamping module 114 obtain fixed clamping voltages Vc1 and Vc2 to set the minimum peak inductor current level, and then monitor the error signal through the burst comparator 117 to Determines when Burst Mode is activated and disabled.
  • the conventional switching converter 100 utilizes the burst mode control circuit 120 to clamp the minimum clamping voltage Vc in the burst mode to charge the inductor L with the minimum peak current.
  • the output The voltage Vout will be charged by the set minimum peak current.
  • the feedback voltage VFB is higher than the reference voltage VREF0, the error signal Vea between the two will drop.
  • the burst mode Burst will be triggered. Shut down the system until the load current increases to a certain value, then release the burst mode, turn on all circuits, and resume normal operation.
  • the disadvantage of the conventional switching converter 100 is that the clamping voltage of the burst mode is fixed inside the switching converter. Since the burst clamp fixes the minimum peak inductor current during each switching cycle, the ripple on the output voltage is also fixed. Efficiency is highest at light loads for higher burst clamp voltages, but at the expense of larger output voltage ripple. For lower burst clamp voltages, the output voltage has less ripple, but its efficiency for light loads decreases again.
  • the current comparator in the conventional switching converter 100 compares the superimposed signal of the current sampling signal Isen and the slope compensation signal Vsaw with the clamping voltage Vc to determine when to turn off the main switching tube,
  • the slope voltage component is a function of the duty cycle D, which can be obtained:
  • the current sampling signal Isen represents the peak current of the inductor. It can be seen from the above formula that under the same clamping voltage Vc, the smaller the duty cycle of the switch, the smaller the slope voltage component, and the comparator flips the corresponding current Isen The larger it is, the larger the ripple of the output voltage will be. Therefore, when the clamping voltage Vc is fixed, the smaller the switching duty cycle D of the traditional switching converter 100 in the burst mode, the larger the ripple of the output voltage, so the traditional switching converter 100 cannot be applied to small Duty cycle application environment.
  • Fig. 5 shows a schematic circuit diagram of a switching converter according to the first embodiment of the present invention.
  • the switching converter 200 in FIG. 5 includes a control circuit 210 and an external power circuit.
  • the power circuit includes one or more switches and filter elements (eg, inductors and capacitors, etc.), and the one or more switches and filters are configured to respond to one or more switch drive signals from the control circuit 210 To regulate the power transfer from the input to the output of the switching converter.
  • one or more switches in the power circuit are integrated with the control circuit 210 to form an integrated circuit chip.
  • the switching converter 200 can be classified according to the topology of the power circuit, and the switching converter can be divided into a buck converter, a boost converter, a flyback converter and a buck-boost converter. Compression type (buck-boost) converter. In the following, a step-down converter is taken as an example for illustration.
  • switching converter 200 can operate and transition in multiple modes, including but not limited to continuous operation mode and burst mode.
  • continuous operation mode the switching converter can reduce noise, RF interference and output voltage ripple.
  • burst mode at least one switch in the power circuit switches at a fixed frequency for a short duration as needed to maintain the output voltage level. These short durations of switching may be referred to as “burson" periods or burst-on intervals. Between short durations, the switches can be disabled and parts of the circuitry in the controller can be temporarily disabled to reduce power loss. These durations of inhibiting switching may be referred to as “burst off” periods or burst off intervals. Switching control in this manner is often referred to as “burst mode” and/or “burst mode”. Higher efficiency can be obtained by reducing the switching loss and quiescent current of the circuit under light load by turning on the switch intermittently under light load.
  • the oscillator 211 When operating in continuous mode of operation, the oscillator 211 or any other suitable device capable of providing switching timing to the circuit (i.e., by generating narrow pulses at a constant frequency) provides switching timing at the beginning of each cycle cycle, which The oscillator 211 provides a narrow pulse clock signal SET to the logic circuit 212, and the logic circuit 212 turns on the main switch M1 and turns off the synchronous switch M2 according to the received clock signal SET.
  • the logic circuit 212 may include a pulse width modulator (PWM) circuit or any other device capable of controlling the duty cycle of the main switch M1 in the power circuit (that is, the amount of time the main switch M1 is turned on is compared with the time of the switching cycle) the appropriate circuit.
  • PWM pulse width modulator
  • the current comparator 213 When the superposition signal Vsum of the current sampling signal Isen and the slope compensation signal Vsaw exceeds the voltage of the non-inverting input terminal of the current comparator 213 (that is, the output of the burst clamp module 217), the current comparator 213 provides a reset signal RST to the logic circuit 212, so that the main switch M1 is turned off and the synchronous switch M2 is turned on. This makes the voltage across the inductor 221 change to -Vout, causing the inductor current to weaken until the next clock signal SET turns on the main switch M1 and turns off the synchronous switch M2 again.
  • any other type of suitable switching element may be used without departing from the principles of the invention.
  • this embodiment is described with a synchronous switching converter, the present invention is not limited thereto. The present invention is also applicable to non-synchronous switching converters. Those skilled in the art may also use rectifier diodes to replace the Synchronous switching tube M2.
  • the burst clamp module 217 fixes the clamp voltage Vc1 at the set reference voltage, thereby setting the minimum peak inductor current level. Specifically, the burst clamping module 217 compares the error signal Vea with the reference voltage VREF1, when the error signal Vea is smaller than the reference voltage VREF1, the clamping voltage Vc1 is equal to the reference voltage VREF1; when the error signal Vea is greater than or equal to the reference voltage VREF1, The clamping voltage Vc1 is equal to the error signal Vea. Error signal Vea is then monitored by burst comparator 214 to determine when to activate and disable burst mode.
  • the burst comparator 214 When the output load current is less than a certain value, the output voltage will be charged by the set minimum peak inductor current, then the feedback voltage VFB is greater than the reference voltage VREF0, and the error signal Vea will decrease.
  • the burst comparator 214 When the error signal Vea is less than the burst threshold voltage VTH, the burst comparator 214 outputs a logic high level to activate the burst mode, so that the main switching tube M1, the synchronous switching tube M2 and the predetermined components of the remaining circuits are all cut off, so that Reduce power consumption. At this time, the load current is completely delivered by the output capacitor 223.
  • the burst comparator 214 When the output voltage drops, the voltage on the error signal Vea increases above the level set by the hysteresis of the burst comparator 214, so that the burst comparator 214 outputs a logic low level, the burst mode is released, all circuits are turned on, and normal operation resumes.
  • the current inverting comparator 215 When working in burst mode, the current inverting comparator 215 is activated and the inductor current is not allowed to be negative.
  • the current inverting comparator 215 monitors the current flowing through the synchronous switch tube M2 and provides a signal to the logic circuit 212 to The synchronous switch M2 is turned off when the inductor current experiences a current reversal state.
  • the current reversal state represents when the current passes through a zero current level.
  • a voltage offset can be applied to the inverting input of the current inverting comparator 215 so that the current inverting comparator 215 turns off the synchronous switch M2 just before the inductor or switch current crosses the zero current threshold.
  • the switching converter 200 of this embodiment further includes an adjustment circuit 230, and the adjustment circuit 230 is used for compensating the reference voltage of the burst clamping module 217 according to the functional relationship between the input voltage Vin and the output voltage Vout, thereby canceling the duty cycle Compared with the influence in the burst mode, the current controlled by the burst mode is always kept constant, thereby reducing the output ripple and noise of the switching converter in the burst mode.
  • the regulating circuit 230 includes a voltage follower 231 , a transconductance amplifier 232 and a resistor R2 connected therebetween.
  • the two input ends of the transconductance amplifier 232 respectively receive the input voltage Vin and the output voltage Vout, and are used to convert the voltage difference between the input voltage Vin and the output voltage Vout into current information, and the current is applied to one end of the resistor R2 superior.
  • the other end of the resistor R2 is connected to the output terminal of the voltage follower 231.
  • the voltage follower 231 is realized by an operational amplifier, and one input terminal thereof is connected to the output terminal. According to the virtual short principle of the operational amplifier, the voltage at the output terminal is equal to the voltage at the input terminal.
  • the other end of the resistor R2 is applied with a voltage equal to the reference voltage VREF1, so that the reference voltage after compensation can be approximately equal to the difference between the product of the reference voltage VREF1 and the current information output by the transconductance amplifier 232 and the resistor R2 ,which is:
  • VREF1' VREF1-(Vin-Vout) ⁇ R2/R1
  • R1 represents the resistance value inside the transconductance amplifier 232
  • R2 represents the resistance value of the resistor R2.
  • the compensated reference voltage in the switching converter 200 of this embodiment carries the voltage difference information between the input voltage and the output voltage.
  • the compensated reference voltage can offset the influence of the duty cycle, so that the controlled current in the burst mode remains relatively constant, effectively reducing the output ripple and noise.
  • FIG. 6 shows a schematic circuit diagram of another switching converter according to the first embodiment of the present invention. Different from FIG. 5, FIG. 6 shows an asynchronous switching converter 300, which differs from the synchronous switching converter in that the synchronous switching tube M2 is replaced by a rectifying diode D1, which can prevent the output capacitor 323 from discharging to ground . Otherwise, the remaining elements in FIG. 6 generally perform the same purpose as the corresponding elements described above in FIG. 5 .
  • FIG. 5 and FIG. 6 each illustrate an embodiment of a buck synchronous switching converter and a buck asynchronous switching converter
  • the present invention is not limited to these embodiments.
  • the advantages of the present invention are also applicable to other types of converters, such as boost synchronous switching converters, boost asynchronous switching converters, buck-boost switching converters and other suitable types of converters.
  • FIG. 7 shows a schematic circuit diagram of a switching converter according to a second embodiment of the present invention.
  • Figure 7 shows a boost synchronous switching converter that uses many of the same components as those found in the buck converter shown in Figure 5 .
  • the control circuit 410 of the boost converter shown in FIG. 7 works as follows.
  • the input voltage Vin is applied to both ends of the inductor 421.
  • the current starts to flow through the inductor 421, and the synchronous switch M2 prevents the output capacitor 423 discharges to ground, and the output capacitor 423 is responsible for delivering current to the load 424 .
  • the main switch M1 is turned off and the synchronous switch M2 is turned on, the output capacitor 423 is charged by the energy stored in the inductor 421 . At this point, additional current begins to flow through the load, causing the output voltage Vout to rise. After a certain period of time, the main switch M1 is turned on again, and the cycle is repeated to maintain the required output voltage level and deliver the required current to the load as required.
  • the remainder of the circuit components in FIG. 7 operate as previously described for the circuit components shown in FIG. 5 .
  • the current comparator 413 compares the superposition signal Vsum of the current sampling signal Isen from the current amplifier 419 and the slope compensation signal Vsaw with the error signal Vea output by the error amplifier 416, and provides a reset signal RST to the logic circuit 412 according to the comparison result, so as to determine what At the same time, the main switching tube M1 is turned off and the synchronous switching tube M2 is turned on.
  • switching converter 400 of FIG. 7 can also operate and transition in multiple modes, including but not limited to continuous operation mode and burst mode.
  • the burst clamp module 417 fixes the clamp voltage Vc2 at the set reference voltage, thereby setting the minimum peak inductor current level.
  • the burst clamping module 417 compares the error signal Vea with the reference voltage VREF2' compensated by the adjustment circuit 430, and when the error signal Vea is smaller than the compensated reference voltage VREF2', the clamping voltage Vc2 is equal to the reference voltage VREF2 '; When the error signal Vea is greater than or equal to the reference voltage VREF2', the clamping voltage Vc1 is equal to the error signal Vea.
  • Error signal Vea is then monitored by burst comparator 414 to determine when to activate and deactivate burst mode.
  • the output load current is less than a certain value, the output voltage will be charged by the set minimum peak inductor current, then the feedback voltage VFB is greater than the reference voltage VREF0, and the error signal Vea will decrease.
  • the burst comparator 414 outputs a logic high level to activate the burst mode, so that the main switching tube M1, the synchronous switching tube M2 and the predetermined components of the remaining circuits are all cut off, so that Reduce power consumption. At this time, the load current is completely delivered by the output capacitor 423.
  • the regulator circuit 430 compensates the reference voltage VREF2 according to the functional relationship between the input voltage Vin and the output voltage Vout, so as to offset the influence of the duty cycle in the burst mode, so that the burst mode control
  • the current is always kept constant to reduce the output ripple and noise of the switching converter in burst mode.
  • the regulating circuit 430 includes a voltage follower 441 , a transconductance amplifier 442 and a resistor R2 connected therebetween.
  • the two input ends of the transconductance amplifier 442 respectively receive the input voltage Vin and the output voltage Vout, and are used to convert the voltage difference between the input voltage Vin and the output voltage Vout into current information, and the current is applied to one end of the resistor R2 superior.
  • the other end of the resistor R2 is connected to the output end of the voltage follower 441.
  • the voltage follower 441 is implemented by an operational amplifier, one input end of which is connected to the reference voltage VREF2, and the other input end is connected to the output end.
  • the voltage at the output end is equal to the voltage at the input end, so the other end of the resistor R2 is applied with a voltage equal to the reference voltage VREF2, so that the reference voltage after compensation is approximately equal to the reference voltage VREF2 and the current information output by the transconductance amplifier 442 and the product of resistor R2, that is:
  • VREF2' VREF2-(Vout-Vin) ⁇ R2/R1
  • R1 represents the resistance value inside the transconductance amplifier 442
  • R2 represents the resistance value of the resistor R2.
  • the reference voltage after compensation in the switching converter 400 of this embodiment carries the voltage difference information between the input voltage and the output voltage.
  • the minimum voltage can be adjusted according to the input voltage and the output voltage.
  • the peak inductor current level so that the influence of the duty cycle on the inductor current can be offset when the input voltage and output voltage fluctuations cause the duty cycle to change, so that the controlled current in the burst mode remains relatively constant, effectively reducing the output ripple and noise.
  • FIG. 8 shows a schematic circuit diagram of another switching converter according to the second embodiment of the present invention. Different from FIG. 7, FIG. 8 shows a non-synchronous switching converter 500. The difference between it and the synchronous switching converter is that the synchronous switching tube M2 is replaced by a rectifying diode D1, which can prevent the output capacitor 523 from discharging to ground. . Otherwise, the remaining elements in FIG. 8 generally perform the same purpose as the corresponding elements described above in FIG. 7 .
  • the buck-type and boost-type switching converters are respectively described in conjunction with FIG. 5 to FIG. 8 , however, it can be understood that the The control circuit can also be used in switching converters of other topologies, including but not limited to buck, boost, buck-boost, non-inverting buck-boost, forward, flyback and other topologies .
  • the switching converter in the embodiment of the present invention includes an adjustment circuit connected to the control circuit, and the adjustment circuit adaptively adjusts the minimum peak value of the switching converter according to the input voltage and the output voltage when the switching converter works in burst mode.
  • Inductor current so that the influence of the duty cycle on the inductor current can be offset when the input voltage and output voltage fluctuations cause the duty cycle to change, so that the controlled current in the burst mode remains relatively constant, effectively reducing the output ripple and noise .

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Abstract

本发明公开了一种开关变换器。包括输入端；连接到负载的输出端；与输入端和输出端连接的功率电路，功率电路采用至少一个主开关管调节提供给所述负载的电流；与功率电路连接的控制电路，控制电路用于根据接收的参考电压控制至少一个主开关管的导通和关断，以便控制流经电感的最小峰值电感电流；以及与控制电路连接的调节电路，用于提供参考电压至控制电路，当开关变换器以突发模式工作时，调节电路用于根据输入电压和输出电压自适应调节参考电压，从而可以在输入电压和输出电压波动导致占空比变化时抵消掉占空比对电感电流的影响，从而使得突发模式下控制的电流保持相对恒定，有效降低了输出纹波和噪声。

Description

开关变换器
本申请要求了申请日为2021年07月22日、申请号为202110831179.5、名称为“开关变换器”的中国发明申请的优先权，并且通过参照上述中国发明申请的全部说明书、权利要求、附图和摘要的方式，将其引用于本申请。
技术领域
本发明涉及电子电路技术领域，更具体地涉及一种开关变换器。
背景技术
现代便携式电子设备通常设置有诸如电池的电源，其用作装置内的各种电子部件的直流(DC)。然而，通常这些部件将具有不同的电压要求，因此这类装置通常采用一个或多个电压转换器，该电压转换器将与电源相关联的标称电压降低到适合于不同电子部件的电压。
现有的电压转换器通常采用线性调节器和开关变换器这两种。在线性调节器中，输出电压通过调节无源元件(例如可变电阻)进行调节，以控制电流从电压源到负载的连续流动。开关变换器通过将电流连通或断开来控制输出电压，通常采用一个或者多个开关以及电感和电容部件来存储和传递能量到负载端，调节器通过控制开关元件的导通和关断，来调节输送到负载端的电压大小，从而控制通过电感以不连续电流脉冲形式输送的电量。所述电感和电容将输送的电流脉冲转化成稳定的负载电流，以便调控负载电压。最终，根据表示输出电压和负载电流的反馈信号通过调节开关元件的导通和关断时间来实现调节输出电压。
以电流模式工作的开关变换器可以提供良好的线性和负载瞬态信号抑制，并且在故障状态(例如输出短路)期间具有很好的限流能力，因此得到了广泛的应用。很多电流模式的DC-DC变换器监控电感电流，并且将它与峰值电感电流作比较，以确定何时断开主开关元件，从而消 除过量电流的输送。
很多电流模式的开关变换器电路包括以下部分：逻辑部分；由逻辑部分控制的输出开关；用于提供周期性定时信号以开启主开关的振荡器；电流放大器，它用于根据电感电流中继感测电压；误差放大器，它根据负载状态调节其输出电压；和电流比较器，当感测电压以预定的方式与来自误差放大器的电压作比较时，所述电流比较器产生一信号，用于控制主开关的关断时机。
开关变换器通常表现出比线性调节器更高的的效率(其中，效率被定义为输出电压与输入电压之比)，从而可以使得不希望的散热显著减少。不过，开关变换器的效率并不总是最大的，并且与负载的大小成比例变化。开关变换器的效率是输出电流的函数，当负载减小时，开关变换器内部的开关损耗和内部损耗并不会减小，因此此时开关变换器的效率是减小的。
上述较轻负载下的效率损失在以强制的连续运行模式工作的开关变换器中是常见的。在强制的连续模式下，开关变换器的主开关不论负载状态如何都会定期的导通和关断，因此其在较轻负载下的效率损失更大。
以强制的连续模式工作的有效替代方案是允许开关变换器进入突发模式工作。当在这种模式下工作时，调节器在负载较轻时可以忽略开关循环，从而降低晶体管的损耗。例如，在突发模式工作时，将主动开关元件和可选的变换器电路其他不需要的部分由于负载电流降到规定值一下儿保持关断状态，从而可以降低电路的内部损耗和开关损耗，提高电路效率。
能够以突发模式工作的开关变换器基本上采用与上述的通常的开关变换器相同的电路，另加突发比较器和电路提供突发阈值水平，当输出负载低于某个值时，输出电压会被设定的输出电流充上去，然后输出反馈电压高于基准电压，触发突发模式切换部分电路，以便减少功率消耗。
现有的以突发模式工作的开关变换器的缺陷在于将采样电流与斜坡补偿的叠加信号与最小钳位电压进行比较来控制电路中主开关的断开时机。其中，斜坡电压分量是占空比的一个函数，当导通时间越长，斜坡 电压分量越大，电感电流峰值越小，导致在突发模式下，当最小钳位电压固定时，开关占空比越小，突发模式的电流越大，输出纹波就越大，在应用时会产生很大的噪声，无法使用。
发明内容
有鉴于此，本发明的目的在于提供一种开关变换器，可在开关变换器以突发模式工作时根据输入电压和输出电压自适应调节开关变换器的最小峰值电感电流，从而使得突发模式下控制的电流保持相对恒定，有效降低了输出纹波和噪声。
根据本发明提供了一种开关变换器，包括：输入端，接收输入电压；连接到负载的输出端，提供输出电压；与所述输入端和输出端连接的功率电路，所述功率电路采用电感以及至少一个主开关管调节提供给所述负载的电流；与所述功率电路连接的控制电路，所述控制电路用于根据接收的参考电压控制所述至少一个主开关管的导通和关断，以便控制流经所述电感的最小峰值电感电流；以及与所述控制电路连接的调节电路，提供所述参考电压至所述控制电路，当所述开关变换器以突发模式工作时，所述调节电路用于根据所述输入电压和所述输出电压自适应调节所述参考电压。
可选的，所述调节电路调节所述参考电压以使得所述参考电压与所述输出电压和所述输入电压之间的差值正相关。
可选的，所述控制电路包括：逻辑电路，用于控制所述至少一个主开关管的导通和关断；振荡器，用于向所述逻辑电路提供时钟信号，所述时钟信号用于控制所述至少一个主开关管的导通时刻。
可选的，所述控制电路还包括：误差放大器，用于将所述输出电压的反馈电压与基准电压进行比较，以获得误差信号，所述误差放大器根据负载状态调节所述误差信号；突发钳位模块，用于将所述误差信号与参考电压进行比较，根据比较结果设定表征最小峰值电感电流的钳位电压；以及电流比较器，用于将电流采样信号和斜坡补偿信号的叠加信号与所述钳位电压进行比较，根据比较结果控制所述至少一个开关管的关 断时刻。
可选的，所述控制电路还包括：突发比较器，用于将所述误差信号与设定的突发阈值电压进行比较，并在所述误差信号小于所述突发阈值电压时，控制所述开关变换器进入突发模式。
可选的，所述突发钳位模块用于在所述误差信号小于所述参考电压时，设置所述钳位电压等于所述参考电压，并在所述误差信号大于等于所述参考电压时，设置所述钳位电压等于所述误差信号。
可选的，所述调节电路包括：跨导放大器，其两个输入端分别接收所述输入电压和输出电压，用于将二者之间的电压差转换成电流信息；第一电阻，其第一端与所述跨导放大器的输出端连接；以及电压跟随器，其一个输入端用于接收所述参考电压，另一输入端和输出端与所述第一电阻的第二端连接，其中，所述跨导放大器与所述第一电阻的连接节点用于将补偿后的参考电压提供至所述突发钳位模块。
可选的，所述控制电路还包括：采样管和采样电阻，用于采样流过所述至少一个主开关管的电感电流；以及电流放大器，通过在所述采样电阻的两端产生感测电压，以获得所述电流采样信号。
可选的，所述功率电路还包括至少一个同步开关管，所述同步开关管的导通和关断与所述主开关管相反。
可选的，所述控制电路还包括：电流反向比较器，当所述开关变换器以突发模式工作时，所述电流反向比较器监控电流流过所述同步开关管，以在电感电流经历电流反向时关断所述同步开关管。
可选的，所述开关变换器是降压变换器、升压变换器或者降压-升压变换器。
可选的，所述开关变换器是同步开关变换器或非同步开关变换器。
本发明实施例的开关变换器包括与控制电路连接的调节电路，该调节电路在开关变换器以突发模式工作时根据输入电压和输出电压自适应调节开关变换器的最小峰值电感电流，从而可以在输入电压和输出电压波动导致占空比变化时抵消掉占空比对电感电流的影响，从而使得突发模式下控制的电流保持相对恒定，有效降低了输出纹波和噪声。
附图说明
通过以下参照附图对本发明实施例的描述，本发明的上述以及其他目的、特征和优点将更为清楚。
图1示出传统的具有突发模式的升降压开关变换器的示意性电路图；
图2示出传统的具有突发模式的升降压开关变换器的原理示意图；
图3示出图1中的突发比较器的工作原理图；
图4示出电流采样信号与斜坡补偿信号的叠加原理图；
图5示出根据本发明第一实施例的一种开关变换器的示意性电路图；
图6示出根据本发明第一实施例的另一种开关变换器的示意性电路图；
图7示出根据本发明第二实施例的一种开关变换器的示意性电路图；
图8示出根据本发明第二实施例的另一种开关变换器的示意性电路图。
具体实施方式
以下将参照附图更详细地描述本发明。在各个附图中，相同的元件采用类似的附图标记来表示。为了清楚起见，附图中的各个部分没有按比例绘制。此外，在图中可能未示出某些公知的部分。
在下文中描述了本发明的许多特定的细节，例如部件的结构、材料、尺寸、处理工艺和技术，以便更清楚地理解本发明。但正如本领域的技术人员能够理解的那样，可以不按照这些特定的细节来实现本发明。
应当理解，在以下的描述中，“电路”是指由至少一个元件或子电路通过电气连接或电磁连接构成的导电回路。当称元件或电路“连接到”另一元件或称元件/电路“连接在”两个节点之间时，它可以直接耦合或连接到另一元件或者可以存在中间元件，元件之间的连接可以是物理上的、逻辑上的、或者其结合。相反，当称元件“直接耦合到”或“直接连接到”另一元件时，意味着两者不存在中间元件。
在本申请的上下文中，当晶体管处于“断开(off)状态”或“断开” 时，晶体管阻挡电流和/或基本不传导电流。相反，当晶体管从处于“导通(on)状态”或“导通”时，晶体管能够显著地传导电流。举例来说，在一个实施例中，高压晶体管包括N沟道金属氧化物半导体(NMOS)场效应晶体管(FET)，其中高压被提供在晶体管的第一端子(即漏极)和第二端子(即源极)之间。在一些实施例中，当调节提供到负载的能量时，可以使用集成控制器电路来驱动功率开关。另外，出于本公开内容的目的，本申请中的“接地”或“接地电势”是指如下参考电压或电势，相对于参考电压或电势来定义或测量电子电路或集成电路(IC)的所有其他电压或电势。
图1示出传统的具有突发模式的升降压开关变换器，如图1所示，开关变换器100的功率电路通过降压-升压拓扑实现，包括晶体管M1-M4以及电感L，晶体管M1、电感L以及晶体管M4依次连接在开关变换器的输入端和输出端之间，晶体管M2连接在晶体管M1和电感L的中间节点和地之间，晶体管M3连接在晶体管M4和电感L的中间节点和地之间。其中，晶体管M1、M2以及电感L构成了开关变换器中的降压电路；晶体管M3、M4以及电感L构成了开关变换器中的升压电路。
控制电路110用来控制功率电路中的晶体管M1-M4的导通和关断，以控制电感以不连续的脉冲输出能量。控制电路110包括振荡器101、逻辑电路102、电流放大器103以及电流反向比较器104。振荡器101用于为电路提供开关定时的内部时钟，同时产生出锯齿波，提供给PWM比较器和为斜坡补偿电路提供补偿信号。逻辑电路102用于实现系统的逻辑控制功能，对控制晶体管M1-M4的工作状态的各个模块的逻辑信号进行处理，产生开关驱动信号提供给晶体管M1-M4。逻辑电路102可以包括脉宽调制器(PWM)电路或任何其他能够控制功率开关M1-M4的占空比的适当电路。
当输入电压Vin高于输出电压Vout时，开关变换器100工作在降压模式下，晶体管M3保持关断，晶体管M4保持导通，晶体管M1和M2交替导通和关断。其中，晶体管M1又被称为主开关管，晶体管M2又被称为同步开关管，当主开关管M1导通时，电流放大器103通过采样 管M0间接对主开关管M1进行采样，电感电流流过采样电阻Rs，并且在其两端产生感测电压，该感测电压大约等于电感电流与采样电阻值的乘积，然后通过电流放大器103放大该电压以得到电流采样信号。当电流采样信号与斜坡补偿信号的叠加信号Vsum超过电流比较器115的正相输入端的电压时(即，第一钳位模块112输出的钳位电压Vc1)，电流比较器115提供第一复位信号RST1给逻辑电路102，以控制主开关管M1关断和同步开关管M2导通。
同理，当输入电压Vin低于输出电压Vout时，开关变换器100工作在升压模式下，晶体管M1保持导通，晶体管M2保持关断，晶体管M3和M4交替导通和关断。同样的，晶体管M3又被称为主开关管，晶体管M4又被称为同步开关管，当主开关管M3导通时，电流比较器116将电流采样信号与斜坡补偿信号的叠加信号Vsum与正相输入端的电压进行比较(即，第二钳位模块114输出的钳位电压Vc2)，当叠加信号Vsum高于正相输入端的电压时，电流比较器116提供第二复位信号RST2至逻辑电路102，以控制主开关管M3关断和同步开关管M4导通。
进一步的，由电平转换模块113对第一钳位模块112输出的钳位电压Vc1进行转换，并将转换后的电压信号提供至第二钳位模块114，第二钳位模块114根据接收的电压信号生成钳位电压Vc2。
图1中的开关变换器具有连续工作模式和突发模式这两种模式。在连续工作模式下，开关变换器可以减少噪音、RF干扰和输出电压的脉动。在突发模式下，电路中的主开关管和同步开关管在轻负载下间歇导通，通过在轻负载下减少电路的开关损耗和静态电流，可以得到更高的效率。
在突发模式下，电流反向比较器104被激活，并且电感电流不允许称为负的。所述电流反向比较器104监控电感电流流过同步开关管M4和提供信号给逻辑电路102，以在电流经历电流反向状态时切断同步开关管M2或M4。
在突发模式期间，第一钳位模块112和第二钳位模块114得到固定的钳位电压Vc1和Vc2，从而设定最小峰值电感电流水平，然后通过突发比较器117监控误差信号，以确定何时激活和禁止突发模式。
参考图2，传统的开关变换器100在突发模式下利用突发模式控制电路120钳位最小的钳位电压Vc来给电感L充入最小的峰值电流，当负载电流小于最小峰值电流，输出电压Vout会被设定的最小峰值电流充上去，当反馈电压VFB高于基准电压VREF0时，二者之间的误差信号Vea会下降，当误差信号Vea小于阈值电压VTH时，触发突发模式Burst将系统关闭，直到负载电流增大到某一值时，再解除突发模式，接通所有电路，并恢复正常工作。
传统的开关变换器100的缺陷是突发模式的钳位电压是开关变换器内部固定的。由于突发钳位在每个开关循环期间固定最小峰值电感电流，因此输出电压的纹波也是固定的。对于较高的突发钳位电压，在轻负载下的效率最高，但是代价是较大的输出电压纹波。对于较低的突发钳位电压，所述输出电压的纹波较小，但是其对于轻负载的效率又会下降。
此外，参考图3和图4，传统的开关变换器100中的电流比较器将电流采样信号Isen与斜坡补偿信号Vsaw的叠加信号与钳位电压Vc进行比较来确定何时关断主开关管，而斜坡电压分量是占空比D的一个函数，由此可以得到：
Vc＝Isen+Vslope(D)
而在开关变换器100处于降压模式的突发模式下时Vc＝VREF1；当开关变换器100处于升压模式的突发模式下时Vc＝VREF2。由此可以得到：
Isen＝VREF1(VREF2)-Vslope(D)
其中，电流采样信号Isen即代表了电感峰值电流，由上式可以看出，在相同的钳位电压Vc下，开关的占空比越小，斜坡电压分量越小，比较器翻转对应的电流Isen就越大，导致输出电压的纹波越大。所以，当钳位电压Vc固定时，传统的开关变换器100在突发模式下的开关占空比D越小，输出电压的纹波就越大，因此传统的开关变换器100无法适用于小占空比的应用环境。
图5示出根据本发明第一实施例的一种开关变换器的示意性电路图。图5中的开关变换器200包括控制电路210以及外部的功率电路。其中， 功率电路包括一个或多个开关和滤波器元件(例如，电感和电容等)，所述一个或多个开关和滤波器被配置为响应于来自控制电路210的一个或多个开关驱动信号来调节开关变换器输入端至输出端的电能传输。在一些实施例中，将功率电路中的一个或多个开关与控制电路210集成在一起以形成集成电路芯片。
开关变换器200可以按照功率电路的拓扑分类，可以将开关变换器分为降压型(buck)变换器、升压型(boost)变换器、反激型(flyback)变换器和降压-升压型(buck-boost)变换器。在下文中以降压型变换器为例进行说明。
根据本实施例的教导，开关变换器200可以在多个模式下运行和在多个模式下转变，所述模式包括但不限于连续工作模式和突发模式。在连续工作模式下，开关变换器可以减少噪音、RF干扰和输出电压的脉动。在突发模式下，功率电路中的至少一个开关根据需要在短持续时间内以固定频率进行开关，以维持输出电压水平。开关的这些短持续时间可以被称为“突发导通(burst on)”时段或突发导通间隔。在短持续时间之间，可以禁止开关并且可以暂时禁用控制器中的部分电路以减少功率损耗。禁止开关的这些持续时间可以被称为“突发断开(burst off)”时段或突发断开间隔。以此方式进行开关控制通常被称为“突发模式”和/或“突发模式”。通过在轻负载下间歇导通开关，来减少轻负载下电路的开关损耗和静态电流，可以得到更高的效率。
在连续工作模式下工作时，振荡器211或者任何其他能够为电路提供开关定时的合适设备(即，通过在恒频下生成窄脉冲)提供了开关定时，在每个周期的循环开始时，该振荡器211向逻辑电路212提供窄脉冲的时钟信号SET，逻辑电路212根据接收到的时钟信号SET使得主开关管M1导通而同步开关管M2关断。逻辑电路212可以包括脉宽调制器(PWM)电路或任何其他能够控制功率电路中的主开关管M1的占空比(即，主开关管M1导通的时间量与开关周期的时间作比较)的适当电路。这使得电感221两端的电压大约为Vin-Vout，通过电感221的电流线性增加，并且较大量的电流被输送给电容223和负载224。当主开 关管M1导通时，通过采样管M0间接采样流过主开关管M1的电感电流，电感电流流过采样电阻Rs，并且在其两端产生感测电压，该电压大约等于电感电流和采样电阻值的乘积。然后通过电流放大器219放大该电压，以得到电流采样信号Isen。当电流采样信号Isen与斜坡补偿信号Vsaw的叠加信号Vsum超过电流比较器213的正相输入端的电压时(即，突发钳位模块217的输出)，电流比较器213提供复位信号RST给逻辑电路212，以使得主开关管M1关断和同步开关管M2导通。这使得电感221两端的电压改变为-Vout，导致电感电流减弱，直到下一个时钟信号SET再次使得主开关管M1导通而同步开关管M2断开。
应当指出，尽管在本实施例中将MOSFET用于开关元件，在不偏离本发明原理的前提下，可以使用任何其他类型的合适开关元件。此外，本实施例虽然以同步开关变换器进行说明，但是，本发明不以此为限制，本发明同样适用于非同步开关变换器，本领域技术人员也可以采用整流二极管代替上述实施例中的同步开关管M2。
在突发模式期间，突发钳位模块217将钳位电压Vc1固定在设定的参考电压上，从而设定最小峰值电感电流水平。具体为，突发钳位模块217将误差信号Vea与参考电压VREF1进行比较，当误差信号Vea小于参考电压VREF1时，钳位电压Vc1等于参考电压VREF1；当误差信号Vea大于等于参考电压VREF1时，钳位电压Vc1等于误差信号Vea。然后通过突发比较器214监控误差信号Vea，以确定何时激活和禁止突发模式。当输出负载电流小于某个值时，输出电压会被设定的最小峰值电感电流充上去，然后反馈电压VFB大于基准电压VREF0，误差信号Vea会降低。当误差信号Vea小于突发阈值电压VTH时，突发比较器214输出逻辑高电平，激活突发模式，使得主开关管M1、同步开关管M2以及余下的电路的预定部件均被切断，以减少功率消耗。此时，负载电流完全由输出电容223输送，当输出电压下降时，误差信号Vea上的电压增加到由突发比较器214的滞后设定的水平之上，使得突发比较器214输出逻辑低电平，解除突发模式，接通所有电路，并且恢复正常工作。
在突发模式工作时，电流反向比较器215被激活，并且电感电流不 允许成为负的，所述电流反向比较器215监控电流流过同步开关管M2和提供信号至逻辑电路212，以在电感电流经历电流反向状态时切断同步开关管M2。在图5的示例中，电流反向状态表示当电流通过零电流水平。不过，本领域技术人员可以理解，本发明的范围包括其他电流反向状态，例如，表示电流接近或已跨过零电流水平。例如，可以在电流反向比较器215的反相输入端施加电压偏移，以便电流反向比较器215恰好在电感或开关电流跨过零电流阈值之前关断同步开关管M2。
本实施例的开关变换器200还包括调节电路230，调节电路230用于根据输入电压Vin和输出电压Vout之间的函数关系对突发钳位模块217的参考电压进行补偿，从而抵消掉占空比在突发模式下的影响，让突发模式控制的电流始终保持恒定，从而降低了开关变换器在突发模式下的输出纹波和噪声。
具体地，如图5所示，调节电路230包括电压跟随器231、跨导放大器232以及连接在二者之间的电阻R2。其中，跨导放大器232的两个输入端分别接收输入电压Vin和输出电压Vout，用于将输入电压Vin和输出电压Vout之间的电压差转换成电流信息，该电流被施加到电阻R2的一端上。电阻R2的另一端与电压跟随器231的输出端连接，电压跟随器231通过运算放大器实现，其一个输入端与输出端连接，根据运算放大器的虚短的原理，其输出端的电压等于输入端的电压，因此电阻R2的另一端被施加了与参考电压VREF1相等的电压，从而可以得到补偿之后的参考电压大约等于参考电压VREF1与跨导放大器232输出的电流信息和电阻R2的乘积之间的差值，即：
VREF1′＝VREF1-(Vin-Vout)×R2/R1
其中，R1表示跨导放大器232内部的电阻值，R2表示电阻R2的电阻值。
从上述公式可以看出，本实施例的开关变换器200中补偿后的参考电压中携带了输入电压和输出电压之间的电压差信息，当输入电压和输出电压波动导致占空比变化时，补偿后的参考电压可以抵消掉占空比的影响，从而使得突发模式下控制的电流保持相对恒定，有效降低了输出 纹波和噪声。
图6示出根据本发明第一实施例的另一种开关变换器的示意性电路图。与图5不同，图6示出了一种非同步开关变换器300，其与同步开关变换器的差别在于，同步开关管M2被整流二极管D1取代，整流二极管D1可以防止输出电容323对地放电。除此之外，图6中的其余元件大体上执行与上文图5中所述的相应元件相同的目的。
应当指出，尽管图5和图6各自示出了降压同步开关变换器和降压非同步开关变换器的实施例，本发明并不局限于这些实施例。本发明的优点同样适用于其他类型的变换器，如升压同步开关变换器、升压非同步开关变换器、降压-升压开关变换器等其他合适类型的变换器。
图7示出根据本发明第二实施例的一种开关变换器的示意性电路图。图7示出了一种升压同步开关变换器，其采用了与图5所示降压型变换器中存在的很多相同的元件。图7所示的升压型变换器的控制电路410按照如下方式工作。
当电路状态导致主开关管M1导通且同步开关管M2关断时，输入电压Vin被施加到电感421两端，在该充电阶段，电流开始流过电感421，同时同步开关管M2阻止输出电容423对地放电，并且输出电容423负责向负载424输送电流。
一旦主开关管M1关断和同步开关管M2导通，输出电容423通过储存在电感421中的能量充电。此时，额外的电流开始流过负载，从而导致输出电压Vout上升。在经过一定的时间周期后，主开关管M1再次导通，该循环再次重复，保持所需的输出电压水平，并且根据需要向负载输送所需的电流。
图7中的电路部件的其余部分如先前图5所示的电路部件说明的那样运行。电流比较器413将来自电流放大器419的电流采样信号Isen和斜坡补偿信号Vsaw的叠加信号Vsum与误差放大器416输出的误差信号Vea做比较，根据比较结果提供复位信号RST给逻辑电路412，以便确定何时关断主开关管M1和导通同步开关管M2。
同样的，图7的开关变换器400也可以在多个模式下运行和在多个 模式下转变，所述模式包括但不限于连续工作模式和突发模式。在突发模式期间，突发钳位模块417将钳位电压Vc2固定在设定的参考电压上，从而设定最小峰值电感电流水平。具体的，突发钳位模块417将误差信号Vea与经调节电路430补偿后的参考电压VREF2’进行比较，当误差信号Vea小于补偿后的参考电压VREF2’时，钳位电压Vc2等于参考电压VREF2’；当误差信号Vea大于等于参考电压VREF2’时，钳位电压Vc1等于误差信号Vea。然后通过突发比较器414监控误差信号Vea，以确定何时激活和禁止突发模式。当输出负载电流小于某个值时，输出电压会被设定的最小峰值电感电流充上去，然后反馈电压VFB大于基准电压VREF0，误差信号Vea会降低。当误差信号Vea小于突发阈值电压VTH时，突发比较器414输出逻辑高电平，激活突发模式，使得主开关管M1、同步开关管M2以及余下的电路的预定部件均被切断，以减少功率消耗。此时，负载电流完全由输出电容423输送，当输出电压下降时，误差信号Vea上的电压增加到由突发比较器414的滞后设定的水平之上，使得突发比较器214输出逻辑低电平，解除突发模式，接通所有电路，并且恢复正常工作。
另外，在突发模式时，调节电路430根据输入电压Vin和输出电压Vout之间的函数关系对参考电压VREF2进行补偿，以抵消掉占空比在突发模式下的影响，让突发模式控制的电流始终保持恒定，以降低开关变换器在突发模式下的输出纹波和噪声。
同样的，调节电路430包括电压跟随器441、跨导放大器442以及连接在二者之间的电阻R2。其中，跨导放大器442的两个输入端分别接收输入电压Vin和输出电压Vout，用于将输入电压Vin和输出电压Vout之间的电压差转换成电流信息，该电流被施加到电阻R2的一端上。电阻R2的另一端与电压跟随器441的输出端连接，电压跟随器441通过运算放大器实现，其一个输入端与参考电压VREF2连接，另一个输入端与输出端连接，根据运算放大器的虚短的原理，其输出端的电压等于输入端的电压，因此电阻R2的另一端被施加了与参考电压VREF2相等的电压，从而可以得到补偿之后的参考电压大约等于参考电压VREF2 与跨导放大器442输出的电流信息和电阻R2的乘积之间的差值，即：
VREF2′＝VREF2-(Vout-Vin)×R2/R1
其中，R1表示跨导放大器442内部的电阻值，R2表示电阻R2的电阻值。
从上述公式可以看出，本实施例的开关变换器400中补偿后的参考电压中携带了输入电压和输出电压之间的电压差信息，通过这种方式，可以根据输入电压和输出电压调节最小峰值电感电流水平，从而可以在输入电压和输出电压波动导致占空比变化时抵消掉占空比对电感电流的影响，从而使得突发模式下控制的电流保持相对恒定，有效降低了输出纹波和噪声。
图8示出根据本发明第二实施例的另一种开关变换器的示意性电路图。与图7不同，图8示出了一种非同步开关变换器500，其与同步开关变换器的差别在于，同步开关管M2被整流二极管D1取代，整流二极管D1可以防止输出电容523对地放电。除此之外，图8中的其余元件大体上执行与上文图7中所述的相应元件相同的目的。
在上述实施例中，尽管结合图5至图8分别描述了降压型和升压型拓扑结构的开关变换器，然而，可以理解，本发明实施例的带突发模式和突发补偿功能的控制电路也可以用于其他拓扑结构的开关变换器中，包括但不限于降压型、升压型、升降压型、非逆变升降压型、正激型、反激型等拓扑结构。
综上所述，本发明实施例的开关变换器包括与控制电路连接的调节电路，该调节电路在开关变换器以突发模式工作时根据输入电压和输出电压自适应调节开关变换器的最小峰值电感电流，从而可以在输入电压和输出电压波动导致占空比变化时抵消掉占空比对电感电流的影响，从而使得突发模式下控制的电流保持相对恒定，有效降低了输出纹波和噪声。
应当说明的是，在本文中，诸如第一和第二等之类的关系术语仅仅用来将一个实体或者操作与另一个实体或操作区分开来，而不一定要求或者暗示这些实体或操作之间存在任何这种实际的关系或者顺序。而且， 术语“包括”、“包含”或者其任何其他变体意在涵盖非排他性的包含，从而使得包括一系列要素的过程、方法、物品或者设备不仅包括那些要素，而且还包括没有明确列出的其他要素，或者是还包括为这种过程、方法、物品或者设备所固有的要素。在没有更多限制的情况下，由语句“包括一个……”限定的要素，并不排除在包括所述要素的过程、方法、物品或者设备中还存在另外的相同要素。
依照本发明的实施例如上文所述，这些实施例并没有详尽叙述所有的细节，也不限制该发明仅为所述的具体实施例。显然，根据以上描述，可作很多的修改和变化。本说明书选取并具体描述这些实施例，是为了更好地解释本发明的原理和实际应用，从而使所属技术领域技术人员能很好地利用本发明以及在本发明基础上的修改使用。本发明仅受权利要求书及其全部范围和等效物的限制。

Claims (12)

  1. 一种开关变换器，包括：
    输入端，接收输入电压；
    连接到负载的输出端，提供输出电压；
    与所述输入端和输出端连接的功率电路，所述功率电路采用电感以及至少一个主开关管调节提供给所述负载的电流；
    与所述功率电路连接的控制电路，所述控制电路用于根据接收的参考电压控制所述至少一个主开关管的导通和关断，以便控制流经所述电感的最小峰值电感电流；以及
    与所述控制电路连接的调节电路，提供所述参考电压至所述控制电路，当所述开关变换器以突发模式工作时，所述调节电路用于根据所述输入电压和所述输出电压自适应调节所述参考电压。
  2. 根据权利要求1所述的开关变换器，其中，所述调节电路调节所述参考电压以使得所述参考电压与所述输出电压和所述输入电压之间的差值正相关。
  3. 根据权利要求1所述的开关变换器，其中，所述控制电路包括：
    逻辑电路，用于控制所述至少一个主开关管的导通和关断；
    振荡器，用于向所述逻辑电路提供时钟信号，所述时钟信号用于控制所述至少一个主开关管的导通时刻。
  4. 根据权利要求3所述的开关变换器，其中，所述控制电路还包括：
    误差放大器，用于将所述输出电压的反馈电压与基准电压进行比较，以获得误差信号，所述误差放大器根据负载状态调节所述误差信号；
    突发钳位模块，用于将所述误差信号与参考电压进行比较，根据比较结果设定表征最小峰值电感电流的钳位电压；以及
    电流比较器，用于将电流采样信号和斜坡补偿信号的叠加信号与所述钳位电压进行比较，根据比较结果控制所述至少一个开关管的关断时刻。
  5. 根据权利要求4所述的开关变换器，其中，所述控制电路还包括：
    突发比较器，用于将所述误差信号与设定的突发阈值电压进行比较，并在所述误差信号小于所述突发阈值电压时，控制所述开关变换器进入突发模式。
  6. 根据权利要求4所述的开关变换器，其中，所述突发钳位模块用于在所述误差信号小于所述参考电压时，设置所述钳位电压等于所述参考电压，并在所述误差信号大于等于所述参考电压时，设置所述钳位电压等于所述误差信号。
  7. 根据权利要求4所述的开关变换器，其中，所述调节电路包括：
    跨导放大器，其两个输入端分别接收所述输入电压和输出电压，用于将二者之间的电压差转换成电流信息；
    第一电阻，其第一端与所述跨导放大器的输出端连接；以及
    电压跟随器，其一个输入端用于接收所述参考电压，另一输入端和输出端与所述第一电阻的第二端连接，
    其中，所述跨导放大器与所述第一电阻的连接节点用于将补偿后的参考电压提供至所述突发钳位模块。
  8. 根据权利要求4所述的开关变换器，其中，所述控制电路还包括：
    采样管和采样电阻，用于采样流过所述至少一个主开关管的电感电流；以及
    电流放大器，通过在所述采样电阻的两端产生感测电压，以获得所述电流采样信号。
  9. 根据权利要求3所述的开关变换器，其中，所述功率电路还包括至少一个同步开关管，所述同步开关管的导通和关断与所述主开关管相反。
  10. 根据权利要求9所述的开关变换器，其中，所述控制电路还包括：
    电流反向比较器，当所述开关变换器以突发模式工作时，所述电流反向比较器监控电流流过所述同步开关管，以在电感电流经历电流反向时关断所述同步开关管。
  11. 根据权利要求1所述的开关变换器，其中，所述开关变换器是 降压变换器、升压变换器或者降压-升压变换器。
  12. 根据权利要求1所述的开关变换器，其中，所述开关变换器是同步开关变换器或非同步开关变换器。
PCT/CN2022/113213 2021-07-22 2022-08-18 开关变换器 WO2023001315A1 (zh)

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