WO2022205975A1 - 偏置电路及射频功率放大器 - Google Patents

偏置电路及射频功率放大器 Download PDF

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WO2022205975A1
WO2022205975A1 PCT/CN2021/133990 CN2021133990W WO2022205975A1 WO 2022205975 A1 WO2022205975 A1 WO 2022205975A1 CN 2021133990 W CN2021133990 W CN 2021133990W WO 2022205975 A1 WO2022205975 A1 WO 2022205975A1
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Prior art keywords
transistor
radio frequency
resistor
power amplifier
bias circuit
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PCT/CN2021/133990
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English (en)
French (fr)
Inventor
彭振飞
徐柏鸣
苏强
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广州慧智微电子股份有限公司
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Publication of WO2022205975A1 publication Critical patent/WO2022205975A1/zh

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32—Modifications of amplifiers to reduce non-linear distortion
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only

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  • the present application relates to the technical field of integrated circuits, and in particular, to a bias circuit and a radio frequency power amplifier.
  • the New Radio (NR, New Radio) system of the fifth generation mobile communication technology 5G, 5th Generation
  • the data transmission rate has higher requirements, and it is necessary to use a modulation signal with a wider bandwidth to transmit data.
  • N41 2490 megahertz (MHz) to 2690MHz
  • the maximum modulation signal bandwidth is extended from 40MHz to 100MHz, which gives The design of RF power amplifiers in mobile terminals presents new challenges, and how to improve the performance of RF power amplifiers has become an urgent problem to be solved.
  • embodiments of the present application provide a bias circuit and a radio frequency power amplifier.
  • the embodiment of the present application provides a bias circuit, including:
  • control loop configured to output a bias current to the input of the RF power amplifier
  • an adjustment circuit configured to adjust the loop bandwidth of the control loop based on the Miller effect under the condition that it is determined that the bias current output by the control loop can satisfy the first condition, so as to stabilize the control loop performance and noise rejection satisfy the second condition;
  • the first condition represents that the bias current can compensate the base current of the radio frequency power amplifier, so that the base current does not decrease with the increase of the power of the input signal;
  • the second condition indicates that the phase margin of the control loop is greater than or equal to 45 degrees, and the noise suppression degree of the control loop can make the noise of the radio frequency power amplifier less than or equal to a noise threshold.
  • the adjustment circuit includes a Miller unit; the adjustment circuit adjusts the loop bandwidth of the control loop based on the Miller effect of the Miller unit.
  • the Miller unit is connected in parallel with the first transistor of the control loop; wherein,
  • the control loop includes the first transistor, the second transistor and the first resistor; the collector of the first transistor is connected to the base of the second transistor; the emitter of the second transistor is connected to the first transistor; One end of a resistor; the other end of the first resistor is connected to the base of the first transistor; an external current source sends to the control loop through the first node between the first transistor and the second transistor input current; the control loop converts the input current into the bias current, and outputs it to the input end of the radio frequency power amplifier through the emitter of the second transistor.
  • the Miller unit includes a first capacitor.
  • the first capacitor is a variable capacitor.
  • the adjustment circuit further includes a second resistor; the first capacitor and the second resistor are connected in parallel with the first transistor; wherein,
  • the adjustment circuit adjusts the transmission zero point of the control loop through the second resistor, so that the stability of the control loop satisfies the third condition; the transmission zero point is the difference between the first transistor and the first transistor. Introduced by the parallel connection of capacitors; the third condition characterizes that the phase margin of the control loop is greater than or equal to 45 degrees.
  • the first capacitor and the second resistor are connected in parallel with the first transistor through the collector and the base of the first transistor.
  • the adjustment circuit further includes a third resistor; the third resistor is connected in series with the first transistor; wherein,
  • the adjustment circuit isolates the radio frequency signal for the first transistor through the third resistor, so that the radio frequency signal cannot be coupled to the first transistor through the first capacitor and the second resistor; the radio frequency The signal is transmitted to the bias circuit through the input terminal of the RF power amplifier.
  • the third resistor is connected in series with the collector of the first transistor.
  • the bias circuit further includes a fourth resistor; the bias current is output to the input end of the radio frequency power amplifier through the fourth resistor;
  • the bias circuit attenuates the radio frequency signal transmitted through the input end of the radio frequency power amplifier through the fourth resistor; and isolates the first transistor through the first resistor after being attenuated by the fourth resistor the radio frequency signal, so that the radio frequency signal attenuated by the fourth resistor is transmitted to the second transistor;
  • the bias current output by the emitter stage of the second transistor is made to follow the radio frequency through the detection effect of the second transistor. varies with the power of the signal.
  • the first resistor is a variable resistor.
  • the bias circuit further includes a second capacitor; one end of the second capacitor is grounded, and the other end is connected to the base of the second transistor; wherein,
  • the bias circuit short-circuits the radio frequency signal transmitted to the second transistor and attenuated by the fourth resistor to ground through the second capacitor.
  • An embodiment of the present application further provides a radio frequency power amplifier, wherein the input end of the radio frequency power amplifier is provided with the bias circuit described in any of the above solutions.
  • the radio frequency power amplifier is a multi-stage cascaded radio frequency power amplifier;
  • the bias circuit is provided at the input end of each stage.
  • the bias circuit includes: a control loop configured to output a bias current to the input end of the radio frequency power amplifier; and a regulation circuit configured to determine the output of the control loop
  • the loop bandwidth of the control loop is adjusted based on the Miller effect, so that the stability and noise suppression of the control loop satisfy the second condition
  • the The first condition characterizes that the bias current can compensate the base current of the RF power amplifier, so that the base current does not decrease with the increase of the power of the input signal
  • the second condition characterizes the control loop
  • the phase margin of the radio frequency power amplifier is greater than or equal to 45 degrees, and the noise suppression degree of the control loop can make the noise of the radio frequency power amplifier less than or equal to the noise threshold; the input end of the radio frequency power amplifier is provided with the bias circuit .
  • the bias circuit includes a loop structure, which enables the bias circuit to have the characteristics of low baseband impedance.
  • the bias circuit provides a low baseband impedance path for the radio frequency power amplifier.
  • the baseband impedance causes The memory effect of the RF power amplifier can be reduced (that is, it can be suppressed), thereby reducing the deterioration of the adjacent channel power ratio (ACPR, Adjacent Channel Power Ratio) of the RF power amplifier output signal caused by the memory effect, reducing the output of the RF power amplifier. Distortion of the signal, thereby improving the performance of the RF power amplifier.
  • ACPR Adjacent Channel Power Ratio
  • the adjusting circuit to adjust the loop bandwidth of the control loop under the condition that the bias current output by the control loop can satisfy the first condition, so as to make the stability of the control loop and the noise suppression In this way, when the bandwidth of the input signal of the radio frequency power amplifier increases, it is ensured that the bias current can compensate the base current of the radio frequency power amplifier, so that the base current does not change with the increase of the power of the input signal.
  • AM-AM distortion refers to the amplitude distortion of the output signal of the RF power amplifier relative to the input signal
  • the stability and noise suppression of the control loop included in the circuit can reduce the influence of circuit oscillation and noise that may be generated by the bias circuit on the sensitivity of the RF power amplifier when it receives signals, and further improve the performance of the RF power amplifier.
  • FIG. 1 is a schematic diagram of asymmetry in the third-order intermodulation of a radio frequency power amplifier in the related art
  • FIG. 2 is a schematic structural diagram of a bias circuit in the related art
  • FIG. 3 is a schematic structural diagram of another bias circuit in the related art.
  • FIG. 4 is a schematic diagram of the frequency response of the impedance of the bias circuit 300 in the entire frequency band in the related art
  • FIG. 5 is a schematic diagram of the loop gain of the bias circuit 300 in the related art
  • FIG. 6 is a schematic diagram of the phase margin of the bias circuit 300 in the related art.
  • FIG. 7 is a schematic structural diagram of a bias circuit according to an embodiment of the present application.
  • FIG. 8 is another schematic structural diagram of a bias circuit according to an embodiment of the present application.
  • FIG. 9 is a schematic diagram illustrating the change of the impedance of the bias circuit 700 with the parallel capacitor C7 according to the embodiment of the present application.
  • FIG. 10 is a schematic diagram of the variation of the loop gain of the bias circuit 700 with the parallel capacitor C7 according to the embodiment of the present application;
  • FIG. 11 is a schematic diagram illustrating the variation of the phase margin of the bias circuit 700 with the parallel capacitor C7 according to the embodiment of the present application;
  • FIG. 12 is a schematic diagram of the radio frequency signal being coupled to T5 through a parallel capacitor C7 and a parallel resistor R4 according to an embodiment of the present application ;
  • FIG. 13 is a schematic structural diagram of a radio frequency power amplifier according to an embodiment of the present application.
  • RF power amplifiers in mobile terminals usually include input matching circuits, output matching circuits, amplifier tubes, and bias circuits.
  • the impedances of frequency response devices such as inductors and capacitors are only consistent within a limited frequency bandwidth.
  • the impedance response of the RF power amplifier at different frequencies is different, which makes the modulation signal superimpose additional amplitude and phase changes, and the third-order intermodulation component IM3 appears asymmetric, that is, the RF power amplifier appears memory effect.
  • the non-frequency power amplifier based on the The expansion of the linear polynomial can be determined: the third-order nonlinear product of the RF power amplifier includes the third-order current vector I 3 , and I 3 contains 2f 1 -f 2 and 2f 2 -f 1 (that is, the two frequencies corresponding to I 3 can be expressed as 2f 1 -f 2 and 2f 2 -f 1 ), the impedance response of the RF power amplifier at the 2f 1 -f 2 frequency point is basically the same as the impedance response at the 2f 2 -f 1 frequency point, and the 2f 1 -f 2
  • the corresponding current vector is similar to the current vector corresponding to 2f 2 -f 1 ; the contributing parts of the second-order nonlinear products of the RF power amplifier mainly include:
  • the third-order intermodulation component IM3 is the superposition of the above-mentioned current vectors I 1 , I 2 , I 3 and I 4 , and the upper sideband IM3_Up and the lower sideband IM3_Low are asymmetrical, that is, the radio frequency power amplifier has a memory effect.
  • the inter-stage impedance and load impedance are designed in the area of ACPR and efficiency, so as to solve the problem of ACPR deterioration of the output signal caused by nonlinear amplification characteristics of the RF power amplifier in the case of a large signal (ie, a signal with higher power) input.
  • the frequency response difference of the impedance of each node in the RF power amplifier will increase as the modulation signal frequency band becomes wider, showing that The memory effect will be more serious, that is, the third-order intermodulation will have obvious asymmetry, causing the ACPR of the output signal to deteriorate.
  • the linear power of the RF power amplifier is increased by reducing the load impedance when designing a narrow-band RF power amplifier (that is, an RF power amplifier designed for a scenario with a narrow modulation signal bandwidth)
  • the third-order current vector I can only be reduced.
  • the distortion brought by 3 the distortion brought by the second-order current vectors I 1 , I 2 and I 4 is weakly improved, which makes the ACPR improvement effect not good, and the power consumption of the RF power amplifier easily becomes unacceptable (such as power consumption). over the budget).
  • a bias network with low baseband impedance can be used to implement a low impedance at the baseband frequency at the input node of the radio frequency power amplifier to suppress the nonlinear contributions of the current vectors I1 and I2 .
  • the RF power amplifier includes: an input matching capacitor C 1 , an amplifying transistor T 1 , output matching inductors L 1 , L 2 and L 3 , capacitors C 2 , C 3 and C 4 , and a bias circuit 200 .
  • the radio frequency signal enters T1 through RFin and C1 for amplification, and then outputs to Rfout through the output matching circuit (including L1, L2, L3 , C2 , C3 and C4 ) .
  • the bias circuit 200 includes an isolation resistor R 1 , transistors T 2 , T 3 and T 4 and a capacitor C 5 ; the bias circuit 200 converts the current Ib1 input by the external current source into a bias current I em3 and outputs it through R 1 To the input end of the RF power amplifier, at this time, the impedance Zin1 of the node A in the bias circuit 200 (ie, the baseband impedance of the bias circuit 200 ) can be expressed by the following formula:
  • Re3 represents the emitter path resistance of T3, which can be calculated by the following formula:
  • VT represents the thermal voltage of T3
  • I em3 represents the bias current of T3.
  • the value of R 1 is 15 ohms (Ohm)
  • VT is about 26 millivolts (mV) at room temperature
  • the bias current of transistor T 1 is 100 milliamps (mA)
  • I em3 is about 0.6 mA
  • Re3 is about 43ohm (ie 26mV/0.6mA)
  • Zin1 is about 58ohm (ie 43ohm+15Ohm). It can be seen that the baseband impedance of the bias circuit 200 is relatively high, which cannot effectively improve the ACPR deterioration of the output signal of the radio frequency power amplifier.
  • the bias circuit 200 may be improved.
  • the bias circuit 300 includes transistors T 5 and T 6 , isolation resistors R 2 and R 3 , and a capacitor C 6 .
  • the bias circuit 300 converts the current Ib1 input by the external current source into a suitable quiescent bias current I em6 and outputs it to the input terminal of the RF power amplifier through the resistor R 2 .
  • the bias circuit 300 blocks most of the radio frequency signal from leaking from the bias circuit through R 2 (that is, attenuates the radio frequency signal transmitted through the input end of the radio frequency power amplifier, R 2 You can take a small value, such as 15Ohm ), and isolate the RF signal after being attenuated by R2 through the base low resistance point where R3 is T5 ( in order to improve the isolation effect of the RF signal, the value of R3 is more critical , usually need to take a larger value, such as 2000Ohm), so that the RF signal attenuated by R 2 is transmitted to T 6 , and through the RF signal attenuated by R 2 , if the RF signal is a large signal (that is, the power is larger signal), the emitter of T 6 will have a certain RF swing, and the detection effect of the RF swing through the BE junction of T 6 reduces the bias voltage of the BE junction, thereby increasing the I em6 output by the T 6 emit
  • C6 can take a small value, such as 3 picofarads (pF)
  • PPF picofarads
  • the bias circuit 300 passes C6 to transmit to T6 by R2
  • the attenuated RF signal is shorted to ground.
  • the impedance Zin2 of the node A in the bias circuit 300 (that is, the baseband impedance of the bias circuit 300 ) can be expressed by the following formula:
  • ⁇ represents the angular frequency
  • Cbe5 represents the base capacitance of T5
  • Cbe6 represents the base capacitance of T6
  • ⁇ represents the current amplification factor of the transistor, usually 80-150
  • g m5 and g m6 represent T5 respectively and the transconductance of T6 .
  • formula (3) can be simplified as the following formula:
  • the baseband impedance of the bias circuit 300 is only 15 Ohm, while the baseband impedance of the bias circuit 200 is 58 Ohm. It can be seen that, compared with the bias circuit 200, the baseband impedance of the bias circuit 300 is further reduced, so that the problem of ACPR deterioration of the output signal of the radio frequency power amplifier can be further improved.
  • the bias circuit 300 can exhibit a very low baseband impedance characteristic within a baseband signal bandwidth of 100 MHz. Since T1 is equivalent to being connected in parallel with node A , when the baseband frequency is within 100MHz, the impedance of node A (that is, the baseband impedance Zin2 of the bias circuit 300 ) is usually in the order of 15Ohm , while the impedance of T1 (can be The magnitude expressed as Zin_T 1 ) is usually around 40 to 50 Ohm. Therefore, most of the second-order nonlinear current vectors I1 and I2 generated by T1 at the input node D as shown in FIG.
  • the bias circuit 300 has the following problems:
  • the loop stability is poor.
  • the loop gain of the bias circuit 300 is shown in FIG. 5
  • the phase margin of the bias circuit 300 is shown in FIG. 6
  • R 3 needs to take a larger value to effectively Isolate the RF signal attenuated by R 2 for the base low resistance point of T 5 (if the value of R 3 is small, the isolation effect of R 3 on the RF signal will become worse, and the detection compensation effect of T 6 will become weaker
  • the RF power amplifier will have obvious AM-AM distortion under the large signal, which will cause the ACPR of the RF power amplifier to deteriorate significantly), and the larger the value of R3 , the resistance-capacitance (RC, Resistor-Capacitance) formed by R3 and Cbe5 .
  • RC Resistor-Capacitance
  • the loop gain of the bias circuit 300 is 0, the phase of the signal fed back to node B will change by -135 degrees compared to the signal input to node C; If the signal phase of C changes by -180 degrees, the loop of the bias circuit 300 will form a positive feedback, which will cause the bias circuit 300 to oscillate; although -135 degrees has a margin of 45 degrees compared to -180 degrees, However, under the conditions of high and low temperature, high and low voltage, or process fluctuations, the change of the signal phase of node B compared to the signal phase of node C can easily exceed the margin of 45 degrees; in other words, the loop of the bias circuit 300 is Stability is poor and there is a risk of circuit oscillation.
  • the noise suppression is poor. Specifically, in the 5G NR system, the baseband bandwidth of the bias circuit 300 is wider, and R3 needs to take a larger value, which makes the noise (that is, thermal noise) generated by R3 in the passband of the bias circuit 300.
  • the noise current is amplified by T5 and fed into node B, and then superimposed to the bias current I em6 through the base of T6 . It can be seen that the noise current will be non-linearly mixed to both ends of the RF signal through T 1 , which will lead to the deterioration of the noise in the receiving frequency band of the RF power amplifier, so that the RF power amplifier cannot meet the requirements of the relevant communication protocol for signal reception sensitivity.
  • the bias circuit includes a control loop and an adjustment circuit, and the bias circuit uses the control loop to output a bias current to the input end of the radio frequency power amplifier, since the bias circuit includes a loop structure , which makes the bias circuit have the characteristics of low baseband impedance.
  • the bias circuit provides a low baseband impedance path for the RF power amplifier, so that the memory effect of the RF power amplifier caused by the baseband impedance can be reduced (that is, it can be obtained. Therefore, the deterioration of the ACPR of the output signal of the radio frequency power amplifier caused by the memory effect can be reduced, the distortion of the output signal of the radio frequency power amplifier can be reduced, and the performance of the radio frequency power amplifier can be improved.
  • the bias circuit uses the adjustment circuit to adjust the loop bandwidth of the control loop under the condition that the bias current output by the control loop can satisfy the first condition, so that the stability of the control loop and the The noise suppression degree satisfies the second condition;
  • the first condition indicates that the bias current can compensate the base current of the radio frequency power amplifier, so that the base current does not decrease with the increase of the power of the input signal;
  • the The second condition indicates that the phase margin of the control loop is greater than or equal to 45 degrees, and the noise suppression degree of the control loop can make the noise of the radio frequency power amplifier less than or equal to the noise threshold;
  • the bandwidth of the input signal increases, it is ensured that the bias current can compensate the base current of the RF power amplifier, so that the base current does not decrease with the increase of the power of the input signal, thereby suppressing the AM-AM distortion of the RF power amplifier
  • it can take into account the stability and noise suppression of the control loop included in the bias circuit, reduce the influence of circuit oscillation and noise that may be generated by the bias
  • the bias circuit 700 includes:
  • control loop 701 configured to output a bias current to the input end of the radio frequency power amplifier
  • the adjustment circuit 702 is configured to adjust the loop bandwidth of the control loop 701 based on the Miller effect under the condition that it is determined that the bias current output by the control loop 701 can satisfy the first condition, so that the control loop The stability and noise suppression of road 701 satisfy the second condition;
  • the first condition indicates that the bias current can compensate the base current of the radio frequency power amplifier, so that the base current does not decrease as the power of the input signal increases;
  • the second condition indicates that the phase margin of the control loop 701 is greater than or equal to 45 degrees, and the noise suppression degree of the control loop 701 can make the noise of the radio frequency power amplifier less than or equal to the noise threshold.
  • control loop 701 is equivalent to the function of the loop (including T 5 , T 6 and R 3 ) of the bias circuit 300 shown in FIG. 3 .
  • the value can be set according to the maximum working bandwidth of the circuit, in other words, the maximum working bandwidth of the control loop 701 is adjusted.
  • the radio frequency power amplifier will cause the baseband impedance to cause The greater the value of the maximum operating bandwidth, the wider the loop bandwidth of the control loop 701, the worse the stability and noise suppression of the control loop 701, and the radio frequency The better the power amplifier is to suppress the memory effect caused by the baseband impedance.
  • the determining that the bias current output by the control loop 701 can satisfy the first condition can be understood as ensuring that the determining that the bias current output by the control loop 701 can satisfy the first condition, that is, In other words, under the condition that it is ensured that the bias current output by the control loop 701 can satisfy the first condition, the adjustment circuit 702 adjusts the loop bandwidth of the control loop 701 based on the Miller effect, so that the The stability and noise suppression of the control loop 701 satisfy the second condition.
  • the user can specifically set the second condition as required, so that the bias
  • the circuit 700 can take into account baseband impedance bandwidth, noise rejection, circuit stability, and radio frequency performance.
  • the phase margin of the control loop 701 may be set to be greater than or equal to 60 degrees, and the noise threshold may be set according to the requirements of the communication protocol in the related art on the signal receiving sensitivity of the radio frequency power amplifier.
  • the protocol requires that the signal receiving sensitivity of the RF power amplifier is less than or equal to -98 decibel millimetres under the 10MHz bandwidth.
  • the second condition may specifically include: the phase margin of the control loop 701 is greater than or equal to 60 degrees, and the noise suppression degree of the control loop 701 can make the noise of the radio frequency power amplifier less than or equal to Equal to -135dBm/Hz.
  • the adjustment circuit 702 may include a Miller unit; the adjustment circuit 702 adjusts the loop bandwidth of the control loop 701 based on the Miller effect of the Miller unit.
  • the Miller unit can be connected in parallel with the first transistor of the control loop 701; wherein,
  • the control loop 701 may include the first transistor (ie, T 5 in FIG. 3 ), a second transistor (ie, T 6 in FIG. 3 ), and a first resistor (ie, R 3 in FIG. 3 );
  • the collector of the first transistor is connected to the base of the second transistor;
  • the emitter of the second transistor is connected to one end of the first resistor;
  • the other end of the first resistor is connected to the base of the first transistor pole;
  • the external current source inputs current to the control loop 701 through the first node between the first transistor and the second transistor (ie, node B in FIG. 3 );
  • the control loop 701 will input current
  • the current is converted into the bias current (ie, I em6 in FIG. 3 ), and is output to the input terminal of the radio frequency power amplifier through the emitter of the second transistor.
  • the Miller unit may include capacitors and/or transistors.
  • the Miller unit includes a first capacitor.
  • the first capacitor may be a variable capacitor (such as a voltage-controlled variable capacitor); or, the first capacitor may include a plurality of different capacitors connected in parallel and/or in series with the switch The capacitance value of the first capacitor can be changed by controlling the turn-on or turn-off of the switch.
  • the adjustment circuit 702 may further include a second resistor; the first capacitor and the second resistor are connected in parallel with the first transistor; wherein,
  • the adjustment circuit 702 adjusts the transmission zero point of the control loop 701 through the second resistor, so that the stability of the control loop 701 satisfies the third condition; the transmission zero point is the connection between the first transistor and all
  • the third condition indicates that the phase margin of the control loop 701 is greater than or equal to 45 degrees.
  • the first capacitor and the second resistor may be connected in parallel with the first transistor through the collector and base of the first transistor.
  • the frequency of the transmission zero introduced by the first capacitor can be calculated by the following formula:
  • ⁇ zero represents the angular frequency of the transmission zero point
  • g m5 represents the transconductance of T 5
  • C 7 represents the first capacitance
  • R4 represents the second resistor.
  • the user can adjust the value of R 4 to make ⁇ zero less than or equal to 0, so as to push the transmission zero point to the left half plane, so that the stability of the control loop 701 is further improved, that is, the bias circuit is further improved. 700 stability.
  • the adjustment circuit 702 may further include a third resistor; the third resistor is connected in series with the first transistor; wherein,
  • the adjustment circuit 702 isolates the radio frequency signal for the first transistor through the third resistor, so that the radio frequency signal cannot be coupled to the first transistor through the first capacitor and the second resistor; the The radio frequency signal is transmitted to the bias circuit 700 through the input terminal of the radio frequency power amplifier.
  • the third resistor may be connected in series with the collector of the first transistor.
  • the radio frequency signal can prevent the base potential of the first transistor from being lowered, thereby preventing the emitter potential of the second transistor from being lowered, and further suppressing the AM-AM distortion of the radio frequency power amplifier, and suppression of ACPR deterioration of the output signal of the radio frequency power amplifier.
  • the bias circuit 700 may further include a fourth resistor (ie, R 2 in FIG. 3 ); the bias current is output to the input end of the radio frequency power amplifier through the fourth resistor; in,
  • the bias circuit 700 attenuates the radio frequency signal transmitted through the input end of the radio frequency power amplifier through the fourth resistor; and isolates the first transistor through the first resistor and is attenuated by the fourth resistor The radio frequency signal after being attenuated by the fourth resistor is transmitted to the second transistor;
  • the bias current output by the emitter stage of the second transistor is made to follow the radio frequency signal through the detection effect of the second transistor. changes with the power.
  • the bias current is a static bias current
  • the greater the power of the radio frequency signal the greater the radio frequency swing of the radio frequency signal after being attenuated by the fourth resistor, so that the bias current is greater, in other words, the The bias current increases as the power of the radio frequency signal increases. In this way, the AM-AM distortion of the radio frequency power amplifier can be suppressed, thereby suppressing the ACPR deterioration of the output signal of the radio frequency power amplifier.
  • the first resistor may be a variable resistor, and the larger the resistance value of the first resistor is, the more the first transistor isolates the radio frequency signal attenuated by the fourth resistor.
  • the first resistor can include a plurality of resistors with different resistance values connected in parallel and/or in series with the switch, and the resistance value of the first resistor can be changed by controlling the on or off of the switch .
  • the bias circuit 700 may further include a second capacitor (ie, C 6 in FIG. 3 ); one end of the second capacitor is grounded, and the other end is connected to the base of the second transistor; wherein ,
  • the bias circuit 700 short-circuits the radio frequency signal transmitted to the second transistor and attenuated by the fourth resistor to ground through the second capacitor.
  • the bias circuit 700 provided by this embodiment of the present application may adopt the structure shown in FIG. 8 .
  • the first capacitor is represented as C 7
  • the second resistor is represented as R 4
  • the The third resistance is denoted R 5 .
  • the impedance Zin3 of node A ie, the baseband impedance of the bias circuit 700
  • the loop gain of the bias circuit 700 changes with the capacitance value of the parallel capacitor C 7 as follows
  • FIG. 10 the variation of the phase margin of the bias circuit 700 with the capacitance value of the parallel capacitor C 7 is shown in FIG. 11 .
  • the resistor R 4 connected in parallel with T 5 can push the transmission zero point introduced by the parallel capacitor C 7 to the left half axis (even if the frequency point of the transmission zero point is shifted to the left), which further improves the stability of the bias circuit 700.
  • the isolation resistor R 5 can Isolate the RF signal of node B from the collector of T5 to prevent the RF signal from coupling to the base of T5 through parallel capacitor C7 and parallel resistor R4 .
  • the RF signal coupled to T5 goes through the BE junction detection effect of T5, which will cause the base potential of T5 to follow the RF signal.
  • the bias circuit 700 provided by the embodiment of the present application includes the adjustment circuit 702 (ie, C 7 , R 4 and R 5 ), compared with the bias circuit 300 , under the same value of the isolation resistance R 3 , the bias circuit 700 provided by the embodiment of the present application can obtain better noise performance and bias circuit stability.
  • the isolation resistor R3 and the parallel capacitor C7 can be switched.
  • R3 can be a variable resistor and C7 can be a variable capacitor.
  • the transistors in the embodiments of the present application may be Heterojunction Bipolar Transistor (HBT, Heterojunction Bipolar Transistor), Metal Oxide Semiconductor (MOS, Metal Oxide Semiconductor) transistor, or Bipolar Junction Transistor (BJT, Bipolar Junction). Transistor).
  • HBT Heterojunction Bipolar Transistor
  • MOS Metal Oxide Semiconductor
  • BJT Bipolar Junction Transistor
  • An embodiment of the present application further provides a radio frequency power amplifier, wherein the input end of the radio frequency power amplifier is provided with the bias circuit described in any of the foregoing embodiments.
  • the radio frequency power amplifier may include: a bias circuit 700 , an input matching circuit 1301 , an amplifying circuit 1302 and an output matching circuit 1303 .
  • the radio frequency power amplifier may be a multi-stage cascaded radio frequency power amplifier; wherein,
  • the bias circuit is provided at the input end of each stage.
  • the bias circuit includes a loop structure, which makes the bias circuit have the characteristics of low baseband impedance.
  • the bias circuit provides a low baseband impedance path for the RF power amplifier.
  • the RF power amplifier caused by the baseband impedance The memory effect of the RF power amplifier can be reduced (that is, suppressed), thereby reducing the ACPR deterioration of the RF power amplifier output signal caused by the memory effect, reducing the distortion of the RF power amplifier output signal, and improving the performance of the RF power amplifier.
  • the bias current can compensate the base current of the RF power amplifier, so that the base current does not increase with the power of the input signal.
  • it can take into account the stability and noise suppression of the control loop contained in the bias circuit, and reduce the circuit oscillation and noise that may be generated by the bias circuit on the RF power.
  • the sensitivity of the amplifier when it receives the signal further improves the performance of the RF power amplifier.
  • the circuit structure is simple, no additional circuit cost is introduced, and the design difficulty of the radio frequency power amplifier is reduced.

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Abstract

本申请公开了一种偏置电路及射频功率放大器。其中，偏置电路包括：控制环路，配置为向射频功率放大器的输入端输出偏置电流；调节电路，配置为在确定所述控制环路输出的偏置电流能够满足第一条件的情况下，基于米勒效应调节所述控制环路的环路带宽，以使所述控制环路的稳定性和噪声抑制度满足第二条件。

Description

偏置电路及射频功率放大器
相关申请的交叉引用
本申请基于申请号为202110337587.5、申请日为2021年03月30日的中国专利申请提出，并要求该中国专利申请的优先权，该中国专利申请的全部内容在此引入本申请作为参考。
技术领域
本申请涉及集成电路技术领域，尤其涉及一种偏置电路及射频功率放大器。
背景技术
与第四代移动通信技术(4G，4th Generation)的长期演进(LTE，Long Term Evolution)系统相比，第五代移动通信技术(5G，5th Generation)的新空口(NR，New Radio)系统对数据传输速率有了更高的要求，需要用带宽更宽的调制信号来传输数据，以频段N41(2490兆赫兹(MHz)至2690MHz)为例，最大调制信号带宽从40MHz扩展到100MHz，这给移动终端中射频功率放大器的设计提出了新的挑战，如何提升射频功率放大器的性能成为亟待解决的问题。
发明内容
为解决相关技术问题，本申请实施例提供一种偏置电路及射频功率放大器。
本申请实施例的技术方案是这样实现的：
本申请实施例提供了一种偏置电路，包括：
控制环路，配置为向射频功率放大器的输入端输出偏置电流；
调节电路，配置为在确定所述控制环路输出的偏置电流能够满足第一条件的情况下，基于米勒效应调节所述控制环路的环路带宽，以使所述控制环路的稳定性和噪声抑制度满足第二条件；其中，
所述第一条件表征所述偏置电流能够补偿所述射频功率放大器的基极电流，使得所述基极电流不随输入信号的功率的增大而降低；
所述第二条件表征所述控制环路的相位裕度大于或等于45度，且所述控制环路的噪声抑制度能够使得所述射频功率放大器的噪声小于或等于噪 声阈值。
上述方案中，所述调节电路包括米勒单元；所述调节电路基于所述米勒单元的米勒效应调节所述控制环路的环路带宽。
上述方案中，所述米勒单元与所述控制环路的第一晶体管并联；其中，
所述控制环路包括所述第一晶体管、第二晶体管和第一电阻；所述第一晶体管的集电极连接所述第二晶体管的基极；所述第二晶体管的发射极连接所述第一电阻的一端；所述第一电阻的另一端连接所述第一晶体管的基极；外部电流源通过所述第一晶体管和所述第二晶体管之间的第一节点向所述控制环路输入电流；所述控制环路将输入的电流转化为所述偏置电流，并通过所述第二晶体管的发射极输出到所述射频功率放大器的输入端。
上述方案中，所述米勒单元包括第一电容。
上述方案中，所述第一电容为可变电容。
上述方案中，所述调节电路还包括第二电阻；所述第一电容和所述第二电阻与所述第一晶体管并联；其中，
所述调节电路通过所述第二电阻调节所述控制环路的传输零点，以使所述控制环路的稳定性满足第三条件；所述传输零点是所述第一晶体管与所述第一电容并联所引入的；所述第三条件表征所述控制环路的相位裕度大于或等于45度。
上述方案中，所述第一电容和所述第二电阻通过所述第一晶体管的集电极和基极与所述第一晶体管并联。
上述方案中，所述调节电路还包括第三电阻；所述第三电阻与所述第一晶体管串联；其中，
所述调节电路通过所述第三电阻为所述第一晶体管隔离射频信号，以使所述射频信号无法通过所述第一电容和所述第二电阻耦合到所述第一晶体管；所述射频信号是通过所述射频功率放大器的输入端传输到所述偏置电路的。
上述方案中，所述第三电阻与所述第一晶体管的集电极串联。
上述方案中，所述偏置电路还包括第四电阻；所述偏置电流通过所述第四电阻输出到所述射频功率放大器的输入端；其中，
所述偏置电路通过所述第四电阻对通过所述射频功率放大器的输入端传输的射频信号进行衰减；并通过所述第一电阻为所述第一晶体管隔离被所述第四电阻衰减后的射频信号，以使被所述第四电阻衰减后的射频信号传输到所述第二晶体管；
在被所述第四电阻衰减后的射频信号的射频摆幅的作用下，通过所述第二晶体管的检波效应，使所述第二晶体管的发射级输出的所述偏置电流随所述射频信号的功率的变化而变化。
上述方案中，所述第一电阻为可变电阻。
上述方案中，所述偏置电路还包括第二电容；所述第二电容的一端接 地，另一端连接所述第二晶体管的基极；其中，
所述偏置电路通过所述第二电容，使传输到所述第二晶体管的被所述第四电阻衰减后的射频信号短路到地。
本申请实施例还提供了一种射频功率放大器，所述射频功率放大器的输入端设置有上述任一方案所述的偏置电路。
上述方案中，所述射频功率放大器为多级级连的射频功率放大器；其中，
每一级的输入端均设置有所述偏置电路。
本申请实施例提供的偏置电路及射频功率放大器，偏置电路包括：控制环路，配置为向射频功率放大器的输入端输出偏置电流；调节电路，配置为在确定所述控制环路输出的偏置电流能够满足第一条件的情况下，基于米勒效应调节所述控制环路的环路带宽，以使所述控制环路的稳定性和噪声抑制度满足第二条件；其中，所述第一条件表征所述偏置电流能够补偿所述射频功率放大器的基极电流，使得所述基极电流不随输入信号的功率的增大而降低；所述第二条件表征所述控制环路的相位裕度大于或等于45度，且所述控制环路的噪声抑制度能够使得所述射频功率放大器的噪声小于或等于噪声阈值；所述射频功率放大器的输入端设置有所述偏置电路。本申请实施例的方案，偏置电路包括环路结构，这使得偏置电路具备低基带阻抗的特性，换句话说，偏置电路为射频功率放大器提供了低基带阻抗通路，如此，基带阻抗引起的射频功率放大器的记忆效应能够减小(即能够得到抑制)，从而能够降低记忆效应造成的射频功率放大器输出信号的邻信道功率比(ACPR，Adjacent Channel Power Ratio)的恶化，减少射频功率放大器输出信号的失真，进而提升射频功率放大器的性能。同时，利用调节电路，在确定所述控制环路输出的偏置电流能够满足第一条件的情况下调节所述控制环路的环路带宽，以使所述控制环路的稳定性和噪声抑制度满足第二条件；如此，射频功率放大器的输入信号的带宽增大时，在保证偏置电流能够补偿射频功率放大器的基极电流，使得所述基极电流不随输入信号的功率的增大而降低，从而抑制射频功率放大器的幅度调制对幅度调制(AM-AM)失真(AM-AM失真是指射频功率放大器的输出信号相对于输入信号在幅度上的失真)的前提下，能够兼顾偏置电路包含的控制环路的稳定性和噪声抑制度，减少偏置电路可能产生的电路振荡及噪声对射频功率放大器接收信号时的灵敏度的影响，进一步提升射频功率放大器的性能。
附图说明
图1为相关技术中射频功率放大器的三阶交调出现不对称的示意图；
图2为相关技术中一种偏置电路的结构示意图；
图3为相关技术中另一种偏置电路的结构示意图；
图4为相关技术中偏置电路300的阻抗在整个频段内的频率响应示意图；
图5为相关技术中偏置电路300的环路增益示意图；
图6为相关技术中偏置电路300的相位裕度示意图；
图7为本申请实施例偏置电路的一种结构示意图；
图8为本申请实施例偏置电路的另一种结构示意图；
图9为本申请实施例偏置电路700的阻抗随并联电容C 7变化的示意图；
图10为本申请实施例偏置电路700的环路增益随并联电容C 7变化的示意图；
图11为本申请实施例偏置电路700的相位裕度随并联电容C 7变化的示意图；
图12为本申请实施例射频信号通过并联电容C 7和并联电阻R 4耦合到T 5的示意图；
图13为本申请实施例射频功率放大器的结构示意图。
具体实施方式
下面结合附图及实施例对本申请再作进一步详细的描述。
移动终端中的射频功率放大器通常包括输入匹配电路、输出匹配电路、放大管以及偏置电路等，其中，电感、电容等频响器件的阻抗仅在有限的频率带宽内表现一致，在5G NR系统中，随着调制信号的带宽变宽，射频功率放大器在不同频率上的阻抗响应不同，使得调制信号叠加了额外的幅度和相位变化，三阶交调分量IM3出现不对称，即射频功率放大器出现记忆效应。
示例性地，在射频功率放大器的输入信号为等幅双音信号(对应的两个频率分别可以表示为f 1和f 2，且f 1<f 2)的情况下，基于射频功率放大器的非线性多项式的展开可以确定：射频功率放大器的三阶非线性产物包括三阶电流矢量I 3，I 3包含2f 1-f 2和2f 2-f 1(即I 3对应的两个频率可以表示为2f 1-f 2和2f 2-f 1)，射频功率放大器在2f 1-f 2频率点上的阻抗响应和在2f 2-f 1频率点上的阻抗响应基本一致，且2f 1-f 2对应的电流矢量和2f 2-f 1对应的电流矢量相近；射频功率放大器的二阶非线性产物中有贡献的部分主要包括：二阶电流矢量I 1、I 2和I 4；其中，I 1和I 2为基带包络信号，分别包含f 1-f 2和f 2-f 1(即I 1和I 2分别对应的两个频率可以表示为f 1-f 2和f 2-f 1)，f 1-f 2对应的电流矢量和f 2-f 1对应的电流矢量幅度相等但相位存在差异；I 4为二阶谐波信号，包含2f 1和2f 2(即I 4对应的两个频率可以表示为2f 1和2f 2)，射频功率放大器在2f 1频率点上的阻抗响应和在2f 2频率点上的阻抗响应基本一致，且2f 1对应的电流矢量和2f 2对应的电流矢量相近。如图1所示，三阶交调 分量IM3为上述电流矢量I 1、I 2、I 3和I 4的叠加，上边带IM3_Up与下边带IM3_Low出现不对称，即射频功率放大器出现记忆效应。
在针对调制信号带宽较窄(即最大调制信号带宽小于或等于20MHz)的场景设计射频功率放大器时，由于射频功率放大器中各节点阻抗的频率响应相对一致，记忆效应微弱，通常可以将射频功率放大器的级间阻抗和负载阻抗设计在兼顾ACPR和效率的区域，从而解决射频功率放大器在大信号(即功率较大的信号)输入的情况下非线性放大特性导致输出信号的ACPR恶化的问题。
而针对调制信号带宽较宽(即最大调制信号带宽≥40MHz)的场景设计射频功率放大器时，由于射频功率放大器中各节点阻抗的频率响应差异会随着调制信号频带变宽而增大，表现出的记忆效应也会越严重，即三阶交调出现明显的不对称，引起输出信号的ACPR恶化。此时，如果采用设计窄带射频功率放大器(即针对调制信号带宽较窄的场景设计的射频功率放大器)时降低负载阻抗的方式来提高射频功率放大器的线性功率，只能减小三阶电流矢量I 3带来的失真，对于二阶电流矢量I 1、I 2和I 4带来的失真改善微弱，使得ACPR改善效果不佳，并且，射频功率放大器的功耗容易变得不可接受(比如功耗超出预算)。
为了降低记忆效应对ACPR的影响，可以考虑从减小二阶电流矢量I 1、I 2和I 4带来的失真入手，通过在射频功率放大器输入输出节点制造基带和二阶频率低阻等方法，降低电流矢量I 1、I 2和I 4流经这些节点阻抗产生的幅度，从而改善射频功率放大器输出信号ACPR恶化的情况。
具体地，可以利用低基带阻抗的偏置网络在射频功率放大器的输入节点实现基带频率的低阻抗，抑制电流矢量I 1和I 2的非线性贡献。如图2所示，射频功率放大器包括：输入匹配电容C 1、放大晶体管T 1、输出匹配电感L 1、L 2和L 3、电容C 2、C 3和C 4以及偏置电路200。射频信号通过RFin以及C 1进入到T 1进行放大，再通过输出匹配电路(包含L 1、L 2、L 3、C 2、C 3和C 4)输出到Rfout。其中，偏置电路200包括隔离电阻R 1、晶体管T 2、T 3和T 4以及电容C 5；偏置电路200将外部电流源输入的电流Ib1转化为偏置电流I em3并通过R 1输出到射频功率放大器的输入端，此时，偏置电路200中的节点A的阻抗Zin1(即偏置电路200的基带阻抗)可以通过以下公式表示：
Zin1＝R 1+R e3   (1)
其中，R e3表示T 3的发射极通路电阻，可以通过以下公式计算：
Figure PCTCN2021133990-appb-000001
其中，VT表示T 3的热电压，I em3表示T 3的偏置电流。示例性地，R 1取值为15欧姆(Ohm)，VT在室温下约为26毫伏(mV)，晶体管T 1的偏置电流为100毫安(mA)，I em3约为0.6mA；此时，R e3约为43ohm(即26mV/0.6mA)，Zin1约为58ohm(即43ohm+15Ohm)。可见，偏置电路200 的基带阻抗较高，并不能有效改善射频功率放大器输出信号ACPR恶化的情况。
为了进一步降低偏置电路的基带阻抗，进而改善射频功率放大器输出信号ACPR恶化的问题，可以对偏置电路200进行改进。如图3所示，偏置电路300包括：晶体管T 5和T 6、隔离电阻R 2和R 3以及电容C 6。在没有射频信号输入射频功率放大器的情况下，偏置电路300将外部电流源输入的电流Ib1转化为合适的静态的偏置电流I em6并通过电阻R 2输出到射频功率放大器的输入端。在有射频信号输入射频功率放大器的情况下，偏置电路300通过R 2阻隔大部分射频信号从偏置电路泄漏(即对通过所述射频功率放大器的输入端传输的射频信号进行衰减，R 2可以取一个较小值，比如15Ohm)，并通过R 3为T 5的基极低阻点隔离被R 2衰减后的射频信号(为了提高对射频信号的隔离效果，R 3的取值比较关键，通常需要取一个较大值，比如2000Ohm)，以使被R 2衰减后的射频信号传输到T 6，通过被R 2衰减后的射频信号，如果该射频信号为大信号(即功率较大的信号)，T 6的发射极会有一定的射频摆幅，射频摆幅经过T 6的BE结的检波效应使BE结的偏置电压变小，从而使T 6发射极输出的I em6增大(即I em6随着射频功率放大器的输入信号的功率的增大而增大)，能够补偿T 1基极电压随输入信号的功率的增大而降低，抑制大信号下射频功率放大器的AM-AM失真，同时，C 6(可以取一个较小值，比如3皮法(pF))在T 6基极形成射频低阻，偏置电路300通过C 6使传输到T 6的被R 2衰减后的射频信号短路到地。此时，偏置电路300中的节点A的阻抗Zin2(即偏置电路300的基带阻抗)可以通过以下公式表示：
Figure PCTCN2021133990-appb-000002
其中，ω表示角频率，C be5表示T 5的基极电容，C be6表示T 6的基极电容，β表示晶体管的电流放大系数，通常为80～150；g m5和g m6分别表示T 5和T 6的跨导。
实际应用时，考虑到C be5、C be6和C 6的电容值较小，在基频频率范围内可以忽略，因此，可以将公式(3)简化为以下公式：
Figure PCTCN2021133990-appb-000003
这样，在R 2的取值与图2中R 1的取值均为15Ohm的情况下，偏置电路300的基带阻抗仅为15Ohm，而偏置电路200的基带阻抗为58ohm。可见，与偏置电路200相比，偏置电路300的基带阻抗进一步降低，从而能够进一步地改善射频功率放大器输出信号ACPR恶化的问题。
实际应用时，如图4所示，偏置电路300可以在100MHz的基带信号带宽内表现出很低的基带阻抗特性。由于T 1相当于和节点A并联，在基带频率为100MHz以内的情况下，节点A的阻抗(即偏置电路300的基带阻 抗Zin2)的量级通常为15Ohm左右，而T 1的阻抗(可以表示为Zin_T 1)的量级通常为40～50Ohm左右。因此，T 1在如图3所示的输入节点D产生的二阶非线性电流矢量I 1和I 2大部分通过阻抗更低的偏置电路300短路到地，减少了进入T 1的二阶非线性电流矢量I 1和I 2的幅度，达到抑制电流矢量I 1和I 2的非线性贡献的目的。
然而，实际应用时，偏置电路300存在以下问题：
第一，环路稳定性较差。具体地，实际应用时，偏置电路300的环路增益如图5所示，同时，偏置电路300的相位裕度如图6所示，由于R 3需要取一个较大值，才能有效地为T 5的基极低阻点隔离被R 2衰减后的射频信号(如果R 3的取值较小，R 3对射频信号的隔离效果会变差，T 6的检波补偿效果会变弱，射频功率放大器在大信号下会出现明显的AM-AM失真，导致射频功率放大器ACPR恶化明显)，而R 3的取值越大，R 3与C be5形成的电阻-电容(RC，Resistor-Capacitance)网络对馈入的信号的移相会越超前。示例性地，在偏置电路300的环路增益为0的情况下，相比输入节点C的信号，反馈到节点B的信号的相位会变化-135度；如果节点B的信号相位相比节点C的信号相位变化了-180度，则偏置电路300的环路会形成正反馈，从而导致偏置电路300出现振荡；虽然-135度相较于-180度还有45度的余量，但在高低温、高低压或工艺波动等情况下，节点B的信号相位相比节点C的信号相位的变化很容易超出这45度的余量；换句话说，偏置电路300的环路的稳定性较差，存在电路振荡的风险。
第二，噪声抑制度较差。具体地，在5G NR系统中，偏置电路300的基带带宽较宽，而R 3需要取一个较大值，这使得R 3产生的噪声(即热噪声)在偏置电路300通带内无法得到很好的抑制，噪声电流通过T 5放大后馈入节点B，再通过T 6的基极叠加到偏置电流I em6。可见，噪声电流会通过T 1非线性混频到射频信号两端，导致射频功率放大器接收频段的噪声恶化，使得射频功率放大器不能满足相关通信协议对信号接收的灵敏度的要求。
基于此，在本申请的各种实施例中，偏置电路包括控制环路和调节电路，偏置电路利用控制环路向射频功率放大器的输入端输出偏置电流，由于偏置电路包括环路结构，这使得偏置电路具备低基带阻抗的特性，换句话说，偏置电路为射频功率放大器提供了低基带阻抗通路，如此，基带阻抗引起的射频功率放大器的记忆效应能够减小(即能够得到抑制)，从而能够降低记忆效应造成的射频功率放大器输出信号的ACPR的恶化，减少射频功率放大器输出信号的失真，进而提升射频功率放大器的性能。同时，偏置电路利用调节电路在确定所述控制环路输出的偏置电流能够满足第一条件的情况下调节所述控制环路的环路带宽，以使所述控制环路的稳定性和噪声抑制度满足第二条件；所述第一条件表征所述偏置电流能够补偿所述射频功率放大器的基极电流，使得所述基极电流不随输入信号的功率的增大而降低；所述第二条件表征所述控制环路的相位裕度大于或等于45度， 且所述控制环路的噪声抑制度能够使得所述射频功率放大器的噪声小于或等于噪声阈值；如此，射频功率放大器的输入信号的带宽增大时，在保证偏置电流能够补偿射频功率放大器的基极电流，使得所述基极电流不随输入信号的功率的增大而降低，从而抑制射频功率放大器的AM-AM失真的前提下，能够兼顾偏置电路包含的控制环路的稳定性和噪声抑制度，减少偏置电路可能产生的电路振荡及噪声对射频功率放大器接收信号时的灵敏度的影响，进一步提升射频功率放大器的性能。
本申请实施例提供了一种偏置电路，如图7所示，该偏置电路700包括：
控制环路701，配置为向射频功率放大器的输入端输出偏置电流；
调节电路702，配置为在确定所述控制环路701输出的偏置电流能够满足第一条件的情况下，基于米勒效应调节所述控制环路701的环路带宽，以使所述控制环路701的稳定性和噪声抑制度满足第二条件；
其中，所述第一条件表征所述偏置电流能够补偿所述射频功率放大器的基极电流，使得所述基极电流不随输入信号的功率的增大而降低；
所述第二条件表征所述控制环路701的相位裕度大于或等于45度，且所述控制环路701的噪声抑制度能够使得所述射频功率放大器的噪声小于或等于噪声阈值。
这里，所述控制环路701的功能相当于图3所示的偏置电路300的环路(包括T 5、T 6和R 3)的功能。
实际应用时，调节电路702基于米勒效应调节所述控制环路701的环路带宽时，可以根据电路的最大工作带宽进行取值，换句话说，调节所述控制环路701的最大工作带宽。其中，最大工作带宽的取值越小，所述控制环路701的环路带宽越窄，所述控制环路701的稳定性和噪声抑制度越好，但所述射频功率放大器对基带阻抗引起的记忆效应的抑制越差；而最大工作带宽的取值越大，所述控制环路701的环路带宽越宽，所述控制环路701的稳定性和噪声抑制度越差，所述射频功率放大器对基带阻抗引起的记忆效应的抑制越好。
实际应用时，所述确定所述控制环路701输出的偏置电流能够满足第一条件，可以理解为保证所述确定所述控制环路701输出的偏置电流能够满足第一条件，也就是说，在保证所述确定所述控制环路701输出的偏置电流能够满足第一条件的情况下，所述调节电路702基于米勒效应调节所述控制环路701的环路带宽，以使所述控制环路701的稳定性和噪声抑制度满足第二条件。这里，基于上述控制环路701的稳定性和噪声抑制度以及射频功率放大器对基带阻抗引起的记忆效应的抑制程度的变化规律，用户可以根据需求对所述第二条件进行具体设置，使得偏置电路700能够兼顾基带阻抗带宽、噪声抑制度、电路稳定性以及射频性能。
示例性地，可以设置所述控制环路701的相位裕度大于或等于60度， 并根据相关技术中的通信协议对射频功率放大器的信号接收灵敏度的要求，设置所述噪声阈值。比如，对于频段N1(上行频段发送(TX)：1920MHz至1980MHz，下行频段接收(RX)：2110MHz至2170MHz)，协议要求在10MHz带宽下，射频功率放大器的信号接收灵敏度小于或等于-98分贝毫瓦(dBm)，结合射频功率放大器输出到N1双工器的RX路径的抑制度，可以计算出射频功率放大器在RX频段内的噪声需要小于或等于-135dBm/Hz。此时，所述第二条件具体可以包括：所述控制环路701的相位裕度大于或等于60度，且所述控制环路701的噪声抑制度能够使得所述射频功率放大器的噪声小于或等于-135dBm/Hz。
在一实施例中，所述调节电路702可以包括米勒单元；所述调节电路702基于所述米勒单元的米勒效应调节所述控制环路701的环路带宽。
具体地，所述米勒单元可以与所述控制环路701的第一晶体管并联；其中，
所述控制环路701可以包括所述第一晶体管(即图3中的T 5)、第二晶体管(即图3中的T 6)和第一电阻(即图3中的R 3)；所述第一晶体管的集电极连接所述第二晶体管的基极；所述第二晶体管的发射极连接所述第一电阻的一端；所述第一电阻的另一端连接所述第一晶体管的基极；外部电流源通过所述第一晶体管和所述第二晶体管之间的第一节点(即图3中的节点B)向所述控制环路701输入电流；所述控制环路701将输入的电流转化为所述偏置电流(即图3中的I em6)，并通过所述第二晶体管的发射极输出到所述射频功率放大器的输入端。
实际应用时，所述米勒单元可以包括电容和/或晶体管。
基于此，在一实施例中，所述米勒单元包括第一电容。
实际应用时，为了增强电路灵活性，所述第一电容可以是可变电容(比如压控可变电容)；或者，所述第一电容可以包括多个与开关并联和/或串联的不同电容值的电容，并可以通过控制开关的导通或断开实现所述第一电容的电容值的变化。
在一实施例中，所述调节电路702还可以包括第二电阻；所述第一电容和所述第二电阻与所述第一晶体管并联；其中，
所述调节电路702通过所述第二电阻调节所述控制环路701的传输零点，以使所述控制环路701的稳定性满足第三条件；所述传输零点是所述第一晶体管与所述第一电容并联所引入的；所述第三条件表征所述控制环路701的相位裕度大于或等于45度。
具体地，实际应用时，所述第一电容和所述第二电阻可以通过所述第一晶体管的集电极和基极与所述第一晶体管并联。
实际应用时，由电路分析可知，加入所述第二电阻前，所述第一电容引入的传输零点的频率可以通过以下公式计算：
Figure PCTCN2021133990-appb-000004
其中，ω zero表示传输零点的角频率，g m5表示T 5的跨导，C 7表示所述第一电容。由于该传输零点存在于右半平面，会减缓环路增益的下降幅度，使环路增益的交点外推，更远离原点，导致所述控制环路701的稳定性降低。而加入所述第二电阻后，该传输零点的频率可以通过以下公式计算：
Figure PCTCN2021133990-appb-000005
其中，R4表示所述第二电阻。用户可以通过调整R 4的取值，使ω zero小于或等于0，从而将所述传输零点推向左半平面，使得所述控制环路701的稳定性得到进一步改善，即进一步改善偏置电路700的稳定性。
在一实施例中，所述调节电路702还可以包括第三电阻；所述第三电阻与所述第一晶体管串联；其中，
所述调节电路702通过所述第三电阻为所述第一晶体管隔离射频信号，以使所述射频信号无法通过所述第一电容和所述第二电阻耦合到所述第一晶体管；所述射频信号是通过所述射频功率放大器的输入端传输到所述偏置电路700的。
具体地，实际应用时，所述第三电阻可以与所述第一晶体管的集电极串联。通过所述第三电阻为所述第一晶体管隔离射频信号，能够避免射频信号造成所述第一晶体管的基极电位降低，从而避免所述第二晶体管的发射极电位降低，进而能够抑制所述射频功率放大器的AM-AM失真，并抑制所述射频功率放大器输出信号的ACPR恶化。
在一实施例中，所述偏置电路700还可以包括第四电阻(即图3中的R 2)；所述偏置电流通过所述第四电阻输出到所述射频功率放大器的输入端；其中，
所述偏置电路700通过所述第四电阻对通过所述射频功率放大器的输入端传输的射频信号进行衰减；并通过所述第一电阻为所述第一晶体管隔离被所述第四电阻衰减后的射频信号，以使被所述第四电阻衰减后的射频信号传输到所述第二晶体管；
在被所述第四电阻衰减后的射频信号的射频摆幅作用下，通过所述第二晶体管的检波效应，使所述第二晶体管的发射级输出的所述偏置电流随所述射频信号的功率的变化而变化。
实际应用时，在没有射频信号输入所述射频功率放大器的情况下，也没有射频信号传输到所述第二晶体管，因此，所述偏置电流为静态的偏置电流；在有射频信号输入所述射频功率放大器的情况下，所述射频信号的功率越大，被所述第四电阻衰减后的射频信号的射频摆幅越大，从而使得所述偏置电流越大，换句话说，所述偏置电流随着所述射频信号的功率的增大而增大。如此，能够抑制所述射频功率放大器的AM-AM失真，从而抑制所述射频功率放大器输出信号的ACPR恶化。
实际应用时，为了增强电路灵活性，所述第一电阻可以是可变电阻， 所述第一电阻的阻值越大，为所述第一晶体管隔离被所述第四电阻衰减后的射频信号的效果越好；所述第一电阻可以包括多个与开关并联和/或串联的不同阻值的电阻，并可以通过控制开关的导通或断开实现所述第一电阻的阻值的变化。
在一实施例中，所述偏置电路700还可以包括第二电容(即图3中的C 6)；所述第二电容的一端接地，另一端连接所述第二晶体管的基极；其中，
所述偏置电路700通过所述第二电容，使传输到所述第二晶体管的被所述第四电阻衰减后的射频信号短路到地。
实际应用时，本申请实施例提供的偏置电路700可以采用图8所示的结构，在图8中，所述第一电容表示为C 7，所述第二电阻表示为R 4，所述第三电阻表示为R 5。另外，节点A的阻抗Zin3(即偏置电路700的基带阻抗)随并联电容C 7电容值的变化如图9所示，偏置电路700的环路增益随并联电容C 7电容值的变化如图10所示，偏置电路700的相位裕度随并联电容C 7电容值的变化如图11所示。T 5并联的电阻R 4可以将并联电容C 7引入的传输零点推向左半轴(即使传输零点的频点左移)，进一步改善偏置电路700的稳定性，同时，隔离电阻R 5可以将节点B的射频信号与T 5的集电极隔离开来，避免射频信号通过并联电容C 7和并联电阻R 4耦合到T 5的基级。如图12所示，如果射频信号通过C 7和R 4耦合到T 5的基级，耦合到T 5的射频信号经过T 5的BE结检波效应，会使T 5基极电位随着射频信号增大而下降，从而使得T 6发射极电位也随之下降，导致射频功率放大器在大信号下出现明显的AM-AM失真，进而导致射频功率放大器ACPR恶化。由此可见，由于本申请实施例提供的偏置电路700包括调节电路702(即C 7、R 4和R 5)，与偏置电路300相比，在相同的隔离电阻R 3的取值下，本申请实施例提供的偏置电路700可以获得更好的噪声性能与偏置电路稳定性。
实际应用时，隔离电阻R 3和并联电容C 7是可以切换的，换句话说，R 3可以是可变电阻，C 7可以是可变电容。通过改变R 3和/或C 7的取值，根据不同的调制带宽信号选择不同的偏置电路基带阻抗带宽以及噪声抑制度，能够达到增加电路的适用性和灵活性的目的。
实际应用时，本申请实施例中的晶体管可以是异质结双极晶体管(HBT，Heterojunction Bipolar Transistor)、金属氧化物半导体(MOS，Metal Oxide Semiconductor)管或双极结型晶体管(BJT，Bipolar Junction Transistor)。
本申请实施例还提供了一种射频功率放大器，所述射频功率放大器的输入端设置有上述任一实施例所述的偏置电路。
示例性地，如图13所示，射频功率放大器可以包括：偏置电路700、输入匹配电路1301、放大电路1302和输出匹配电路1303。
在一实施例中，所述射频功率放大器可以是多级级连的射频功率放大器；其中，
每一级的输入端均设置有所述偏置电路。
本申请实施例提供的偏置电路和射频功率放大器，具有以下优点：
第一，偏置电路包括环路结构，这使得偏置电路具备低基带阻抗的特性，换句话说，偏置电路为射频功率放大器提供了低基带阻抗通路，如此，基带阻抗引起的射频功率放大器的记忆效应能够减小(即能够得到抑制)，从而能够降低记忆效应造成的射频功率放大器输出信号的ACPR的恶化，减少射频功率放大器输出信号的失真，进而提升射频功率放大器的性能。
第二，在5G NR系统中，当射频功率放大器的输入信号的带宽增大时，在保证偏置电流能够补偿射频功率放大器的基极电流，使得所述基极电流不随输入信号的功率的增大而降低，从而抑制射频功率放大器的AM-AM失真的前提下，能够兼顾偏置电路包含的控制环路的稳定性和噪声抑制度，减少偏置电路可能产生的电路振荡及噪声对射频功率放大器接收信号时的灵敏度的影响，进一步提升射频功率放大器的性能。
第三，电路结构简单，不会引入额外的电路成本，降低了射频功率放大器的设计难度。
需要说明的是：“第一”、“第二”等是用于区别类似的对象，而不必用于描述特定的顺序或先后次序。
另外，本申请实施例所记载的技术方案之间，在不冲突的情况下，可以任意组合。
以上所述，仅为本申请的较佳实施例而已，并非用于限定本申请的保护范围。

Claims (14)

  1. 一种偏置电路，包括：
    控制环路，配置为向射频功率放大器的输入端输出偏置电流；
    调节电路，配置为在确定所述控制环路输出的偏置电流能够满足第一条件的情况下，基于米勒效应调节所述控制环路的环路带宽，以使所述控制环路的稳定性和噪声抑制度满足第二条件；其中，
    所述第一条件表征所述偏置电流能够补偿所述射频功率放大器的基极电流，使得所述基极电流不随输入信号的功率的增大而降低；
    所述第二条件表征所述控制环路的相位裕度大于或等于45度，且所述控制环路的噪声抑制度能够使得所述射频功率放大器的噪声小于或等于噪声阈值。
  2. 根据权利要求1所述的偏置电路，其中，所述调节电路包括米勒单元；所述调节电路基于所述米勒单元的米勒效应调节所述控制环路的环路带宽。
  3. 根据权利要求2所述的偏置电路，其中，所述米勒单元与所述控制环路的第一晶体管并联；其中，
    所述控制环路包括所述第一晶体管、第二晶体管和第一电阻；所述第一晶体管的集电极连接所述第二晶体管的基极；所述第二晶体管的发射极连接所述第一电阻的一端；所述第一电阻的另一端连接所述第一晶体管的基极；外部电流源通过所述第一晶体管和所述第二晶体管之间的第一节点向所述控制环路输入电流；所述控制环路将输入的电流转化为所述偏置电流，并通过所述第二晶体管的发射极输出到所述射频功率放大器的输入端。
  4. 根据权利要求3所述的偏置电路，其中，所述米勒单元包括第一电容。
  5. 根据权利要求4所述的偏置电路，其中，所述第一电容为可变电容。
  6. 根据权利要求4所述的偏置电路，其中，所述调节电路还包括第二电阻；所述第一电容和所述第二电阻与所述第一晶体管并联；其中，
    所述调节电路通过所述第二电阻调节所述控制环路的传输零点，以使所述控制环路的稳定性满足第三条件；所述传输零点是所述第一晶体管与所述第一电容并联所引入的；所述第三条件表征所述控制环路的相位裕度大于或等于45度。
  7. 根据权利要求6所述的偏置电路，其中，所述第一电容和所述第二电阻通过所述第一晶体管的集电极和基极与所述第一晶体管并联。
  8. 根据权利要求6所述的偏置电路，其中，所述调节电路还包括第 三电阻；所述第三电阻与所述第一晶体管串联；其中，
    所述调节电路通过所述第三电阻为所述第一晶体管隔离射频信号，以使所述射频信号无法通过所述第一电容和所述第二电阻耦合到所述第一晶体管；所述射频信号是通过所述射频功率放大器的输入端传输到所述偏置电路的。
  9. 根据权利要求8所述的偏置电路，其中，所述第三电阻与所述第一晶体管的集电极串联。
  10. 根据权利要求3至9任一项所述的偏置电路，其中，所述偏置电路还包括第四电阻；所述偏置电流通过所述第四电阻输出到所述射频功率放大器的输入端；其中，
    所述偏置电路通过所述第四电阻对通过所述射频功率放大器的输入端传输的射频信号进行衰减；并通过所述第一电阻为所述第一晶体管隔离被所述第四电阻衰减后的射频信号，以使被所述第四电阻衰减后的射频信号传输到所述第二晶体管；
    在被所述第四电阻衰减后的射频信号的射频摆幅的作用下，通过所述第二晶体管的检波效应，使所述第二晶体管的发射级输出的所述偏置电流随所述射频信号的功率的变化而变化。
  11. 根据权利要求3至9任一项所述的偏置电路，其中，所述第一电阻为可变电阻。
  12. 根据权利要求10所述的偏置电路，其中，所述偏置电路还包括第二电容；所述第二电容的一端接地，另一端连接所述第二晶体管的基极；其中，
    所述偏置电路通过所述第二电容，使传输到所述第二晶体管的被所述第四电阻衰减后的射频信号短路到地。
  13. 一种射频功率放大器，所述射频功率放大器的输入端设置有权利要求1至12任一项所述的偏置电路。
  14. 根据权利要求13所述的射频功率放大器，其中，所述射频功率放大器为多级级连的射频功率放大器；其中，
    每一级的输入端均设置有所述偏置电路。
PCT/CN2021/133990 2021-03-30 2021-11-29 偏置电路及射频功率放大器 WO2022205975A1 (zh)

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CN113489461A (zh) * 2021-07-28 2021-10-08 电子科技大学 一种射频预失真线性化电路及射频功率放大器
CN116781045A (zh) * 2022-03-11 2023-09-19 长鑫存储技术有限公司 偏置信号生成电路与时钟输入电路
CN114944819B (zh) * 2022-05-16 2023-02-10 广东工业大学 一种用于射频功率放大器的偏置电路
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