WO2022105068A1 - 并网逆变器的控制方法和装置 - Google Patents

并网逆变器的控制方法和装置 Download PDF

Info

Publication number
WO2022105068A1
WO2022105068A1 PCT/CN2021/075979 CN2021075979W WO2022105068A1 WO 2022105068 A1 WO2022105068 A1 WO 2022105068A1 CN 2021075979 W CN2021075979 W CN 2021075979W WO 2022105068 A1 WO2022105068 A1 WO 2022105068A1
Authority
WO
WIPO (PCT)
Prior art keywords
grid
connected inverter
current
moment
value
Prior art date
Application number
PCT/CN2021/075979
Other languages
English (en)
French (fr)
Inventor
何炜琛
但志敏
高锦凤
侯贻真
Original Assignee
江苏时代新能源科技有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 江苏时代新能源科技有限公司 filed Critical 江苏时代新能源科技有限公司
Priority to ES21745203T priority Critical patent/ES2936309T3/es
Priority to EP21745203.6A priority patent/EP4030611B1/en
Priority to US17/485,554 priority patent/US11251720B1/en
Publication of WO2022105068A1 publication Critical patent/WO2022105068A1/zh

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L55/00—Arrangements for supplying energy stored within a vehicle to a power network, i.e. vehicle-to-grid [V2G] arrangements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/28—Arrangements for balancing of the load in a network by storage of energy
    • H02J3/32—Arrangements for balancing of the load in a network by storage of energy using batteries with converting means
    • H02J3/322—Arrangements for balancing of the load in a network by storage of energy using batteries with converting means the battery being on-board an electric or hybrid vehicle, e.g. vehicle to grid arrangements [V2G], power aggregation, use of the battery for network load balancing, coordinated or cooperative battery charging
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/38—Arrangements for parallely feeding a single network by two or more generators, converters or transformers
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0016—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters
    • H02M1/0019—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters the disturbance parameters being load current fluctuations
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0025—Arrangements for modifying reference values, feedback values or error values in the control loop of a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/493—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode the static converters being arranged for operation in parallel
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current

Definitions

  • the present application relates to the field of electric power technology, and in particular, to a method and device for controlling a grid-connected inverter.
  • grid-connected inverters have been widely used as key equipment for grid-connected operation of new energy power generation systems such as wind power and photovoltaics. How to optimize the control strategy of grid-connected inverters and improve the operational reliability of grid-connected inverters has become a current research focus.
  • model predictive control has become the main research direction in the field of predictive control of grid-connected inverters because of its advantages of fast dynamic response.
  • the present application provides a method and device for controlling a grid-connected inverter, which improves the power transmission efficiency of the system.
  • a control method for a grid-connected inverter comprising: acquiring an output voltage of the grid-connected inverter and an output current value at the current moment; calculating an output current value based on the output current value at the current moment N output current values at the next moment of the grid-connected inverter, the N output current values at the next moment are in one-to-one correspondence with the N switch states of the grid-connected inverter, and N is greater than or equal to 2; obtaining a first reference current based on the output voltage; obtaining a second reference current based on the output current of the grid-connected inverter and the first reference current; determining a first switch state among the N switch states , the difference between the output current value at the next moment corresponding to the first switch state and the value of the second reference current at the next moment is the N output current values at the next moment and the second reference current The minimum value among the differences of the values at the next moment; and controlling the grid-connected inverter to perform power
  • a control device for a grid-connected inverter including an acquisition unit and a processing unit.
  • the obtaining unit is used to obtain the output voltage of the grid-connected inverter and the output current value at the current moment.
  • the processing unit is configured to: calculate N output current values of the grid-connected inverter at the next moment based on the output current value at the current moment, and the N output current values at the next moment are the same as the grid-connected inverter.
  • N is greater than or equal to 2
  • the first switch state is determined among the N switch states, and the difference between the output current value at the next moment corresponding to the first switch state and the value of the second reference current at the next moment is the N
  • the minimum value of the difference between the output current value at the next moment and the value of the second reference current at the next moment is the N
  • controlling the grid-connected inverter to perform the next moment in the first switching state Power transmission.
  • the obtained second reference current is a reference current including a supplementary disturbance amount.
  • the above processing unit may be configured to: establish a third-order state model based on the output current of the grid-connected inverter; and obtain the second-order state model based on the established third-order state model and the first reference current A reference current, the second reference current includes a compensation value for a system disturbance.
  • the third-order state model is:
  • X 1 represents the output current ig of the grid-connected inverter
  • X 2 is the differential of X 1
  • X 3 is the differential of X 2 and represents the system disturbance
  • the second reference current is calculated by the following formula Get:
  • u represents the second reference current
  • i ref1 is the first reference current
  • k d 2w c
  • w c is the cut-off frequency of the grid-connected inverter
  • L is the inductance in the grid-connected inverter
  • C is the capacitance in the grid-connected inverter.
  • the above processing unit may be configured to: obtain N switch output voltages based on the N switch states and the input voltage of the grid-connected inverter, respectively; and based on the output current value at the current moment, the The N output current values at the next moment are obtained from the value of the N switch output voltages at the current moment and the value of the output voltage of the grid-connected inverter at the current moment.
  • k represents the current moment
  • T s is the control frequency of the grid-connected inverter
  • r is the internal resistance of the grid-connected inverter
  • L is the inductance in the grid-connected inverter
  • i g ( k) is the output current value of the N current moments
  • U 0 (k) is the value of the N switch output voltages at the current moment
  • V g (k) is the output voltage of the grid-connected inverter at The value of the current moment.
  • the processing unit is configured to: obtain a grid phase of the grid-connected inverter based on the output voltage; and obtain the first reference current based on the grid phase.
  • a grid-connected inverter system which includes: a grid-connected inverter; and the control device as described above.
  • the actual output current of the grid-connected inverter can be closer to the expected output current.
  • an electric vehicle-to-grid (Vehicle-to-Grid, V2G) system including the above grid-connected inverter system.
  • the V2G system adopts the grid-connected inverter system as described above, it can more stably transmit the excess electric energy on the electric vehicle to the public grid.
  • a computer-readable storage medium stores instructions, and when the instructions are executed by a processor, the above-mentioned control of the grid-connected inverter can be realized.
  • FIG. 1 is a flowchart of a control method of a grid-connected inverter according to the present application
  • FIG. 2 is a working principle diagram exemplarily showing the control method of the grid-connected inverter according to the present application
  • FIG. 3 is a schematic diagram of a topology structure of an embodiment of a grid-connected inverter according to the present application
  • FIG. 4 is a flowchart of an embodiment of a control method for a grid-connected inverter according to the present application
  • FIG. 5 is a flowchart of another embodiment of the control method of the grid-connected inverter according to the present application.
  • FIG. 6 is a flowchart of another embodiment of a control method for a grid-connected inverter according to the present application.
  • FIG. 7 is a schematic diagram showing how to obtain the grid phase based on the output voltage of the grid-connected inverter
  • FIG. 8 is a structural block diagram of a control device for a grid-connected inverter according to the present application.
  • FIG. 9 is a structural block diagram showing a grid-connected inverter system according to the present application.
  • FIG. 10 is a block diagram showing the structure of the V2G system according to the present application.
  • Fig. 11(a) and Fig. 11(b) are respectively a comparison of the output signals obtained by using the conventional method to perform model predictive control on the grid-connected inverter and using the control method of the present application to perform the model predictive control on the grid-connected inverter.
  • Figures 12(a) to 12(e) show the model predictive control of the grid-connected inverter by using the conventional method and the model predictive control of the grid-connected inverter by using the control method of the present application when a voltage jump occurs.
  • FIGS 13(a) to 13(d) show the results obtained by using the conventional method to perform model predictive control on the grid-connected inverter and using the control method of the present application to perform model predictive control on the grid-connected inverter when the reference current suddenly changes.
  • a control method of a grid-connected inverter is provided.
  • control method 100 of a grid-tied inverter may include steps 110 to 160 .
  • step 110 the output voltage of the grid-connected inverter and the output current value at the current moment are obtained.
  • the output of the grid-connected inverter is three-phase alternating current, the output voltage is represented by a function V g , and the output current is represented by a function ig, and the output current value at the current moment is i g ( k), where k represents the current moment.
  • step 120 based on the output current value at the current moment, calculate N output current values at the next moment of the grid-connected inverter, and the N output current values at the next moment are the same as the grid-connected inverter.
  • N is a natural number greater than or equal to 2.
  • a first reference current is obtained based on the output voltage.
  • the first reference current is the current expected to be output by the grid-connected inverter, so it is also a three-phase alternating current like the output current, which is represented by a function i_ref1.
  • a second reference current is obtained based on the output current of the grid-connected inverter and the first reference current.
  • the second reference current is represented by a function i_ref2, which actually adds compensation to the first reference current i_ref1. Because the obtaining of the second reference current is also based on the output current of the grid-connected inverter, which is equivalent to feeding back the error and disturbance of the system to the input end, and making the final reference current contain the error and disturbance that can cancel the error and disturbance. compensation.
  • a first switch state is determined among the N switch states, and the difference between the output current value at the next moment corresponding to the first switch state and the value of the second reference current at the next moment is equal to The minimum value among the differences between the N output current values at the next moment and the value of the second reference current at the next moment.
  • step 160 the grid-connected inverter is controlled to perform power transmission at the next moment in the first switching state.
  • the above control method adopts the value of the second reference current with compensation at the next moment, and the obtained output current value at the next moment is more accurate.
  • the output current value at each next moment is predicted by the above control method, and the switch of the grid-connected inverter is set according to the switch state corresponding to the predicted output current value at the next moment. In this way, The output stability of the grid-connected inverter can be improved.
  • control method 100 will be further described below with reference to the working principle diagram and the schematic diagram of the topology structure of the grid-connected inverter.
  • the output of the grid-connected inverter is three-phase alternating current.
  • the output voltage is the voltage across the capacitor in Figure 2, available from the sensor and expressed as a function V g .
  • the output current is the branch ig in Fig. 2, that is, the difference between the grid-connected current flowing through the inductor in Fig. 2 and the current flowing through the capacitor in Fig. 2, which can be obtained through the sensor, and is represented by the function ig , the current moment's
  • the output current value is i g (k), where k represents the current moment.
  • the output voltage V g of the grid-connected inverter and the output current value ig (k) at the current moment are acquired through the sensor.
  • N output current values ig (k+1) of the grid-connected inverter at the next moment are calculated based on the obtained output current value i g (k) at the current moment.
  • the N output current values i g (k+1) at the next moment correspond one-to-one with the N switching states of the grid-connected inverter.
  • the topology of an embodiment shown in FIG. 3 is used as an example below. Describe how to obtain the N output current values ig (k+1) at the next moment.
  • FIG. 3 exemplarily shows a schematic diagram of a topology structure of an embodiment of the grid-connected inverter of the present application.
  • the grid-connected inverter is a single-phase grid-connected inverter, which includes four switches S1 , S2 , S3 and S4 , which may be, for example, MOS transistors.
  • the grid-connected inverter of the present application is not limited to such a form, and correspondingly, the switch combinations it has are not limited to the above-mentioned S1, S2, S3 and S4.
  • the grid-tied inverter of the present application may also be a three-phase grid-tied inverter, which has six switches.
  • the four switches S1, S2, S3 and S4 have a closed state (denoted as “1") and an open state (denoted as "0"), respectively.
  • the following four switch states of the grid-connected inverter can be obtained.
  • the output current value i g (k+1) at the next moment calculated based on the output current value i g (k) at the current moment is different from each other.
  • the above step 120 may include sub-step 1202 and sub-step 1204 as shown in FIG. 4 .
  • N switch output voltages are obtained respectively based on the N switch states and the input voltage of the grid-connected inverter.
  • U dc is the voltage at both ends of the battery, that is, the input voltage of the DC side of the grid-connected inverter
  • U 0 is the output voltage of the switch output after the switch combination
  • S1, S2, S3 and S4 are the closing of the corresponding switches /Off state ("closed” is recorded as “1”, “open” is recorded as “0").
  • sub-step 1204 based on the output current value i g (k) at the current moment, the value U 0 (k) of the output voltages of the N switches at the current moment, and the output voltage V g of the grid-connected inverter at the current moment.
  • the value V g (k) at the moment obtains N output current values ig (k+1) at the next moment.
  • equation (2) can be obtained.
  • L is the inductance size of the grid-connected inverter
  • i is the output current at the current moment
  • r is the internal resistance of the grid-connected inverter
  • d is a differential function.
  • the output current values i g (k+1) of the N next moments can be calculated from the above formula (4).
  • the values of U0 at the current moment (U dc , 0, -U dc and 0) corresponding to the switch states 1 to 4 listed in Table 2 can be respectively Substitute into formula (4) to obtain four output current values at the next moment i g (k+1)_1, i g (k+1)_2, i g (k+1)_3 and i g (k+1 )_4.
  • a first reference current i_ref1 is also obtained based on the output voltage V g , that is, corresponding to step 130 in the control method 100 .
  • the obtaining of the first reference current i_ref1 also needs to consider the grid phase of the grid-connected inverter, so the output voltage V g is used.
  • the above step 130 may include sub-step 1302 and sub-step 1304 as shown in FIG. 5 .
  • the grid phase of the grid-connected inverter is obtained based on the output voltage V g .
  • it can be implemented by a phase-locked loop.
  • a phase-locked loop based on a generalized second-order integrator needs to be used.
  • Figure 7 shows the phase-locked loop and the generalized second-order integrator, respectively, where the single-phase voltage component V g passes through the generalized second-order integrator to generate two quadrature voltage phasors V gsin and V gcos , which pass through the left side of dq Transformation, the voltage phasor is converted to the dq axis, and the q axis reactive power component tends to 0 through the pi controller to achieve phase locking, and the output ⁇ is the grid phase.
  • the desired output current value i is expected to be, for example, a value in the range of 0-30A.
  • a second reference current i_ref2 is obtained based on the output current i g of the grid-connected inverter and the first reference current i_ref1 , that is, corresponding to the control Step 140 in method 100 .
  • the above-mentioned step 140 may include sub-step 1402 and sub-step 1404 as shown in FIG. 6 .
  • a third-order state model is established based on the output current ig of the grid-connected inverter.
  • the LC filter included in the grid-connected inverter shown in Figure 2 is a second-order system, and the third-order state model is established on the basis of the second-order LC system. That is, the estimated system disturbance is obtained.
  • a second reference current is obtained based on the established third-order state model and the first reference current. That is, the output of the disturbance quantity obtained above is compensated to the first reference current, and the second reference current obtained thereby includes the compensation value of the disturbance quantity of the system. Feeding back the compensated reference current (ie, the second reference current) into the predictive control of the above method can significantly reduce both the THD value and the DC component injection (direct current input, DCI) value of the grid-connected current. Improve the stability and robustness of predictive control.
  • the above-mentioned expanded system model is a third-order state model, because its corresponding grid-connected inverter includes a second-order LC system, but this application is not limited to this, but can also be based on The configuration of the specific grid-connected inverter is used to establish other expansion state models. For example, if the grid-connected inverter includes a third-order LCL system, then a fourth-order state model needs to be established based on the output current i g . Similarly, the expanded system model is the disturbance of the system.
  • a second-order LC system model can be established:
  • L represents the size of the inductance in Figure 2
  • i l is the current flowing through the inductance (ie, grid-connected current)
  • S represents the closed/open state of the corresponding switch in Figure 2 ("closed” is denoted as “1”, “disconnected” is recorded as "0", if it is the topology shown in Figure 3, S can be expressed as (S1 ⁇ S3-S2 ⁇ S4)
  • U dc is the voltage across the battery in Figure 3 (also That is, the DC side input voltage of the grid-connected inverter)
  • U ac is the voltage across the capacitor in Figure 3 (that is, the AC side output voltage of the grid-connected inverter)
  • C represents the size of the capacitor in Figure 3
  • i g is the output current of the grid-connected inverter
  • d is a differential function.
  • the above-expanded X 3 is the system disturbance.
  • the output current i g can be directly regarded as the grid-connected current i l , that is, from the output current i g i g can directly obtain the third-order state model as follows:
  • u represents the control quantity for the expanded state model, that is, the grid-connected current i l flowing through the inductor, and f represents the system disturbance.
  • the output y in the above expansion state model is the reference current in the ideal state, that is, the first reference current i_ref1 without considering the system disturbance.
  • the calculated u represents the second reference current i_ref2 considering the system disturbance.
  • third-order state model and the functional expression of the second reference current i_ref2 calculated on the basis thereof are not unique. According to different grid-connected inverter structures, other forms of third-order state models can be established, and different functional expressions of the second reference current i_ref2 can be calculated accordingly.
  • the N next time can Among the N switch states corresponding to the output current value i g (k+1) at a moment, determine the corresponding switch state with the smallest difference between i g (k+1) and i_ref2 (k+1), that is, Find the switch state when i g (k+1) is closest to i_ref2(k+1).
  • switch state 1 For the grid-tied inverter example shown in Figure 3, as shown in Table 1 above, there are four switching states. According to the four switch states listed in Table 1, namely, switch state 1, switch state 2, switch state 3 and switch state 4, calculate the output current values ig (k+1) _1 , ig at the next moment respectively (k+1)_2, i g (k+1)_3 and i g (k+1)_4, and then calculate i g (k+1)_1 and i_ref2(k+1), i g (k+1) respectively )_2 and i_ref2 (k+1), ig (k+1)_3 and i_ref2 (k+1), and the difference between ig (k+1)_4 and i_ref2(k+1). Assuming that the difference between i g (k+1)_2 and i_ref2(k+1) is the smallest after calculation, switch state 2 is the switch state to be determined.
  • the switch state to be determined can be obtained by setting a cost function and optimizing the cost function.
  • the cost function can be
  • the above cost function G is optimized so that the obtained value is the smallest, and the switch state corresponding to i g (k+1) when G is optimized to the smallest value is the switch state to be determined.
  • the above cost function G amplifies the difference between the output current value i g (k+1) at the next moment in each switching state and the value i_ref2 (k+1) of the second reference current at the next moment by squaring, so that all The determined switching states more precisely match the current output by the desired grid-tied inverter.
  • cost function G is just an example, and the control method of the present application can adopt various forms of cost functions, as long as i g (k+1) which is closest to i_ref2(k+1) can be found through this cost function ) can be used.
  • switch state 2 is the switch state to be determined, then refer to Table 1 to set the four switches S1 to S4 to "closed” correspondingly at the next moment. ", "open”, “close”, “open”, thereby controlling the power transmission of the grid-connected inverter at the next moment.
  • the obtained output current value i g (k+1) at the next moment is more accurate.
  • the output current value i g (k+1) at each next moment is predicted by the above control method, and the switches of the grid-connected inverter are adjusted according to the predicted output current value i g (k+1) at the next moment. 1)
  • the corresponding switch states are set.
  • Fig. 11(a) shows the waveform simulation diagram of the output current and output voltage obtained by using the conventional method to perform model predictive control of the grid-connected inverter
  • Fig. 11(b) shows the control method of the present application to control the parallel
  • the waveform simulation diagram of the output current and output voltage obtained by the model predictive control of the grid inverter It can be clearly seen from this that the obtained output current and output voltage are more stable by using the control method of the present application.
  • Figures 12(a) to 12(e) exemplarily show the comparison of the output currents obtained by using the conventional method and the control method of the present application to perform model predictive control of the grid-connected inverter when voltage jumps occur .
  • Figure 12(a) shows that the DC side input voltage of the grid-connected inverter jumps from 380V to 500V in 0.5 seconds.
  • FIGS. 13( a ) to 13 ( d ) exemplarily show that the conventional method is used to perform model prediction control on the grid-connected inverter and the control method of the present application is used to perform model prediction on the grid-connected inverter when the reference current is abruptly changed. Controls the comparison of the THD value of the resulting output current.
  • Fig. 13(a) shows the reference current waveform when the reference current value suddenly changes from 10A to 20A at a certain moment in the conventional method
  • Fig. 13(b) shows the control method of the present application when a certain moment The reference current waveform when the reference current value (corresponding to the first reference current in this application) suddenly changes from 10A to 20A
  • Fig. 13(a) shows the reference current waveform when the reference current value suddenly changes from 10A to 20A
  • Fig. 13(b) shows the control method of the present application when a certain moment
  • the reference current waveform when the reference current value (corresponding to the first reference current in this application) suddenly changes from 10A to 20
  • control method may be implemented by executing computer instructions by a processor, and the instructions may be stored in a computer-readable storage medium.
  • the computer-readable storage medium may include hard disk drives, floppy disk drives, compact disk read/write (CD-R/W) drives, digital versatile disk (DVD) drives, flash drives, and/or solid state storage devices, among others.
  • a control device for a grid-connected inverter is also provided accordingly.
  • a control device 800 of a grid-connected inverter according to the present application is shown, which includes an acquisition unit 810 and a processing unit 820 .
  • the obtaining unit 810 is configured to obtain the output voltage of the grid-connected inverter and the output current value at the current moment.
  • the output voltage is the voltage across the capacitor in FIG. 2 , which can be obtained by a sensor and is represented by the function V g .
  • the output current is the branch ig in FIG. 2 , that is, the output current flows through the inductor in FIG.
  • the difference between the grid-connected current and the current flowing through the capacitor in Figure 2 can be obtained through the sensor, and is represented by the function i g .
  • the output current value at the current moment is i g (k), where k represents the current moment.
  • the processing unit 820 is configured to implement the following function: based on the output current value i g (k) at the current moment, calculate the output current values i g (k+1)_1, i of the grid-connected inverter at the next moment of N g (k+1)_2...i g (k+1)_N, the N output current values at the next moment correspond one-to-one with the N switching states of the grid-connected inverter, and the N is greater than equal to 2; obtain a first reference current i_ref1 based on the output voltage V g , and obtain a second reference current i_ref2 based on the output current i g of the grid-connected inverter and the first reference current i_ref1; at the N A first switch state is determined from each switch state, the output current value i g (k+1) at the next moment corresponding to the first switch state and the value i_ref2 (k+1) of the second reference current at the next moment The difference is the output current value (
  • the processing unit 820 may be configured to: establish a third-order state model based on the output current i g of the grid-connected inverter; and obtain the obtained third-order state model based on the established third-order state model and the first reference current i_ref1.
  • the second reference current i_ref2 includes the compensation value of the system disturbance.
  • the processing unit 820 may be configured to obtain N switch output voltages U 0 _1 , U 0 _2 . . . U 0 based on the N switch states and the input voltage U dc of the grid-connected inverter, respectively. _N; and based on the output current value ig (k) at the current moment, the values U 0 (k) _1 , U 0 (k)_2 . . .
  • the N output current values ig (k+1)_1, ig (k+1)_2 . . . ig (k+1) _N can be calculated by the above formula (10) get.
  • the processing unit 820 is configured to: obtain the grid phase ⁇ of the grid-connected inverter based on the output voltage V g ; and obtain the first reference current i_ref1 based on the grid phase ⁇ .
  • control device 800 can implement the control method according to the present application as described above. Many of the above-mentioned design concepts and details applicable to the control method of the present application are also applicable to the above-mentioned control device 800, and the same beneficial technical effects can be obtained, which will not be repeated here.
  • a grid-connected inverter system 900 is also provided, as shown in FIG. 9 , which includes a grid-connected inverter 910 and a control device 920 .
  • This control device 920 corresponds exactly to the control device 800 described above.
  • the acquisition unit 922 in the control device 920 corresponds to the acquisition unit 810 as described above, and is used to acquire the output voltage V g , the output current ig and the output current value ig (k) of the grid-connected inverter 910 at the current moment.
  • the processing unit 924 in the control device 920 determines the switching state that the grid-connected inverter 910 should be in at the next moment through a series of processing and operations, and makes the grid-connected inverter The controller 910 performs power transmission at the next moment in this switch state.
  • the grid-connected inverter system 900 has excellent robustness and stability, and the THD value and DCI value of the grid-connected current are significantly reduced compared with the prior art.
  • a V2G system 1000 is also provided, as shown in FIG. 10 , which includes a grid-connected inverter system 1010 , and the grid-connected inverter system 1010 corresponds to the above-mentioned grid-connected inverter system.
  • the controller system 900 is implemented, so that the power transmission from the car battery 1100 to the public grid 1200 can be realized more stably and accurately.

Abstract

本申请提供了一种并网逆变器的控制方法和装置。所述控制方法包括：获取并网逆变器的输出电压和当前时刻的输出电流值；基于当前时刻的输出电流值计算与N个开关状态一一对应的N个下一时刻的输出电流值，N为大于等于2的自然数；基于输出电压获取第一参考电流；基于输出电流和第一参考电流获取第二参考电流；确定第一开关状态，所述第一开关状态对应的下一时刻的输出电流值与第二参考电流在下一时刻的值的差值为所述N个下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值中的最小值；以及控制所述并网逆变器在所述第一开关状态下进行下一时刻的电能传输。

Description

并网逆变器的控制方法和装置
相关申请的交叉引用
本申请要求享有于2020年11月17日提交的名称为“并网逆变器的控制方法和装置”的中国专利申请202011284302.8的优先权，该申请的全部内容通过引用并入本文中。
技术领域
本申请涉及电力技术领域，特别地涉及对并网逆变器进行控制的方法和装置。
背景技术
近年来，作为风电、光伏等新能源发电系统并网运行的关键设备，并网逆变器已得到广泛应用。如何优化并网逆变器的控制策略并提高并网逆变器的运行可靠性，已成为当前的研究热点。
随着数字信号处理器运算速度的提高，出现了一些新型的智能控制方法，如模糊控制、自适应控制、滑动模式控制、模型预测控制等。其中，模型预测控制因为具有动态响应快等优点，已成为当今并网逆变器预测控制领域的主要研究方向。
然而，现网中总会不可避免地出现扰动，从而导致电能传输效率不高。
发明内容
本申请提供一种对并网逆变器进行控制的方法和装置，提高了系统的电能传输效率。
根据本申请的第一方面，提供一种并网逆变器的控制方法，包括：获取并网逆变器的输出电压和当前时刻的输出电流值；基于所述当前时刻的输出电流值计算所述并网逆变器的N个下一时刻的输出电流值，所述N个下一时刻的输出电流值与所述并网逆变器的N个开关状态一一对应，所述N大于等于2；基于所述输出电压获取第一参考电流；基于所述并网逆变器的输出电流和所述第一参考电流获取第二参考电流；在所述N个开关状态中确定第一开关状态，所述第一开关状态对应的下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值为所述N个下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值中的最小值；以及控制所述并网逆变器在所述第一开关状态下进行下一时刻的电能传输。
根据本申请的第二方面，提供一种并网逆变器的控制装置，包括获取单元和处理单元。所述获取单元用于获取并网逆变器的输出电压和当前时刻的输出电流值。所述处理单元配置为：基于当前时刻的输出电流值计算并网逆变器的N个下一时刻的输出电流值，所述N个下一时刻的输出电流值与所述并网逆变器的N个开关状态一一对应，所述N大于等于2；基于所述输出电压获取第一参考电流，并基于所述并网逆变器的输出电流和所述第一参考电流获取第二参考电流；在所述N个开关状态中确定第一开关状态，所述第一开关状态对应的下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值为所述N个下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值中的最小值；和控制所述并网逆变器在所述第一开关状态下进行下一时刻的电能传输。
由于上述用来确定第一开关状态的第二参考电流在原来第一参考电流的基础上还考虑了并网逆变器的输出电流，这相当于将带扰动量的输出重新反馈回控制系统，因此所得到的第二参考电流是包含了对扰动量进行补充的参考电流。这样确定的第一开关状态无疑可以更加精确地对并网逆变器的输出电流进行控制，从而大大提高了整个控制系统的鲁棒性。
可选地，上述处理单元可被配置为：基于所述并网逆变器的输出电流建立三阶状态模型；以及基于所建立的三阶状态模型和所述第一参考电流获取所述第二参考电流，所述第二参考电流包含系统扰动量的补偿值。
所述三阶状态模型为：
Figure PCTCN2021075979-appb-000001
其中，X 1表示所述并网逆变器的输出电流i g，X 2为X 1的微分，X 3为X 2的微分且表示所述系统扰动量，所述第二参考电流通过以下公式求得：
Figure PCTCN2021075979-appb-000002
其中，u表示所述第二参考电流，i ref1为所述第一参考电流，
Figure PCTCN2021075979-appb-000003
k d＝2w c，w c为所述并网逆变器的截止频率，
Figure PCTCN2021075979-appb-000004
L为所述并网逆变器中的电感，C为所述并网逆变器中的电容。
可选地，上述处理单元可被配置为：基于所述N个开关状态和所述并网逆变器的输入电压分别得到N个开关输出电压；以及基于所述当前时刻的输出电流值、所述N个开关输出电压在当前时刻的值和所述并网逆变器的输出电压在当前时刻的值得到所述N个下一时刻的输出电流值。
所述N个下一时刻的输出电流值i g(k+1)由以下公式计算得到：
Figure PCTCN2021075979-appb-000005
其中，k代表当前时刻，T s为所述并网逆变器的控制频率，r为所述并网逆变器的内阻，L为所述并网逆变器中的电感，i g(k)为所述N个当前时刻的输出电流值， U 0(k)为所述N个开关输出电压在当前时刻的值，V g(k)为所述并网逆变器的输出电压在当前时刻的值。
可选地，所述处理单元被配置为：基于所述输出电压得到所述并网逆变器的电网相位；以及基于所述电网相位获取所述第一参考电流。
根据本申请的第三方面，提供一种并网逆变器系统，其包括：并网逆变器；和如上所述的控制装置。
所述并网逆变器系统因为用如上所述的控制装置对并网逆变器进行控制，因此可使并网逆变器实际输出的电流更接近所期望输出的电流。
根据本申请的第四方面，提供一种电动汽车到电网(Vehicle-to-Grid，V2G)系统，其包括如上所述的并网逆变器系统。
所述V2G系统因为采用了如上所述的并网逆变器系统，因此可以更稳定地将电动汽车上多余的电能传输给公共电网。
根据本申请的第五方面，提供一种计算机可读存储介质，所述计算机可读存储介质存储有指令，所述指令被处理器执行时可实现如上所述的对并网逆变器进行控制的方法。
通过下面结合附图的详细描述，本申请的上述诸个方面的特征将变得更为清楚。
附图说明
为了更清楚地说明本申请实施例的技术方案，下面将对本申请实施例中所需要使用的附图作简单地介绍，显而易见地，下面所描述的附图仅仅是本申请的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据附图获得其他的附图。
图1为根据本申请的并网逆变器的控制方法的流程图；
图2为示例性地示出根据本申请的并网逆变器的控制方法的工作原理图；
图3为根据本申请的并网逆变器的一个实施方式的拓扑结构示意图；
图4为根据本申请的并网逆变器的控制方法的一个实施方式的流程图；
图5为根据本申请的并网逆变器的控制方法的另一个实施方式的流程图；
图6为根据本申请的并网逆变器的控制方法的另一个实施方式的流程图；
图7为示出了如何基于并网逆变器的输出电压得到电网相位的原理图；
图8为根据本申请的并网逆变器的控制装置的结构框图；
图9为示出了根据本申请的并网逆变器系统的结构框图；
图10为示出了根据本申请的V2G系统的结构框图；
图11(a)和图11(b)分别为采用常规方法对并网逆变器进行模型预测控制和采用本申请的控制方法对并网逆变器进行模型预测控制所得到的输出信号的比较示意图；
图12(a)至图12(e)示出了发生电压跳变时采用常规方法对并网逆变器进行模型预测控制和采用本申请的控制方法对并网逆变器进行模型预测控制所得到的输出 电流的比较示意图；以及
图13(a)至图13(d)示出了参考电流突变时采用常规方法对并网逆变器进行模型预测控制和采用本申请的控制方法对并网逆变器进行模型预测控制所得到的输出电流的总谐波失真(total harmonic distortion,THD)值的比较示意图。
具体实施方式
以下将描述本申请的具体实施方式。在本申请中，如果没有特别的说明，本文所提到的所有实施方式以及优选实施方式可以相互组合形成新的技术方案，本文所提到的所有技术特征以及优选特征可以相互组合形成新的技术方案。
此外，如果没有特别的说明，本文所提到的“包括”和“包含”表示开放式，也可以是封闭式。例如，所述“包括”和“包含”可以表示还可以包括或包含没有列出的其他元件、步骤或组分，也可以仅包括或包含列出的元件、步骤或组分。在本文的描述中，除非另有说明，“以上”、“以下”为包含本数，“一种或几种”中“几种”的含义是两种及两种以上。
根据本申请的实施例，提供一种并网逆变器的控制方法。
参考图1，其中示出了根据本申请的并网逆变器的控制方法100。该控制方法100可包括步骤110至步骤160。
如图1所示，在步骤110，获取并网逆变器的输出电压和当前时刻的输出电流值。所述并网逆变器的输出为三相交流电，输出电压以函数V g表示，输出电流以函数i g表示，当前时刻的输出电流值即为i g(k)，其中k代表当前时刻。
在步骤120，基于所述当前时刻的输出电流值计算所述并网逆变器的N个下一时刻的输出电流值，所述N个下一时刻的输出电流值与所述并网逆变器的N个开关状态一一对应，其中N为大于等于2的自然数。
在步骤130，基于所述输出电压获取第一参考电流。该第一参考电流就是期望所述并网逆变器输出的电流，因此其也跟输出电流一样，是一个三相交流电，以函数i_ref1表示。
在步骤140，基于所述并网逆变器的输出电流和所述第一参考电流获取第二参考电流。所述第二参考电流以函数i_ref2表示，其实际上是在第一参考电流i_ref1的基础上加入了补偿。因为该第二参考电流的得到还要基于并网逆变器的输出电流，这相当于将系统的误差和扰动反馈回输入端，并使最后得到的参考电流包含了能抵消所述误差和扰动的补偿。
在步骤150，在所述N个开关状态中确定第一开关状态，所述第一开关状态对应的下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值为所述N个下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值中的最小值。
在步骤160，控制所述并网逆变器在所述第一开关状态下进行下一时刻的电能传输。
上述控制方法在预测下一时刻的输出电流值时，采用了带有补偿的第二参考电流在下一时刻的值，由此得到的下一时刻的输出电流值更加精确。通过上述控制方法对每一个下一时刻的输出电流值进行预测，并将并网逆变器的开关按照与所预测的下一时刻的输出电流值相对应的开关状态进行设置，以此方式，可以提高并网逆变器的输出稳定性。
以下将结合并网逆变器的工作原理图以及拓扑结构示意图来进一步描述根据本申请的控制方法100。
如图2中的并网逆变器的工作原理图所示，所述并网逆变器的输出为三相交流电。输出电压为图2中电容两端的电压，可通过传感器获得，以函数V g表示。输出电流为图2中的分支i g，也即流过图2中电感的并网电流与流过图2中电容的电流的差值，可通过传感器获得，以函数i g表示，当前时刻的输出电流值即为i g(k)，其中k代表当前时刻。在步骤110，通过传感器获取并网逆变器的输出电压V g和当前时刻的输出电流值i g(k)。
在步骤120，基于上述获得的当前时刻的输出电流值i g(k)计算所述并网逆变器的N个下一时刻的输出电流值i g(k+1)。所述N个下一时刻的输出电流值i g(k+1)与所述并网逆变器的N个开关状态一一对应，以下以图3所示的一个实施方式的拓扑结构来举例说明如何得到所述N个下一时刻的输出电流值i g(k+1)。
图3示例性地示出了本申请并网逆变器的一个实施方式的拓扑结构示意图。该并网逆变器为单相并网逆变器，其包括四个开关S1、S2、S3和S4，它们可以比如是MOS管。需要特别说明的是，本申请的并网逆变器并不仅限于这样的形式，相应地，所具有的开关组合也并不仅限于上述的S1、S2、S3和S4。例如，本申请的并网逆变器也可以是三相并网逆变器，其具有六个开关。
对于图3所示的并网逆变器，其中的四个开关S1、S2、S3和S4分别具有闭合状态(记为“1”)和断开状态(记为“0”)，将这些状态进行排列组合，可以得到并网逆变器的如下四种开关状态。每一种开关状态下，基于当前时刻的输出电流值i g(k)所计算出的下一时刻的输出电流值i g(k+1)是彼此不同的。
表1：开关状态
编号 S1 S2 S3 S4
开关状态1 1 0 0 1
开关状态2 1 0 1 0
开关状态3 0 1 1 0
开关状态4 0 1 0 1
可选地，上述步骤120可如图4所示地包括子步骤1202和子步骤1204。
在子步骤1202，基于所述N个开关状态和所述并网逆变器的输入电压分别得到N个开关输出电压。
参考图3，对于该图所示的并网逆变器的拓扑结构，开关状态、并网逆变器的输入电压和开关输出电压的关系如下：
U 0＝(S1×S3-S2×S4)×U dc    公式(1)
其中，U dc为电池两端的电压，也即，并网逆变器的直流侧输入电压，U 0为经过开关组合后输出的开关输出电压，S1、S2、S3和S4为各对应开关的闭合/断开状态(“闭合”记为“1”，“断开”记为“0”)。由此可得到如下表2所示的关系：
表2
编号 S1 S2 S3 S4 U0
开关状态1 1 0 0 1 U dc
开关状态2 1 0 1 0 0
开关状态3 0 1 1 0 -U dc
开关状态4 0 1 0 1 0
可以理解的是，上述表2的关系是基于图3所示的拓扑结构。如果并网逆变器采用的是其它形式的拓扑结构，那么开关状态与开关输出电压的关系将会不同。
在子步骤1204，基于当前时刻的输出电流值i g(k)、所述N个开关输出电压在当前时刻的值U 0(k)和所述并网逆变器的输出电压V g在当前时刻的值V g(k)得到N个下一时刻的输出电流值i g(k+1)。
可选地，根据基尔霍夫电压定律，可以得到公式(2)。
Figure PCTCN2021075979-appb-000006
其中，L为所述并网逆变器中的电感大小，i为当前时刻的输出电流，r为所述并网逆变器的内阻，d为微分函数。
再将根据欧拉公式得到的式(3)代入上述公式(2)，便可得到并网逆变器在下一时刻的输出电流值i g(k+1)，其中，T s为所述并网逆变器的控制频率。
Figure PCTCN2021075979-appb-000007
所述N个下一时刻的输出电流值i g(k+1)可由上述公式(4)计算得到。具体地，比如对于图3所示的拓扑结构，可分别将上述表2中所列出的开关状态1～4所对应的U0在当前时刻的值(U dc、0、-U dc和0)代入公式(4)，从而得到四个下一时刻的输出电流值i g(k+1)_1、i g(k+1)_2、i g(k+1)_3和i g(k+1)_4。
在计算得到所述并网逆变器的N个下一时刻的输出电流值i g(k+1)的同时，如图2所示，还基于所述输出电压V g获取第一参考电流i_ref1，也即对应于控制方法100中的步骤130。该第一参考电流i_ref1的获得还需要考虑到并网逆变器的电网相位，因此要用到所述输出电压V g。
可选地，上述步骤130可如图5所示地包括子步骤1302和子步骤1304。
在子步骤1302，基于所述输出电压V g得到所述并网逆变器的电网相位。例如，可通过锁相环来实现。可选地，对于图3所示的单相并网逆变器，因为输出的并网电压只有一个电压分量，需要采用基于广义二阶积分器的锁相环。图7中分别示出了锁相环和广义二阶积分器，其中，单相电压分量V g经过广义二阶积分器，生成两个正交的电 压相量V gsin和V gcos，通过dq左边变换，电压相量转换到dq轴，经过pi控制器将q轴无功分量趋为0，实现锁相，输出θ即为电网相位。
在子步骤1304，基于所述电网相位获取所述第一参考电流。具体地，将上述得到的电网相位θ与所述并网逆变器期望输出的电流值i 期望相乘，即可得到所述第一参考电流i_ref1＝i 期望*sinθ。所述期望输出的电流值i 期望可例如为在0～30A范围内的值。
在获得了所述第一参考电流之后，如图2所示，基于所述并网逆变器的输出电流i g和所述第一参考电流i_ref1获取第二参考电流i_ref2，即，对应于控制方法100中的步骤140。
可选地，上述步骤140可如图6所示地包括子步骤1402和子步骤1404。
在子步骤1402，基于并网逆变器的输出电流i g建立三阶状态模型。图2所示的并网逆变器包含的LC滤波器为二阶系统，所述三阶状态模型在该二阶LC系统的基础上建立，扩张出的系统模型即为系统的扰动量，也即得到了估计的系统扰动量。
在子步骤1404，基于所建立的三阶状态模型和第一参考电流获取第二参考电流。也即，将上述得到的扰动量输出补偿给第一参考电流，由此得到的第二参考电流包含了系统扰动量的补偿值。将补偿后的参考电流(即，第二参考电流)反馈到上述方法的预测控制中，可以使得并网电流的THD值和直流分量注入(direct current input，DCI)值都显著减小，从而大大提高预测控制的稳定性和鲁棒性。
需要特别说明的是，上述扩张后的系统模型为三阶状态模型，是因为其对应的并网逆变器包含的是二阶LC系统，但本申请并不局限于此，而是还可以根据具体的并网逆变器的配置来建立其它的扩张状态模型。例如，如果并网逆变器包含的是三阶LCL系统，那么相应地就需要基于输出电流i g建立四阶状态模型，同样地，该扩张出的系统模型即为系统的扰动量。
可选地，根据图2所示的并网逆变器，可以建立二阶LC系统模型：
Figure PCTCN2021075979-appb-000008
上述公式中，L表示图2中电感的大小，i l为流过所述电感的电流(即，并网电流)，S表示图2中对应开关的闭合/断开状态(“闭合”记为“1”，“断开”记为“0”，如果是图3所示的拓扑结构，S可以表述为(S1×S3-S2×S4)，U dc为图3中电池两端的电压(也即，并网逆变器的直流侧输入电压)，U ac为图3中电容两端的电压(也即，并网逆变器的交流侧输出电压)，C表示图3中电容的大小，i g为所述并网逆变器的输出电流，d为微分函数。
在上述公式(5)的基础上进行模型扩张，特别令并网电流i l为X 1，X 2为X 1的微分，X 3为X 2的微分，由此得到：
Figure PCTCN2021075979-appb-000009
上述扩张出的X 3即为系统扰动量。
由于图2中流过电容的电流远远小于并网逆变器的输出电流i g，甚至可以忽略不计，因此可以直接将输出电流i g看作是并网电流i l，即从所述输出电流i g就可以直接得到三阶状态模型如下：
Figure PCTCN2021075979-appb-000010
同时，对上述公式(6)进行矩阵变换后得到扩张状态的模型：
Figure PCTCN2021075979-appb-000011
其中，
Figure PCTCN2021075979-appb-000012
u表示对所述扩张状态模型的控制量，即，流过电感的并网电流i l，f表示系统扰动。
上述扩张状态模型中的输出y就是理想状态下的参考电流，即，不考虑系统扰动的第一参考电流i_ref1。同时，如前面所述地，由于图2中流过电容的电流远远小于并网逆变器的输出电流i g，甚至可以忽略不计，因此可以直接将输出电流i g看作是并网电流i l。由此，当p＝1时，y＝X 1＝i_ref1，经过计算可得到
Figure PCTCN2021075979-appb-000013
其中，
Figure PCTCN2021075979-appb-000014
k d＝2w c，w c为所述并网逆变器的截止频率。该计算出的u即表示考虑了系统扰动的第二参考电流i_ref2。
需要特别说明的是，上述三阶状态模型和在此基础上计算出的第二参考电流i_ref2的函数表达式并不是唯一的。根据不同的并网逆变器的结构，可以建立其它形式的三阶状态模型，并相应地计算出不同的第二参考电流i_ref2的函数表达式。
由图2可知，在获得了所述N个下一时刻的输出电流值i g(k+1)和所述第二参考电流i_ref2的函数表达式之后，接下来就可以在与这N个下一时刻的输出电流值i g(k+1)相对应的N个开关状态中确定i g(k+1)与i_ref2(k+1)的差值为最小的那个对应的开关状态，即，找出i g(k+1)与i_ref2(k+1)最接近时的开关状态。
对于图3所示的并网逆变器示例，如前面表1所示，其共有四个开关状态。按照表1所列出的四个开关状态，即，开关状态1、开关状态2、开关状态3和开关状态4，分别计算下一时刻的输出电流值i g(k+1)_1、i g(k+1)_2、i g(k+1)_3和i g(k+1)_4，再分别计算i g(k+1)_1与i_ref2(k+1)、i g(k+1)_2与i_ref2(k+1)、i g(k+1)_3与i_ref2(k+1)以及i g(k+1)_4与i_ref2(k+1)的差值。假设最后经计算确定i g(k+1)_2与i_ref2(k+1)的差值最小，那么开关状态2就是所要确定的开关状态。
可选地，可通过设置一个代价函数并对该代价函数进行优化来得到所要确定的开关状态。具体地，该代价函数可以为
G＝(i g(k+1)-i_ref2(k+1)) 2    公式(9)
上对上述代价函数G进行优化以使得到的值为最小，G被优化为最小值时的i g(k+1)所对应的开关状态即为所要确定的开关状态。上述代价函数G通过平方将各个开关状态下的下一时刻的输出电流值i g(k+1)与第二参考电流在下一时刻的值i_ref2(k+1)的差放大，从而可以使得所确定的开关状态更精确地匹配期望并网逆变器输出的电流。需要特别说明的是，上述代价函数G只是一个示例，本申请的控制方法可以采用多种形式的代价函数，只要通过这个代价函数可以找到最接近i_ref2(k+1)的i g(k+1)即可。
对于图3所示的并网逆变器示例，按照前面所假设地，开关状态2就是所要确定的开关状态，那么可参考表1在下一时刻将四个开关S1至S4对应地设置为“闭合”、“断开”、“闭合”、“断开”，由此控制所述并网逆变器在下一时刻的电能传输。
至此，描述了根据本申请的并网逆变器的控制方法。一直以来，对于常规的模型预测控制，由于现网中总会不可避免地出现扰动，包括系统中元器件的给定值与实际值之间存在的误差，以及温度升高所引起的参数变化等外扰，这些都会使模型预测控制系统中的模型参数不准确，出现模型失配的情况，从而导致控制不精确、输出电流质量和电能传输效率都不高的结果。但本申请的并网逆变器的控制方法在预测下一时刻的输出电流值i g(k+1)时，采用了带有补偿的第二参考电流在下一时刻的值i_ref2(k+1)，由此得到的下一时刻的输出电流值i g(k+1)更加精确。通过上述控制方法对每一个下一时刻的输出电流值i g(k+1)进行预测，并将并网逆变器的开关按照与所预测的下一时刻的输出电流值i g(k+1)相对应的开关状态进行设置，以此方式，可以克服常规模型预测控制中由于扰动和误差所带来的各种问题，从而提高控制的精确度以及并网逆变器输出电流的质量，进而提高并网逆变器的输出稳定性。
图11(a)示出了采用常规方法对并网逆变器进行模型预测控制所得到的输出电流和输出电压的波形仿真图，图11(b)示出了采用本申请的控制方法对并网逆变器进行模型预测控制所得到的输出电流和输出电压的波形仿真图。从中可以清楚地看到，采用本申请的控制方法，所得到的输出电流和输出电压更加稳定。
图12(a)至图12(e)示例性地示出了当发生电压跳变时采用常规方法和采用本申请的控制方法对并网逆变器进行模型预测控制所得到的输出电流的比较。其中，图12(a)显示在0.5秒并网逆变器的直流侧输入电压由380V跳变到500V。在此情况下，从图12(b)(采用常规方法对并网逆变器进行模型预测控制)和图12(c)(采用本申请的控制方法对并网逆变器进行模型预测控制)所示的输出电流的波形仿真图可以清楚地看到，采用本申请的控制方法所得到的输出电流的变化范围小，稳定性好，鲁棒性强。同时，从图12(d)和图12(e)所示的THD值可以清楚地看到，采用本申请的控制方法所得到的并网电流的THD明显减小。
图13(a)至图13(d)示例性地示出了参考电流突变时采用常规方法对并网逆变器进行模型预测控制和采用本申请的控制方法对并网逆变器进行模型预测控制所得到的输出电流的THD值的比较。其中，图13(a)示出了常规方法中当某一时刻参考电流值从10A突变到20A时的参考电流波形，图13(b)示出了在本申请的控制方法中当某一时刻参考电流值(对应于本申请中的第一参考电流)从10A突变到20A时的参考电流波形，图13(c)为图13(a)情况下对应的THD波形，图13(d)为图13 (b)情况下对应的THD波形。从中可以清楚地看到，采用本申请的控制方法对并网逆变器进行模型预测控制，因为真正用来进行预测控制的是根据上述突变产生的带补偿的参考电流，所以在发生同等参考电流突变的条件下，相比常规方法，通过本申请的控制方法所得到的输出电流的THD显著减小，输出波形的质量也更好。
此外，上述根据本申请的控制方法可以通过由处理器执行计算机指令来实现，该指令可存储在计算机可读存储介质中。所述计算机可读存储介质可以包括硬盘驱动器、软盘驱动器、光盘读/写(CD-R/W)驱动器、数字通用磁盘(DVD)驱动器、闪存驱动器和/或固态存储装置等。
根据本申请的实施例，还相应地提供一种并网逆变器的控制装置。
参考图8，其中示出了根据本申请的并网逆变器的控制装置800，其包括获取单元810和处理单元820。
获取单元810用于获取并网逆变器的输出电压和当前时刻的输出电流值。如上所述地，所述输出电压为图2中电容两端的电压，可通过传感器获得，以函数V g表示，所述输出电流为图2中的分支i g，也即流过图2中电感的并网电流与流过图2中电容的电流的差值，可通过传感器获得，以函数i g表示，当前时刻的输出电流值即为i g(k)，其中k代表当前时刻。
处理单元820则被配置为实现以下的功能：基于当前时刻的输出电流值i g(k)计算并网逆变器的N个下一时刻的输出电流值i g(k+1)_1、i g(k+1)_2……i g(k+1)_N，所述N个下一时刻的输出电流值与所述并网逆变器的N个开关状态一一对应，所述N大于等于2；基于所述输出电压V g获取第一参考电流i_ref1，并基于所述并网逆变器的输出电流i g和所述第一参考电流i_ref1获取第二参考电流i_ref2；在所述N个开关状态中确定第一开关状态，所述第一开关状态对应的下一时刻的输出电流值i g(k+1)与所述第二参考电流在下一时刻的值i_ref2(k+1)的差值为所述N个下一时刻的输出电流值(i g(k+1)_1、i g(k+1)_2……i g(k+1)_N)与所述第二参考电流在下一时刻的值i_ref2(k+1)的差值中的最小值；以及控制所述并网逆变器在所述第一开关状态下进行下一时刻的电能传输。
可选地，处理单元820可被配置为：基于所述并网逆变器的输出电流i g建立三阶状态模型；以及基于所建立的三阶状态模型和所述第一参考电流i_ref1获取所述第二参考电流i_ref2，所述第二参考电流包含系统扰动量的补偿值。
同样可选地，所述三阶状态模型可以为上述公式(7)，所述第二参考电流i_ref2可通过
Figure PCTCN2021075979-appb-000015
其中，
Figure PCTCN2021075979-appb-000016
k d＝2w c，w c为所述并网逆变器的截止频率，
Figure PCTCN2021075979-appb-000017
L为所述并网逆变器中的电感，C为所述并网逆变器中的电容。
可选地，处理单元820可被配置为：基于所述N个开关状态和所述并网逆变器的输入电压U dc分别得到N个开关输出电压U 0_1、U 0_2……U 0_N；以及基于所述当前时刻的输出电流值i g(k)、所述N个开关输出电压在当前时刻的值U 0(k)_1、U 0(k)_2……U 0(k)_N和所述并网逆变器的输出电压在当前时刻的值V g(k)得到所述N个下一时刻的输出电流值i g(k+1)_1、i g(k+1)_2……i g(k+1)_N。
同样可选地，所述N个下一时刻的输出电流值i g(k+1)_1、i g(k+1)_2……i g(k+1)_N可由上述公式(10)计算得到。
可选地，处理单元820被配置为：基于所述输出电压V g得到所述并网逆变器的电网相位θ；以及基于所述电网相位θ获取所述第一参考电流i_ref1。
上述控制装置800可以实现如前面所述地根据本申请的控制方法。上述在本申请的控制方法中适用的很多设计构思和细节同样适用于上述控制装置800，且可以得到相同的有益技术效果，此处不再赘述。
根据本申请的实施例，还提供一种并网逆变器系统900，如图9所示，其包括并网逆变器910和控制装置920。该控制装置920正对应于如上所述的控制装置800。控制装置920中的获取单元922正对应于如上所述的获取单元810，用于获取并网逆变器910的输出电压V g、输出电流i g及当前时刻的输出电流值i g(k)，再通过控制装置920中的处理单元924(对应于如上所述的处理单元820)经过一系列处理和运算确定下一时刻并网逆变器910应处于的开关状态，并使并网逆变器910在这一开关状态下进行下一时刻的电能传输。该并网逆变器系统900具有极佳的鲁棒性和稳定性，并网电流的THD值和DCI值相比现有技术都显著减小。
根据本申请的实施例，还提供一种V2G系统1000，如图10所示，其包括并网逆变器系统1010，该并网逆变器系统1010正对应于如上所述的并网逆变器系统900，从而可以更加稳定且精确地实现从汽车电池1100向公共电网1200的电力传输。
以上通过一些示例性实施例对本申请的进行了描述。然而，应该理解的是，在不脱离本申请精神和范围的情况下，还可以对上述示例性实施例做出各种修改。例如，如果所描述的技术以不同的顺序执行和/或如果所描述的系统、架构、设备或电路中的组件以不同方式被组合和/或被另外的组件或其等同物替代或补充，也可以实现合适的结果，那么相应地，这些修改后的其它实施方式也落入权利要求书的保护范围内。

Claims (12)

  1. 一种并网逆变器的控制方法，包括：
    获取并网逆变器的输出电压和当前时刻的输出电流值；
    基于所述当前时刻的输出电流值计算所述并网逆变器的N个下一时刻的输出电流值，所述N个下一时刻的输出电流值与所述并网逆变器的N个开关状态一一对应，所述N为大于等于2的自然数；
    基于所述输出电压得到所述并网逆变器的电网相位，并基于所述电网相位获取第一参考电流；
    基于所述并网逆变器的输出电流建立三阶状态模型；
    基于所述三阶状态模型和所述第一参考电流获取第二参考电流，所述第二参考电流包含系统扰动量的补偿值；
    在所述N个开关状态中确定第一开关状态，所述第一开关状态对应的下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值为所述N个下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值中的最小值；以及
    控制所述并网逆变器在所述第一开关状态下进行下一时刻的电能传输。
  2. 如权利要求1所述的控制方法，其特征在于，所述三阶状态模型为：
    Figure PCTCN2021075979-appb-100001
    其中，X 1表示所述并网逆变器的输出电流i g，X 2为X 1的微分，X 3为X 2的微分且表示所述系统扰动量，
    所述第二参考电流通过以下公式求得：
    Figure PCTCN2021075979-appb-100002
    其中，u表示所述第二参考电流，i ref1为所述第一参考电流，
    Figure PCTCN2021075979-appb-100003
    k d＝2w c，w c为所述并网逆变器的截止频率，
    Figure PCTCN2021075979-appb-100004
    L为所述并网逆变器中的电感，C为所述并网逆变器中的电容。
  3. 如权利要求1或2所述的控制方法，其特征在于，所述基于所述当前时刻的输出电流值计算所述并网逆变器的N个下一时刻的输出电流值包括：
    基于所述N个开关状态和所述并网逆变器的输入电压分别得到N个开关输出电压；以及
    基于所述当前时刻的输出电流值、所述N个开关输出电压在当前时刻的值和所述并网逆变器的输出电压在当前时刻的值得到所述N个下一时刻的输出电流值。
  4. 如权利要求3所述的控制方法，其特征在于，所述N个下一时刻的输出电流值i g(k+1)由以下公式计算得到：
    Figure PCTCN2021075979-appb-100005
    其中，k代表当前时刻，T s为所述并网逆变器的控制频率，r为所述并网逆变器的内阻，L为所述并网逆变器中的电感，i g(k)为所述N个当前时刻的输出电流值，U 0(k)为所述N个开关输出电压在当前时刻的值，V g(k)为所述并网逆变器的输出电压在当前时刻的值。
  5. 一种并网逆变器的控制装置，包括：
    获取单元，用于获取并网逆变器的输出电压和当前时刻的输出电流值；以及
    处理单元，配置为：
    基于当前时刻的输出电流值计算并网逆变器的N个下一时刻的输出电流值，所述N个下一时刻的输出电流值与所述并网逆变器的N个开关状态一一对应，所述N为大于等于2的自然数；
    基于所述输出电压得到所述并网逆变器的电网相位，并基于所述电网相位获取第一参考电流；
    基于所述并网逆变器的输出电流建立三阶状态模型，并基于所述三阶状态模型和所述第一参考电流获取第二参考电流，所述第二参考电流包含系统扰动量的补偿值；
    在所述N个开关状态中确定第一开关状态，所述第一开关状态对应的下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值为所述N个下一时刻的输出电流值与所述第二参考电流在下一时刻的值的差值中的最小值；和
    控制所述并网逆变器在所述第一开关状态下进行下一时刻的电能传输。
  6. 如权利要求5所述的控制装置，其特征在于，所述三阶状态模型为：
    Figure PCTCN2021075979-appb-100006
    其中，X 1表示所述并网逆变器的输出电流i g，X 2为X 1的微分，X 3为X 2的微分且表示所述系统扰动量，
    所述第二参考电流通过以下公式求得：
    Figure PCTCN2021075979-appb-100007
    其中，u表示所述第二参考电流，i ref1为所述第一参考电流，
    Figure PCTCN2021075979-appb-100008
    k d＝2w c，w c为所述并网逆变器的截止频率，
    Figure PCTCN2021075979-appb-100009
    L为所述并网逆变器中的电感，C为所述并网逆变器中的电容。
  7. 如权利要求5或6所述的控制装置，其特征在于，所述处理单元被配置为：
    基于所述N个开关状态和所述并网逆变器的输入电压分别得到N个开关输出电压；以及
    基于所述当前时刻的输出电流值、所述N个开关输出电压在当前时刻的值和所述并网逆变器的输出电压在当前时刻的值得到所述N个下一时刻的输出电流值。
  8. 如权利要求7所述的控制装置，其特征在于，所述N个下一时刻的输出电流值i g(k+1)由以下公式计算得到：
    Figure PCTCN2021075979-appb-100010
    其中，k代表当前时刻，T s为所述并网逆变器的控制频率，r为所述并网逆变器的内阻，L为所述并网逆变器中的电感，i g(k)为所述N个当前时刻的输出电流值，U 0(k)为所述N个开关输出电压在当前时刻的值，V g(k)为所述并网逆变器的输出电压在当前时刻的值。
  9. 如权利要求5所述的控制装置，其特征在于，所述处理单元被配置为：
    基于所述输出电压得到所述并网逆变器的电网相位；以及
    基于所述电网相位获取所述第一参考电流。
  10. 一种并网逆变器系统，包括：
    并网逆变器；和
    如权利要求5至9中任一项所述的控制装置。
  11. 一种V2G系统，包括如权利要求10所述的并网逆变器系统。
  12. 一种计算机可读存储介质，所述计算机可读存储介质存储有指令，所述指令被处理器执行时实现如权利要求1至4中任一项所述的方法。
PCT/CN2021/075979 2020-11-17 2021-02-08 并网逆变器的控制方法和装置 WO2022105068A1 (zh)

Priority Applications (3)

Application Number Priority Date Filing Date Title
ES21745203T ES2936309T3 (es) 2020-11-17 2021-02-08 Método y dispositivo de control para un inversor conectado a red
EP21745203.6A EP4030611B1 (en) 2020-11-17 2021-02-08 Control method and device for grid-connected inverter
US17/485,554 US11251720B1 (en) 2020-11-17 2021-09-27 Method and apparatus for controlling grid-tie inverter

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
CN202011284302.8A CN112104244B (zh) 2020-11-17 2020-11-17 并网逆变器的控制方法和装置
CN202011284302.8 2020-11-17

Related Child Applications (1)

Application Number Title Priority Date Filing Date
US17/485,554 Continuation US11251720B1 (en) 2020-11-17 2021-09-27 Method and apparatus for controlling grid-tie inverter

Publications (1)

Publication Number Publication Date
WO2022105068A1 true WO2022105068A1 (zh) 2022-05-27

Family

ID=73784671

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/CN2021/075979 WO2022105068A1 (zh) 2020-11-17 2021-02-08 并网逆变器的控制方法和装置

Country Status (4)

Country Link
CN (1) CN112104244B (zh)
HU (1) HUE061082T2 (zh)
PT (1) PT4030611T (zh)
WO (1) WO2022105068A1 (zh)

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN115133802A (zh) * 2022-06-07 2022-09-30 深圳市京泉华科技股份有限公司 一种逆变器模型预测控制方法
CN117240126A (zh) * 2023-11-15 2023-12-15 通达电磁能股份有限公司 一种有限集模型预测控制方法、系统、终端及存储介质

Families Citing this family (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP4030611B1 (en) 2020-11-17 2022-12-21 Jiangsu Contemporary Amperex Technology Limited Control method and device for grid-connected inverter
CN112104244B (zh) * 2020-11-17 2021-03-26 江苏时代新能源科技有限公司 并网逆变器的控制方法和装置

Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN101710797A (zh) * 2009-12-07 2010-05-19 哈尔滨工业大学 Z源型并网逆变器的电流预测无差拍控制方法及其控制装置
CN201947196U (zh) * 2011-04-12 2011-08-24 中国科学院广州电子技术研究所 一种基于最大功率点跟踪的光伏并网逆变器
CN104393620A (zh) * 2014-12-01 2015-03-04 江西仪能新能源微电网协同创新有限公司 一种预测电流光伏并网逆变器控制方法及装置
US20190052097A1 (en) * 2017-08-14 2019-02-14 Young Cheol SHIN Grid-connected inverter system having seamless switching
CN110138253A (zh) * 2019-06-28 2019-08-16 盐城正邦环保科技有限公司 一种多谐振pr和pi联合控制的光伏并网逆变器控制方法
CN112104244A (zh) * 2020-11-17 2020-12-18 江苏时代新能源科技有限公司 并网逆变器的控制方法和装置

Family Cites Families (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN109787495A (zh) * 2019-03-22 2019-05-21 福州大学 一种两电平逆变器的无模型预测电流控制方法
CN110045610A (zh) * 2019-04-18 2019-07-23 中国地质大学（武汉） 逆变器改进型多步模型预测控制方法、设备及存储设备
CN112018809B (zh) * 2020-08-14 2022-03-08 长安大学 一种单相并网逆变器定频模型预测电流控制方法

Patent Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN101710797A (zh) * 2009-12-07 2010-05-19 哈尔滨工业大学 Z源型并网逆变器的电流预测无差拍控制方法及其控制装置
CN201947196U (zh) * 2011-04-12 2011-08-24 中国科学院广州电子技术研究所 一种基于最大功率点跟踪的光伏并网逆变器
CN104393620A (zh) * 2014-12-01 2015-03-04 江西仪能新能源微电网协同创新有限公司 一种预测电流光伏并网逆变器控制方法及装置
US20190052097A1 (en) * 2017-08-14 2019-02-14 Young Cheol SHIN Grid-connected inverter system having seamless switching
CN110138253A (zh) * 2019-06-28 2019-08-16 盐城正邦环保科技有限公司 一种多谐振pr和pi联合控制的光伏并网逆变器控制方法
CN112104244A (zh) * 2020-11-17 2020-12-18 江苏时代新能源科技有限公司 并网逆变器的控制方法和装置

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN115133802A (zh) * 2022-06-07 2022-09-30 深圳市京泉华科技股份有限公司 一种逆变器模型预测控制方法
CN115133802B (zh) * 2022-06-07 2023-03-24 深圳市京泉华科技股份有限公司 一种逆变器模型预测控制方法
CN117240126A (zh) * 2023-11-15 2023-12-15 通达电磁能股份有限公司 一种有限集模型预测控制方法、系统、终端及存储介质
CN117240126B (zh) * 2023-11-15 2024-01-23 通达电磁能股份有限公司 一种有限集模型预测控制方法、系统、终端及存储介质

Also Published As

Publication number Publication date
CN112104244A (zh) 2020-12-18
HUE061082T2 (hu) 2023-05-28
CN112104244B (zh) 2021-03-26
PT4030611T (pt) 2023-01-19

Similar Documents

Publication Publication Date Title
WO2022105068A1 (zh) 并网逆变器的控制方法和装置
Ahmed et al. Sliding mode based adaptive linear neuron proportional resonant control of Vienna rectifier for performance improvement of electric vehicle charging system
US11251720B1 (en) Method and apparatus for controlling grid-tie inverter
Zhou et al. High-frequency resonance mitigation for plug-in hybrid electric vehicles’ integration with a wide range of grid conditions
CN106936134A (zh) 有源阻尼lcl滤波器、有源阻尼控制装置、方法和系统
CN106936157A (zh) 并网变流系统的控制方法和控制装置
CN111221253A (zh) 适用于三相并网逆变器的鲁棒模型预测控制方法
CN102709938B (zh) Lcl滤波并网逆变器单进网电流采样的电流控制方法
CN113013926B (zh) 一种分布式并网发电系统序阻抗聚合方法及系统
CN115276439A (zh) 适应弱电网阻抗变化的lcl型并网逆变器谐振抑制方法
Zhang et al. Composite Fast Terminal Sliding Mode Control of DC-DC Converters for Renewable Energy Systems
Yang et al. Improved weighted average current control of LCL grid‐connected inverter and analysis of its order reduction characteristics
Liu et al. High performance controller design with PD feedback inner loop for three-phase four-leg inverter
CN113014250A (zh) 一种可消除直流偏移电压的锁相环及其锁相控制方法
Yang et al. Current PIλ Control of the Single-Phase Grid Inverter
Guo et al. A model-free direct predictive grid-current control strategy for grid-connected converter with an inductance-capacitance-inductance filter
Wang et al. Enhanced robustness with damping interval widening strategy of LCL-type converter under weak grid condition
WO2022262066A1 (zh) T型三电平电压型的逆变器的输出控制方法及相关设备
CN111812984B (zh) 一种用于逆变器控制系统基于模型的鲁棒滤波方法
Devarashetti et al. Design and simulation of hybrid active power filter using the adaptive fuzzy dividing frequency-control method
Yu et al. Compensation Function Observer Based Model Predictive Control for Interleaved Boost Converter
Li et al. Harmonic current forecasting method for hybrid active power filter based on optimal linear prediction theory
Nguyen et al. Disturbance observer-based transformer current estimation for bidirectional dual-active-bridge DC-DC converter using LMI-based optimization
CN113904578B (zh) 单相级联h桥变流器的无权重系数模型预测控制方法
Aapro Factors in active damping design to mitigate grid interactions in three-phase grid-connected photovoltaic inverters

Legal Events

Date Code Title Description
ENP Entry into the national phase

Ref document number: 2021745203

Country of ref document: EP

Effective date: 20210802

NENP Non-entry into the national phase

Ref country code: DE