WO2022095843A1 - 电子设备、信号处理方法以及计算机可读存储介质 - Google Patents

电子设备、信号处理方法以及计算机可读存储介质 Download PDF

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Publication number
WO2022095843A1
WO2022095843A1 PCT/CN2021/128116 CN2021128116W WO2022095843A1 WO 2022095843 A1 WO2022095843 A1 WO 2022095843A1 CN 2021128116 W CN2021128116 W CN 2021128116W WO 2022095843 A1 WO2022095843 A1 WO 2022095843A1
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pilot signal
signal
received
phase
electronic device
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PCT/CN2021/128116
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English (en)
French (fr)
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何东轩
王昭诚
曹建飞
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索尼集团公司
何东轩
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Priority to CN202180073694.XA priority Critical patent/CN116458127A/zh
Publication of WO2022095843A1 publication Critical patent/WO2022095843A1/zh

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/22—Demodulator circuits; Receiver circuits
    • G—PHYSICS
    • G06—COMPUTING; CALCULATING OR COUNTING
    • G06N—COMPUTING ARRANGEMENTS BASED ON SPECIFIC COMPUTATIONAL MODELS
    • G06N3/00—Computing arrangements based on biological models
    • G06N3/02—Neural networks
    • G06N3/08—Learning methods
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0014—Carrier regulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0014—Carrier regulation
    • H04L2027/0024—Carrier regulation at the receiver end
    • H04L2027/0026—Correction of carrier offset

Definitions

  • the present application relates to the technical field of wireless communication, and more particularly, to an electronic device, a signal processing method and a non-transitory computer-readable storage medium suitable for demodulating a signal with a high transmission data rate such as a terahertz signal.
  • Terahertz communication has a high data rate, so ideally, it is expected that the receiver has a high-speed analog-to-digital conversion device.
  • the terahertz frequency band is located between the microwave frequency band and the optical frequency band.
  • the communication devices in this frequency band are difficult to manufacture, and there are nonlinear effects such as power amplifier nonlinear effects, in-phase (in-phase) branch (I branch) and quadrature (quadrature) branch. (Q branch) unbalance, carrier phase noise and other hardware mismatch effects.
  • Such hardware mismatch effect will cause mixed distortion of the received signal, resulting in the degradation of the communication performance, thereby further affecting the accuracy of the demodulation decision.
  • the present disclosure proposes an electronic device, a signal processing method, and a non-transitory computer-readable storage medium suitable for processing high transmission data rate signals such as terahertz signals, which can utilize pilot-based frequency signals at a receiving end
  • the phase estimation and demodulation neural network obtained by the signal improves the conversion accuracy of low-resolution analog-to-digital conversion units (eg, single-bit analog-to-digital conversion units) and compensates for mixing distortion, thereby improving the accuracy of demodulation decisions.
  • an electronic device including a processing circuit configured to: phase adjust a received complex signal according to a compensation phase estimated using a pilot signal; The analog-to-digital conversion process obtains the received bit sequence based on the phase-adjusted complex signal; and demodulates the received bit sequence based on the demodulation neural network obtained by training the pilot signal to obtain modulation symbols of the received complex signal.
  • a signal processing method comprising: performing phase adjustment on a received complex signal according to a compensation phase estimated by using a pilot signal;
  • the phase-adjusted complex signal obtains a received bit sequence; and based on the demodulation neural network obtained by training the pilot signal, the received bit sequence is demodulated to obtain modulation symbols of the received complex signal.
  • a non-transitory computer-readable storage medium storing executable instructions, the executable instructions, when executed by a processor, cause the processor to execute the above signal processing method or electronic device of each function.
  • the compensation phase obtained based on the pilot signal can be used at the receiving end to improve the conversion accuracy of the low-resolution analog-to-digital conversion unit, and the demodulation neural network obtained by training based on the pilot signal can be implemented. Compensation for mixing distortion, so that accurate demodulation decisions can also be achieved for high transmission data rate signals such as terahertz signals.
  • FIG. 1 is a schematic diagram illustrating the distortion of a constellation diagram of a received signal caused by mixing distortion in the prior art
  • FIG. 2 is a schematic block diagram showing one configuration example of an electronic device according to an embodiment of the present disclosure
  • FIG. 3 is a schematic block diagram illustrating an example circuit implementation of an electronic device according to an embodiment of the present disclosure
  • FIG. 4 schematically shows an example of a demodulation neural network that can be adopted by an electronic device according to an embodiment of the present disclosure
  • FIG. 5 is a schematic diagram for illustrating a data structure of a complex signal received by an electronic device according to an embodiment of the present disclosure
  • FIG. 6 is a schematic diagram illustrating an example of an information interaction process according to an embodiment of the present disclosure
  • FIG. 7 is an explanatory diagram of an example simulation result for illustrating performance degradation due to phase noise
  • FIG. 8 is a diagram illustrating an example simulation result of the performance of demodulation processing by an electronic device according to an embodiment of the present disclosure
  • FIG. 9 is a flowchart illustrating a process example of a signal processing method according to an embodiment of the present disclosure.
  • FIG. 10 is a block diagram illustrating a first example of a schematic configuration of an eNB to which techniques of this disclosure may be applied;
  • FIG. 11 is a block diagram illustrating a second example of a schematic configuration of an eNB to which techniques of the present disclosure may be applied;
  • FIG. 12 is a block diagram illustrating an example of a schematic configuration of a smartphone to which the techniques of the present disclosure may be applied;
  • FIG. 13 is a block diagram showing an example of a schematic configuration of a car navigation apparatus to which the technology of the present disclosure can be applied.
  • Example embodiments are provided so that this disclosure will be thorough, and will fully convey the scope to those skilled in the art. Numerous specific details are set forth such as examples of specific components, devices, and methods in order to provide a thorough understanding of embodiments of the present disclosure. It will be apparent to those skilled in the art that specific details need not be employed, that example embodiments may be embodied in many different forms, and that neither should be construed to limit the scope of the disclosure. In some example embodiments, well-known processes, well-known structures and well-known technologies are not described in detail.
  • Terahertz communication has a high data rate, so ideally, it is expected that the receiver has a high-speed analog-to-digital conversion device.
  • the receiver has a high-speed analog-to-digital conversion device.
  • only single-bit receivers and other equipment can be used to complete acquisition with low-precision analog-to-digital conversion devices.
  • single-bit receivers for low-frequency signals generally use likelihood information to determine and complete signal demodulation.
  • the accurate expression of the likelihood information of the received signal is not easy to obtain, so the traditional method cannot be used to complete the accurate judgment of the terahertz single-bit receiver. This makes it difficult for the receiving end to make an accurate decision on the received signal, thus affecting the accuracy of the demodulation decision.
  • the terahertz frequency band is located between the microwave frequency band and the optical frequency band.
  • the communication devices in this frequency band are difficult to manufacture, and there are nonlinear effects such as power amplifier nonlinear effects, in-phase (in-phase) branch (I branch) and quadrature (quadrature) branch. (Q branch) unbalance, carrier phase noise and other hardware mismatch effects.
  • Such hardware mismatch effect will cause mixed distortion of the received signal, resulting in the degradation of communication performance, thereby further affecting the accuracy of the demodulation decision.
  • a bit group (also referred to as a "modulation symbol” in this paper) is represented by the I-channel carrier signal cos(2 ⁇ f c t) and the Q-channel carrier signal sin(2 ⁇ f c t) (where f c is the carrier frequency, t represents time) and then modulate and add them to obtain a QPSK modulated transmission symbol or transmission symbol.
  • each transmission symbol corresponds to a constellation point
  • the angle between the vector formed by each constellation point and the origin and the coordinate axis is 45 degrees
  • the angle between each constellation point and the coordinate axis is equal to distance.
  • the transmitter may have IQ path imbalance.
  • ⁇ T and ⁇ T can be used to represent the unbalance factor of the amplitude and phase of the IQ channel respectively, then the carrier signal of the I channel becomes (1+ ⁇ T )cos(2 ⁇ f c t- ⁇ T ), and the carrier signal of the Q channel becomes ( 1- ⁇ T )sin(2 ⁇ f c t+ ⁇ T ).
  • the ideal transmitted signal (complex signal) s after QPSK modulation will become the following form:
  • ⁇ T cos ⁇ T -j ⁇ T sin ⁇ T
  • v T ⁇ T cos ⁇ T -jsin ⁇ T
  • v T s * is the image interference term caused by IQ imbalance.
  • phase noise ⁇ T may have the form of a block walk model, i.e. ⁇ T is a fixed value ⁇ k within the kth transport block, and differs between adjacent transport blocks by a random walk with a Gaussian distribution term ⁇ k .
  • the phase noise ⁇ k+1 of the k+1th transport block can be expressed as:
  • QPSK modulated signal at the transmitter It will be amplified by a Power Amplifier.
  • the nonlinear nature of the power amplifier causes the signal further distortion.
  • a memoryless polynomial model can be used to represent the nonlinear characteristics of the power amplifier, and the signal output by the power amplifier can be expressed in the following form:
  • 2K-1 represents the polynomial order.
  • the above-mentioned signal s PA output by the power amplifier of the transmitter will be transmitted to the receiver side through the channel. Since terahertz communication usually uses antennas with extremely high directional gain at the transceiver end, there is only one effective transmission path in the channel, and it can be represented by a flat fading channel model. For example, the signal received at the receiver side can be expressed as:
  • h is the channel fading factor
  • AWGN additive white Gaussian noise
  • CSCG circularly symmetric complex Gaussian
  • the ideal processing at the receiver is to multiply it by the I-channel carrier signal cos(2 ⁇ f c t) and the Q-channel carrier signal sin(2 ⁇ f c t), respectively, in order to obtain the complex signal
  • the real and imaginary parts of also known as I-channel signal and Q-channel signal.
  • IQ path imbalance and phase noise can also occur at the receiver.
  • ⁇ R and ⁇ R represent the amplitude imbalance factor and phase imbalance factor of the IQ channel respectively, then the I channel carrier signal at the receiver becomes (1+ ⁇ R )cos(2 ⁇ f c t- ⁇ R ), the Q channel carrier signal becomes (1+ ⁇ R )cos(2 ⁇ f c t- ⁇ R ), becomes (1- ⁇ R )sin(2 ⁇ f c t+ ⁇ R ).
  • the final received signal can be expressed as the following form:
  • ⁇ R cos ⁇ R +j ⁇ R sin ⁇ R
  • v R ⁇ R cos ⁇ R -jsin ⁇ R .
  • the compensation (or predistortion) algorithm at the sending end and the compensation algorithm at the receiving end are generally required to work together.
  • the pre-distortion algorithm at the transmitting end can handle the nonlinearity of the power amplifier and the imbalance of the IQ branch at the transmitting end
  • the compensation algorithm at the receiving end can handle the IQ branch imbalance and phase noise at the receiving end.
  • hardware in the terahertz frequency band is difficult to manufacture and has a high cost.
  • the compensation link at the sending end often requires a high-precision and high-speed sampling link, which is difficult to design and difficult to implement; only using The compensation technology at the receiving end cannot effectively compensate the mixed distortion of the terahertz channel.
  • the present disclosure provides an electronic device, a signal processing method, and a non-transitory computer-readable storage medium suitable for processing high transmission data rate signals such as terahertz signals, which can utilize pilot-based
  • the phase estimation and demodulation neural network obtained by the signal improves the conversion accuracy of low-resolution analog-to-digital conversion units (eg, single-bit analog-to-digital conversion units) and compensates for mixing distortion, thereby improving the accuracy of demodulation decisions.
  • the electronic device according to the present disclosure may be an electronic device on the user equipment side or an electronic device on the network side, as long as it can function as a receiving end device and perform corresponding signal processing.
  • the electronic equipment on the user equipment side may be implemented as various user equipment such as mobile terminals such as smart phones, tablet personal computers (PCs), notebook PCs, portable game terminals, portable/dongle-type mobile routers, and digital cameras Or an in-vehicle terminal (such as a car navigation device).
  • the user equipment described above may also be implemented as a terminal (also referred to as a machine type communication (MTC) terminal) performing machine-to-machine (M2M) communication.
  • the user equipment may include a wireless communication module (such as an integrated circuit module including a single die) or the like mounted on each of the above-mentioned terminals.
  • the electronic device on the network side may be the base station device itself, such as an eNB (evolved node B), a gNB, and any type of TRP (transmit and receive port).
  • the TRP may have sending and receiving functions, for example, it may receive information from user equipment and base station equipment, and may also send information to user equipment and base station equipment.
  • the TRP may serve user equipment and be controlled by base station equipment. That is, the base station equipment can provide services to the user equipment through the TRP.
  • the base station device is used as an example of the electronic device on the network side for description, but the present disclosure is not limited to this, but can be appropriately applied to the situation of electronic devices with similar functions.
  • FIG. 2 is a block diagram showing one configuration example of an electronic device according to an embodiment of the present disclosure.
  • the electronic device 200 may include a phase adjustment unit 210 , a low-resolution analog-to-digital conversion unit 220 , a demodulation unit 230 and an optional compensation phase estimation unit 240 .
  • each unit of the electronic device 200 may be included in the processing circuit.
  • the electronic device 200 may include either one processing circuit or multiple processing circuits.
  • the processing circuit may include various discrete functional units to perform various different functions and/or operations. It should be noted that these functional units may be physical entities or logical entities, and units with different names may be implemented by the same physical entity.
  • the phase adjustment unit 210 of the electronic device 200 may perform phase adjustment on the received complex signal according to the compensation phase estimated by using the pilot signal.
  • the low-resolution analog-to-digital conversion unit 220 may obtain a received bit sequence based on the phase-adjusted complex signal through a low-resolution analog-to-digital conversion process.
  • the demodulation unit 230 may demodulate the received bit sequence based on the demodulation neural network obtained by training the pilot signal to obtain the modulation symbol of the received complex signal.
  • the complex signal and the pilot signal may each comprise a QPSK modulated signal, and eg a high transmission data rate signal such as a terahertz signal.
  • QPSK modulation e.g., those skilled in the art can understand that in the context of the present disclosure, the QPSK modulation involved may cover relative phase-shifted QPSK (offset QPSK, OQPSK) and differential DQPSK (differential QPSK, DQPSK) modulation etc.
  • the electronic device 200 receives, for example, the form of equation (6) complex signal Problems such as the distortion of the constellation diagram shown in Figure 1 may occur. Therefore, the electronic device 200 can perform phase adjustment on the received complex signal through the phase adjustment unit 210 according to the compensation phase ⁇ * estimated by using the pilot signal, so as to obtain the phase-adjusted complex signal in the following form:
  • the above-described phase adjustment process can properly rotate the constellation diagram of the received signal to, for example, maximize the minimum distance between the constellation points and different coordinate axes (including the x-axis and the y-axis), for example, such that the diagram The smallest distances in d 1 , d 2 , d 3 and d 4 in 1 are maximized.
  • Such phase adjustment will increase the probability of correct decision of the received signal. The details of how to determine the compensation phase will be described later in the "Determination of the Compensation Phase" section.
  • the phase adjustment unit 210 may be configured to control a local oscillator (not shown in the figure) to perform phase rotation on the received complex signal according to the compensation phase, so as to realize the above-mentioned phase adjustment.
  • a local oscillator included in or connected to the electronic device 200 can be used to generate the I-channel carrier signal and the Q-channel carrier signal at the electronic device 200 at the receiving end for obtaining the real and imaginary parts of the complex signal department.
  • the phase adjustment unit 210 realizes the phase adjustment by controlling the local oscillator, it actually directly adjusts the phase of the carrier signal, and then uses such a carrier signal to multiply the received complex signal, so that the phase-adjusted complex signal can be directly obtained.
  • phase-adjusted complex signal can be changed from equation (6) to the following form:
  • the complex signal after the phase adjustment is performed on the phase adjustment unit 210 The low-resolution analog-to-digital conversion unit 220 may perform low-resolution analog-to-digital conversion processing to obtain the received bit sequence.
  • the low-resolution analog-to-digital conversion processing unit 220 may be configured to obtain a real-part received bit sequence and an imaginary-part received bit sequence based on the real part and imaginary part of the phase-adjusted complex signal, respectively, through low-resolution analog-to-digital conversion processing .
  • the low-resolution analog-to-digital conversion processing unit 220 may obtain a real part received bit sequence and an imaginary part received bit sequence of each transmission symbol of the received complex signal by over-sampling the real part and the imaginary part .
  • the phase adjustment unit 210 can perform single-bit analog signal acquisition for the I/Q branch, and the real part of the acquired signal each time and the imaginary part can be expressed as the following
  • the correct probability of the received signal decision depends on the real part of each transmitted symbol collected and the imaginary part Comparison with 0, thus depending on the minimum distance of each constellation point to different coordinate axes (including x-axis and y-axis) (eg the minimum distances in d 1 , d 2 , d 3 and d 4 in Figure 1) , so increasing the minimum distance after phase adjustment will improve the probability of correct decision of the received signal (that is, increase the real part of each transmitted symbol and the imaginary part the correct probability of the result of the comparison with 0).
  • N is the oversampling multiple of the signal, for example, it can be 10.
  • Such real part and imaginary part received bit sequence can also be uniformly expressed as received bit sequence
  • the demodulation unit 230 may demodulate the demodulation neural network obtained by training the pilot signal to obtain the modulation symbol of the transmission symbol.
  • the demodulation neural network is obtained by training, for example, a pilot signal marked with modulation symbols, and can characterize the mapping relationship between the received bit sequence of the transmission symbols of the received complex signal and the corresponding modulation symbols, and later Further details will be described in the "Training of the Demodulation Neural Network" section.
  • the compensation phase obtained based on the pilot signal can be used at the receiving end to improve the conversion accuracy of the low-resolution analog-to-digital conversion unit, and the demodulation neural network obtained by training based on the pilot signal can be implemented. Compensation for mixing distortion, so that accurate demodulation decisions can also be achieved for high transmission data rate signals such as terahertz signals.
  • FIG. 3 is a schematic block diagram illustrating an example circuit implementation 300 of an electronic device according to an embodiment of the present disclosure, showing the phase adjustment unit 210 , the low-resolution analog-to-digital conversion unit 220 , the demodulation unit 220 and the demodulation unit in FIG. 2 , respectively.
  • Unit 230 phase adjustment unit 310 which is an example of compensation phase estimation unit 240, ADCs 320a and 320b as single-bit flip-flops (also collectively referred to as ADC 320 when no distinction is required), demodulation unit 330, and compensation phase estimation unit 340.
  • FIG. 3 is a schematic block diagram illustrating an example circuit implementation 300 of an electronic device according to an embodiment of the present disclosure, showing the phase adjustment unit 210 , the low-resolution analog-to-digital conversion unit 220 , the demodulation unit 220 and the demodulation unit in FIG. 2 , respectively.
  • Unit 230 phase adjustment unit 310 which is an example of compensation phase estimation unit 240
  • FIG. 3 also shows an antenna 350, a local oscillator LO, a phase shifter 360, and multiplying circuits 370a and 370b (also collectively referred to as multiplying circuits 370 when no distinction is required) as the basic configuration of the electronic device at the receiving end.
  • These components may be optional additional parts of the example circuit implementation 300, or may be additional circuit parts connected to the example circuit implementation 300, the present There is no limit to the disclosure.
  • the respective units 310 to 340 of FIG. 3 may be used to implement the functions of the corresponding units 210 to 240 of the electronic device 200 previously described with reference to FIG. 2 .
  • the phase adjustment unit 310 controls the local oscillator LO to perform phase rotation on the local oscillator LO according to, for example, the compensation phase ⁇ * estimated by the compensation phase estimation unit 340 using the pilot signal, which is equivalent to changing the carrier signals cos(2 ⁇ f c t) and sin(2 ⁇ f c t)
  • the multiplying circuits 370a and 370b can use the corresponding phase-adjusted carrier signals cos(2 ⁇ f c t+ ⁇ * ) and sin(2 ⁇ f c t+ ⁇ * ) and the complex signal multiplied to obtain the phase-adjusted complex signal directly the real part of and the imaginary part
  • the example circuit implementation of an electronic device of an embodiment of the present disclosure such as the example circuit implementation of FIG.
  • ADCs 320a and 320b which are examples of low-resolution analog-to-digital conversion processing units, can respectively and the imaginary part Single-bit oversampling to get the received complex signal
  • the real bit sequence for each transmitted symbol of and imaginary bit sequence Among them, N is the oversampling multiple of the signal, for example, it can be 10.
  • the low-resolution analog-to-digital conversion processing unit that performs single-bit oversampling can be implemented by using an analog signal collector such as a high-speed flip-flop.
  • DI's HMC729LC3C high-speed flip-flop can be used as the ADC 320 in this example.
  • Received complex signals by ADCs 320a and 320b The real part bit sequence of each transmitted symbol of and imaginary bit sequence is input to the demodulation unit 330 to obtain the received complex signal The demodulated symbol n of each transmitted symbol of .
  • the demodulation unit 330 may use a deep feedforward neural network (DFNN) to perform demodulation processing, which uses the deep feedforward neural network's ability to fit the mapping relationship with arbitrary precision to fit the received bits.
  • DFNN deep feedforward neural network
  • the mapping relationship between the sequence and the modulation symbol is used to complete the demodulation decision.
  • Figure 4 schematically shows an example of a demodulation neural network 400 that may be employed.
  • the network includes an input layer 410 , a hidden layer 420 , and an output layer 430 .
  • the input layer 410 has, for example, 2N input channels (N is an oversampling factor, and is, for example, 10) to input, for each transmission symbol of the received complex signal, a sequence of bits consisting of the real part of the transmission symbol.
  • the demodulation neural network can, for example, characterize the mapping relationship between the received bit sequence of each transmission symbol of the received complex signal and the corresponding modulation symbol by the following equation:
  • DFNN( ) represents the input-output relationship of the demodulation neural network
  • n represents the received bit sequence obtained by the demodulation neural network and the current transmission symbol
  • the corresponding modulation symbol of the transmission symbol which can be, for example, one of the dibit groups 11, 10, 00, 01.
  • the pilot signal needs to be used to complete the determination of the compensation phase and the training of the demodulation neural network.
  • the phase adjustment unit 310, ADCs 320a and 320b as examples of low-resolution analog-to-digital conversion units, etc.
  • the pilot signal is processed similarly to the data signal to obtain a received bit sequence of the pilot signal, and the received bit sequence is used to determine the compensation phase, for example, by the process of the compensation phase estimation unit 340 or the like or the process of the demodulation unit 440 etc. Train a demodulation neural network.
  • the electronic device in the embodiment of the present disclosure may, for example, perform processing such as the compensation phase estimation unit 340 of FIG. Bit sequence to estimate the offset phase.
  • the transmitting end sends the first pilot signal with the following complex signal form
  • p 1I and p 1Q are the real and imaginary parts of the first pilot signal, respectively, which may be 1 or -1, respectively, depending on the current transmission symbol of the pilot signal.
  • the transmitting end can obtain the above first pilot signal by performing QPSK modulation on a bit sequence including multiple double-bit groups ab, wherein each double-bit group ab obtains a transmission symbol in the first pilot signal after modulation.
  • Each transmission symbol in the first pilot signal may correspond to at least two adjacent constellation points in the QPSK constellation, and may preferably correspond to only two adjacent constellation points in the QPSK constellation. Due to the symmetry of the constellation points in the QPSK constellation diagram, the distance between any two adjacent constellation points and the coordinate axis (such as d 1 , d 2 , d 3 and d 4 shown in FIG. 1 ) can characterize all four The distance between the constellation point and the coordinate axis.
  • QPSK modulation can be performed by using a bit sequence capable of obtaining transmission symbols corresponding to the selected two adjacent constellation points, so as to obtain 1+j, 1-j, -1-j, -1+j are the first pilot signals of the corresponding two.
  • each transmission symbol in the first pilot signal may alternately correspond to two adjacent constellation points in the QPSK constellation.
  • the first The pilot signal can be represented as [1+j, 1-j, . . . , 1+j, 1-j].
  • the manner in this example can simplify the generation of the first pilot signal.
  • the electronic device at the receiving end can obtain the received signal of the above-mentioned first pilot signal p 1 through the antenna 350 in FIG. 3 . (hereinafter also referred to as the received first pilot signal or simply the first pilot signal when appropriate ).
  • the electronic device may perform phase adjustment through, for example, the phase adjustment unit 310 of FIG. 3 , and may obtain the phase adjustment based on the phase-adjusted, received first pilot signal through, for example, low-resolution analog-to-digital conversion processing performed by the ADC 320 of FIG. 3 .
  • the first received bit sequence is referred to as the received first pilot signal or simply the first pilot signal when appropriate.
  • the phase adjustment unit 310 uses the compensation phase ⁇ (t) to be determined, which varies continuously over time, for example, between 0 and ⁇ , and performs phase rotation on it by the local oscillator LO, that is, performs phase rotation on the carrier signal, and the multiplication circuit 370a and 370b use the corresponding phase-adjusted carrier signals cos(2 ⁇ f c t + ⁇ (t)) and sin(2 ⁇ f c t + ⁇ (t)) with the first pilot signal multiplied to obtain the phase-adjusted received first pilot signal the real part of and the imaginary part
  • the ADCs 320a and 320b can respectively adjust the phase of the first pilot signal the real part of and the imaginary part Single-bit oversampling to obtain the real bit sequence for each transmitted symbol of the received first pilot signal and imaginary bit sequence Among them, N is the oversampling multiple of the signal, for example, it can be 10.
  • the above real part bit sequence and imaginary bit sequence may be collectively referred to as the first received bit sequence
  • the electronic device can also compensate for the processing of the phase estimation unit 340 according to the first received bit sequence ( and ) with respect to the error of the transmission symbol of the first pilot signal p 1 , estimating the compensation phase; and the received first pilot signal may be phase-adjusted according to the estimated compensation phase.
  • the processing circuit is further configured to determine the compensation phase that will minimize the error as the final compensation phase.
  • the compensation phase estimation unit 340 may be based on the received real part bit sequence of each transmission symbol of the first pilot signal and imaginary bit sequence
  • the optimal compensation phase is determined by comparison with the transmitted symbols of the first pilot signal p 1 (ie, the real part p 1I and the imaginary part p 1Q of the first pilot signal p 1 ). More specifically, while the compensation phase ⁇ (t) is continuously changed (for example, continuously increased or decreased), it can be obtained according to the phase adjustment obtained by using the supplementary phase. and Each bit (also referred to as a sampling point) in p 1I and p 1Q makes an optimal phase decision with respect to the number of p 1I and p 1Q decision errors.
  • the compensation phase estimation unit 340 After each phase change, the compensation phase estimation unit 340 re-determines the first received bit sequence of the current transmission symbol of the first pilot signal The number of erroneous sampling points in accuracy. In this way, it is possible, for example, to make the first received bit sequence The compensation phase with the smallest number of error sampling points is determined as the final compensation phase. Therefore, although the values of the distances d 1 , d 2 , d 3 and d 4 between each constellation point and the coordinate axis under the current compensation phase in the constellation diagram shown in FIG. 1 cannot be directly calculated, it can still be determined that the The smallest distance among d 1 , d 2 , d 3 and d 4 maximizes the optimal compensation phase for this effect.
  • the electronic device may, for example, use the second pilot signal obtained based on the received second pilot signal and marked with the modulation symbol of the second pilot signal through the processing of the demodulation unit 330 in FIG. 3 , for example. Receive the bit sequence and obtain the demodulation neural network through training.
  • the transmitting end sends the second pilot signal having the following complex signal form
  • p 2I and p 2Q are the real and imaginary parts of the second pilot signal, respectively, which may be 1 or -1, respectively, depending on the current transmission symbol of the pilot signal.
  • the transmitting end obtains the above second pilot signal by performing QPSK modulation on a bit sequence including multiple double-bit groups ab, wherein each double-bit group ab obtains one transmission symbol in the second pilot signal after modulation.
  • the second pilot signal p 2 includes all transmission symbols corresponding to different bis-groups (ie, different modulation symbols), in order to use this pilot signal to train a demodulation neural network to obtain the received complex signal
  • the mapping relationship between the bit sequence of each transmission symbol and each modulation symbol may correspond to four constellation points in the QPSK constellation.
  • a bit sequence capable of obtaining all four constellation points (eg, using bits including four different double-bit groups or modulation symbols 11, 10, 00, and 01) can be used. sequence) to perform QPSK modulation, thereby obtaining a second pilot signal including 1+j, 1-j, -1-j, -1+j.
  • each transmission symbol in the second pilot signal p2 may randomly correspond to four constellation points in the QPSK constellation.
  • the second pilot signal may be represented as [1+j,-1+j,1-j,-1-j,-1-j,1+j,...,1-j,1+j].
  • the electronic device at the receiving end can obtain the received signal of the above-mentioned second pilot signal p 2 through the antenna 350 in FIG. 3 .
  • the received second pilot signal or simply the second pilot signal as appropriate .
  • the electronic device may perform phase adjustment through, for example, the phase adjustment unit 310 of FIG. 3 , and may obtain the phase adjustment based on the phase-adjusted, received second pilot signal through, for example, low-resolution analog-to-digital conversion processing performed by the ADC 320 of FIG. 3 .
  • the second received bit sequence is the received signal of the above-mentioned second pilot signal p 2 through the antenna 350 in FIG. 3 .
  • the phase adjustment unit 310 uses the compensation phase ⁇ * determined by the compensation phase estimation unit to perform phase rotation on it through the local oscillator LO, that is, performs phase rotation on the carrier signal, and the multiplication circuits 370a and 370b use the corresponding phase-adjusted carrier signal cos (2 ⁇ f c t+ ⁇ * ) and sin(2 ⁇ f c t+ ⁇ * ) and the second pilot signal multiplied to obtain the phase-adjusted received second pilot signal the real part of and the imaginary part
  • the ADCs 320a and 320b can respectively adjust the phase of the second pilot signal the real part of and the imaginary part Single-bit oversampling to obtain the real bit sequence for each transmitted symbol of the received second pilot signal and imaginary bit sequence Among them, N is the oversampling multiple of the signal, for example, it can be 10.
  • the above real part bit sequence and imaginary bit sequence may be collectively referred to as the second received bit sequence
  • the second received bit sequence of the current transmission symbol of the second pilot signal obtained by ADC 320 is input to the demodulation unit 330, and the demodulation unit 330 can convert the second received bit sequence Input into the demodulation neural network as shown in Figure 4, and obtain the output of the demodulation neural network the output is a modulation symbol or a double-bit group of the current transmission symbol of the second pilot signal obtained by demodulating the neural network, which may be, for example, one of the double-bit groups 11, 10, 00, and 01.
  • the demodulation unit 330 may demodulate the output result of the neural network based on each modulation symbol of the second pilot signal
  • the difference between the loss function is constructed, and the iterative training is carried out in any suitable way, such as gradient descent, to determine the optimal value of the parameters of the demodulation neural network, for example, when the loss function reaches a minimum value or no longer changes. value.
  • suitable way such as gradient descent
  • those skilled in the art can implement the training of the demodulation neural network in any appropriate manner, which will not be repeated here.
  • FIG. 5 is a schematic diagram for explaining a data structure of a complex signal received by an electronic device according to an embodiment of the present disclosure.
  • the complex signal received by the electronic device according to the embodiment of the present disclosure may include, for example, three parts in sequence, namely, a first pilot signal used for estimating the compensation phase, and a second pilot signal used for training the demodulation neural network. pilot signal, and data signal.
  • the first pilot signal here may have the example form of the first pilot signal described above in the "Determination of Compensation Phase” section
  • the second pilot signal may have the above in "Training of the Demodulation Neural Network”
  • An example form of the second pilot signal described in the section, the complex signal as the data signal can be any QPSK modulated signal.
  • a complex signal including the above three parts in sequence can be sent from the sender to the receiver.
  • FIG. 6 is a schematic diagram illustrating an example of an information exchange process according to an embodiment of the present disclosure, in which a sending end and a receiving end are schematically shown (here, the receiving end may be, for example, the electronic device 200 described earlier). or its example circuit implementation 300) and information interaction between the two.
  • FIG. 6 shows, for example, an example of information exchange between the sender and the receiver in one communication, in which, for example, three parts of the complex signal shown in FIG. 5 are transmitted.
  • the transmitting end sends a first pilot signal to the receiving end.
  • the electronic device at the receiving end determines the compensation phase according to the transmission symbol of the first pilot signal and the first received bit sequence obtained based on the received first pilot signal.
  • the transmitting end sends a second pilot signal to the receiving end.
  • the electronic device at the receiving end obtains the demodulation neural network through training by using the second received bit sequence obtained based on the received second pilot signal and marked with the modulation symbol of the second pilot signal.
  • the transmitting end transmits the complex signal as the data signal to the receiving end.
  • the electronic device at the receiving end may perform phase adjustment on the received complex signal according to the previously estimated compensation phase using the first pilot signal. Then, the electronic device at the receiving end can obtain the received bit sequence based on the phase-adjusted complex signal through low-resolution analog-to-digital conversion processing. Next, the electronic device at the receiving end may demodulate the received bit sequence based on the demodulation neural network obtained by training the second pilot signal to obtain the modulation symbol of the received complex signal.
  • the example flow shown in FIG. 6 can be implemented by the electronic device 200 or its example circuit implementation 300 at the receiving end described above with reference to FIGS. 2 to 4 and the electronic device at the transmitting end that communicates with it, so the configuration examples of the above electronic devices can be obtained.
  • the advantages and benefits described in are not further described here.
  • the transmitting end (or the electronic device of the transmitting end) involved here can only generate and transmit the corresponding first pilot signal, second pilot signal and data signal and can communicate with the electronic device 200 of the receiving end or its example circuit to realize 300 You can cooperate to communicate. Therefore, various electronic devices having a transmitter function in the prior art can be used to implement the electronic device at the transmitting end through proper configuration.
  • the electronic device on the receiving end is a network-side device such as a base station
  • the electronic device on the sending end can be a user equipment that can communicate with it
  • the electronic device on the receiving end is a user equipment
  • the electronic device on the sending end can be a user equipment that can communicate with it.
  • Network side devices such as base stations for communication will not be repeated here.
  • Equation (2) set the variance of the walk term of the phase noise ⁇ T and ⁇ R Since the walk terms of the phase noise ⁇ T and ⁇ R conform to the Gaussian distribution, the total phase noise ⁇ T + ⁇ R generated by the influence of the transceiver is also a random variable with a mean value of 0, and its distribution can be determined according to Equation (2) .
  • AWGN additive white Gaussian noise
  • FIG. 7 is an explanatory diagram illustrating an example simulation result of performance degradation due to phase noise.
  • FIG. 7 shows that in the electronic device shown in FIG. 2 , in the case of no compensation phase estimation or phase adjustment (that is, the functions of the phase adjustment unit 210 and the compensation phase estimation unit 240 in FIG. 2 are removed) ), the effect on demodulation performance caused by the total phase noise caused by the influence of the transceiver, where the horizontal axis represents the total phase noise ⁇ T + ⁇ R caused by the effect of the transceiver, and the vertical axis represents the signal-to-noise ratio at different Bit Error Rate (BER) at (E S /N 0 ).
  • BER Bit Error Rate
  • ⁇ T + ⁇ R determined according to equation (2) is also a random variable with mean 0, and ⁇ is set in the example shown in Figure 7 T + ⁇ R varies randomly in the range of (-0.2 ⁇ , 0.2 ⁇ ).
  • the simulation results shown in Fig. 7 are obtained by simulating 1000 communications for each phase noise within the range of phase noise ⁇ T + ⁇ R determined from the distributions of ⁇ T and ⁇ R set above, where, In each communication, a second pilot signal with a length of 10 5 and a data signal with a length of 10 6 are received in sequence.
  • the electronic device such as the embodiment of the present disclosure shown in FIG.
  • the demodulation neural network performs demodulation processing on the data signal.
  • the bit error rate (BER) varies with the phase at different signal-to-noise ratios (E S /N 0 ). Therefore, it is desirable to adjust the phase of the received complex signal with a properly determined compensation phase, so as to optimize the demodulation performance.
  • FIG. 8 is a diagram illustrating an example simulation result of the performance of demodulation processing performed by an electronic device according to an embodiment of the present disclosure.
  • FIG. 8 respectively illustrates a conventional hard-decision (single-bit quantization of a received analog signal, and converting the quantization result into an output result of a modulation symbol) method and an electronic device using an embodiment of the present disclosure such as that shown in FIG. 2 , respectively.
  • Bit error rate performance of demodulating a received complex signal where the horizontal axis represents the signal-to-noise ratio (E S /N 0 ) and the vertical axis represents the corresponding bit error rate (BER).
  • E S /N 0 the signal-to-noise ratio
  • BER bit error rate
  • the traditional hard decision method receives a data signal with a length of 10 6
  • the electronic device in this embodiment of the present disclosure sequentially receives a first pilot signal with a length of 10 3 and a second pilot signal with a length of 10 5 in sequence signal and a data signal of length 10 6 .
  • the electronic device according to the embodiment of the present disclosure uses the first pilot signal to first determine the compensation phase in each communication, and then uses the second pilot signal to obtain a demodulation neural network through training, and then phase-phases the data signal.
  • the traditional hard decision method cannot effectively achieve demodulation, while the processing performed by the electronic device of the embodiment of the present disclosure is performed by real-time phase compensation in each communication. Or adjust and demodulate the accurate demodulation of the neural network, solve the channel mixing distortion problem and realize the accurate demodulation of the terahertz QPSK signal, and when the signal-to-noise ratio is 10dB, the bit error rate can reach 10 -2 .
  • FIG. 9 is a flowchart illustrating a process example of a signal processing method according to an embodiment of the present disclosure, which may be implemented, for example, by the electronic device 200 or its example circuit implementation 300 described with reference to FIGS. 2 to 4 .
  • step S901 phase adjustment is performed on the received complex signal according to the compensation phase estimated by using the pilot signal.
  • step S902 a received bit sequence is obtained based on the phase-adjusted complex signal through a low-resolution analog-to-digital conversion process.
  • step S903 the received bit sequence is demodulated based on the demodulation neural network obtained by training the pilot signal to obtain the modulation symbol of the received complex signal.
  • the complex and pilot signals here may include QPSK modulated signals.
  • the local oscillator may be controlled to perform phase rotation on the received complex signal according to the compensation phase, so as to realize phase adjustment.
  • a real part received bit sequence and an imaginary part received bit sequence may be obtained based on the real part and the imaginary part of the phase-adjusted complex signal through a low-resolution analog-to-digital conversion process, respectively.
  • the real part and the imaginary part may be oversampled to obtain a real-part received bit sequence and an imaginary-part received bit sequence for each transmission symbol of the received complex signal.
  • the method may further include: before step S901, estimating compensation according to the transmission symbols of the first pilot signal and the first received bit sequence obtained based on the received first pilot signal phase.
  • the method may include the following processing for estimating the compensation phase: obtaining a first received bit sequence based on the phase-adjusted, received first pilot signal by a low-resolution analog-to-digital conversion process; A compensation phase is estimated according to an error of the first received bit sequence relative to the transmission symbols of the first pilot signal; and a phase adjustment is performed on the received first pilot signal according to the estimated compensation phase.
  • the compensation phase that minimizes the error may be determined as the final compensation phase.
  • the first pilot signal may include a plurality of transmission symbols corresponding to at least two adjacent constellation points in the QPSK constellation.
  • each transmission symbol in the first pilot signal may correspond to two adjacent constellation points in the QPSK constellation.
  • each transmission symbol in the first pilot signal may alternately correspond to the two adjacent constellation points.
  • the method may further include: before step S901, using a second received bit sequence obtained based on the received second pilot signal marked with the modulation symbol of the second pilot signal , the demodulation neural network is obtained by training.
  • the method may include the following processes for training the demodulation neural network: phasing the received second pilot signal according to the compensated phase estimated using the first pilot signal; and A resolution analog-to-digital conversion process to obtain a second received bit sequence based on the phase-adjusted second pilot signal.
  • the transmission symbols in the second pilot signal may correspond to four constellation points in the QPSK constellation.
  • each transmission symbol in the second pilot signal may randomly correspond to the four constellation points.
  • the subject performing the above method may be the electronic device 200 or its example circuit implementation 300 according to the embodiment of the present disclosure, so the foregoing is about the electronic device 200 or its example circuit implementation 300 and its functional unit embodiments A variety of aspects apply here.
  • the electronic device 200 may be implemented as any type of base station device, such as macro eNB and small eNB, and may also be implemented as any type of gNB (base station in a 5G system).
  • Small eNBs may be eNBs covering cells smaller than macro cells, such as pico eNBs, micro eNBs, and home (femto) eNBs.
  • the base station may be implemented as any other type of base station, such as NodeB and base transceiver station (BTS).
  • a base station may include: a subject (also referred to as a base station device) configured to control wireless communications; and one or more remote radio heads (RRHs) disposed at a different location than the subject.
  • RRHs remote radio heads
  • the electronic device 200 may also be implemented as any type of TRP.
  • the TRP can have sending and receiving functions, for example, it can receive information from user equipment and base station equipment, and can also send information to user equipment and base station equipment.
  • the TRP can serve the user equipment and be controlled by the base station equipment.
  • the TRP may have a structure similar to that of the base station equipment, or may only have the structure related to sending and receiving information in the base station equipment.
  • the electronic device 200 may also be various user devices, which may be implemented as mobile terminals such as smart phones, tablet personal computers (PCs), notebook PCs, portable game terminals, portable/dongle-type mobile routers, and digital cameras device) or an in-vehicle terminal (such as a car navigation device).
  • the user equipment may also be implemented as a terminal performing machine-to-machine (M2M) communication (also referred to as a machine type communication (MTC) terminal).
  • M2M machine-to-machine
  • MTC machine type communication
  • the user equipment may be a wireless communication module (such as an integrated circuit module comprising a single die) mounted on each of the above-mentioned user equipments.
  • eNB 1800 includes one or more antennas 1810 and base station equipment 1820.
  • the base station apparatus 1820 and each antenna 1810 may be connected to each other via an RF cable.
  • Each of the antennas 1810 includes a single or multiple antenna elements (such as multiple antenna elements included in a multiple-input multiple-output (MIMO) antenna), and is used by the base station apparatus 1820 to transmit and receive wireless signals.
  • the eNB 1800 may include multiple antennas 1810.
  • multiple antennas 1810 may be compatible with multiple frequency bands used by eNB 1800.
  • 10 shows an example in which the eNB 1800 includes multiple antennas 1810, the eNB 1800 may also include a single antenna 1810.
  • the base station apparatus 1820 includes a controller 1821 , a memory 1822 , a network interface 1823 , and a wireless communication interface 1825 .
  • the controller 1821 may be, for example, a CPU or a DSP, and operates various functions of a higher layer of the base station apparatus 1820 .
  • the controller 1821 generates data packets from the data in the signal processed by the wireless communication interface 1825, and communicates the generated packets via the network interface 1823.
  • the controller 1821 may bundle data from a plurality of baseband processors to generate a bundled packet, and deliver the generated bundled packet.
  • the controller 1821 may have logical functions to perform controls such as radio resource control, radio bearer control, mobility management, admission control, and scheduling. This control may be performed in conjunction with nearby eNB or core network nodes.
  • the memory 1822 includes RAM and ROM, and stores programs executed by the controller 1821 and various types of control data such as a terminal list, transmission power data, and scheduling data.
  • the network interface 1823 is a communication interface for connecting the base station apparatus 1820 to the core network 1824 .
  • Controller 1821 may communicate with core network nodes or further eNBs via network interface 1823 .
  • the eNB 1800 and core network nodes or other eNBs may be connected to each other through logical interfaces such as S1 interface and X2 interface.
  • the network interface 1823 may also be a wired communication interface or a wireless communication interface for wireless backhaul. If the network interface 1823 is a wireless communication interface, the network interface 1823 may use a higher frequency band for wireless communication than the frequency band used by the wireless communication interface 1825 .
  • Wireless communication interface 1825 supports any cellular communication scheme, such as Long Term Evolution (LTE) and LTE-Advanced, and provides wireless connectivity to terminals located in cells of eNB 1800 via antenna 1810.
  • the wireless communication interface 1825 may generally include, for example, a baseband (BB) processor 1826 and RF circuitry 1827 .
  • the BB processor 1826 may perform, for example, encoding/decoding, modulation/demodulation, and multiplexing/demultiplexing, and performs layers such as L1, Medium Access Control (MAC), Radio Link Control (RLC), and Packet Data Convergence Protocol (PDCP)) various types of signal processing.
  • the BB processor 1826 may have some or all of the above-described logical functions.
  • the BB processor 1826 may be a memory storing a communication control program, or a module including a processor and associated circuitry configured to execute the program.
  • the update procedure may cause the functionality of the BB processor 1826 to change.
  • the module may be a card or blade that is inserted into a slot in the base station device 1820. Alternatively, the module can also be a chip mounted on a card or blade.
  • the RF circuit 1827 may include, for example, a mixer, a filter, and an amplifier, and transmit and receive wireless signals via the antenna 1810 .
  • the wireless communication interface 1825 may include a plurality of BB processors 1826.
  • multiple BB processors 1826 may be compatible with multiple frequency bands used by eNB 1800.
  • the wireless communication interface 1825 may include a plurality of RF circuits 1827 .
  • multiple RF circuits 1827 may be compatible with multiple antenna elements.
  • FIG. 10 shows an example in which the wireless communication interface 1825 includes multiple BB processors 1826 and multiple RF circuits 1827 , the wireless communication interface 1825 may also include a single BB processor 1826 or a single RF circuit 1827 .
  • the phase adjustment unit 210, the low-resolution analog-to-digital conversion unit 220, and the compensated phase estimation unit 240 in the electronic device 200 previously described with reference to FIG. 2 can be implemented through the wireless communication interface 1825 or the like.
  • a high-speed flip-flop or the like may be provided in the wireless communication interface 1825 to function as the low-resolution analog-to-digital conversion unit 220 .
  • the demodulation unit 230 in the electronic device 200 can be implemented by, for example, the controller 1821 or the like.
  • the controller 1821 may perform at least a part of the functions of the demodulation unit 230 and the like by executing the instructions stored in the memory 1822, which will not be repeated here.
  • eNB 1930 includes one or more antennas 1940, base station equipment 1950, and RRH 1960.
  • the RRH 1960 and each antenna 1940 may be connected to each other via RF cables.
  • the base station apparatus 1950 and the RRH 1960 may be connected to each other via high-speed lines such as fiber optic cables.
  • Each of the antennas 1940 includes a single or multiple antenna elements (such as multiple antenna elements included in a MIMO antenna) and is used by the RRH 1960 to transmit and receive wireless signals.
  • the eNB 1930 may include multiple antennas 1940.
  • multiple antennas 1940 may be compatible with multiple frequency bands used by eNB 1930.
  • 11 shows an example in which the eNB 1930 includes multiple antennas 1940, the eNB 1930 may also include a single antenna 1940.
  • the base station apparatus 1950 includes a controller 1951 , a memory 1952 , a network interface 1953 , a wireless communication interface 1955 , and a connection interface 1957 .
  • the controller 1951 , the memory 1952 and the network interface 1953 are the same as the controller 1821 , the memory 1822 and the network interface 1823 described with reference to FIG. 10 .
  • the network interface 1953 is a communication interface for connecting the base station apparatus 1950 to the core network 1954 .
  • Wireless communication interface 1955 supports any cellular communication scheme, such as LTE and LTE-Advanced, and provides wireless communication via RRH 1960 and antenna 1940 to terminals located in a sector corresponding to RRH 1960.
  • the wireless communication interface 1955 may generally include, for example, a BB processor 1956.
  • the BB processor 1956 is the same as the BB processor 1826 described with reference to FIG. 10, except that the BB processor 1956 is connected to the RF circuit 1964 of the RRH 1960 via the connection interface 1957.
  • the wireless communication interface 1955 may include multiple BB processors 1956 .
  • multiple BB processors 1956 may be compatible with multiple frequency bands used by eNB 1930.
  • FIG. 11 shows an example in which the wireless communication interface 1955 includes multiple BB processors 1956 , the wireless communication interface 1955 may also include a single BB processor 1956 .
  • connection interface 1957 is an interface for connecting the base station apparatus 1950 (the wireless communication interface 1955 ) to the RRH 1960.
  • the connection interface 1957 may also be a communication module for communication in the above-mentioned high-speed line connecting the base station device 1950 (the wireless communication interface 1955) to the RRH 1960.
  • the RRH 1960 includes a connection interface 1961 and a wireless communication interface 1963.
  • connection interface 1961 is an interface for connecting the RRH 1960 (the wireless communication interface 1963 ) to the base station apparatus 1950.
  • the connection interface 1961 may also be a communication module for communication in the above-mentioned high-speed line.
  • the wireless communication interface 1963 transmits and receives wireless signals via the antenna 1940 .
  • Wireless communication interface 1963 may typically include RF circuitry 1964, for example.
  • RF circuitry 1964 may include, for example, mixers, filters, and amplifiers, and transmit and receive wireless signals via antenna 1940 .
  • the wireless communication interface 1963 may include a plurality of RF circuits 1964 .
  • multiple RF circuits 1964 may support multiple antenna elements.
  • FIG. 11 shows an example in which the wireless communication interface 1963 includes multiple RF circuits 1964 , the wireless communication interface 1963 may include a single RF circuit 1964 .
  • the phase adjustment unit 210, the low-resolution analog-to-digital conversion unit 220, and the compensated phase estimation unit 240 in the electronic device 200 described earlier with reference to FIG. 2 can be implemented through the wireless communication interface 1963.
  • a high-speed flip-flop or the like may be provided in the wireless communication interface 1963 to function as the low-resolution analog-to-digital conversion unit 220 .
  • the demodulation unit 230 in the electronic device 200 may be implemented by the controller 1951 or the like.
  • the controller 1951 may perform at least a part of the functions of the demodulation unit 230 and the like by executing the instructions stored in the memory 1952, which will not be repeated here.
  • FIG. 12 is a block diagram showing an example of a schematic configuration of a smartphone 2000 to which the technology of the present disclosure can be applied.
  • Smartphone 2000 includes processor 2001, memory 2002, storage device 2003, external connection interface 2004, camera device 2006, sensor 2007, microphone 2008, input device 2009, display device 2010, speaker 2011, wireless communication interface 2012, one or more Antenna switch 2015, one or more antennas 2016, bus 2017, battery 2018, and auxiliary controller 2019.
  • the processor 2001 may be, for example, a CPU or a system on a chip (SoC), and controls the functions of the application layer and further layers of the smartphone 2000 .
  • the memory 2002 includes RAM and ROM, and stores data and programs executed by the processor 2001 .
  • the storage device 2003 may include a storage medium such as a semiconductor memory and a hard disk.
  • the external connection interface 2004 is an interface for connecting external devices such as memory cards and Universal Serial Bus (USB) devices to the smartphone 2000 .
  • the camera 2006 includes an image sensor such as a charge coupled device (CCD) and a complementary metal oxide semiconductor (CMOS), and generates a captured image.
  • Sensors 2007 may include a set of sensors, such as measurement sensors, gyroscope sensors, geomagnetic sensors, and acceleration sensors.
  • the microphone 2008 converts the sound input to the smartphone 2000 into an audio signal.
  • the input device 2009 includes, for example, a touch sensor, a keypad, a keyboard, buttons, or switches configured to detect a touch on the screen of the display device 2010, and receives operations or information input from a user.
  • the display device 2010 includes a screen such as a liquid crystal display (LCD) and an organic light emitting diode (OLED) display, and displays an output image of the smartphone 2000.
  • the speaker 2011 converts the audio signal output from the smartphone 2000 into sound.
  • the wireless communication interface 2012 supports any cellular communication scheme, such as LTE and LTE-Advanced, and performs wireless communication.
  • Wireless communication interface 2012 may typically include, for example, BB processor 2013 and RF circuitry 2014.
  • the BB processor 2013 can perform, for example, encoding/decoding, modulation/demodulation, and multiplexing/demultiplexing, and perform various types of signal processing for wireless communication.
  • the RF circuit 2014 may include, for example, mixers, filters, and amplifiers, and transmit and receive wireless signals via the antenna 2016 .
  • the wireless communication interface 2012 may be a chip module on which the BB processor 2013 and the RF circuit 2014 are integrated. As shown in FIG. 12 , the wireless communication interface 2012 may include multiple BB processors 2013 and multiple RF circuits 2014 .
  • FIG. 12 shows an example in which the wireless communication interface 2012 includes multiple BB processors 2013 and multiple RF circuits 2014
  • the wireless communication interface 2012 may include a single BB processor 2013 or a single RF circuit 2014 .
  • the wireless communication interface 2012 may support additional types of wireless communication schemes, such as short-range wireless communication schemes, near field communication schemes, and wireless local area network (LAN) schemes.
  • the wireless communication interface 2012 may include a BB processor 2013 and an RF circuit 2014 for each wireless communication scheme.
  • Each of the antenna switches 2015 switches the connection destination of the antenna 916 between a plurality of circuits included in the wireless communication interface 2012 (eg, circuits for different wireless communication schemes).
  • Each of the antennas 2016 includes a single or multiple antenna elements (such as multiple antenna elements included in a MIMO antenna), and is used for the wireless communication interface 2012 to transmit and receive wireless signals.
  • smartphone 2000 may include multiple antennas 2016 .
  • FIG. 12 shows an example in which the smartphone 2000 includes multiple antennas 2016
  • the smartphone 2000 may also include a single antenna 2016 .
  • the smartphone 2000 may include an antenna 2016 for each wireless communication scheme.
  • the antenna switch 2015 can be omitted from the configuration of the smartphone 2000 .
  • the bus 2017 connects the processor 2001, the memory 2002, the storage device 2003, the external connection interface 2004, the camera device 2006, the sensor 2007, the microphone 2008, the input device 2009, the display device 2010, the speaker 2011, the wireless communication interface 2012, and the auxiliary controller 2019 to each other connect.
  • the battery 2018 provides power to the various blocks of the smartphone 2000 shown in FIG. 12 via feeders, which are partially shown in phantom in the figure.
  • the auxiliary controller 2019 operates the minimum necessary functions of the smartphone 2000, eg, in a sleep mode.
  • the phase adjustment unit 210 , the low-resolution analog-to-digital conversion unit 220 and the compensated phase estimation unit 240 in the electronic device 200 described earlier with reference to FIG. 2 may be implemented through the wireless communication interface 2012 .
  • a high-speed flip-flop or the like may be provided in the wireless communication interface 2012 to function as the low-resolution analog-to-digital conversion unit 220 .
  • the demodulation unit 230 in the electronic device 200 may be implemented by the processor 2001 or the auxiliary controller 2019 .
  • the processor 2001 or the auxiliary controller 2019 may execute at least a part of the functions of the demodulation unit 230 and the like by executing the instructions stored in the memory 2002 or the storage device 2003 , which will not be repeated here.
  • FIG. 13 is a block diagram showing an example of a schematic configuration of a car navigation apparatus 2120 to which the technology of the present disclosure can be applied.
  • the car navigation device 2120 includes a processor 2121, a memory 2122, a global positioning system (GPS) module 2124, a sensor 2125, a data interface 2126, a content player 2127, a storage medium interface 2128, an input device 2129, a display device 2130, a speaker 2131, a wireless A communication interface 2133, one or more antenna switches 2136, one or more antennas 2137, and a battery 2138.
  • GPS global positioning system
  • the processor 2121 may be, for example, a CPU or a SoC, and controls the navigation function and other functions of the car navigation device 2120 .
  • the memory 2122 includes RAM and ROM, and stores data and programs executed by the processor 2121 .
  • the GPS module 2124 measures the position (such as latitude, longitude, and altitude) of the car navigation device 2120 using GPS signals received from GPS satellites.
  • Sensors 2125 may include a set of sensors such as gyroscope sensors, geomagnetic sensors, and air pressure sensors.
  • the data interface 2126 is connected to, for example, the in-vehicle network 2141 via a terminal not shown, and acquires data generated by the vehicle, such as vehicle speed data.
  • the content player 2127 reproduces content stored in storage media such as CDs and DVDs, which are inserted into the storage media interface 2128 .
  • the input device 2129 includes, for example, a touch sensor, a button, or a switch configured to detect a touch on the screen of the display device 2130, and receives an operation or information input from a user.
  • the display device 2130 includes a screen such as an LCD or OLED display, and displays an image of a navigation function or reproduced content.
  • the speaker 2131 outputs the sound of the navigation function or the reproduced content.
  • the wireless communication interface 2133 supports any cellular communication scheme such as LTE and LTE-Advanced, and performs wireless communication.
  • Wireless communication interface 2133 may generally include, for example, BB processor 2134 and RF circuitry 2135.
  • the BB processor 2134 may perform, for example, encoding/decoding, modulation/demodulation, and multiplexing/demultiplexing, and perform various types of signal processing for wireless communication.
  • the RF circuit 2135 may include, for example, mixers, filters, and amplifiers, and transmit and receive wireless signals via the antenna 2137 .
  • the wireless communication interface 2133 can also be a chip module on which the BB processor 2134 and the RF circuit 2135 are integrated. As shown in FIG.
  • the wireless communication interface 2133 may include a plurality of BB processors 2134 and a plurality of RF circuits 2135 .
  • FIG. 13 shows an example in which the wireless communication interface 2133 includes multiple BB processors 2134 and multiple RF circuits 2135
  • the wireless communication interface 2133 may include a single BB processor 2134 or a single RF circuit 2135 .
  • the wireless communication interface 2133 may support another type of wireless communication scheme, such as a short-range wireless communication scheme, a near field communication scheme, and a wireless LAN scheme.
  • the wireless communication interface 2133 may include the BB processor 2134 and the RF circuit 2135 for each wireless communication scheme.
  • Each of the antenna switches 2136 switches the connection destination of the antenna 2137 among a plurality of circuits included in the wireless communication interface 2133, such as circuits for different wireless communication schemes.
  • Each of the antennas 2137 includes a single or multiple antenna elements (such as multiple antenna elements included in a MIMO antenna), and is used for the wireless communication interface 2133 to transmit and receive wireless signals.
  • the car navigation device 2120 may include a plurality of antennas 2137 .
  • FIG. 13 shows an example in which the car navigation device 2120 includes a plurality of antennas 2137 , the car navigation device 2120 may also include a single antenna 2137 .
  • the car navigation device 2120 may include an antenna 2137 for each wireless communication scheme.
  • the antenna switch 2136 may be omitted from the configuration of the car navigation device 2120.
  • the battery 2138 provides power to the various blocks of the car navigation device 2120 shown in FIG. 13 via feeders, which are partially shown as dashed lines in the figure.
  • the battery 2138 accumulates power supplied from the vehicle.
  • the phase adjustment unit 210 , the low-resolution analog-to-digital conversion unit 220 and the compensated phase estimation unit 240 in the electronic device 200 described earlier with reference to FIG. 2 can be implemented through the wireless communication interface 2133 .
  • a high-speed flip-flop or the like may be provided in the wireless communication interface 2133 to function as the low-resolution analog-to-digital conversion unit 220 .
  • the demodulation unit 230 in the electronic device 200 may be implemented by the processor 2121 .
  • the processor 2121 may perform at least a part of the functions of the demodulation unit 230 by executing the instructions stored in the memory 2122, which will not be repeated here.
  • the techniques of this disclosure may also be implemented as an in-vehicle system (or vehicle) 2140 that includes one or more blocks of a car navigation device 2120 , an in-vehicle network 2141 , and a vehicle module 2142 .
  • the vehicle module 2142 generates vehicle data such as vehicle speed, engine speed, and fault information, and outputs the generated data to the in-vehicle network 2141.
  • the units shown in dotted boxes in the functional block diagram shown in the drawings all indicate that the functional units are optional in the corresponding device, and each optional functional unit can be combined in an appropriate manner to achieve the required functions .
  • a plurality of functions included in one unit in the above embodiments may be implemented by separate devices.
  • multiple functions implemented by multiple units in the above embodiments may be implemented by separate devices, respectively.
  • one of the above functions may be implemented by multiple units. Needless to say, such a configuration is included in the technical scope of the present disclosure.
  • the steps described in the flowcharts include not only processing performed in time series in the stated order, but also processing performed in parallel or individually rather than necessarily in time series. Furthermore, even in the steps processed in time series, needless to say, the order can be appropriately changed.

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Abstract

提供了电子设备、信号处理方法和非暂态计算机可读存储介质。电子设备包括处理电路，该处理电路被配置为：根据利用导频信号估计的补偿相位，对所接收的复信号进行相位调整；通过低分辨率模数转换处理，基于经相位调整的复信号获得接收比特序列；以及基于利用导频信号训练获得的解调神经网络，对接收比特序列进行解调，以获得所接收的复信号的调制符号。根据本公开的实施例的至少一方面，能够在接收端利用基于导频信号获得的补偿相位改进低分辨率模数转换单元的转换精度，并通过基于导频信号训练获得的解调神经网络实现对混合失真的补偿，从而针对诸如太赫兹信号的高传输数据率信号也能够实现准确的解调判决。

Description

电子设备、信号处理方法以及计算机可读存储介质
本申请要求于2020年11月9日提交中国专利局、申请号为202011239774.1、发明名称为“电子设备、信号处理方法以及计算机可读存储介质”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本申请涉及无线通信技术领域，更具体地，涉及一种适合于对诸如太赫兹信号的高传输数据率信号进行解调的电子设备、信号处理方法以及非暂态计算机可读存储介质。
背景技术
太赫兹通信传输数据率高，因此理想情况下期望接收机具有高速率的模数转换器件。实际应用中，由于成本和技术限制，只能使用单比特接收机等设备以低精度的模数转换器件完成采集，这导致接收端难以对接收信号进行准确判决，因而影响了解调判决的准确性。此外，太赫兹频段位于微波频段和光频段之间，此频段的通信器件制作难度高，存在诸如功放非线性效应、同相(in-phase)支路(I支路)和正交(quadrature)支路(Q支路)失衡、载波相位噪声等硬件失配效应。这样的硬件失配效应将会引起接收信号的混合失真，造成通信性能的下降，从而进一步影响解调判决的准确性。
因此，现有技术中，无法实现诸如太赫兹信号的高传输数据率信号的准确解调。
发明内容
在下文中给出了关于本公开的简要概述，以便提供关于本公开的某些方面的基本理解。但是，应当理解，这个概述并不是关于本公开的穷举性概述。它并不是意图用来确定本公开的关键性部分或重要部分，也不是意图用来限定本公开的范围。其目的仅仅是以简化的形式给出关于本公开的某些概念，以此作为稍后给出的更详细描述的前序。
鉴于上述问题，本公开提出一种适合于对诸如太赫兹信号的高传输数据率 信号进行处理的电子设备、信号处理方法以及非暂态计算机可读存储介质，其能够在接收端利用基于导频信号获得的相位估计和解调神经网络改进低分辨率模数转换单元(例如单比特模数转换单元)的转换精度并对混合失真进行补偿，从而提高了解调判决的准确性。
根据本公开的一方面，提供了一种电子设备，其包括处理电路，该处理电路被配置为：根据利用导频信号估计的补偿相位，对所接收的复信号进行相位调整；通过低分辨率模数转换处理，基于经相位调整的复信号获得接收比特序列；以及基于利用导频信号训练获得的解调神经网络，对接收比特序列进行解调，以获得所接收的复信号的调制符号。
根据本公开的再一方面，提供了一种信号处理方法，其包括：根据利用导频信号估计的补偿相位，对所接收的复信号进行相位调整；通过低分辨率模数转换处理，基于经相位调整的复信号获得接收比特序列；以及基于利用导频信号训练获得的解调神经网络，对接收比特序列进行解调，以获得所接收的复信号的调制符号。
根据本公开的再一方面，还提供了一种存储有可执行指令的非暂态计算机可读存储介质，该可执行指令当由处理器执行时，使得处理器执行上述信号处理方法或电子设备的各个功能。
根据本公开的其它方面，还提供了用于实现上述根据本公开的信号处理方法的计算机程序代码和计算机程序产品。
根据本公开的实施例的至少一方面，能够在接收端利用基于导频信号获得的补偿相位改进低分辨率模数转换单元的转换精度，并通过基于导频信号训练获得的解调神经网络实现对混合失真的补偿，从而针对诸如太赫兹信号的高传输数据率信号也能够实现准确的解调判决。
在下面的说明书部分中给出本公开实施例的其它方面，其中，详细说明用于充分地公开本公开实施例的优选实施例，而不对其施加限定。
附图说明
在此描述的附图只是为了所选实施例的示意的目的而非全部可能的实施，并且不旨在限制本公开的范围。在附图中：
图1是示出现有技术中混合失真导致的接收信号的星座图扭曲的示意图；
图2是示出根据本公开实施例的电子设备的一个配置示例的示意性框图；
图3是用于说明根据本公开实施例的电子设备的示例电路实现的示意性框图；
图4示意性地示出了根据本公开实施例的电子设备能够采用的解调神经网络的示例；
图5是用于说明根据本公开实施例的电子设备所接收的复信号的数据结构的示意图；
图6是示出根据本公开实施例的信息交互过程的示例的示意图；
图7是用于说明相位噪声导致的性能下降的示例仿真结果的说明图；
图8是用于说明根据本公开实施例的电子设备所进行的解调处理的性能的示例仿真结果的图；
图9是示出根据本公开实施例的信号处理方法的过程示例的流程图；
图10是示出可以应用本公开内容的技术的eNB的示意性配置的第一示例的框图；
图11是示出可以应用本公开内容的技术的eNB的示意性配置的第二示例的框图；
图12是示出可以应用本公开内容的技术的智能电话的示意性配置的示例的框图；
图13是示出可以应用本公开内容的技术的汽车导航设备的示意性配置的示例的框图。
虽然本公开容易经受各种修改和替换形式，但是其特定实施例已作为例子在附图中示出，并且在此详细描述。然而应当理解的是，在此对特定实施例的描述并不打算将本公开限制到公开的具体形式，而是相反地，本公开目的是要覆盖落在本公开的精神和范围之内的所有修改、等效和替换。要注意的是，贯穿几个附图，相应的标号指示相应的部件。
具体实施方式
现在参考附图来更加充分地描述本公开的例子。以下描述实质上只是示例性的，而不旨在限制本公开、应用或用途。
提供了示例实施例，以便本公开将会变得详尽，并且将会向本领域技术人员充分地传达其范围。阐述了众多的特定细节如特定部件、装置和方法的例子，以提供对本公开的实施例的详尽理解。对于本领域技术人员而言将会明显的是，不需要使用特定的细节，示例实施例可以用许多不同的形式来实施，它们都不应当被解释为限制本公开的范围。在某些示例实施例中，没有详细地描述众所周知的过程、众所周知的结构和众所周知的技术。
将按照以下顺序进行描述：
1.问题的描述
2.电子设备的配置示例
2.1配置示例
2.2示例电路实现
2.3补偿相位的确定
2.4解调神经网络的训练
3.信息交互过程的示例
4.仿真示例
5.方法实施例
6.应用示例
<1.问题的描述>
太赫兹通信传输数据率高，因此理想情况下期望接收机具有高速率的模数转换器件。实际应用中，由于成本和技术限制，只能使用单比特接收机等设备以低精度的模数转换器件完成采集。目前，针对低频段信号的单比特接收机一般利用似然信息判决完成信号的解调。然而，对于太赫兹通信，由于太赫兹硬件失配的存在，接收信号的似然信息的准确表达不易获取，因此无法利用传统方法完成太赫兹单比特接收机的准确判决。这导致接收端难以对接收信号进行准确判决，因而影响了解调判决的准确性。
此外，太赫兹频段位于微波频段和光频段之间，此频段的通信器件制作难度高，存在诸如功放非线性效应、同相(in-phase)支路(I支路)和正交(quadrature)支路(Q支路)失衡、载波相位噪声等硬件失配效应。这样的硬件失配效应将会引起接收信号的混合失真，造成通信性能的下降，从而进一 步影响解调判决的准确性。
更具体地，对于采用正交相移键控(quadrature phase shift keying，QPSK)调制的太赫兹通信，其理想情况是在发射机处针对基带信号(二进制比特序列)的每两个比特构成的双比特组(在本文中，也将其称为“调制符号”)，分别以I路载波信号cos(2πf ct)和Q路载波信号sin(2πf ct)(其中，f c为载波频率，t表示时间)进行调制之后进行相加，即得到QPSK调制的一个传输符号或发射符号(transmission symbol)。作为示例，对于四个不同的双比特组或调制符号11,10,00,01，QPSK调制后可以分别得到1+j,1-j,-1-j,-1+j的传输符号。在QPSK星座图中，每个传输符号对应于一个星座点，并且由每个星座点和原点构成的矢量与坐标轴之间的角度是45度，每个星座点与坐标轴之间具有相等的距离。
然而，在实际应用中，发射机可能存在IQ路失衡。例如，可以以∈ T，φ T分别表示IQ路幅度和相位的失衡因子，则I路载波信号变为(1+∈ T)cos(2πf ct-φ T)，Q路载波信号变为(1-∈ T)sin(2πf ct+φ T)。同时，进一步考虑发射机处的相位噪声θ T，则QPSK调制后理想的发射信号(复信号)s将变为下述形式：
Figure PCTCN2021128116-appb-000001
其中，μ T＝cosφ T-j∈ Tsinφ T,v T＝∈ Tcosφ T-jsinφ T，v Ts *为IQ失衡引起的镜像干扰项。
这里，作为示例，相位噪声θ T可以具有块游走模型的形式，即θ T在第k个传输块内为固定值θ k，并且在相邻传输块之间相差一个高斯分布的随机游走项Δθ k。第k+1个传输块的相位噪声θ k+1可表示为下述形式：
Figure PCTCN2021128116-appb-000002
其中，
Figure PCTCN2021128116-appb-000003
表示均值为0，方差为
Figure PCTCN2021128116-appb-000004
的正态分布，
Figure PCTCN2021128116-appb-000005
也称为游走项方差。
发射机处进行QPSK调制后的信号
Figure PCTCN2021128116-appb-000006
将会通过功率放大器(Power Amplifiter)进行放大。功率放大器的非线性特性会导致信号
Figure PCTCN2021128116-appb-000007
的进一步失真。例如可以采用无记忆多项式模型表示功率放大器的非线性特性，并且功率放大器输出的信号可表示为下述形式：
Figure PCTCN2021128116-appb-000008
其中，2K-1表示多项式阶次。
发射机的功率放大器输出的上述信号s PA将会经过信道传输至接收机一侧。由于太赫兹通信通常在收发端均采用极高方向增益的天线，因此信道中的有效传输路径可认为只有一条，并且可以采用平衰落信道模型表示。例如，接收机侧接收到的信号可表示为下述形式：
y＝hs PA+w…等式(4)
其中，h为信道衰落因子，
Figure PCTCN2021128116-appb-000009
为加性高斯白噪声(Additive white Gaussian noise，AWGN)，
Figure PCTCN2021128116-appb-000010
表示均值为0，方差为
Figure PCTCN2021128116-appb-000011
的循环对称复高斯分布(circularly symmetric complex Gaussian，CSCG)。
对于上述接收信号(复信号)y，接收机处的理想处理是将其分别与I路载波信号cos(2πf ct)和Q路载波信号sin(2πf ct)相乘，以便获得该复信号的实部和虚部(也可以称为I路信号和Q路信号)。然而，与发射机类似地，接收机处也可能出现IQ路失衡以及相位噪声。以∈ R，φ R分别表示IQ路的幅度失衡因子和相位失衡因子，则接收机处的I路载波信号变为(1+∈ R)cos(2πf ct-φ R)，Q路载波信号变为(1-∈ R)sin(2πf ct+φ R)。同时，进一步考虑接收机处的相位噪声θ R，则最终接收信号可表示为下述形式：
Figure PCTCN2021128116-appb-000012
其中，μ R＝cosφ R+j∈ Rsinφ R,v R＝∈ Rcosφ R-jsinφ R。
上述等式(5)形式的接收信号可以表示为下述的复信号的形式：
Figure PCTCN2021128116-appb-000013
其中，
Figure PCTCN2021128116-appb-000014
和
Figure PCTCN2021128116-appb-000015
分别为复信号
Figure PCTCN2021128116-appb-000016
的实部和虚部。
对于涉及以上讨论的各种失真的诸如等式(6)形式的复信号，以现有技术的线性均衡策略在接收端获得的
Figure PCTCN2021128116-appb-000017
和
Figure PCTCN2021128116-appb-000018
所构成的星座点的示意图如图1所示，其中，星座点和原点构成的矢量与坐标轴的夹角θ不再是理想情况下的45度，并且星座点与原点的距离d 1、d 2、d 3、d 4不再相等，即，发生了星座图的扭曲。
从以上关于失真的讨论可以看出，导致诸如图1所示的星座图的扭曲的原 因有很多，为了解决这种问题，一般需要发送端的补偿(或预失真)算法和接收端的补偿算法来共同解决，其中，发送端的预失真算法可以处理功放非线性和发送端IQ支路失衡，接收端的补偿算法可以处理接收端的IQ支路失衡和相位噪声。然而，太赫兹频段的硬件制作难度大、成本高，为进行硬件失配效应研究，发送端的补偿链路常需要高精度高速率的采样链路，其设计难度较大而很难实现；仅使用接收端的补偿技术则无法对太赫兹信道的混合失真进行有效补偿。
为此，本公开提供了一种适合于对诸如太赫兹信号的高传输数据率信号进行处理的电子设备、信号处理方法和非暂态计算机可读存储介质，其能够在接收端利用基于导频信号获得的相位估计和解调神经网络改进低分辨率模数转换单元(例如单比特模数转换单元)的转换精度并对混合失真进行补偿，从而提高了解调判决的准确性。
根据本公开的电子设备可以是用户设备侧的电子设备，也可以是网络侧的电子设备，只要其能够作用接收端设备并进行相应的信号处理即可。
用户设备侧的电子设备可以被实现为各种用户设备，例如移动终端(诸如智能电话、平板个人计算机(PC)、笔记本式PC、便携式游戏终端、便携式/加密狗型移动路由器和数字摄像装置)或者车载终端(诸如汽车导航设备)。上述用户设备还可以被实现为执行机器对机器(M2M)通信的终端(也称为机器类型通信(MTC)终端)。此外，用户设备可以包括安装在上述终端中的每个终端上的无线通信模块(诸如包括单个晶片的集成电路模块)等。
网络侧的电子设备可以是基站设备本身，例如可以是eNB(演进型节点B)，也可以是gNB，并且也可以是任何类型的TRP(发送和接收端口)。该TRP可以具备发送和接收功能，例如可以从用户设备和基站设备接收信息，也可以向用户设备和基站设备发送信息。在一个示例中，TRP可以为用户设备提供服务，并且受基站设备的控制。也就是说，基站设备可以通过TRP向用户设备提供服务。在下文的一些具体实施例或示例中，以基站设备作为网络侧的电子设备的示例进行描述，但本公开不限于此，而是可以适当地适用于具有类似功能的电子设备的情形。
<2.电子设备的配置示例>
[2.1配置示例]
图2是示出根据本公开的实施例的电子设备的一个配置示例的框图。
如图2所示，电子设备200可以包括相位调整单元210、低分辨率模数转换单元220、解调单元230以及可选的补偿相位估计单元240。
这里，电子设备200的各个单元都可以包括在处理电路中。需要说明的是，电子设备200既可以包括一个处理电路，也可以包括多个处理电路。进一步，处理电路可以包括各种分立的功能单元以执行各种不同的功能和/或操作。需要说明的是，这些功能单元可以是物理实体或逻辑实体，并且不同称谓的单元可能由同一个物理实体实现。
根据本公开的实施例，电子设备200的相位调整单元210可以根据利用导频信号估计的补偿相位，对所接收的复信号进行相位调整。低分辨率模数转换单元220可以通过低分辨率模数转换处理，基于经相位调整的复信号获得接收比特序列。解调单元230可以基于利用导频信号训练获得的解调神经网络，对接收比特序列进行解调，以获得所接收的复信号的调制符号。
作为示例，复信号和导频信号各自可以包括QPSK调制信号，并且例如是诸如太赫兹信号的高传输数据率信号。这里，尽管使用了QPSK调制的术语，但本领域技术人员可以理解，在本公开的上下文中，所涉及的QPSK调制可以涵盖相对移相QPSK(offset QPSK,OQPSK)和差分DQPSK(differentialQPSK,DQPSK)调制等。
如此前在“问题的描述”部分所描述的，对于采用QPSK调制的诸如太赫兹信号的高传输数据率信号，由于各种失真导致的影响，电子设备200接收的例如具有等式(6)形式的复信号
Figure PCTCN2021128116-appb-000019
可能出现例如图1所示的星座图扭曲的问题。因此，电子设备200可以通过相位调整单元210根据利用导频信号估计的补偿相位θ *，对所接收的复信号进行相位调整，以得到下述形式的相位调整后的复信号：
Figure PCTCN2021128116-appb-000020
利用适当确定的补偿相位θ *，上述相位调整处理可以使接收信号的星座图适当旋转，从而例如最大化星座点与不同坐标轴(包括x轴和y轴)之间的最小距离，例如使得图1中的d 1、d 2、d 3和d 4中的最小距离最大化。这样的相位调整会提高接收信号正确判决的概率。稍后将会在“补偿相位的确定”部分描述如何确定补偿相位的细节。
作为示例，相位调整单元210可以被配置为根据补偿相位，控制本地振 荡器(图中未示出)对所接收的复信号进行相位旋转，以实现上述相位调整。例如包括在电子设备200中的或者连接到电子设备200的本地振荡器可以用于产生接收端的电子设备200处的I路载波信号和Q路载波信号，以用于获得复信号的实部和虚部。当相位调整单元210通过控制本地振荡器实现相位调整时，实际上是直接调整了载波信号的相位，再利用这样的载波信号与所接收的复信号相乘，从而可以直接获得相位调整后的复信号
Figure PCTCN2021128116-appb-000021
的实部
Figure PCTCN2021128116-appb-000022
和虚部
Figure PCTCN2021128116-appb-000023
相位调整后的复信号可以从等式(6)变为下述形式：
Figure PCTCN2021128116-appb-000024
针对相位调整单元210进行相位调整后的复信号
Figure PCTCN2021128116-appb-000025
低分辨率模数转换单元220可以进行低分辨率模数转换处理，以获得接收比特序列。这里，低分辨率模数转换处理单元220可以被配置为通过低分辨率模数转换处理，分别基于相位调整后的复信号的实部和虚部获得实部接收比特序列和虚部接收比特序列。例如，低分辨率模数转换处理单元220可以通过对所述实部和所述虚部进行过采样以得到所接收的复信号的每个传输符号的实部接收比特序列和虚部接收比特序列。换言之，相位调整单元210可以针对I/Q支路进行单比特模拟信号采集，每次采集到的信号的实部
Figure PCTCN2021128116-appb-000026
和虚部
Figure PCTCN2021128116-appb-000027
可以表示为如下形式
Figure PCTCN2021128116-appb-000028
由于在存在噪声的情况下，接收信号判决的正确概率取决于所采集的各个传输符号的实部
Figure PCTCN2021128116-appb-000029
和虚部
Figure PCTCN2021128116-appb-000030
与0之间的比较、从而取决于各个星座点到不同坐标轴(包括x轴和y轴)的最小距离(例如图1中的d 1、d 2、d 3和d 4中的最小距离)，因此相位调整后增加了该最小距离将会提高接收信号正确判决的概率(即，增加了各个传输符号的实部
Figure PCTCN2021128116-appb-000031
和虚部
Figure PCTCN2021128116-appb-000032
与0之间的比较结果的正确概率)。
按照以上方式，对于所接收的复信号的每个传输符号，例如通过低分辨率模数转换单元220进行单比特过采样可以得到实部比特序列
Figure PCTCN2021128116-appb-000033
和虚部比特序列
Figure PCTCN2021128116-appb-000034
其中，N是信号的过采样倍数，例如可以为10。这样的实部和虚部接收比特序列也可以统一表示为接收比特序列
Figure PCTCN2021128116-appb-000035
对于通过低分辨率模数转换单元220获得的所接收的复信号的当前传输符号的接收比特序例如
Figure PCTCN2021128116-appb-000036
解调单元230可以基于利用导频信号训练获得的解调神经网络对其进行进行解调，以获得该传输符号的调制符号。该解调神经网络是通过例如标记好调制符号的导频信号进行训练而获得的，并且能够表征所接收的复信号的传输符号的接收比特序列与相应的调制符号之间的映射关系，稍后将会在“解调神经网络的训练”部分描述进一步的细节。
以上描述了根据本公开的实施例的电子设备的配置示例。利用根据本公开的实施例的电子设备，能够在接收端利用基于导频信号获得的补偿相位改进低分辨率模数转换单元的转换精度，并通过基于导频信号训练获得的解调神经网络实现对混合失真的补偿，从而针对诸如太赫兹信号的高传输数据率信号也能够实现准确的解调判决。
[2.2示例电路实现]
接下来，将参照图3描述图2所示的电子设备200的示例电路实现。图3是用于说明根据本公开实施例的电子设备的示例电路实现300的示意性框图，其中示出了分别作为图2中的相位调整单元210、低分辨率模数转换单元220、解调单元230、补偿相位估计单元240的示例的相位调整单元310、作为单比特触发器的ADC 320a和320b(在无需区分时也统称为ADC 320)、解调单元330以及补偿相位估计单元340。此外，图3还示出了作为接收端的电子设备的基本配置的天线350、本地振荡器LO、移相器360以及乘法电路370a和370b(在无需区分时也统称为乘法电路370)。这些部件(天线350、本地振荡器LO、移相器360、乘法电路370)可以是示例电路实现300的可选的附加部分，也可以是与示例电路实现300相连接的另外的电路部分，本公开对此不进行限制。
图3的各个单元310至340可以用于实现此前参照图2描述的电子设备 200的相应单元210至240的功能。例如，对于通过天线350接收的复信号
Figure PCTCN2021128116-appb-000037
相位调整单元310根据例如补偿相位估计单元340利用导频信号估计的补偿相位θ *控制本地振荡器LO对其进行相位旋转、相当于对载波信号cos(2πf ct)和sin(2πf ct)进行相位旋转，乘法电路370a和370b可以利用相应的相位调整后的载波信号cos(2πf ct+θ *)和sin(2πf ct+θ *)与复信号
Figure PCTCN2021128116-appb-000038
相乘，从而直接获得相位调整后的复信号
Figure PCTCN2021128116-appb-000039
的实部
Figure PCTCN2021128116-appb-000040
和虚部
Figure PCTCN2021128116-appb-000041
以此方式，在本公开实施例的电子设备的诸如图3的示例电路实现中，可以通过模拟器件实现对接收信号的相位补偿。
这里，作为低分辨率模数转换处理单元的示例的ADC 320a和320b可以分别对实部
Figure PCTCN2021128116-appb-000042
和虚部
Figure PCTCN2021128116-appb-000043
进行单比特过采样，以得到所接收的复信号
Figure PCTCN2021128116-appb-000044
的每个传输符号的实部比特序列
Figure PCTCN2021128116-appb-000045
和虚部比特序列
Figure PCTCN2021128116-appb-000046
其中，N是信号的过采样倍数，例如可以为10。进行单比特过采样的低分辨率模数转换处理单元可以利用模拟信号采集器例如高速触发器等实现。例如，可以采用DI公司的HMC729LC3C高速触发器作为本示例中的ADC 320。
通过ADC 320a和320b得到的所接收的复信号
Figure PCTCN2021128116-appb-000047
的各个传输符号的实部比特序列
Figure PCTCN2021128116-appb-000048
和虚部比特序列
Figure PCTCN2021128116-appb-000049
被输入到解调单元330，以得到所接收的复信号
Figure PCTCN2021128116-appb-000050
的各个传输符号的解调符号n。
作为示例，解调单元330可以利用深度前馈神经网络(Deep feedforward neural network，DFNN)进行解调处理，其利用深度前馈神经网络以任意精度拟合映射关系的拟合能力，拟合接收比特序列与调制符号之间的映射关系，进而完成解调判决。
图4示意性地示出了可以采用的解调神经网络400的示例。该网络包括输入层410、隐含层420、输出层430。输入层410例如具有2N个输入通道(N为过采样倍数，并且例如为10)，以针对所接收的复信号的每个传输符号来输入该传输符号的由实部比特序列
Figure PCTCN2021128116-appb-000051
和虚部比特序列
Figure PCTCN2021128116-appb-000052
构成的接收比特序列
Figure PCTCN2021128116-appb-000053
隐含层420的层数例如可以为3，各隐含层的神经元个数可以为(10,10,10)，并且隐含层的激活函数为双曲正切S型(Tan-Sigmoid,Tansig)函数，即f(x)＝2/(1+e -2x)-1。输出层430的激活 函数为线性整流(Rectified Linear Unit，ReLU)函数，即f(x)＝max(x,0)。
解调神经网络可以例如以下述等式表征所接收的复信号的每个传输符号的接收比特序列与相应的调制符号之间的映射关系：
Figure PCTCN2021128116-appb-000054
其中，DFNN(·)表示解调神经网络的输入输出关系，n表示通过解调神经网络得到的与当前传输符号的接收比特序列
Figure PCTCN2021128116-appb-000055
对应的该传输符号的调制符号，其例如可以为双比特组11,10,00,01中的一者。
如前所述，在本公开实施例的电子设备接收数据信号并进行相应处理之前，需要利用导频信号完成补偿相位的确定以及解调神经网络的训练。相应地，在图3的示例电路实现中，如果天线350所接收的不是数据信号而是导频信号，则通过相位调整单元310、作为低分辨率模数转换单元的示例的ADC 320a和320b等将对导频信号进行与数据信号类似的处理，以获得导频信号的接收比特序列，并利用该接收比特序列例如通过补偿相位估计单元340等的处理确定补偿相位或者解调单元440等的处理训练解调神经网络。
接下来，将继续结合图3的示例电路实现描述本公开实施例的电子设备可以实现的关于补偿相位的确定以及解调神经网络的训练的处理。
[2.3补偿相位的确定]
作为示例，本公开实施例的电子设备例如可以通过例如图3的补偿相位估计单元340等的处理，根据第一导频信号的传输符号以及基于所接收的第一导频信号获得的第一接收比特序列，估计补偿相位。
这里，假设发送端发送具有下述复信号形式的第一导频信号
p 1＝p 1I+jp 1Q….等式(11)
p 1I和p 1Q分别第一导频信号的实部和虚部部分，其根据导频信号的当前传输符号而可以分别为1或-1。
发送端例如可以通过对包括多个双比特组ab的比特序列进行QPSK调制，获得以上第一导频信号，其中，每个双比特组ab在调制后获得第一导频信号中的一个传输符号。第一导频信号中的各个传输符号可以对应于QPSK星座图中的至少两个相邻星座点，并且优选地可以仅对应于QPSK星座图中的两个相 邻星座点。由于QPSK星座图中的星座点的对称性，任意两个相邻的星座点与坐标轴的距离(诸如图1所示的d 1、d 2、d 3和d 4)即可以表征全部四个星座点与坐标轴的距离。
相应地，在发送端发送第一导频信号时，可以利用能够获得与所选择的两个相邻星座点对应的传输符号的比特序列进行QPSK调制，从而获得包括1+j,1-j,-1-j,-1+j中相应的两者的第一导频信号。作为示例，第一导频信号中的各个传输符号可以交替地对应于QPSK星座图中的两个相邻星座点。例如，如果选择两个双比特组11和10，第一导频信号的传输符号包括1+j和1-j(对应于QPSK星座图中右侧的两个相邻星座点)，则第一导频信号可以表示为[1+j,1-j,...,1+j,1-j]。相较于采用全部四个传输符号构成第一导频信号的方式，本示例的方式可以简化第一导频信号的产生。
接收端的电子设备例如可以通过图3的天线350获得上述第一导频信号p 1的接收信号
Figure PCTCN2021128116-appb-000056
(下文中也称为所接收的第一导频信号或在适当时简称为第一导频信号
Figure PCTCN2021128116-appb-000057
)。对于所接收的第一导频信号
Figure PCTCN2021128116-appb-000058
电子设备例如可以通过图3的相位调整单元310进行相位调整，并且可以通过例如图3的ADC 320进行的低分辨率模数转换处理，基于经过相位调整的、所接收的第一导频信号获得第一接收比特序列。
例如，对于通过天线350接收的第一导频信号
Figure PCTCN2021128116-appb-000059
相位调整单元310利用随着时间例如在0至π之间连续变化的、待确定的补偿相位θ(t)通过本地振荡器LO对其进行相位旋转、即对载波信号进行相位旋转，乘法电路370a和370b利用相应的相位调整后的载波信号cos(2πf ct+θ(t))和sin(2πf ct+θ(t))与第一导频信号
Figure PCTCN2021128116-appb-000060
相乘，从而获得相位调整后的所接收的第一导频信号
Figure PCTCN2021128116-appb-000061
的实部
Figure PCTCN2021128116-appb-000062
和虚部
Figure PCTCN2021128116-appb-000063
这里，ADC 320a和320b可以分别对相位调整后的第一导频信号
Figure PCTCN2021128116-appb-000064
的实部
Figure PCTCN2021128116-appb-000065
和虚部
Figure PCTCN2021128116-appb-000066
进行单比特过采样，以得到所接收的第一导频信号的每个传输符号的实部比特序列
Figure PCTCN2021128116-appb-000067
和虚部比特序列
Figure PCTCN2021128116-appb-000068
其中，N是信号的过采样倍数，例如可以为10。以上实部比特序列
Figure PCTCN2021128116-appb-000069
和虚部比特序列
Figure PCTCN2021128116-appb-000070
可以统称为第一接收比特序列
Figure PCTCN2021128116-appb-000071
电子设备还可以通过补偿相位估计单元340的处理，根据第一接收比特序 列(
Figure PCTCN2021128116-appb-000072
和
Figure PCTCN2021128116-appb-000073
)相对于第一导频信号p 1的传输符号的误差，估计补偿相位；并且可以根据所估计的补偿相位，对所接收的第一导频信号进行相位调整。例如，所述处理电路还被配置为将使所述误差最小的补偿相位确定为最终的补偿相位。
作为示例，补偿相位估计单元340可以根据所接收的第一导频信号的每个传输符号的实部比特序列
Figure PCTCN2021128116-appb-000074
和虚部比特序列
Figure PCTCN2021128116-appb-000075
与第一导频信号p 1的传输符号(即，第一导频信号p 1的实部p 1I和虚部p 1Q)之间的比较，确定最优的补偿相位。更具体地，这里可以在使补偿相位θ(t)连续变化(例如持续增加或减小)的同时，根据利用该补充相位进行相位调整后所获得的
Figure PCTCN2021128116-appb-000076
和
Figure PCTCN2021128116-appb-000077
中的每个比特(也可称为采样点)关于p 1I和p 1Q判决错误的个数进行最优相位判决。每一次相位改变以后，补偿相位估计单元340重新判决第一导频信号的当前传输符号的第一接收比特序列
Figure PCTCN2021128116-appb-000078
中的错误采样点的个数，如果错误采样点的个数减少，则说明该补偿相位有利于提高解调准确性，如果错误采样点的个数增加，则说明该补偿相位不利于提高解调准确性。以此方式，例如可以将使第一接收比特序列
Figure PCTCN2021128116-appb-000079
的错误采样点的个数最小的补偿相位确定为最终的补偿相位。因此，尽管无法直接计算例如图1所示的星座图中当前补偿相位下的各星座点与坐标轴之间的距离d 1、d 2、d 3和d 4的值，但仍然可以确定能够实现d 1、d 2、d 3和d 4中的最小距离最大化这一效果的最优补偿相位。
[2.4解调神经网络的训练]
作为示例，本公开实施例的电子设备例如可以通过例如图3的解调单元330的处理，利用以第二导频信号的调制符号标记的、基于所接收的第二导频信号获得的第二接收比特序列，通过训练获得解调神经网络。
这里，假设发送端发送具有下述复信号形式的第二导频信号
p 2＝p 2I+jp 2Q….等式(12)
p 2I和p 2Q分别为第二导频信号的实部和虚部部分，其根据导频信号的当前传输符号而可以分别为1或-1。发送端例如通过对包括多个双比特组ab的比 特序列进行QPSK调制，获得以上第二导频信号，其中，每个双比特组ab在调制后获得第二导频信号中的一个传输符号。
优选地，第二导频信号p 2包括与不同的双比特组(即，不同的调制符号)对应的全部传输符号，以便利用该导频信号训练解调神经网络来获得所接收到的复信号的各个传输符号的比特序列与各个调制符号之间的映射关系。换言之，第二导频信号p 2中的传输符号可以对应于QPSK星座图中的四个星座点。
相应地，在发送端发送第二导频信号时，可以利用能够获得全部四个星座点的比特序列(例如，利用包括四个不同的双比特组或调制符号11,10,00,01的比特序列)进行QPSK调制，从而获得包括1+j,1-j,-1-j,-1+j的第二导频信号。作为示例，第二导频信号p 2中的各个传输符号可以随机地对应于QPSK星座图中的四个星座点。例如，第二导频信号可以表示为[1+j,-1+j,1-j,-1-j,-1-j,1+j,…,1-j,1+j]。
接收端的电子设备例如可以通过图3的天线350获得上述第二导频信号p 2的接收信号
Figure PCTCN2021128116-appb-000080
(下文中也称为所接收的第二导频信号或在适当时简称为第二导频信号
Figure PCTCN2021128116-appb-000081
)。对于所接收的第二导频信号
Figure PCTCN2021128116-appb-000082
电子设备例如可以通过图3的相位调整单元310进行相位调整，并且可以通过例如图3的ADC 320进行的低分辨率模数转换处理，基于经过相位调整的、所接收的第二导频信号获得第二接收比特序列。
例如，对于通过天线350接收的第二导频信号
Figure PCTCN2021128116-appb-000083
相位调整单元310利用补偿相位估计单元所确定的补偿相位θ *通过本地振荡器LO对其进行相位旋转、即对载波信号进行相位旋转，乘法电路370a和370b利用相应的相位调整后的载波信号cos(2πf ct+θ *)和sin(2πf ct+θ *)与第二导频信号
Figure PCTCN2021128116-appb-000084
相乘，从而获得相位调整后的所接收的第二导频信号
Figure PCTCN2021128116-appb-000085
的实部
Figure PCTCN2021128116-appb-000086
和虚部
Figure PCTCN2021128116-appb-000087
这里，ADC 320a和320b可以分别对相位调整后的第二导频信号
Figure PCTCN2021128116-appb-000088
的实部
Figure PCTCN2021128116-appb-000089
和虚部
Figure PCTCN2021128116-appb-000090
进行单比特过采样，以得到所接收的第二导频信号的每个传输符号的实部比特序列
Figure PCTCN2021128116-appb-000091
和虚部比特序列
Figure PCTCN2021128116-appb-000092
其中，N是信号的过采样倍数，例如可以为10。以上实部比特序列
Figure PCTCN2021128116-appb-000093
和虚部比特序列
Figure PCTCN2021128116-appb-000094
可以统称为第二接收比特序列
Figure PCTCN2021128116-appb-000095
例如通过ADC 320得到的第二导频信号的当前传输符号的第二接收比特序列
Figure PCTCN2021128116-appb-000096
被输入到解调单元330，解调单元330可以将第二接收比特序列
Figure PCTCN2021128116-appb-000097
输入到如图4所示的解调神经网络中，并且获得解调神经网络的输出结果
Figure PCTCN2021128116-appb-000098
该输出结果
Figure PCTCN2021128116-appb-000099
是解调神经网络得到的第二导频信号的当前传输符号的调制符号或双比特组，其例如可以为双比特组11,10,00,01中的一者。解调单元330可以基于第二导频信号的各个调制符号与解调神经网络的输出结果
Figure PCTCN2021128116-appb-000100
之间的差异来构建损失函数，并例如通过梯度下降法等任意适当方式进行迭代训练，以例如在该损失函数取得最小值或不再变化时，确定解调神经网络的各个参数的最优取值。在本公开内容所构建的解调神经网络的基础上，本领域技术人员可以采用任意适当方式实现该解调神经网络的训练，这里不再赘述。
<3.信息交互过程的示例>
接下来，将参照图5和图6描述本公开实施例的信息交互过程的示例。
首先参照图5，图5是用于说明根据本公开实施例的电子设备所接收的复信号的数据结构的示意图。如图5所示，根据本公开实施例的电子设备所接收的复信号例如可以依次包括三个部分，即用于估计补偿相位的第一导频信号、用于训练解调神经网络的第二导频信号、以及数据信号。作为示例，这里的第一导频信号可以具有以上在“补偿相位的确定”部分中描述的第一导频信号的示例形式，第二导频信号可以具有以上在“解调神经网络的训练”部分中描述的第二导频信号的示例形式，作为数据信号的复信号可以是任意的QPSK调制信号。可以在每次通信时，从发送端向接收端发送依次包括以上三个部分的复信号。
接下来参照图6，图6是示出本公开实施例的信息交互过程的示例的示意图，其中示意性地示出了发送端和接收端(这里，接收端例如可以是此前描述的电子设备200或其示例电路实现300)以及两者之间的信息交互。这里，图6示出了例如一次通信中的发送端与接收端之间的信息交互的示例，其中传输了例如图5所示的复信号的三个部分。如图6所示，首先，发送端向接收端发送第一导频信号。接收端的电子设备根据第一导频信号的传输符号以及基于所接收的第一导频信号获得的第一接收比特序列，确定补偿相位。接着，发送端向接收端发送第二导频信号。接收端的电子设备利用以第二导频信号的调制符号标记的、基于所接收的第二导频信号获得的第二接收比特序列，通过训练获得解调神经网络。
然后，发送端向接收端发送作为数据信号的复信号。接收端的电子设备可以根据此前利用第一导频信号估计的补偿相位，对所接收的复信号进行相位调整。然后，接收端的电子设备可以通过低分辨率模数转换处理，基于经相位调整的复信号获得接收比特序列。接着，接收端的电子设备可以基于利用第二导频信号训练获得的解调神经网络，对接收比特序列进行解调，以获得所接收的复信号的调制符号。
图6所示的示例流程可以通过以上参照图2至图4描述的接收端的电子设备200或其示例电路实现300以及与之通信的发送端的电子设备来实现，因此可以获得以上电子设备的配置示例中描述的优点和益处，在此不再展开描述。
注意，这里所涉及的发送端(或发送端的电子设备)只要能够产生并发送相应的第一导频信号、第二导频信号和数据信号并且能够与接收端的电子设备200或其示例电路实现300配合进行通信即可。因此，可以采用各种现有技术的具有发射机功能的电子设备通过适当配置来实现发送端的电子设备。举例而言，当接收端的电子设备为基站等网络侧设备时，发送端的电子设备可以是能够与之通信的用户设备；当接收端的电子设备为用户设备时，发送端的电子设备可以是能够与之通信的基站等网络侧设备，这里不再赘述。
<4.仿真示例>
接下来，将参照图7和图8描述关于本公开实施例的电子设备所进行的解调处理的仿真结果。
在图7和图8的示例中，所接收和处理的复信号为太赫兹QPSK信号，并且采用了此前在“问题的描述”部分中描述的混合失真的模型。更具体地，在图7和图8的示例中，发送端和接收端的(幅度及相位)IQ失衡参数为∈ T＝∈ R＝0.2，φ T＝φ R＝2°，发送端和接收端的相位噪声θ T和θ R例如根据等式(2)由各自的游走项
Figure PCTCN2021128116-appb-000101
和
Figure PCTCN2021128116-appb-000102
确定。这里，设置相位噪声θ T和θ R的游走项方差
Figure PCTCN2021128116-appb-000103
由于相位噪声θ T和θ R的游走项符合高斯分布，因此收发端影响产生的总的相位噪声θ T+θ R也是一个均值为0的随机变量，其分布可以根据等式(2)确定。另外，对于等式(3)中的功率放大器输出的信号模型，采用了三项(K＝3)的无记忆多项式模型，其中，多项式系数b 2k-1(k＝1，2，3)为b 1＝1.0108+j0.0858， b 3＝0.0879-j0.1583，b 5＝-1.0992-j0.8991。对于等式(4)中的信道模型，设置信道衰落因子h＝1，加性高斯白噪声(AWGN)的方差
Figure PCTCN2021128116-appb-000104
在仿真中根据所需的信噪比而适当设置。
首先参照图7，图7是用于说明相位噪声导致的性能下降的示例仿真结果的说明图。图7示出了在图2所示的电子设备中，在没有进行补偿相位估计或相位调整的情况下(即，相当于去除了图2中的相位调整单元210和补偿相位估计单元240的功能)，由收发端影响产生的总的相位噪声导致的对解调性能的影响，其中，横轴表示收发端影响产生的总的相位噪声θ T+θ R，纵轴表示在不同的信噪比(E S/N 0)下的误码率(BER)。如前所述，由于相位噪声的游走项符合高斯分布，因此根据等式(2)确定的θ T+θ R也是一个均值为0的随机变量，在图7所示的示例中设置了θ T+θ R在(-0.2π，0.2π)的范围内随机变化。图7所示的仿真结果是针对根据以上设置的θ T和θ R的分布而确定的相位噪声θ T+θ R的范围内的每个相位噪声进行1000次通信的仿真而得到的，其中，每次通信中依次接收长度为10 5的第二导频信号和长度为10 6的数据信号。在本示例中，诸如图2所示的本公开实施例的电子设备没有进行补偿相位估计或相位调整，而在每次通信中仅利用第二导频信号通过训练获得了解调神经网络，并基于该解调神经网络对数据信号进行了解调处理。从图7可以看出，对于利用这样的电子设备进行的解调处理，在不同的信噪比(E S/N 0)下，误码率(BER)随相位的变化而变化。因此，希望采用适当确定的补偿相位对所接收的复信号进行相位调整，以实现解调性能的最优化。
接下来参照图8，图8是用于说明根据本公开实施例的电子设备所进行的解调处理的性能的示例仿真结果的图。图8分别示出了传统的硬判决(对所接收的模拟信号进行单比特量化，并将量化结果转化为调制符号的输出结果)方法以及利用诸如图2所示的本公开实施例的电子设备对所接收的复信号进行解调处理的误码率性能，其中，横轴表示信噪比(E S/N 0)，纵轴表示相应的误码率(BER)。图8所示的仿真结果是在相位噪声θ T+θ R根据以上设置的分布而根据等式(2)随机变化的情况下，针对每个信噪比进行1000次通信的仿真而得到的，其中，每次通信中，传统的硬判决方法接收长度为10 6的数据信号，本公开实施例的电子设备依次接收长度为10 3的第一导频信号、长度为10 5的第二导频信号和长度为10 6的数据信号。在本示例中，本公开实施例的电子设备在每次通信中先利用第一导频信号确定了补偿相位，再利用第二导频信号通过训练获得了解调神经网络，之后对数据信号进行相位调整以及相应的解调处 理。从图8可以看出，对于涉及不断变动的相位噪声的情况，传统的硬判决方法不能有效实现解调，而本公开实施例的电子设备所执行的处理则通过每次通信中实时的相位补偿或调整以及解调神经网络的准确解调，解决了信道混合失真问题并实现了太赫兹QPSK信号的准确解调，并且在信噪比为10dB时，可使误码率达到10 -2。
<5.方法实施例>
接下来将详细描述根据本公开实施例的电子设备中执行的方法。注意，这些方法实施与以上参照图2至图4描述的装置配置示例相对应，因此，以上装置配置示例的各个细节及益处适当地适用于以下方法实施例。
图9是示出根据本公开实施例的信号处理方法的过程示例的流程图，其例如可以由参照图2至图4描述的电子设备200或其示例电路实现300来实现。
如图9所示，首先，步骤S901中，根据利用导频信号估计的补偿相位，对所接收的复信号进行相位调整。接下来，在步骤S902中，通过低分辨率模数转换处理，基于经相位调整的复信号获得接收比特序列。接着，在步骤S903中，基于利用导频信号训练获得的解调神经网络，对接收比特序列进行解调，以获得所接收的复信号的调制符号。
作为示例，这里的复信号和导频信号可以包括QPSK调制信号。
作为示例，在步骤S901中，可以根据补偿相位，控制本地振荡器对所接收的复信号进行相位旋转，以实现相位调整。
作为示例，在步骤S902中，可以通过低分辨率模数转换处理，分别基于相位调整后的复信号的实部和虚部获得实部接收比特序列和虚部接收比特序列。例如，可以对所述实部和所述虚部进行过采样以得到所接收的复信号的每个传输符号的实部接收比特序列和虚部接收比特序列。
此外，尽管图中未示出，但该方法还可以包括：在步骤S901之前，根据第一导频信号的传输符号以及基于所接收的第一导频信号获得的第一接收比特序列，估计补偿相位。
例如，可选地，该方法可以包括用于估计补偿相位的下述处理：通过低分辨率模数转换处理，基于经相位调整的、所接收的第一导频信号获得第一接收比特序列；根据第一接收比特序列相对于第一导频信号的传输符号的误差，估计补偿相位；以及根据所估计的补偿相位，对所接收的第一导频信号进行相位 调整。
可选地，在该方法所包括的用于估计补偿相位的处理中，可以将使所述误差最小的补偿相位确定为最终的补偿相位。
作为示例，第一导频信号可以包括与QPSK星座图中的至少两个相邻星座点对应的多个传输符号。例如，第一导频信号中的各个传输符号可以对应于QPSK星座图中的两个相邻星座点。例如，第一导频信号中的各个传输符号可以交替地对应于所述两个相邻星座点。
此外，尽管图中未示出，但该方法还可以包括：在步骤S901之前，利用以第二导频信号的调制符号标记的、基于所接收的第二导频信号获得的第二接收比特序列，通过训练获得解调神经网络。
例如，可选地，该方法可以包括用于训练解调神经网络的下述处理：根据利用第一导频信号估计的补偿相位，对所接收的第二导频信号进行相位调整；以及通过低分辨率模数转换处理，基于经相位调整的第二导频信号获得第二接收比特序列。
作为示例，第二导频信号中的传输符号可以对应于QPSK星座图中的四个星座点。例如，第二导频信号中的各个传输符号可以随机地对应于所述四个星座点。
根据本公开的实施例，执行上述方法的主体可以是根据本公开实施例的电子设备200或其示例电路实现300，因此前文中关于电子设备200或其示例电路实现300及其功能单元的实施例的各种方面均适用于此。
<6.应用示例>
本公开内容的技术能够应用于各种产品。
例如，电子设备200可以被实现为任何类型的基站设备，诸如宏eNB和小eNB，还可以被实现为任何类型的gNB(5G系统中的基站)。小eNB可以为覆盖比宏小区小的小区的eNB，诸如微微eNB、微eNB和家庭(毫微微)eNB。代替地，基站可以被实现为任何其他类型的基站，诸如NodeB和基站收发台(BTS)。基站可以包括：被配置为控制无线通信的主体(也称为基站设备)；以及设置在与主体不同的地方的一个或多个远程无线头端(RRH)。
另外，电子设备200还可以被实现为任何类型的TRP。该TRP可以具备发送和接收功能，例如可以从用户设备和基站设备接收信息，也可以向用户设 备和基站设备发送信息。在典型的示例中，TRP可以为用户设备提供服务，并且受基站设备的控制。进一步，TRP可以具备与的基站设备类似的结构，也可以仅具备基站设备中与发送和接收信息相关的结构。
此外，电子设备200也可以为各种用户设备，其可以被实现为移动终端(诸如智能电话、平板个人计算机(PC)、笔记本式PC、便携式游戏终端、便携式/加密狗型移动路由器和数字摄像装置)或者车载终端(诸如汽车导航设备)。用户设备还可以被实现为执行机器对机器(M2M)通信的终端(也称为机器类型通信(MTC)终端)。此外，用户设备可以为安装在上述用户设备中的每个用户设备上的无线通信模块(诸如包括单个晶片的集成电路模块)。
[关于基站的应用示例]
(第一应用示例)
图10是示出可以应用本公开内容的技术的eNB的示意性配置的第一示例的框图。eNB 1800包括一个或多个天线1810以及基站设备1820。基站设备1820和每个天线1810可以经由RF线缆彼此连接。
天线1810中的每一个均包括单个或多个天线元件(诸如包括在多输入多输出(MIMO)天线中的多个天线元件)，并且用于基站设备1820发送和接收无线信号。如图10所示，eNB 1800可以包括多个天线1810。例如，多个天线1810可以与eNB 1800使用的多个频带兼容。虽然图10示出其中eNB 1800包括多个天线1810的示例，但是eNB 1800也可以包括单个天线1810。
基站设备1820包括控制器1821、存储器1822、网络接口1823以及无线通信接口1825。
控制器1821可以为例如CPU或DSP，并且操作基站设备1820的较高层的各种功能。例如，控制器1821根据由无线通信接口1825处理的信号中的数据来生成数据分组，并经由网络接口1823来传递所生成的分组。控制器1821可以对来自多个基带处理器的数据进行捆绑以生成捆绑分组，并传递所生成的捆绑分组。控制器1821可以具有执行如下控制的逻辑功能：该控制诸如为无线资源控制、无线承载控制、移动性管理、接纳控制和调度。该控制可以结合附近的eNB或核心网节点来执行。存储器1822包括RAM和ROM，并且存储由控制器1821执行的程序和各种类型的控制数据(诸如终端列表、传输功率数据以及调度数据)。
网络接口1823为用于将基站设备1820连接至核心网1824的通信接口。控制器1821可以经由网络接口1823而与核心网节点或另外的eNB进行通信。在此情况下，eNB 1800与核心网节点或其他eNB可以通过逻辑接口(诸如S1接口和X2接口)而彼此连接。网络接口1823还可以为有线通信接口或用于无线回程线路的无线通信接口。如果网络接口1823为无线通信接口，则与由无线通信接口1825使用的频带相比，网络接口1823可以使用较高频带用于无线通信。
无线通信接口1825支持任何蜂窝通信方案(诸如长期演进(LTE)和LTE-先进)，并且经由天线1810来提供到位于eNB 1800的小区中的终端的无线连接。无线通信接口1825通常可以包括例如基带(BB)处理器1826和RF电路1827。BB处理器1826可以执行例如编码/解码、调制/解调以及复用/解复用，并且执行层(例如L1、介质访问控制(MAC)、无线链路控制(RLC)和分组数据汇聚协议(PDCP))的各种类型的信号处理。代替控制器1821，BB处理器1826可以具有上述逻辑功能的一部分或全部。BB处理器1826可以为存储通信控制程序的存储器，或者为包括被配置为执行程序的处理器和相关电路的模块。更新程序可以使BB处理器1826的功能改变。该模块可以为插入到基站设备1820的槽中的卡或刀片。可替代地，该模块也可以为安装在卡或刀片上的芯片。同时，RF电路1827可以包括例如混频器、滤波器和放大器，并且经由天线1810来传送和接收无线信号。
如图10所示，无线通信接口1825可以包括多个BB处理器1826。例如，多个BB处理器1826可以与eNB 1800使用的多个频带兼容。如图10所示，无线通信接口1825可以包括多个RF电路1827。例如，多个RF电路1827可以与多个天线元件兼容。虽然图10示出其中无线通信接口1825包括多个BB处理器1826和多个RF电路1827的示例，但是无线通信接口1825也可以包括单个BB处理器1826或单个RF电路1827。
在图10所示的eNB 1800中，此前参照图2描述的电子设备200中的相位调整单元210、低分辨率模数转换单元220和补偿相位估计单元240可以通过无线通信接口1825等实现。这里，尽管未示出，但例如无线通信接口1825中可以设置有高速触发器等，以用作低分辨率模数转换单元220。电子设备200中的解调单元230例如可以通过控制器1821等实现。例如，控制器1821可以通过执行存储器1822中存储的指令而执行解调单元230等的至少一部分功能，这里不再赘述。
(第二应用示例)
图11是示出可以应用本公开内容的技术的eNB的示意性配置的第二示例的框图。eNB 1930包括一个或多个天线1940、基站设备1950和RRH 1960。RRH 1960和每个天线1940可以经由RF线缆而彼此连接。基站设备1950和RRH 1960可以经由诸如光纤线缆的高速线路而彼此连接。
天线1940中的每一个均包括单个或多个天线元件(诸如包括在MIMO天线中的多个天线元件)并且用于RRH 1960发送和接收无线信号。如图11所示，eNB 1930可以包括多个天线1940。例如，多个天线1940可以与eNB 1930使用的多个频带兼容。虽然图11示出其中eNB 1930包括多个天线1940的示例，但是eNB 1930也可以包括单个天线1940。
基站设备1950包括控制器1951、存储器1952、网络接口1953、无线通信接口1955以及连接接口1957。控制器1951、存储器1952和网络接口1953与参照图10描述的控制器1821、存储器1822和网络接口1823相同。网络接口1953为用于将基站设备1950连接至核心网1954的通信接口。
无线通信接口1955支持任何蜂窝通信方案(诸如LTE和LTE-先进)，并且经由RRH 1960和天线1940来提供到位于与RRH 1960对应的扇区中的终端的无线通信。无线通信接口1955通常可以包括例如BB处理器1956。除了BB处理器1956经由连接接口1957连接到RRH 1960的RF电路1964之外，BB处理器1956与参照图10描述的BB处理器1826相同。如图11所示，无线通信接口1955可以包括多个BB处理器1956。例如，多个BB处理器1956可以与eNB 1930使用的多个频带兼容。虽然图11示出其中无线通信接口1955包括多个BB处理器1956的示例，但是无线通信接口1955也可以包括单个BB处理器1956。
连接接口1957为用于将基站设备1950(无线通信接口1955)连接至RRH 1960的接口。连接接口1957还可以为用于将基站设备1950(无线通信接口1955)连接至RRH 1960的上述高速线路中的通信的通信模块。
RRH 1960包括连接接口1961和无线通信接口1963。
连接接口1961为用于将RRH 1960(无线通信接口1963)连接至基站设备1950的接口。连接接口1961还可以为用于上述高速线路中的通信的通信模块。
无线通信接口1963经由天线1940来传送和接收无线信号。无线通信接口1963通常可以包括例如RF电路1964。RF电路1964可以包括例如混频器、滤波器和放大器，并且经由天线1940来传送和接收无线信号。如图11所示，无线通信接口1963可以包括多个RF电路1964。例如，多个RF电路1964可以支持多个天线元件。虽然图11示出其中无线通信接口1963包括多个RF电路1964的示例，但是无线通信接口1963也可以包括单个RF电路1964。
在图11所示的eNB 1930中，此前参照图2描述的电子设备200中的相位调整单元210、低分辨率模数转换单元220和补偿相位估计单元240可以通过无线通信接口1963实现。这里，尽管未示出，但例如无线通信接口1963中可以设置有高速触发器等，以用作低分辨率模数转换单元220。电子设备200中的解调单元230可以通过控制器1951等实现。例如，控制器1951可以通过执行存储器1952中存储的指令而执行解调单元230等的至少一部分功能，这里不再赘述。
[关于用户设备的应用示例]
(第一应用示例)
图12是示出可以应用本公开内容的技术的智能电话2000的示意性配置的示例的框图。智能电话2000包括处理器2001、存储器2002、存储装置2003、外部连接接口2004、摄像装置2006、传感器2007、麦克风2008、输入装置2009、显示装置2010、扬声器2011、无线通信接口2012、一个或多个天线开关2015、一个或多个天线2016、总线2017、电池2018以及辅助控制器2019。
处理器2001可以为例如CPU或片上系统(SoC)，并且控制智能电话2000的应用层和另外层的功能。存储器2002包括RAM和ROM，并且存储数据和由处理器2001执行的程序。存储装置2003可以包括存储介质，诸如半导体存储器和硬盘。外部连接接口2004为用于将外部装置(诸如存储卡和通用串行总线(USB)装置)连接至智能电话2000的接口。
摄像装置2006包括图像传感器(诸如电荷耦合器件(CCD)和互补金属氧化物半导体(CMOS))，并且生成捕获图像。传感器2007可以包括一组传感器，诸如测量传感器、陀螺仪传感器、地磁传感器和加速度传感器。麦克风2008将输入到智能电话2000的声音转换为音频信号。输入装置2009包括例如被配置为检测显示装置2010的屏幕上的触摸的触摸传感器、小键盘、键盘、按钮或开关，并且接收从用户输入的操作或信息。显示装置2010包括屏幕(诸 如液晶显示器(LCD)和有机发光二极管(OLED)显示器)，并且显示智能电话2000的输出图像。扬声器2011将从智能电话2000输出的音频信号转换为声音。
无线通信接口2012支持任何蜂窝通信方案(诸如LTE和LTE-先进)，并且执行无线通信。无线通信接口2012通常可以包括例如BB处理器2013和RF电路2014。BB处理器2013可以执行例如编码/解码、调制/解调以及复用/解复用，并且执行用于无线通信的各种类型的信号处理。同时，RF电路2014可以包括例如混频器、滤波器和放大器，并且经由天线2016来传送和接收无线信号。无线通信接口2012可以为其上集成有BB处理器2013和RF电路2014的一个芯片模块。如图12所示，无线通信接口2012可以包括多个BB处理器2013和多个RF电路2014。虽然图12示出其中无线通信接口2012包括多个BB处理器2013和多个RF电路2014的示例，但是无线通信接口2012也可以包括单个BB处理器2013或单个RF电路2014。
此外，除了蜂窝通信方案之外，无线通信接口2012可以支持另外类型的无线通信方案，诸如短距离无线通信方案、近场通信方案和无线局域网(LAN)方案。在此情况下，无线通信接口2012可以包括针对每种无线通信方案的BB处理器2013和RF电路2014。
天线开关2015中的每一个在包括在无线通信接口2012中的多个电路(例如用于不同的无线通信方案的电路)之间切换天线916的连接目的地。
天线2016中的每一个均包括单个或多个天线元件(诸如包括在MIMO天线中的多个天线元件)，并且用于无线通信接口2012传送和接收无线信号。如图12所示，智能电话2000可以包括多个天线2016。虽然图12示出其中智能电话2000包括多个天线2016的示例，但是智能电话2000也可以包括单个天线2016。
此外，智能电话2000可以包括针对每种无线通信方案的天线2016。在此情况下，天线开关2015可以从智能电话2000的配置中省略。
总线2017将处理器2001、存储器2002、存储装置2003、外部连接接口2004、摄像装置2006、传感器2007、麦克风2008、输入装置2009、显示装置2010、扬声器2011、无线通信接口2012以及辅助控制器2019彼此连接。电池2018经由馈线向图12所示的智能电话2000的各个块提供电力，馈线在图中被部分地示为虚线。辅助控制器2019例如在睡眠模式下操作智能电话2000 的最小必需功能。
在图12所示的智能电话2000中，此前参照图2描述的电子设备200中的相位调整单元210、低分辨率模数转换单元220和补偿相位估计单元240可以通过无线通信接口2012实现。这里，尽管未示出，但例如无线通信接口2012中可以设置有高速触发器等，以用作低分辨率模数转换单元220。电子设备200中的解调单元230可以由处理器2001或辅助控制器2019实现。例如，处理器2001或辅助控制器2019可以通过执行存储器2002或存储装置2003中存储的指令而执行解调单元230等的至少一部分功能，这里不再赘述。
(第二应用示例)
图13是示出可以应用本公开内容的技术的汽车导航设备2120的示意性配置的示例的框图。汽车导航设备2120包括处理器2121、存储器2122、全球定位系统(GPS)模块2124、传感器2125、数据接口2126、内容播放器2127、存储介质接口2128、输入装置2129、显示装置2130、扬声器2131、无线通信接口2133、一个或多个天线开关2136、一个或多个天线2137以及电池2138。
处理器2121可以为例如CPU或SoC，并且控制汽车导航设备2120的导航功能和另外的功能。存储器2122包括RAM和ROM，并且存储数据和由处理器2121执行的程序。
GPS模块2124使用从GPS卫星接收的GPS信号来测量汽车导航设备2120的位置(诸如纬度、经度和高度)。传感器2125可以包括一组传感器，诸如陀螺仪传感器、地磁传感器和空气压力传感器。数据接口2126经由未示出的终端而连接到例如车载网络2141，并且获取由车辆生成的数据(诸如车速数据)。
内容播放器2127再现存储在存储介质(诸如CD和DVD)中的内容，该存储介质被插入到存储介质接口2128中。输入装置2129包括例如被配置为检测显示装置2130的屏幕上的触摸的触摸传感器、按钮或开关，并且接收从用户输入的操作或信息。显示装置2130包括诸如LCD或OLED显示器的屏幕，并且显示导航功能的图像或再现的内容。扬声器2131输出导航功能的声音或再现的内容。
无线通信接口2133支持任何蜂窝通信方案(诸如LTE和LTE-先进)，并且执行无线通信。无线通信接口2133通常可以包括例如BB处理器2134和RF电路2135。BB处理器2134可以执行例如编码/解码、调制/解调以及复用/解复用，并且执行用于无线通信的各种类型的信号处理。同时，RF电路2135 可以包括例如混频器、滤波器和放大器，并且经由天线2137来传送和接收无线信号。无线通信接口2133还可以为其上集成有BB处理器2134和RF电路2135的一个芯片模块。如图13所示，无线通信接口2133可以包括多个BB处理器2134和多个RF电路2135。虽然图13示出其中无线通信接口2133包括多个BB处理器2134和多个RF电路2135的示例，但是无线通信接口2133也可以包括单个BB处理器2134或单个RF电路2135。
此外，除了蜂窝通信方案之外，无线通信接口2133可以支持另外类型的无线通信方案，诸如短距离无线通信方案、近场通信方案和无线LAN方案。在此情况下，针对每种无线通信方案，无线通信接口2133可以包括BB处理器2134和RF电路2135。
天线开关2136中的每一个在包括在无线通信接口2133中的多个电路(诸如用于不同的无线通信方案的电路)之间切换天线2137的连接目的地。
天线2137中的每一个均包括单个或多个天线元件(诸如包括在MIMO天线中的多个天线元件)，并且用于无线通信接口2133传送和接收无线信号。如图13所示，汽车导航设备2120可以包括多个天线2137。虽然图13示出其中汽车导航设备2120包括多个天线2137的示例，但是汽车导航设备2120也可以包括单个天线2137。
此外，汽车导航设备2120可以包括针对每种无线通信方案的天线2137。在此情况下，天线开关2136可以从汽车导航设备2120的配置中省略。
电池2138经由馈线向图13所示的汽车导航设备2120的各个块提供电力，馈线在图中被部分地示为虚线。电池2138累积从车辆提供的电力。
在图13示出的汽车导航设备2120中，此前参照图2描述的电子设备200中的相位调整单元210、低分辨率模数转换单元220和补偿相位估计单元240可以通过无线通信接口2133实现。这里，尽管未示出，但例如无线通信接口2133中可以设置有高速触发器等，以用作低分辨率模数转换单元220。电子设备200中的解调单元230可以由处理器2121实现。例如，处理器2121可以通过执行存储器2122中存储的指令而执行解调单元230的至少一部分功能，这里不再赘述。
本公开内容的技术也可以被实现为包括汽车导航设备2120、车载网络2141以及车辆模块2142中的一个或多个块的车载系统(或车辆)2140。车辆模块2142生成车辆数据(诸如车速、发动机速度和故障信息)，并且将所生成 的数据输出至车载网络2141。
以上参照附图描述了本公开的优选实施例，但是本公开当然不限于以上示例。本领域技术人员可在所附权利要求的范围内得到各种变更和修改，并且应理解这些变更和修改自然将落入本公开的技术范围内。
例如，附图所示的功能框图中以虚线框示出的单元均表示该功能单元在相应装置中是可选的，并且各个可选的功能单元可以以适当的方式进行组合以实现所需功能。
例如，在以上实施例中包括在一个单元中的多个功能可以由分开的装置来实现。替选地，在以上实施例中由多个单元实现的多个功能可分别由分开的装置来实现。另外，以上功能之一可由多个单元来实现。无需说，这样的配置包括在本公开的技术范围内。
在该说明书中，流程图中所描述的步骤不仅包括以所述顺序按时间序列执行的处理，而且包括并行地或单独地而不是必须按时间序列执行的处理。此外，甚至在按时间序列处理的步骤中，无需说，也可以适当地改变该顺序。
以上虽然结合附图详细描述了本公开的实施例，但是应当明白，上面所描述的实施方式只是用于说明本公开，而并不构成对本公开的限制。对于本领域的技术人员来说，可以对上述实施方式作出各种修改和变更而没有背离本公开的实质和范围。因此，本公开的范围仅由所附的权利要求及其等效含义来限定。

Claims (20)

  1. 一种电子设备，包括：
    处理电路，被配置为：
    根据利用导频信号估计的补偿相位，对所接收的复信号进行相位调整；
    通过低分辨率模数转换处理，基于经相位调整的复信号获得接收比特序列；以及
    基于利用导频信号训练获得的解调神经网络，对接收比特序列进行解调，以获得所接收的复信号的调制符号。
  2. 如权利要求1所述的电子设备，其中，复信号和导频信号包括QPSK调制信号。
  3. 如权利要求2所述的电子设备，其中，所述处理电路还被配置为：
    根据第一导频信号的传输符号以及基于所接收的第一导频信号获得的第一接收比特序列，估计补偿相位。
  4. 如权利要求3所述的电子设备，其中，所述处理电路还被配置为：
    通过低分辨率模数转换处理，基于经相位调整的、所接收的第一导频信号获得第一接收比特序列；
    根据第一接收比特序列相对于第一导频信号的传输符号的误差，估计补偿相位；以及
    根据所估计的补偿相位，对所接收的第一导频信号进行相位调整。
  5. 如权利要求4所述的电子设备，其中，所述处理电路还被配置为将使所述误差最小的补偿相位确定为最终的补偿相位。
  6. 如权利要求3所述的电子设备，其中，第一导频信号包括与QPSK星座图中的至少两个相邻星座点对应的多个传输符号。
  7. 如权利要求3所述的电子设备，其中，第一导频信号中的各个传输符号对应于QPSK星座图中的两个相邻星座点。
  8. 如权利要求7所述的电子设备，其中，第一导频信号中的各个传输符号交替地对应于所述两个相邻星座点。
  9. 如权利要求1所述的电子设备，其中，所述处理电路还被配置为：根据补偿相位，控制本地振荡器对所接收的复信号进行相位旋转，以实现相位调整。
  10. 如权利要求1所述的电子设备，其中，所述处理电路还被配置为利用以第二导频信号的调制符号标记的、基于所接收的第二导频信号获得的第二接收比特序列，通过训练获得解调神经网络。
  11. 如权利要求10所述的电子设备，其中，所述处理电路还被配置为：
    根据利用第一导频信号估计的补偿相位，对所接收的第二导频信号进行相位调整；以及
    通过低分辨率模数转换处理，基于经相位调整的第二导频信号获得第二接收比特序列。
  12. 如权利要求10所述的电子设备，其中，第二导频信号中的传输符号对应于QPSK星座图中的四个星座点。
  13. 如权利要求12所述的电子设备，其中，第二导频信号中的各个传输符号随机地对应于所述四个星座点。
  14. 如权利要求1至13中任一项所述的电子设备，其中，所述处理电路还被配置为通过低分辨率模数转换处理，分别基于相位调整后的复信号的实部和虚部获得实部接收比特序列和虚部接收比特序列。
  15. 如权利要求14所述的电子设备，其中，所述处理电路还被配置为对所述实部和所述虚部进行过采样以得到所接收的复信号的每个传输符号的实部接收比特序列和虚部接收比特序列。
  16. 一种信号处理方法，包括：
    根据利用导频信号估计的补偿相位，对所接收的复信号进行相位调整；
    通过低分辨率模数转换处理，基于经相位调整的复信号获得接收比特序列；以及
    基于利用导频信号训练获得的解调神经网络，对接收比特序列进行解调，以获得所接收的复信号的调制符号。
  17. 如权利要求16所述的信号处理方法，其中，复信号和导频信号包括QPSK调制信号。
  18. 如权利要求16所述的信号处理方法，还包括：
    根据第一导频信号的传输符号以及基于所接收的第一导频信号获得的第一接收比特序列，估计补偿相位。
  19. 如权利要求16所述的信号处理方法，还包括：
    利用以第二导频信号的调制符号标记的、基于所接收的第二导频信号获得的第二接收比特序列，通过训练获得解调神经网络。
  20. 一种存储有程序的非暂态计算机可读存储介质，所述程序当由处理器执行时，使得所述处理器执行根据权利要求17至19中任一项所述的方法。
PCT/CN2021/128116 2020-11-09 2021-11-02 电子设备、信号处理方法以及计算机可读存储介质 WO2022095843A1 (zh)

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2024007299A1 (en) * 2022-07-08 2024-01-11 Huawei Technologies Co., Ltd. A signal processing device and method for a non-stationary dynamic environment

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN115102815B (zh) * 2022-08-25 2022-11-04 北京智芯微电子科技有限公司 射频信号解调方法、装置、电子设备、存储介质及芯片

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN109194595A (zh) * 2018-09-26 2019-01-11 东南大学 一种基于神经网络的信道环境自适应ofdm接收方法
CN109743268A (zh) * 2018-12-06 2019-05-10 东南大学 基于深度神经网络的毫米波信道估计和压缩方法
US20190207683A1 (en) * 2016-08-17 2019-07-04 Nec Corporation Phase noise compensation apparatus, demodulation apparatus, communication apparatus, communication system, and phase noise compensation method
WO2020092391A1 (en) * 2018-10-29 2020-05-07 Board Of Regents, The University Of Texas System Low resolution ofdm receivers via deep learning

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20190207683A1 (en) * 2016-08-17 2019-07-04 Nec Corporation Phase noise compensation apparatus, demodulation apparatus, communication apparatus, communication system, and phase noise compensation method
CN109194595A (zh) * 2018-09-26 2019-01-11 东南大学 一种基于神经网络的信道环境自适应ofdm接收方法
WO2020092391A1 (en) * 2018-10-29 2020-05-07 Board Of Regents, The University Of Texas System Low resolution ofdm receivers via deep learning
CN109743268A (zh) * 2018-12-06 2019-05-10 东南大学 基于深度神经网络的毫米波信道估计和压缩方法

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2024007299A1 (en) * 2022-07-08 2024-01-11 Huawei Technologies Co., Ltd. A signal processing device and method for a non-stationary dynamic environment

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