WO2021227231A1 - 一种适用于不同输入电网的dcdc架构及其控制方法 - Google Patents

一种适用于不同输入电网的dcdc架构及其控制方法 Download PDF

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Publication number
WO2021227231A1
WO2021227231A1 PCT/CN2020/101136 CN2020101136W WO2021227231A1 WO 2021227231 A1 WO2021227231 A1 WO 2021227231A1 CN 2020101136 W CN2020101136 W CN 2020101136W WO 2021227231 A1 WO2021227231 A1 WO 2021227231A1
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Prior art keywords
transformer
conversion module
dcdc
primary
different input
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PCT/CN2020/101136
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English (en)
French (fr)
Inventor
刘钧
冯颖盈
姚顺
徐金柱
张远昭
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深圳威迈斯新能源股份有限公司
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Publication of WO2021227231A1 publication Critical patent/WO2021227231A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/44—Conversion of dc power input into dc power output with intermediate conversion into ac by combination of static with dynamic converters; by combination of dynamo-electric with other dynamic or static converters
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L53/00—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles
    • B60L53/20—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles characterised by converters located in the vehicle
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J7/00—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
    • H02J7/02—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries for charging batteries from ac mains by converters
    • H02J7/04—Regulation of charging current or voltage
    • H02J7/06—Regulation of charging current or voltage using discharge tubes or semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/70—Energy storage systems for electromobility, e.g. batteries
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/7072—Electromobility specific charging systems or methods for batteries, ultracapacitors, supercapacitors or double-layer capacitors
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/72—Electric energy management in electromobility
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/80—Technologies aiming to reduce greenhouse gasses emissions common to all road transportation technologies
    • Y02T10/92—Energy efficient charging or discharging systems for batteries, ultracapacitors, supercapacitors or double-layer capacitors specially adapted for vehicles
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T90/00—Enabling technologies or technologies with a potential or indirect contribution to GHG emissions mitigation
    • Y02T90/10—Technologies relating to charging of electric vehicles
    • Y02T90/14—Plug-in electric vehicles

Definitions

  • the invention belongs to the technical field of power supplies, and specifically relates to a DCDC architecture suitable for different input power grids and a control method thereof.
  • the current OBC design composition is shown in Figure 2. It consists of two stages of PFC and DCDC in series.
  • the high-power output is based on the scenario where the input AC voltage is three-phase, and the vehicle-mounted OBC often needs to be compatible with single-phase and three-phase inputs.
  • the PFC output voltage is usually 800V, that is, the DCDC input voltage is 800V; in the single-phase input, because the DCDC transformer turns ratio is fixed, it is necessary to increase the PFC input voltage to 800V during single-phase input. It will increase the loss of PFC, resulting in low efficiency.
  • the present invention proposes a DCDC architecture and its control method suitable for different input power grids.
  • the technical scheme adopted by the present invention is to design a DCDC architecture suitable for different input power grids, including a primary side conversion module, a first transformer T1, a secondary side conversion module, a controller, a second transformer T2, and a secondary side rectifier module connected in sequence , wherein the primary winding W5 of the second transformer T2 is connected in series with the primary winding W1 of the first transformer T1 and then connected to the output end of the primary conversion module, and the secondary winding W6 of the second transformer T2 is connected to the secondary The input terminal of the rectifier module, the output terminal of the secondary side rectifier module and the output terminal of the secondary side conversion module are connected in parallel.
  • the primary side or the secondary side of the second transformer T2 is provided with a switch K; the controller converts the module according to the bus voltage of the primary side The switch K is controlled to put the second transformer T2 and the secondary side rectifier module into operation or out of operation.
  • the switch K may be connected in parallel to both ends of the primary winding W5 of the second transformer T2.
  • the switch K can also be connected in parallel to both ends of the secondary winding W6 of the second transformer T2.
  • the DCDC architecture includes a charging mode and an inverter mode; in the charging mode, when the bus voltage is higher than the threshold M, the controller controls the switch K to turn off; in the charging mode, when the bus voltage is not higher than the threshold M, control The controller controls the switch K to close; in the inverter mode, the controller controls the switch K to close.
  • the threshold value M is 600 volts.
  • the ratio of the primary winding W1 to the secondary winding W2 of the first transformer T1 is equal to the ratio of the primary winding W5 to the secondary winding W6 of the second transformer T2; the primary winding W1 of the first transformer T1 and the primary winding of the second transformer T2 are equal W5 has the same number of turns and the same wire diameter; the first transformer T1 secondary winding W2 and the second transformer T2 secondary winding W6 have the same number of turns and wire diameters.
  • the power switch in the secondary side conversion module adopts an active device, and the power switch in the secondary side rectifier module adopts a passive device.
  • the secondary side rectifier module adopts a bridge rectifier module, which includes a first diode D1, a second diode D2, a third diode D2, and a fourth diode D4.
  • the primary-side conversion module adopts a full-bridge structure, including a first power switch Q1, a second power switch Q2, a third power switch Q3, and a fourth power switch Q4; the secondary-side conversion module adopts a full-bridge structure, including a fifth power switch.
  • the secondary winding W2 of the first transformer T1 is connected to the secondary conversion module through a DC blocking capacitor C2; the secondary winding W6 of the second transformer T2 is directly connected to the secondary rectification module.
  • a resonant capacitor C1 is connected in series between the primary winding of the first transformer T1 and the primary winding of the second transformer T2.
  • the primary winding of the first transformer T1 is connected in series with the first resonant inductor Lr1.
  • the primary winding of the second transformer T2 is connected in series with a second resonant inductor Lr2.
  • the first transformer T1 and the second transformer T2 are integrated on the same magnetic core.
  • the present invention also designs a control method of DCDC architecture suitable for different input power grids, and the DCDC architecture adopts the above-mentioned DCDC architecture suitable for different input power grids. During control, only the primary side conversion module and the secondary side conversion module are controlled, and the secondary side rectifier module is not controlled, which simplifies the control method and reduces the cost of parts.
  • the specific control method is: there is a timing difference ⁇ between the control signal for driving the power switch in the primary side conversion module and the control signal for driving the power switch in the secondary side conversion module, and the controller controls the lead or lag of the timing difference ⁇ ;
  • the timing difference ⁇ leading can increase the gain of the DCDC architecture and increase the output power of the secondary side conversion module;
  • the timing difference ⁇ lag can reduce the gain of the DCDC architecture and reduce the output power of the secondary side conversion module.
  • the controller includes a collector that collects the input voltage (Vin) of the primary side conversion module, and a collector that collects the common output voltage (VoHV) and output current (IoHV) of the secondary side conversion module and the secondary side rectifier module.
  • Vin input voltage
  • VoHV common output voltage
  • IoHV output current
  • the controller performs sampling and calibration on the output current (IoHV) and output voltage (VoHV) respectively, and obtains the output power through power calculation (Power Calculation); the output current (IoHV) after sampling and calibration is the same as the output current
  • the reference value (IrefHV) is subjected to difference calculation, the difference between the two is subjected to loop compensation, and the obtained compensation value and the pre-set voltage loop preset value (VsetHV) are calculated to be smaller, and the smaller value is taken as the voltage Loop reference value (VrefHV); use the sampled and calibrated output voltage (VoHV) to perform the difference calculation with the voltage loop reference value (VrefHV), perform loop compensation on the difference between the two, and use the obtained compensation value to compare with the
  • the output power calculation obtains the timing difference ⁇ , and then the power switches in the primary side conversion module and the secondary side conversion module are driven by a PWM operation (PWM Generator).
  • the controller includes a collector that collects the input voltage (Vin) of the primary side conversion module, and collects the common output voltage (VoHV) and output current (IoHV) of the secondary side conversion module and the secondary side rectifier module.
  • the controller samples and calibrates the input voltage (Vin), and then performs a difference operation with the input current reference value (VrefVin), and performs loop compensation on the difference between the two to obtain the timing difference ⁇ , and then
  • the power switches in the primary side conversion module and the secondary side conversion module are driven by PWM operation (PWM Generator).
  • the invention solves the problem of multi-path parallel current sharing of high-power vehicle-mounted OBC rear-stage DCDC and compatibility with single-phase and three-phase input voltages without increasing the cost; it has the advantages of small number of components, simplicity, and easy implementation.
  • Fig. 1 is a principle block diagram of parallel operation of multiple DCDCs in the prior art
  • FIG. 2 is a functional block diagram of the charger
  • Figure 3 is a circuit diagram of a resonant inductor connected in series according to the present invention.
  • Figure 4 is a circuit diagram of the present invention in which two resonant inductors are connected in series;
  • Figure 5 is a circuit diagram of the combined first and second transformers of the present invention.
  • Figure 6 is a control waveform diagram of the primary side conversion module
  • Figure 7 is a control waveform diagram of the secondary side conversion module
  • Figure 8 is a comparison diagram of the control wave timing difference between the primary side conversion module and the secondary side conversion module
  • Figure 9 is a comparison diagram of the control waveform of the secondary side conversion module and the waveform of the midpoint voltage of the bridge arm;
  • Figure 10 is a comparison diagram of the respective output currents and total output current waveforms of the first and second conversion modules on the secondary side;
  • 11 is a comparison diagram of the output current of the primary side conversion module, the primary side conversion module bridge arm midpoint voltage, and the secondary side first and second conversion module bridge arms midpoint voltage waveform comparison diagram;
  • Figure 12 is a schematic diagram of the timing difference waveform of the control wave between the primary side conversion module and the secondary side conversion module;
  • Figure 13 is the second conversion module bridge arm midpoint voltage V_EF, the first conversion module bridge arm midpoint voltage V_CD, primary conversion module output voltage V_AB, primary conversion module output current Ip, and DCDC architecture output current when connected to a single-phase power grid IoHV waveform comparison chart;
  • Figure 14 is a block diagram of the controller control principle in charging mode
  • Figure 15 is a block diagram of the controller control principle in inverter mode.
  • the present invention discloses a DCDC architecture suitable for different input power grids, including a primary side conversion module, a first transformer T1, a secondary side conversion module, a controller, a second transformer T2, and a secondary side rectifier module connected in sequence, wherein the first The primary winding W5 of the second transformer T2 is connected in series with the primary winding W1 of the first transformer T1 and then connected to the output end of the primary conversion module, and the secondary winding W6 of the second transformer T2 is connected to the input end of the secondary rectification module ,
  • the output terminal of the secondary side rectifier module is connected in parallel with the output terminal of the secondary side conversion module, the primary side or the secondary side of the second transformer T2 is provided with a switch K; the controller controls the switch K according to the bus voltage of the primary side conversion module
  • the second transformer T2 and the secondary side rectifier module are put into operation or out of operation.
  • the front end of the DCDC architecture is connected to an AC input terminal, an EMI filter, and a PFC circuit in order to provide DC power for the DCDC circuit.
  • the AC input terminal is connected to different external input power grids, which can be a three-phase power grid or a single-phase power grid.
  • This application is applied to a vehicle-mounted charger, and the output end of the DCDC circuit is mainly connected to the high-voltage battery in the vehicle.
  • the switch K is connected in parallel to both ends of the primary winding W5 of the second transformer T2. In some other embodiments, the switch K is connected in parallel with both ends of the secondary winding W6 of the second transformer T2 (the circuit diagram is not shown).
  • the DCDC architecture includes a charging mode and an inverter mode; in the charging mode, when the bus voltage is higher than the threshold M, the controller controls the switch K to turn off; in the charging mode, when the bus voltage is not higher than the threshold M, control The controller controls the switch K to close; in the inverter mode, the controller controls the switch K to close.
  • the switch K adopts one of a two-way switch or a relay.
  • the threshold M is 600 volts.
  • the bus voltage is higher than the threshold value M, which means that the charger is connected to a three-phase power grid.
  • the bus voltage is not higher than the threshold value M, which means that the charger is connected to a single-phase power grid.
  • the ratio of the primary winding W1 to the secondary winding W2 of the first transformer T1 is equal to the ratio of the primary winding W5 to the secondary winding W6 of the second transformer T2; the primary winding W1 of the first transformer T1
  • the number of turns and the wire diameter of the primary winding W5 of the second transformer T2 are equal; the number of turns and the wire diameter of the secondary winding W2 of the first transformer T1 and the secondary winding W6 of the second transformer T2 are the same.
  • the power switch in the secondary side conversion module adopts an active device, and the power switch in the secondary side rectifier module adopts a passive device.
  • the secondary side rectifier module adopts a bridge rectifier module, which includes a first diode D1, a second diode D2, a third diode D2, and a fourth diode D4.
  • the primary side conversion module adopts a full-bridge structure, including a first power switch Q1, a second power switch Q2, a third power switch Q3, and a fourth power switch Q4;
  • the side conversion module adopts a full bridge structure, and includes a fifth power switch Q5, a sixth power switch Q6, a seventh power switch Q7, and an eighth power switch Q8.
  • the control signal and input voltage of the secondary side conversion module are shown in Figure 9.
  • the first power switch Q1, the second power switch Q2, the third power switch Q3, the fourth power switch Q4, the fifth power switch Q5, the sixth power switch Q6, the seventh power switch Q7, and the eighth power switch Q8 are adopted :
  • the secondary winding W2 of the first transformer T1 is connected to the secondary conversion module through a DC blocking capacitor C2; the secondary winding W6 of the second transformer T2 is directly connected to the secondary rectifier Module. As shown in Figures 3, 4, and 5.
  • a resonant capacitor C1 is connected in series between the primary winding of the first transformer T1 and the primary winding of the second transformer T2. As shown in Figures 3, 4, and 5.
  • the primary winding of the first transformer T1 is connected in series with the first resonant inductor Lr1. That is, only one vibrating inductor is connected in series on the entire primary side.
  • the primary winding of the second transformer T2 is connected in series with the second resonant inductor Lr2. That is, two vibrating inductors are connected in series on the entire primary side.
  • the first transformer T1 and the second transformer T2 are integrated on the same magnetic core. That is, the first transformer T1 and the second transformer T2 can be installed separately or combined.
  • the resonant inductor setting may be one resonant inductor, or two separate resonant inductors, or it may be integrated in the same magnetic core.
  • the resonant inductor can be an independent element or the leakage inductance of the transformer.
  • FIG. 3 taking the use of the present invention in a charger as an example.
  • the charging mode is connected to the three-phase grid:
  • switch K is disconnected, and Q1 and Q3 in the primary side conversion module form the first leg of the primary side, and the midpoint of the bridge arm is A;
  • Q2 and Q4 are composed
  • the resonant inductor Lr1, the transformer T1 winding W1, the resonant capacitor C1, the transformer T2 winding W5, and the resonant capacitor Lr2 are connected in series, and one end is connected to the midpoint A of the first leg of the primary side , The other end is connected to the middle point B of the second bridge arm of the primary side, forming a structure in which the primary sides of the transformer T1 and T2 are connected in series.
  • Q5 and Q7 form the first bridge arm of the secondary side, with the midpoint C of the bridge arm;
  • Q6 and Q8 form the second bridge arm of the secondary side, with the midpoint D of the bridge arm;
  • the transformer T1 winding W2 is connected in series with the DC blocking capacitor C2, One end is connected to the midpoint C of the first bridge arm of the secondary side, and the other end is connected to the midpoint D of the second bridge arm of the secondary side.
  • the output capacitor C4 is connected in parallel to form the secondary side conversion module output HV1;
  • the third bridge arm on the side, the midpoint E of the bridge arm; D2 and D4 form the fourth bridge arm of the secondary side, the midpoint F of the bridge arm; one end of the transformer T2 winding W6 is connected to the midpoint E of the third bridge arm of the secondary side, and the other end is connected to At the midpoint F of the fourth bridge arm of the secondary side, the output capacitor C5 is connected in parallel to form the secondary side rectifier module to output HV2.
  • the positive output terminal of HV1 and the positive output terminal of HV2 are connected together.
  • HV2 is connected in parallel to form a high-voltage HV output.
  • transformer T1 winding W1 W2 transformer T2 winding W5: W6 turns ratio and number of turns, winding wire diameter are the same.
  • Control method In the topology of Figure 3, the controller realizes the control of HV voltage and current by driving the power switches Q1—Q4 in the primary side conversion module and the power switches Q5—Q8 in the secondary side conversion module.
  • the primary side Conversion module Q1 and Q4 drive the same, both are 50% duty cycle; Q2 and Q3 drive the same, both are 50% duty cycle, Q1, Q4 and Q2, Q3 drive completely opposite, as shown in Figure 6;
  • secondary side Conversion module Q5 and Q8 drive the same, both are 50% duty cycle;
  • Q6 and Q7 drive the same, both are 50% duty cycle, Q5, Q8 and Q6, Q7 drives are completely opposite, as shown in Figure 7;
  • the 50% duty cycle is specifically implemented in order to prevent the upper and lower switching tubes of a pair of bridge arms from being turned on at the same time, and a dead time needs to be subtracted.
  • the 50% duty cycle mentioned here is collectively referred to as Contains dead time.
  • the currents coupled to the secondary side of the transformers T1 and T2 are also the same, namely The output current Io1 of the secondary side conversion module and the output current Io2 of the secondary side rectifier module are equal.
  • the total output current is controlled by controlling the drive output of the primary side and the secondary side conversion module, and the output current of the secondary side rectifier module automatically and the secondary side conversion module The output currents are equal. Since the number of turns of the primary and secondary windings of the transformers T1 and T2 are the same, the output of the secondary conversion module and the output of the secondary rectification module are automatically balanced, and no additional current sharing is required.
  • the secondary side rectifier module is a diode rectifier, and the diode conduction is determined by the zero-crossing of the primary side current.
  • the side rectifier module can omit the DC blocking capacitor, that is, the C2 capacitor is not needed in the secondary side rectifier module.
  • the primary winding W5 of transformer T2 is short-circuited, and the input voltage is applied to the resonant cavity and the primary winding W1 of transformer T1.
  • the cavity parameters and resonance points are the same.
  • the resonance points of single-phase and three-phase input are the following formulas 1 shown;
  • the resonance point remains unchanged.
  • the gain of a single transformer T1 is still the same, compared to three-phase Input, the single-phase power is reduced by half, and the primary current is the same.
  • the turns ratio of the transformer T1 and the design of the winding wire diameter are the same, which will not cause over-design due to single-phase and three-phase compatibility.
  • Inverter mode (inverter mode is not divided into three-phase or single-phase grid):
  • the switch K is closed, that is, the winding W5 of the transformer T2 is short-circuited.
  • HV is the input terminal of energy
  • the output side is Vin, which is opposite to the charging mode.
  • the primary side is the energy output side
  • the secondary HV is the energy input side.
  • Control method The controller realizes the control of energy output by driving the power switches Q1—Q4 of the primary side conversion module and the power switches Q5—Q8 of the secondary side conversion module.
  • the primary-side conversion modules Q1 and Q4 are driven in the same way and both have a 50% duty cycle;
  • Q2 and Q3 are driven in the same way and both have a 50% duty cycle.
  • the drives of Q1, Q4 and Q2, Q3 are completely opposite, as shown in Figure 6. Show.
  • Q5 and Q8 drive the same, both are 50% duty cycle;
  • Q6 and Q7 drive the same, both are 50% duty cycle, Q5, Q8 and Q6, Q7 drive completely opposite, as shown in Figure 7.
  • the 50% duty cycle mentioned above is specifically implemented in order to prevent the upper and lower switch tubes of a pair of bridge arms from being turned on at the same time, and a dead time needs to be subtracted.
  • the 50% duty cycle It is collectively referred to as including dead time.
  • FIG. 10 shows the simulation results of Table 2, and the waveform comparison of the respective output currents and total output currents of the first and second conversion modules on the secondary side. It can be seen from Table 2 that there is almost no deviation between the output of the secondary conversion module and the output of the secondary rectifier module, which proves the feasibility of this control method.
  • Figure 11 shows the output current of the primary side conversion module, the bridge arm midpoint voltage of the primary side conversion module, and the comparison of the bridge arm midpoint voltage waveforms of the first and second secondary side conversion modules (i.e., the secondary side conversion module and the secondary side rectifier module) picture.
  • the present invention also discloses a control method of DCDC architecture suitable for different input power grids.
  • the DCDC architecture adopts the above-mentioned DCDC architecture suitable for different input power grids; in order to adjust the output power of the DCDC architecture, drive the power in the primary conversion module
  • the controller controls the lead or lag of the timing difference ⁇ ; the timing difference ⁇ leading can increase the gain of the DCDC architecture and increase the output power of the secondary side conversion module; the timing difference ⁇ lag can reduce the gain of the DCDC architecture and reduce the secondary side The output power of the conversion module.
  • the timing difference ⁇ has left and right shifts, as follows: the secondary side conversion module drives the primary side conversion module to drive to the right, and the secondary side conversion module drives the lagging of the primary side conversion module to drive to the left. shift.
  • the timing difference ⁇ leading can increase the gain of the DCDC architecture and increase the output power of the secondary side conversion module; the timing difference ⁇ lag can reduce the gain of the DCDC architecture and reduce the output power of the secondary side conversion module.
  • the controller includes a collector that collects the input voltage (Vin) of the primary-side conversion module, and a collector that collects the common output voltage (VoHV) and output current (IoHV) of the secondary-side conversion module and the secondary-side rectification module; in the charging mode
  • the controller performs sampling and calibration on the output current (IoHV) and output voltage (VoHV) respectively, and obtains the output power through Power Calculation; the output current (IoHV) after sampling and calibration is the same as the output current
  • the reference value (IrefHV) performs the difference calculation, the loop compensation is performed on the difference between the two, and the obtained compensation value and the pre-set voltage loop preset value (VsetHV) are used for the smaller calculation, and the smaller value is taken as the voltage Loop reference value (VrefHV); use the sampled and calibrated output voltage (VoHV) to perform the difference calculation with the voltage loop reference value (VrefH
  • the controller moves to the right to increase The timing difference ⁇ increases the gain. Conversely, when the secondary side conversion module side current IoHV sampled by the controller is greater than the set current reference value refHV or the secondary side conversion module side voltage VoHV sampled is greater than the set voltage loop reference value VrefHV, the controller moves to the left The movement increases the timing difference ⁇ and decreases the gain.
  • the controller includes a collector that collects the input voltage (Vin) of the primary-side conversion module, and a collector that collects the common output voltage (VoHV) and output current (IoHV) of the secondary-side conversion module and the secondary-side rectification module; in inverter mode
  • the controller samples and calibrates the input voltage (Vin), then performs a difference operation with the input voltage reference value (VrefVin), and performs loop compensation on the difference between the two to obtain the timing difference ⁇ , and then
  • PWM operation PWM Generator
  • timing difference ⁇ between the primary side conversion module and the secondary side conversion module drive that is, there is a timing difference ⁇ between the primary side conversion modules Q1, Q4 and the secondary side conversion modules Q5, Q8, the primary side conversion modules Q2, Q3 and the secondary side
  • ⁇ between the conversion modules Q6 and Q7, as shown in Figs. 8 and 12.
  • the timing difference ⁇ has left and right shifts, as follows: the secondary side conversion module drives the primary side conversion module to drive to the right, and the secondary side conversion module drives the lagging of the primary side conversion module to drive to the left. shift.
  • shifting ⁇ to the right can reduce the gain (that is, reducing the output power of the primary conversion module), and shifting ⁇ to the left can increase the gain (that is, increasing the output power of the primary conversion module).
  • Z(fs) is the change of equivalent reactance of Lr1, Lr2 and C1 as fs changes.
  • fs is controlled to change Z(fs), and then the optimal matching characteristic is obtained.

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Abstract

本发明公开了一种适用于不同输入电网的DCDC架构及其控制方法，DCDC架构包括原边转换模块、第一变压器T1、副边转换模块，控制器、第二变压器T2和副边整流模块，第二变压器T2的原边绕组W5与第一变压器T1的原边绕组W1串联后连接原边转换模块的输出端，第二变压器T2的副边绕组W6连接副边整流模块的输入端，副边整流模块的输出端与副边转换模块的输出端并联，第二变压器T2的原边或副边设置切换开关K；控制器根据原边转换模块的母线电压控制切换开关K将第二变压器T2和副边整流模块投入或退出运行；本发明解决了大功率车载OBC后级DCDC多路并联均流的问题以及兼容单相和三相输入电压的问题；具有器件数量少、简单、容易实现等优点。

Description

一种适用于不同输入电网的DCDC架构及其控制方法 技术领域
本发明属于电源技术领域，具体涉及一种适用于不同输入电网的DCDC架构及其控制方法。
背景技术
随着社会的发展，环境污染和能源紧缺问题得到越来越多的关注，大力发展新能源汽车是解决上述两大问题的一个有效途径。随着新能源汽车技术的发展，续航里程越来越高，动力电池的容量的要求越来越高，对电池充电时间也要求越来越短，使得对车载充电机(简称：OBC)的功率急需提升。当前大功率的OBC都是采用多路并联的设计，如图1所示。在当前的设计中，由于元器件的参数都有容差，会出现不均流的问题，导致每一路的设计都要留比较的余量以及增加额外的硬件处理保证均流，造成过设计，使得成本过高。此外，当前OBC的设计组成如图2所示，由PFC和DCDC两级串联组成，大功率输出都是基于输入交流电压为三相的场景，而车载OBC往往都需要兼容单相和三相输入。在三相输入，PFC输出电压通常为800V,即DCDC的输入电压为800V；在单相输入，因为DCDC的变压器匝比是固定的，需要在单相输入时将PFC输入电压也升到800V，会增大PFC的损耗，导致效率低。
因此，如何设计一种在不增加额外的硬件成本下解决大功率的场景多路并联均流的问题，解决单相输入和三相输入不同母线电压DCDC变压器原副边匝比兼容的问题，是业界亟待解决的技术问题。
发明内容
为了解决现有技术中存在的上述缺陷，本发明提出一种适用于不同输入电网的DCDC架构及其控制方法。
本发明采用的技术方案是设计一种适用于不同输入电网的DCDC架构，包括依次连接的原边转换模块、第一变压器T1、副边转换模块、控制器、第二变压器T2和副边整流模块，其中所述第二变压器T2的原边绕组W5与第一变 压器T1的原边绕组W1串联后连接原边转换模块的输出端，所述第二变压器T2的副边绕组W6连接所述副边整流模块的输入端，副边整流模块的输出端与副边转换模块的输出端并联，所述第二变压器T2的原边或副边设置切换开关K；控制器根据原边转换模块的母线电压控制切换开关K将第二变压器T2和副边整流模块投入运行或退出运行。
所述切换开关K可以并联在所述第二变压器T2原边绕组W5的两端。
所述切换开关K也可以并联在所述第二变压器T2副边绕组W6的两端。
所述DCDC架构包括充电模式和逆变模式；在充电模式中所述母线电压高于阈值M时，控制器控制开关K断开；在充电模式中所述母线电压不高于阈值M时，控制器控制开关K闭合；在逆变模式中，控制器控制开关K闭合。
所述阈值M为600伏。
所述第一变压器T1原边绕组W1与副边绕组W2的比值和第二变压器T2原边绕组W5与副边绕组W6的比值相等；第一变压器T1原边绕组W1与二变压器T2原边绕组W5的匝数相等、线径相等；第一变压器T1副边绕组W2与二变压器T2副边绕组W6的匝数相等、线径相等。
所述副边转换模块中的功率开关采用有源器件，所述副边整流模块中的功率开关采用有无源器件。
所述副边整流模块采用桥式整流模块，包括第一二极管D1、第二二极管D2、第三二极管D2、第四二极管D4。
所述原边转换模块采用全桥结构，包括第一功率开关Q1、第二功率开关Q2、第三功率开关Q3、第四功率开关Q4；所述副边转换模块采用全桥结构，包括第五功率开关Q5、第六功率开关Q6、第七功率开关Q7、第八功率开关Q8。
所述第一变压器T1的副边绕组W2通过隔直电容C2后连接所述副边转换模块；所述第二变压器T2的副边绕组W6直接连接所述副边整流模块。
所述第一变压器T1的原边绕组与第二变压器T2的原边绕组之间串接谐振电容C1。
所述第一变压器T1的原边绕组串接第一谐振电感Lr1。
所述第二变压器T2的原边绕组串接第二谐振电感Lr2。
所述第一变压器T1和第二变压器T2集成在同一个磁芯上。
本发明还设计一种适用于不同输入电网的DCDC架构的控制方法，所述DCDC架构采用上述的适用于不同输入电网的DCDC架构。控制时只控制原边转换模块和副边转换模块，对副边整流模块不予控制，简化了控制方式降低了零部件的成本。具体控制方式是：驱动所述原边转换模块中功率开关的控制信号与驱动副边转换模块中功率开关的控制信号之间存在时序差Φ，所述控制器控制时序差Φ的超前或滞后；时序差Φ超前可增大DCDC架构的增益，增大副边转换模块的输出功率；时序差Φ滞后可降低DCDC架构的增益，减小副边转换模块的输出功率。
在充电模式中，所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边转换模块和副边整流模块共同的输出电压(VoHV)和输出电流(IoHV)的采集器；所述控制器对输出电流(IoHV)和输出电压(VoHV)分别进行采样定标、并通过功率运算(Power Calculation)得出输出功率；用采样定标后的输出电流(IoHV)同输出电流基准值(IrefHV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电压环预设值(VsetHV)做取小运算、将取其小值作为电压环基准值(VrefHV)；用采样定标后的输出电压(VoHV)同所述电压环基准值(VrefHV)进行差值运算，对两者的差值进行环路补偿，用所得补偿值与所述输出功率运算得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边转换模块中的功率开关。
在逆变模式中，所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边转换模块和副边整流模块共同的输出电压(VoHV)和输出电流(IoHV)的采集器；所述控制器对输入电压(Vin)进行采样定标、然后同输入电流基准值(VrefVin)进行差值运算，对两者的差值进行环路补偿得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边转换模块中的功率开关。
本发明提供的技术方案的有益效果是：
本发明在不增加成本的情况下解决了大功率车载OBC后级DCDC多路并联 均流的问题以及兼容单相和三相输入电压的问题；具有器件数量少、简单、容易实现等优点。
附图说明
下面结合实施例和附图对本发明进行详细说明，其中：
图1是现有技术中多路DCDC并联运行的原理框图；
图2是充电机原理框图；
图3是本发明串联一个谐振电感的电路图；
图4是本发明串联两个谐振电感的电路图；
图5是本发明第一和第二变压器合并的电路图；
图6是原边转换模块控制波形图；
图7是副边转换模块控制波形图；
图8是原边转换模块和副边转换模块控制波时序差对照图；
图9是副边转换模块控制波形与桥臂中点电压波形对照图；
图10是副边第一和第二转换模块各自输出电流和总输出电流波形对照图；
图11是原边转换模块输出电流、原边转换模块桥臂中点电压、副边第一和第二换模块桥臂中点电压波形对照图；
图12是原边转换模块和副边转换模块控制波时序差波形示意图；
图13是接单相电网时第二转换模块桥臂中点电压V_EF、第一转换模块桥臂中点电压V_CD、原边转换模块输出电压V_AB、原边转换模块输出电流Ip、DCDC架构输出电流IoHV的波形对照图；
图14是充电模式时控制器控制原理框图；
图15是逆变模式时控制器控制原理框图。
具体实施方式
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明作进一步详细说明。应当理解，此处所描述的具体实施例仅仅用于解释本发明，并不用于限定本发明。
本发明公开一种适用于不同输入电网的DCDC架构，包括依次连接的原边转换模块、第一变压器T1、副边转换模块、控制器、第二变压器T2和副边整流模块，其中所述第二变压器T2的原边绕组W5与第一变压器T1的原边绕组W1串联后连接原边转换模块的输出端，所述第二变压器T2的副边绕组W6连接所述副边整流模块的输入端，副边整流模块的输出端与副边转换模块的输出端并联，所述第二变压器T2的原边或副边设置切换开关K；控制器根据原边转换模块的母线电压控制切换开关K将第二变压器T2和副边整流模块投入运行或退出运行。
参看图2示出的充电机原理框图，在较佳实施例中DCDC架构的前端依次连接交流输入端、EMI滤波器、PFC电路构成，为DCDC电路提供直流电源。交流输入端连接外部不同输入电网，可以是三相电网，也可以是单相电网。本申请运用到车载充电机中，DCDC电路输出端主要连接车内高压电池。
在较佳实施例中(参看图3)，所述切换开关K并联在所述第二变压器T2原边绕组W5的两端。在另一些实施例中，所述切换开关K并联在所述第二变压器T2副边绕组W6的两端(未给出电路图)。
所述DCDC架构包括充电模式和逆变模式；在充电模式中所述母线电压高于阈值M时，控制器控制开关K断开；在充电模式中所述母线电压不高于阈值M时，控制器控制开关K闭合；在逆变模式中，控制器控制开关K闭合。所述开关K采用双向开关或继电器中的一种。
在较佳实施例中所述阈值M为600伏。本发明应用在充电机中时，母线电压高于阈值M，代表充电机连接三相电网。母线电压不高于阈值M，代表充电机连接单相电网。
在较佳实施例中，所述第一变压器T1原边绕组W1与副边绕组W2的比值和第二变压器T2原边绕组W5与副边绕组W6的比值相等；第一变压器T1原边绕组W1与二变压器T2原边绕组W5的匝数相等、线径相等；第一变压器T1副边绕组W2与二变压器T2副边绕组W6的匝数相等、线径相等。
所述副边转换模块中的功率开关采用有源器件，所述副边整流模块中的功率开关采用有无源器件。
在较佳实施例中，所述副边整流模块采用桥式整流模块，包括第一二极 管D1、第二二极管D2、第三二极管D2、第四二极管D4。
参看图3示出的较佳实施例，所述原边转换模块采用全桥结构，包括第一功率开关Q1、第二功率开关Q2、第三功率开关Q3、第四功率开关Q4；所述副边转换模块采用全桥结构，包括第五功率开关Q5、第六功率开关Q6、第七功率开关Q7、第八功率开关Q8。副边转换模块控制信号和输入电压如图9所示。所述第一功率开关Q1、第二功率开关Q2、第三功率开关Q3、第四功率开关Q4、第五功率开关Q5、第六功率开关Q6、第七功率开关Q7、第八功率开关Q8采用：MOSFET、SiC MOSFET、IGBT并联二极管、GAN HEMT中的一种。
在较佳实施例中，所述第一变压器T1的副边绕组W2通过隔直电容C2后连接所述副边转换模块；所述第二变压器T2的副边绕组W6直接连接所述副边整流模块。如图3、4、5所示。
在较佳实施例中，所述第一变压器T1的原边绕组与第二变压器T2的原边绕组之间串接谐振电容C1。如图3、4、5所示。
参看图3示出的实施例，所述第一变压器T1的原边绕组串接第一谐振电感Lr1。即整个原边侧只串联一个振电感。
参看图4示出的实施例，所述第二变压器T2的原边绕组串接第二谐振电感Lr2。即整个原边侧串联两个振电感。
参看图5示出的实施例，所述第一变压器T1和第二变压器T2集成在同一个磁芯上。即第一变压器T1和第二变压器T2可以分开设置，也可以合并设置。
需要指出，谐振电感设置在不同的实施例中有不同的设置方法，可以一个谐振电感，也可以是两个分立的谐振电感，还可以是集成在同一个磁芯。谐振电感可以是独立原件，也可以是变压器的漏感。
下面结合图3以本发明使用在充电机中为例，对本发明作说明。
充电模式连接三相电网：
三相输入时，母线电压为800V，即图3中的Vin＝800V，开关K断开，原边转换模块中Q1和Q3组成原边第一桥臂，桥臂中点A；Q2和Q4组成原边第二桥臂，桥臂中点B；谐振电感Lr1、变压器T1绕组W1、谐振电容C1、变压 器T2绕组W5，谐振电容Lr2串联接在一起，一端接在原边第一桥臂中点A，另一端接在原边第二桥臂中点B，组成变压器T1和T2原边串联的结构。副边转换模块中Q5和Q7组成副边第一桥臂，桥臂中点C；Q6和Q8组成副边第二桥臂，桥臂中点D；变压器T1绕组W2与隔直电容C2串联，一端接在副边第一桥臂中点C，另一端接在副边第二桥臂中点D，并联输出电容C4，组成副边转换模块输出HV1；副边整流模块中D1和D3组成副边第三桥臂，桥臂中点E；D2和D4组成副边第四桥臂，桥臂中点F；变压器T2绕组W6一端接在副边第三桥臂中点E，另一端接在副边第四桥臂中点F，并联输出电容C5组成副边整流模块输出HV2，HV1输出正端和HV2输出正端接在一起，HV1输出负端和HV2输出负端接在一起，HV1和HV2并联组成高压HV输出。按照上述连接，变压器T1绕组W2和变压器T2绕组W6组成并联连接。其中变压器T1绕组W1：W2变压器T2绕组W5：W6的匝比以及匝数，绕线线径都是相同的。
控制方式：图3的拓扑结构中，控制器通过驱动原边转换模块中功率开关Q1—Q4、以及副边转换模块中功率开关Q5—Q8实现对HV电压和电流的控制，具体的，原边转换模块：Q1和Q4驱动一致，都是50％占空比；Q2和Q3驱动一致，都是50％占空比，Q1、Q4和Q2、Q3驱动完全相反，如图6所示；副边转换模块：Q5和Q8驱动一致，都是50％占空比；Q6和Q7驱动一致，都是50％占空比，Q5、Q8和Q6、Q7驱动完全相反，如图7所示；上述所说的50％占空比在具体实施的为防止一对桥臂的上下两个开关管同时导通出现短路，都需要减去一个死区时间，在这所说的50％占空比是统称包含有死区时间。
并联均流原理：两个变压器T1原副边匝比W1：W2和T2原副边匝比W5：W6相等，且变压器原边是串联的，输入电压Vin的电压加在谐振腔、变压器T1原边W1和变压器T2原边W5上，流过两个变压器原边绕组的电流是一样的,即使谐振参数Lr，谐振电容Cr参数出现偏差，流过两个变压器的电流都是一样的。两个变压器原副边匝比，匝数都是一样的，且两路输出是短接在一起的，两路的输出电压一样，则变压器T1和T2耦合到副边的电流也是一样的，即副边转换模块输出电流Io1和副边整流模块输出电流Io2是相等的， 通过控制原边和副边转换模块输出的驱动控制总的输出电流，而副边整流模块输出电流自动和副边转换模块输出电流相等。由于变压器T1和T2的原副边绕组匝数一样，副边转换模块输出和副边整流模块输出的电流自动均衡，不需要额外作均流处理。此外，副边整流模块是二极管整流，二极管导通由原边电流过零决定，不存在副边转换模块是有源器件由于驱动的轻微差异导致的变压器偏磁的问题，在本控制方式上副边整流模块可以省略隔直电容，即C2电容在副边整流模块是不需要的。
充电模式连接单相电网：
单相输入时，母线电压降低一半，即Vin＝400V，开关K闭合。原边转换模块中，变压器T2原边绕组W5短路，输入电压加在谐振腔和变压器T1原边绕组W1上，谐振腔参数以及谐振点一样，单相和三相输入的谐振点都是如下公式1所示；
Figure PCTCN2020101136-appb-000001
由于输入电压降低了一半，同时原边转换模块的变压器也减少一半(短路T2原边绕组W5)，谐振点不变，在单相输入时，单个变压器T1增益还是一样的，则相比三相输入，单相的功率减少一半，同时原边电流也是一样的，变压器T1的匝比以及绕线线径的设计都是一样的，不会造成因为单相和三相兼容造成过设计。
逆变模式(逆变模式不分三相或单相电网)：
在图3的拓扑图中，开关K闭合，即变压器T2绕组W5短路。HV为能量的输入端，输出侧为Vin，与充电模式相反。原边侧为能量输出侧，副边HV为能量输入侧。
控制方式：控制器通过驱动原边转换模块功率开关Q1—Q4，以及副边转换模块功率开关Q5—Q8实现对能量输出的控制。具体地，原边转换模块Q1和Q4驱动一致，都是50％占空比；Q2和Q3驱动一致，都是50％占空比，Q1、Q4和Q2、Q3驱动完全相反，如图6所示。副边转换模块中Q5和Q8驱动一致，都是50％占空比；Q6和Q7驱动一致，都是50％占空比，Q5、Q8和Q6、Q7驱动完全相反，如图7所示。上述所说的50％占空比在具体实施的为防止 一对桥臂的上下两个开关管同时导通出现短路，都需要减去一个死区时间，在这所说的50％占空比是统称包含有死区时间。
具体实施举例：
运用本发明的硬件框架和控制方式；工作在三相时，Vin＝800V,高压输出300V，HV输出功率21KW，即：高压HV输出总电流70A，对比副边转换模块输出和副边整流模块输出的均流性。
表1：仿真参数
Figure PCTCN2020101136-appb-000002
仿真结果如表2所示，副边第一和第二转换模块各自输出电流和总输出电流波形对照如图10所示。从表2中可以看出，副边转换模块输出和副边整流模块输出的电流几乎无偏差，证明本控制方式的可行性。图11示出了原边转换模块输出电流、原边转换模块桥臂中点电压、副边第一和第二换模块(即副边转换模块和副边整流模块)桥臂中点电压波形对照图。
表2：仿真结果
项目 结果
副边转换模块输出电流(IoHV1) 35.019A
副边整流模块输出电流(IoHV2) 35.025A
副边输出总电流(IoHV) 70.045A
图13是接单相电网时，Vin＝400V，继电器K闭合将变压器T2原边绕组W5短路状态下第二转换模块桥臂中点电压V_EF、第一转换模块桥臂中点电 压V_CD、原边转换模块输出电压V_AB、原边转换模块输出电流Ip、DCDC架构输出电流IoHV的波形对照图。从图中可以看到：(1)在单相输入时高压输出总电流相比三相输入时降低一半，即：输出功率降低一半；(2)副边整流模块输出桥臂中点E、F电压V_EF为0，说明所有的输出功率来自副边转换模块输出；(3)单相和三相输入时原边电流Ip有效值是一样的。
表3：单相和三相输入电流对比
  三相输入 单相输入
原边电流Ip 45.42A 45.501A
本发明还公开了一种适用于不同输入电网的DCDC架构的控制方法，所述DCDC架构采用上述的适用于不同输入电网的DCDC架构；为调节DCDC架构的输出功率，驱动原边转换模块中功率开关的控制信号与驱动副边转换模块中功率开关的控制信号之间存在时序差Φ，如图8和图12所示。即原边转换模块中Q1、Q4和副边转换模块中Q5、Q8之间存在时序差Φ，原边转换模块中Q2、Q3和边第一转换模块中Q6、Q7之间存在时序差Φ。所述控制器控制时序差Φ的超前或滞后；时序差Φ超前可增大DCDC架构的增益，增大副边转换模块的输出功率；时序差Φ滞后可降低DCDC架构的增益，减小副边转换模块的输出功率。以原边转换模块驱动为参考，时序差Φ有左移和右移，具体如下：副边转换模块驱动超前原边转换模块驱动为右移，副边转换模块驱动滞后原边转换模块驱动为左移。时序差Φ超前可增大DCDC架构的增益，增大副边转换模块的输出功率；时序差Φ滞后可降低DCDC架构的增益，减小副边转换模块的输出功率。
参看图14示出的充电模式时控制器控制原理框图。所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边转换模块和副边整流模块共同的输出电压(VoHV)和输出电流(IoHV)的采集器；在充电模式中，所述控制器对输出电流(IoHV)和输出电压(VoHV)分别进行采样定标、并通过功率运算(Power Calculation)得出输出功率；用采样定标后的输出电流(IoHV)同输出电流基准值(IrefHV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电压环预设值(VsetHV)做取小运算、 将取其小值作为电压环基准值(VrefHV)；用采样定标后的输出电压(VoHV)同所述电压环基准值(VrefHV)进行差值运算，对两者的差值进行环路补偿，用所得补偿值与所述输出功率运算得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边转换模块中的功率开关。
当控制器采样到的副边转换模块电流IoHV比实际设定的IrefHV小或者采样到的副边转换模块侧电压VoHV比设定的电压环基准值(VrefHV)小，控制器往右移动增大时序差Φ增大增益。反之，当控制器采样到的副边转换模块侧电流IoHV比设定的电流基准值refHV大或者采样到的副边转换模块侧电压VoHV比设定的电压环基准值VrefHV大，控制器往左移动增大时序差Φ降低增益。
参看图15示出的逆变模式时控制器控制原理框图。所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边转换模块和副边整流模块共同的输出电压(VoHV)和输出电流(IoHV)的采集器；在逆变模式中，所述控制器对输入电压(Vin)进行采样定标、然后同输入电压基准值(VrefVin)进行差值运算，对两者的差值进行环路补偿得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边转换模块中的功率开关。
原边转换模块和副边转换模块驱动之间存在时序差Φ，即原边转换模块Q1、Q4和副边转换模块Q5、Q8之间存在时序差Φ，原边转换模块Q2、Q3和副边转换模块Q6、Q7之间存在时序差Φ，如图8和图12所示。以原边转换模块驱动为参考，时序差Φ有左移和右移，具体如下：副边转换模块驱动超前原边转换模块驱动为右移，副边转换模块驱动滞后原边转换模块驱动为左移。逆变模式Φ右移可以降低增益(即减小原边转换模块输出功率)，Φ左移可以增大增益(即增大原边转换模块输出功率)。
在控制Φ的同时，也控制原边转换模块和副边转换模块功率开关开关周期fs。其意义在于，由于原边转换模块中存在电感Lr1、Lr2和电容C1，两者组成一个可以随开关周期变化而等效电抗发生变化的网络，其数学表达式为：
Figure PCTCN2020101136-appb-000003
其中Z(fs)便是随着fs变化，Lr1、Lr2和C1等效电抗的变化。实际应用中，为了提高效率，避免无功能量过多，会通过控制fs，从而改变Z(fs)，进而得出最优匹配特性。
以上实施例仅为举例说明，非起限制作用。任何未脱离本申请精神与范畴，而对其进行的等效修改或变更，均应包含于本申请的权利要求范围之中。

Claims (17)

  1. 一种适用于不同输入电网的DCDC架构，包括依次连接的原边转换模块、第一变压器T1、副边转换模块，以及控制器，其特征在于：还包括第二变压器T2和副边整流模块，其中所述第二变压器T2的原边绕组W5与第一变压器T1的原边绕组W1串联后连接原边转换模块的输出端，所述第二变压器T2的副边绕组W6连接所述副边整流模块的输入端，副边整流模块的输出端与副边转换模块的输出端并联，所述第二变压器T2的原边或副边设置切换开关K；控制器根据原边转换模块的母线电压控制切换开关K将第二变压器T2和副边整流模块投入运行或退出运行。
  2. 如权利要求1所述的适用于不同输入电网的DCDC架构，其特征在于：所述切换开关K并联在所述第二变压器T2原边绕组W5的两端。
  3. 如权利要求1所述的适用于不同输入电网的DCDC架构，其特征在于：所述切换开关K并联在所述第二变压器T2副边绕组W6的两端。
  4. 如权利要求2或3所述的适用于不同输入电网的DCDC架构，其特征在于：所述DCDC架构包括充电模式和逆变模式；
    在充电模式中所述母线电压高于阈值M时，控制器控制开关K断开；
    在充电模式中所述母线电压不高于阈值M时，控制器控制开关K闭合；
    在逆变模式中，控制器控制开关K闭合。
  5. 如权利要求4所述的适用于不同输入电网的DCDC架构，其特征在于：所述阈值M为600伏。
  6. 如权利要求4所述的适用于不同输入电网的DCDC架构，其特征在于：所述第一变压器T1原边绕组W1与副边绕组W2的比值和第二变压器T2原边绕组W5与副边绕组W6的比值相等；第一变压器T1原边绕组W1与二变压器T2原边绕组W5的匝数相等、线径相等；第一变压器T1副边绕组W2与二变压器T2副边绕组W6的匝数相等、线径相等。
  7. 如权利要求1所述的适用于不同输入电网的DCDC架构，其特征在于：所述副边转换模块中的功率开关采用有源器件，所述副边整流模块中的功率开关采用有无源器件。
  8. 如权利要求7所述的适用于不同输入电网的DCDC架构，其特征在于： 所述副边整流模块采用桥式整流模块，包括第一二极管D1、第二二极管D2、第三二极管D2、第四二极管D4。
  9. 如权利要求7所述的适用于不同输入电网的DCDC架构，其特征在于：所述原边转换模块采用全桥结构，包括第一功率开关Q1、第二功率开关Q2、第三功率开关Q3、第四功率开关Q4；所述副边转换模块采用全桥结构，包括第五功率开关Q5、第六功率开关Q6、第七功率开关Q7、第八功率开关Q8。
  10. 如权利要求1所述的适用于不同输入电网的DCDC架构，其特征在于：所述第一变压器T1的副边绕组W2通过隔直电容C2后连接所述副边转换模块；所述第二变压器T2的副边绕组W6直接连接所述副边整流模块。
  11. 如权利要求1所述的适用于不同输入电网的DCDC架构，其特征在于：所述第一变压器T1的原边绕组与第二变压器T2的原边绕组之间串接谐振电容C1。
  12. 如权利要求1所述的适用于不同输入电网的DCDC架构，其特征在于：所述第一变压器T1的原边绕组串接第一谐振电感Lr1。
  13. 如权利要求11所述的适用于不同输入电网的DCDC架构，其特征在于：所述第二变压器T2的原边绕组串接第二谐振电感Lr2。
  14. 如权利要求1所述的适用于不同输入电网的DCDC架构，其特征在于：所述第一变压器T1和第二变压器T2集成在同一个磁芯上。
  15. 一种适用于不同输入电网的DCDC架构的控制方法，其特征在于：所述DCDC架构采用权利要求1至14任一项所述的适用于不同输入电网的DCDC架构；驱动所述原边转换模块中功率开关的控制信号与驱动副边转换模块中功率开关的控制信号之间存在时序差Φ，所述控制器控制时序差Φ的超前或滞后；时序差Φ超前可增大DCDC架构的增益，增大副边转换模块的输出功率；时序差Φ滞后可降低DCDC架构的增益，减小副边转换模块的输出功率。
  16. 如权利要求15所述的适用于不同输入电网的DCDC架构，其特征在于：所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边转换模块和副边整流模块共同的输出电压(VoHV)和输出电流(IoHV)的采集器；
    在充电模式中，所述控制器对输出电流(IoHV)和输出电压(VoHV)分 别进行采样定标、并通过功率运算(Power Calculation)得出输出功率；用采样定标后的输出电流(IoHV)同输出电流基准值(IrefHV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电压环预设值(VsetHV)做取小运算、将取其小值作为电压环基准值(VrefHV)；用采样定标后的输出电压(VoHV)同所述电压环基准值(VrefHV)进行差值运算，对两者的差值进行环路补偿，用所得补偿值与所述输出功率运算得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边转换模块中的功率开关。
  17. 如权利要求15所述的适用于不同输入电网的DCDC架构，其特征在于：所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边转换模块和副边整流模块共同的输出电压(VoHV)和输出电流(IoHV)的采集器；
    在逆变模式中，所述控制器对输入电压(Vin)进行采样定标、然后同输入电流基准值(VrefVin)进行差值运算，对两者的差值进行环路补偿得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边转换模块中的功率开关。
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