WO2021227230A1 - 一种兼容型大功率双端输出车载充电机及其控制方法 - Google Patents

一种兼容型大功率双端输出车载充电机及其控制方法 Download PDF

Info

Publication number
WO2021227230A1
WO2021227230A1 PCT/CN2020/101135 CN2020101135W WO2021227230A1 WO 2021227230 A1 WO2021227230 A1 WO 2021227230A1 CN 2020101135 W CN2020101135 W CN 2020101135W WO 2021227230 A1 WO2021227230 A1 WO 2021227230A1
Authority
WO
WIPO (PCT)
Prior art keywords
voltage
conversion module
power switch
transformer
power
Prior art date
Application number
PCT/CN2020/101135
Other languages
English (en)
French (fr)
Inventor
刘钧
冯颖盈
姚顺
徐金柱
张远昭
Original Assignee
深圳威迈斯新能源股份有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 深圳威迈斯新能源股份有限公司 filed Critical 深圳威迈斯新能源股份有限公司
Publication of WO2021227230A1 publication Critical patent/WO2021227230A1/zh

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J7/00—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
    • H02J7/02—Circuit arrangements for charging or depolarising batteries or for supplying loads from batteries for charging batteries from ac mains by converters
    • H02J7/04—Regulation of charging current or voltage
    • H02J7/06—Regulation of charging current or voltage using discharge tubes or semiconductor devices
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L53/00—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles
    • B60L53/20—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles characterised by converters located in the vehicle
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L53/00—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles
    • B60L53/20—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles characterised by converters located in the vehicle
    • B60L53/22—Constructional details or arrangements of charging converters specially adapted for charging electric vehicles
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/44—Conversion of dc power input into dc power output with intermediate conversion into ac by combination of static with dynamic converters; by combination of dynamo-electric with other dynamic or static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/008—Plural converter units for generating at two or more independent and non-parallel outputs, e.g. systems with plural point of load switching regulators
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/70—Energy storage systems for electromobility, e.g. batteries
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/7072—Electromobility specific charging systems or methods for batteries, ultracapacitors, supercapacitors or double-layer capacitors
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/72—Electric energy management in electromobility
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/80—Technologies aiming to reduce greenhouse gasses emissions common to all road transportation technologies
    • Y02T10/92—Energy efficient charging or discharging systems for batteries, ultracapacitors, supercapacitors or double-layer capacitors specially adapted for vehicles
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T90/00—Enabling technologies or technologies with a potential or indirect contribution to GHG emissions mitigation
    • Y02T90/10—Technologies relating to charging of electric vehicles
    • Y02T90/14—Plug-in electric vehicles

Definitions

  • the invention belongs to the technical field of power supplies, and specifically relates to a compatible high-power dual-end output vehicle-mounted charger and a control method thereof.
  • the current OBC design composition is shown in Figure 2. It consists of two stages of PFC and DCDC in series.
  • the high-power output is based on the scenario where the input AC voltage is three-phase, and the vehicle-mounted OBC often needs to be compatible with single-phase and three-phase inputs.
  • the PFC output voltage is usually 800V, that is, the DCDC input voltage is 800V; in the single-phase input, because the DCDC transformer turns ratio is fixed, it is necessary to increase the PFC input voltage to 800V during single-phase input. It will increase the loss of PFC, resulting in low efficiency.
  • the present invention proposes a compatible high-power dual-end output vehicle-mounted charger and a control method thereof.
  • the technical scheme adopted by the present invention is to design a compatible high-power dual-end output vehicle-mounted charger, which includes connecting the primary side conversion module, the first transformer T1, the secondary side high-voltage conversion module, the controller, and the secondary side of the first transformer T1 in sequence.
  • the secondary low-voltage conversion module of the winding, and the second transformer T2 and the secondary rectifier module wherein the primary winding W5 of the second transformer T2 and the primary winding W1 of the first transformer T1 are connected in series and then connected to the output of the primary conversion module Terminal, the secondary winding W6 of the second transformer T2 is connected to the input terminal of the secondary rectifier module, the output terminal of the secondary rectifier module is connected in parallel with the output terminal of the secondary high-voltage conversion module, and the principle of the second transformer T2
  • the switching switch K is set on the side or the secondary side; the controller controls the switching switch K according to the bus voltage of the primary side conversion module to put the second transformer T2 and the secondary side rectifier module into operation or out of operation.
  • the switch K may be connected in parallel to both ends of the primary winding W5 of the second transformer T2.
  • the switch K can also be connected in parallel to both ends of the secondary winding W6 of the second transformer T2.
  • the charger includes a charging mode, an inverter mode, and a DCDC mode; when the bus voltage is higher than the threshold M in the charging mode, the controller controls the switch K to turn off; in the charging mode, the bus voltage is not higher than the threshold M When the time, the controller controls the switch K to close; in the inverter mode, the controller controls the switch K to close; in the DCDC mode, the controller controls the switch K to close.
  • the threshold value M is 600 volts.
  • the ratio of the primary winding W1 of the first transformer T1 to the second secondary winding W2 is equal to the ratio of the primary winding W5 to the secondary winding W6 of the second transformer T2; the primary winding W1 of the first transformer T1 and the primary winding W1 of the second transformer T2 are equal
  • the number of turns of the side winding W5 is the same, and the wire diameter is the same; the number of turns and the wire diameter of the second secondary winding W2 of the first transformer T1 and the secondary winding W6 of the second transformer T2 are the same.
  • the power switch in the secondary side high voltage conversion module adopts active devices, and the power switch in the secondary side rectifier module adopts passive devices.
  • the secondary side rectifier module adopts a bridge rectifier module, which includes a first diode D1, a second diode D2, a third diode D2, and a fourth diode D4.
  • the primary-side conversion module adopts a full-bridge structure, including a first power switch Q1, a second power switch Q2, a third power switch Q3, and a fourth power switch Q4;
  • the secondary-side high-voltage conversion module adopts a full-bridge structure, including a first Five power switch Q5, sixth power switch Q6, seventh power switch Q7, eighth power switch Q8; among them, the fifth power switch Q5 and the seventh power switch Q7 are a pair of bridge arms, and the sixth power switch Q6 and the eighth power switch
  • the switch Q8 is a pair of bridge arms, the fifth power switch Q5 and the sixth power switch Q6 are upper bridge arms, and the seventh power switch Q7 and the eighth power switch Q8 are lower bridge arms.
  • the first transformer T1 includes a second secondary winding W2, a third secondary winding W3, and a fourth secondary winding W4.
  • the second secondary winding W2 is connected to the secondary high-voltage conversion module; the secondary low-voltage
  • the conversion module includes a ninth power switch Q9, a tenth power switch Q10, and an eleventh power switch Q11; the drain of the ninth power switch Q9 is connected to the end of the same name of the fourth secondary winding W4, and the tenth power switch Q10
  • the drain of is connected to the synonymous end of the third secondary winding W3, the synonymous end of the fourth secondary winding W4 and the same end of the third secondary winding W3 are connected in series, the eleventh power switch Q11 and the output inductor L0 are connected in series, and then connected
  • the positive output terminal of the low-voltage conversion module on the secondary side, the source of the ninth power switch Q9 and the source of the tenth power switch Q10 are grounded.
  • the second secondary winding W2 of the first transformer T1 is connected to the secondary high-voltage conversion module through a DC blocking capacitor C2; the secondary winding W6 of the second transformer T2 is directly connected to the secondary rectifier module.
  • a resonant capacitor C1 is connected in series between the primary winding of the first transformer T1 and the primary winding of the second transformer T2.
  • the primary winding of the first transformer T1 is connected in series with the first resonant inductor Lr1.
  • the primary winding of the second transformer T2 is connected in series with a second resonant inductor Lr2.
  • the first transformer T1 and the second transformer T2 are integrated on the same magnetic core.
  • the present invention also designs a control method for a compatible high-power dual-end output vehicle-mounted charger, the charger adopts the above-mentioned compatible high-power dual-end output vehicle-mounted charger;
  • the controller controls the lead or lag of the timing difference ⁇ ; the lead of the timing difference ⁇ can increase the gain of the charger and increase the secondary The output power of the high-voltage conversion module on the side; the timing difference ⁇ lag can reduce the gain of the charger and reduce the output power of the high-voltage conversion module on the secondary side.
  • the fifth power switch Q5 and the eighth power switch Q8 in the secondary side high voltage conversion module are turned on, corresponding to the tenth power switch Q10 being turned on, and the sixth power switch Q6 and the seventh power switch Q7 are turned on.
  • Turning on corresponds to the turning on of the ninth power switch Q9, the switching period of the eleventh power switch Q11 is twice the switching period of the fifth power switch Q5, the turn-off edge of the eleventh power switch Q11 and the turn-off edge of the fifth power switch Q5 It is aligned with the turn-off edge of the sixth power switch Q6; in the DCDC mode, the eleventh power switch Q11 maintains a normally-on state.
  • the controller includes a collector that collects the input voltage (Vin) of the primary-side conversion module, and a collector that collects the high-voltage output voltage (VoHV) and the high-voltage output current (IoHV) common to the secondary-side high-voltage conversion module and the secondary-side rectifier module; A collector for low-voltage output voltage (VoLV) and low-voltage output current (IoLV) of the secondary-side low-voltage conversion module; in the charging mode, the controller samples and calibrates the low-voltage output voltage (VoLV) and the low-voltage output current (IoLV) respectively , And obtain the output power through Power Calculation; use the sampled and calibrated low-voltage output voltage (VoLV) to perform the difference calculation with the low-voltage output voltage reference value (VrefLV), and perform loop compensation for the difference between the two.
  • Vin input voltage
  • VrefLV low-voltage output voltage reference value
  • the obtained compensation value and the preset current loop preset value (IsetLV) are calculated to be small, and the small value is used as the current loop reference value (IrefLV), and then the low voltage output current (IoLV) after sampling and calibration is performed Difference calculation, loop compensation calculation is performed on the difference between the two to obtain the duty cycle, and then the eleventh power switch Q11 is driven by the PWM operation (PWM Generator); the high-voltage output current after sampling and calibration (IoHV ) Perform difference calculation with the high-voltage output current reference value (IrefHV), perform loop compensation on the difference between the two, and take the smaller calculation between the obtained compensation value and the preset voltage loop preset value (VsetHV), and take The small value is used as the voltage loop reference value (VrefHV), and then the difference calculation is performed with the sampled and calibrated high-voltage output current (VoHV), and the difference between the two is subjected to loop compensation, and the obtained loop compensation value is compared with the The output power calculation obtains the timing difference
  • the controller includes a collector that collects the input voltage (Vin) of the primary-side conversion module, and a collector that collects the high-voltage output voltage (VoHV) and the high-voltage output current (IoHV) common to the secondary-side high-voltage conversion module and the secondary-side rectifier module; The collector of the low-voltage output voltage (VoLV) and low-voltage output current (IoLV) of the secondary side low-voltage conversion module; in the inverter mode, the controller samples and sets the low-voltage output voltage (VoLV) and the low-voltage output current (IoLV) respectively.
  • the output power is obtained by power calculation (Power Calculation); the difference calculation between the sampled and calibrated low-voltage output voltage (VoLV) and the low-voltage output voltage reference value (VrefLV) is performed, and the difference between the two is subjected to loop compensation ,
  • the obtained compensation value and the pre-set current loop preset value (IsetLV) are calculated to be small, and the small value is used as the current loop reference value (IrefLV), and then the low voltage output current after sampling and calibration (IoLV) Perform difference calculation, perform loop compensation calculation on the difference between the two to obtain the duty cycle, and then drive the eleventh power switch Q11 through the PWM operation (PWM Generator); use the sampled and calibrated input voltage (Vin ) And the voltage reference value (VrefVo) for difference calculation, loop compensation for the difference between the two, and the obtained compensation value and the output power to calculate the timing difference ⁇ , and then through the PWM operation (PWM Generator) Drive the power switches in the primary side conversion module and the secondary side high voltage conversion
  • the invention solves the problem of multi-channel parallel current sharing of high-power vehicle-mounted OBC rear-stage DCDC and compatibility with single-phase and three-phase input voltages without increasing the cost; it has the advantages of small number of components, simplicity, and easy implementation.
  • Fig. 1 is a principle block diagram of parallel operation of multiple DCDCs in the prior art
  • FIG. 2 is a functional block diagram of the charger
  • Figure 3 is a circuit diagram of a resonant inductor connected in series according to the present invention.
  • Figure 4 is a circuit diagram of the present invention in which two resonant inductors are connected in series;
  • Figure 5 is a circuit diagram of the combined first and second transformers of the present invention.
  • Figure 6 is a control waveform diagram of the primary side conversion module
  • Figure 7 is a control waveform diagram of the secondary side high voltage conversion module
  • Figure 8 is a comparison diagram of the control wave timing difference between the primary side conversion module and the secondary side high voltage conversion module
  • Figure 9 is a comparison diagram of the control waveform of the secondary side high voltage conversion module and the waveform of the midpoint voltage of the bridge arm;
  • Figure 10 is a comparison diagram of the respective output currents and total output current waveforms of the first and second conversion modules on the secondary side;
  • 11 is a comparison diagram of the output current of the primary side conversion module, the primary side conversion module bridge arm midpoint voltage, and the secondary side first and second conversion module bridge arms midpoint voltage waveform comparison diagram;
  • Figure 12 is a schematic diagram of the timing difference waveform of the control wave between the primary side conversion module and the secondary side high voltage conversion module;
  • Figure 13 shows the second conversion module bridge arm midpoint voltage V_EF, the first conversion module bridge arm midpoint voltage V_CD, primary conversion module output voltage V_AB, primary conversion module output current Ip, and charger output current when connected to a single-phase power grid. IoHV waveform comparison chart;
  • Figure 14 is a block diagram of the controller control principle in charging mode
  • Figure 15 is a block diagram of the controller control principle in inverter mode and DCDC mode
  • Figure 16 is a comparison diagram of the control waveforms of the secondary side high and low voltage conversion modules in the charging mode
  • Figure 17 is a control waveform comparison diagram of the low-voltage conversion module in DCDC mode
  • Figure 18 is a schematic diagram of the flow direction of charging mode capabilities
  • Figure 19 is a schematic diagram of the flow direction of the inverter mode capability
  • Figure 20 is a schematic diagram of the flow direction of DCDC mode capabilities.
  • the invention discloses a compatible high-power double-terminal output vehicle-mounted charger, which includes a secondary side connected to a primary side conversion module, a first transformer T1, a secondary side high-voltage conversion module, a controller, and a secondary side winding of the first transformer T1 in sequence Low-voltage conversion module, as well as a second transformer T2 and a secondary rectifier module, wherein the primary winding W5 of the second transformer T2 and the primary winding W1 of the first transformer T1 are connected in series to the output end of the primary conversion module,
  • the secondary winding W6 of the second transformer T2 is connected to the input terminal of the secondary rectification module, the output terminal of the secondary rectification module is connected in parallel with the output terminal of the secondary high-voltage conversion module, and the primary or secondary side of the second transformer T2 Set the switch K; the controller controls the switch K according to the bus voltage of the primary side conversion module to put the second transformer T2 and the secondary side rectifier module into operation or out of operation.
  • the front end of the charger is sequentially connected to an AC input terminal, an EMI filter, and a PFC circuit to provide DC power for the DCDC circuit.
  • the AC input terminal is connected to different external input power grids, which can be a three-phase power grid or a single-phase power grid.
  • the secondary side high-voltage conversion module is connected to the high-voltage power battery, and the secondary side low-voltage conversion module is connected to the low-voltage battery and the entire vehicle electrical equipment.
  • the switch K can be connected in parallel to both ends of the primary winding W5 of the second transformer T2. In some other embodiments, the switch K is connected in parallel with both ends of the secondary winding W6 of the second transformer T2 (the circuit diagram is not shown).
  • the charger includes a charging mode, an inverter mode, and a DCDC mode;
  • the charging mode refers to the transfer of energy from the AC grid to the high-voltage power battery and the low-voltage side battery and the entire vehicle electrical equipment
  • the inverter mode (Refer to Figure 19) means that the energy is taken from the high-voltage power battery, and the alternating current is inverted to the consumer and to the low-voltage side battery and the electrical equipment at the same time.
  • the DCDC mode (refer to Figure 20) means that the energy is taken from the high-voltage power battery to the whole Low-voltage battery on the low-voltage side of the vehicle and electrical appliances for the entire vehicle.
  • the controller controls the switch K to open; in the charging mode, when the bus voltage is not higher than the threshold M, the controller controls the switch K to close; in the inverter mode, The controller controls the switch K to close; in the DCDC mode, the controller controls the switch K to close.
  • the switch K adopts one of a two-way switch or a relay.
  • the threshold M is 600 volts.
  • the bus voltage is higher than the threshold value M, which means that the charger is connected to a three-phase power grid.
  • the bus voltage is not higher than the threshold value M, which means that the charger is connected to a single-phase power grid.
  • the ratio of the primary winding W1 of the first transformer T1 to the second secondary winding W2 is equal to the ratio of the primary winding W5 to the secondary winding W6 of the second transformer T2; the primary winding of the first transformer T1
  • the winding W1 and the primary winding W5 of the second transformer T2 have the same number of turns and the same wire diameter; the first transformer T1 and the second secondary winding W2 and the second transformer T2 have the same number of turns and the same wire diameter.
  • the power switch in the secondary side high voltage conversion module adopts active devices, and the power switch in the secondary side rectifier module adopts passive devices.
  • the secondary side rectifier module adopts a bridge rectifier module, which includes a first diode D1, a second diode D2, a third diode D2, and a fourth diode D4.
  • the primary side conversion module adopts a full-bridge structure, including a first power switch Q1, a second power switch Q2, a third power switch Q3, and a fourth power switch Q4;
  • the side high voltage conversion module adopts a full bridge structure, including a fifth power switch Q5, a sixth power switch Q6, a seventh power switch Q7, and an eighth power switch Q8;
  • the fifth power switch Q5 and the seventh power switch Q7 are a pair of bridges Arm
  • the sixth power switch Q6 and the eighth power switch Q8 are a pair of bridge arms
  • the fifth power switch Q5 and the sixth power switch Q6 are upper bridge arms
  • the seventh power switch Q7 and the eighth power switch Q8 are lower bridge arms. arm.
  • the control signal and input voltage of the secondary side high voltage conversion module are shown in Figure 9.
  • the first power switch Q1, the second power switch Q2, the third power switch Q3, the fourth power switch Q4, the fifth power switch Q5, the sixth power switch Q6, the seventh power switch Q7, and the eighth power switch Q8 are adopted One of MOSFET, SiC MOSFET, IGBT parallel diode, and GAN HEMT.
  • the first transformer T1 includes a second secondary winding W2, a third secondary winding W3, and a fourth secondary winding W4, and the second secondary winding W2 is connected to the Secondary side high voltage conversion module;
  • the secondary side low voltage conversion module includes a ninth power switch Q9, a tenth power switch Q10, and an eleventh power switch Q11;
  • the drain of the ninth power switch Q9 is connected to the fourth secondary winding W4
  • the drain of the tenth power switch Q10 is connected to the synonymous end of the third secondary winding W3, and the synonymous end of the fourth secondary winding W4 is connected to the same end of the third secondary winding W3 and then connected in series for the tenth
  • a power switch Q11 and an output inductor L0 are then connected to the positive output terminal of the low-voltage conversion module on the secondary side, and the sources of the ninth power switch Q9 and the tenth power switch Q10 are grounded.
  • the second secondary winding W2 of the first transformer T1 is connected to the secondary high-voltage conversion module through a DC blocking capacitor C2; the secondary winding W6 of the second transformer T2 is directly connected to the Secondary side rectifier module. As shown in Figures 3, 4, and 5.
  • a resonant capacitor C1 is connected in series between the primary winding of the first transformer T1 and the primary winding of the second transformer T2. As shown in Figures 3, 4, and 5.
  • the primary winding of the first transformer T1 is connected in series with the first resonant inductor Lr1. That is, only one vibrating inductor is connected in series on the entire primary side.
  • the primary winding of the second transformer T2 is connected in series with the second resonant inductor Lr2. That is, two vibrating inductors are connected in series on the entire primary side.
  • the first transformer T1 and the second transformer T2 are integrated on the same magnetic core. That is, the first transformer T1 and the second transformer T2 can be installed separately or combined.
  • the resonant inductor setting may be one resonant inductor, or two separate resonant inductors, or it may be integrated in the same magnetic core.
  • the resonant inductor can be an independent component or the leakage inductance of the transformer.
  • FIG. 3 taking the use of the present invention in a charger as an example.
  • the charging mode is connected to the three-phase grid:
  • switch K is disconnected, and Q1 and Q3 in the primary side conversion module form the first leg of the primary side, and the midpoint of the bridge arm is A;
  • Q2 and Q4 are composed
  • the resonant inductor Lr1, the transformer T1 winding W1, the resonant capacitor C1, the transformer T2 winding W5, and the resonant capacitor Lr2 are connected in series, and one end is connected to the midpoint A of the first leg of the primary side , The other end is connected to the middle point B of the second bridge arm of the primary side, forming a structure in which the primary sides of the transformer T1 and T2 are connected in series.
  • Q5 and Q7 form the first bridge arm of the secondary side, with the midpoint C of the bridge arm;
  • Q6 and Q8 form the second bridge arm of the secondary side, with the midpoint D of the bridge arm;
  • the transformer T1 winding W2 is connected in series with the DC blocking capacitor C2 One end is connected to the midpoint C of the first bridge arm of the secondary side, and the other end is connected to the midpoint D of the second bridge arm of the secondary side.
  • the output capacitor C4 is connected in parallel to form the secondary side high voltage conversion module to output HV1; D1 and D3 in the secondary side rectifier module It forms the third bridge arm of the secondary side and the midpoint E of the bridge arm; D2 and D4 form the fourth bridge arm of the secondary side, the midpoint F of the bridge arm; one end of the transformer T2 winding W6 is connected to the midpoint E of the third bridge arm of the secondary side and the other end Connected to the middle point F of the fourth bridge arm of the secondary side, parallel output capacitor C5 forms the secondary side rectifier module to output HV2, the positive output terminal of HV1 and the positive output terminal of HV2 are connected together, and the negative output terminal of HV1 and the negative output terminal of HV2 are connected together.
  • HV1 and HV2 are connected in parallel to form a high-voltage HV output.
  • the winding W2 of the transformer T1 and the winding W6 of the transformer T2 are connected in parallel.
  • transformer T1 winding W1: W2 transformer T2 winding W5: W6 turns ratio and number of turns, winding wire diameter are the same.
  • transformer T1 windings W3 and W4 are connected in series to form a middle tap form, and the tenth power switch Q10, D5, Lo, C3 forms a buck circuit to output LV voltage; the LV output voltage is stabilized by controlling the duty cycle of Q11.
  • Control method In the topological structure of Figure 3, the controller realizes the control of HV voltage and current by driving the power switches Q1—Q4 in the primary side conversion module and the power switches Q5—Q8 in the secondary side high voltage conversion module.
  • the original Edge conversion module Q1 and Q4 drive the same, both are 50% duty cycle;
  • Q2 and Q3 drive the same, both are 50% duty cycle, Q1, Q4 and Q2, Q3 drive are completely opposite, as shown in Figure 6;
  • Side high voltage conversion module Q5 and Q8 drive the same, both are 50% duty cycle;
  • Q6 and Q7 drive the same both are 50% duty cycle, Q5, Q8 and Q6, Q7 drive completely opposite, as shown in Figure 7;
  • the 50% duty cycle mentioned above is specifically implemented in order to prevent the upper and lower switch tubes of a pair of bridge arms from being turned on at the same time, and a dead time needs to be subtracted.
  • the 50% duty cycle It is collectively referred to as including dead time.
  • the secondary side low voltage conversion module (LV) output control the voltage of the integrated transformer T1 is clamped by the secondary side high voltage conversion module output, and the transformer voltage inversion is determined by the drive of Q5/Q8 and Q6/Q7, as shown in Figure 9.
  • the conduction of the synchronous rectification Q9 and Q10 of the secondary-side low-voltage conversion module is determined by the power switch output by the secondary-side high-voltage conversion module. That is, Q5/Q8 conduction corresponds to synchronous rectifier Q10 conduction, and Q6/Q7 conduction corresponds to Q9 conduction.
  • the power switch Q11 driven by the LV output must complete a switching cycle when Q5/Q8 or Q6/Q7 is turned on, that is, the switching cycle of Q11 is Q5/Q6/Q7/Q8. 2 times, as shown in Figure 16, the turn-off edge of Q11 is aligned with the turn-off edge of the first high-voltage switch Q5-Q8. Stabilize the LV output voltage and current by controlling the duty cycle of Q11. With this control method, current sharing is realized by controlling the drive output of the high-voltage conversion module, which reduces the complexity of control, and at the same time solves the problem of automatic distribution of HV and LV power in magnetic integration.
  • the currents coupled to the secondary side of the transformers T1 and T2 are also the same, namely The output current Io1 of the secondary side high voltage conversion module and the output current Io2 of the secondary side rectifier module are equal.
  • the total output current is controlled by controlling the drive output of the primary side and the secondary side high voltage conversion module, and the output current of the secondary side rectifier module is automatically and the secondary side
  • the output currents of the high-voltage conversion modules are equal. Since the number of turns of the primary and secondary windings of the transformers T1 and T2 are the same, the output of the secondary high-voltage conversion module and the output of the secondary rectifier module are automatically balanced, and no additional current sharing is required.
  • the secondary side rectifier module is a diode rectifier, and the diode conduction is determined by the zero-crossing of the primary side current.
  • the secondary side rectifier module can omit the DC blocking capacitor, that is, the C2 capacitor is not needed in the secondary side rectifier module.
  • the primary winding W5 of transformer T2 is short-circuited, and the input voltage is applied to the resonant cavity and the primary winding W1 of transformer T1.
  • the cavity parameters and resonance points are the same.
  • the resonance points of single-phase and three-phase input are the following formulas 1 shown;
  • the resonance point remains unchanged.
  • the gain of a single transformer T1 is still the same, compared to three-phase Input, the single-phase power is reduced by half, and the primary current is the same.
  • the turns ratio of the transformer T1 and the design of the winding wire diameter are the same, which will not cause over-design due to single-phase and three-phase compatibility.
  • Inverter mode (inverter mode is not divided into three-phase or single-phase grid):
  • the switch K is closed, that is, the winding W5 of the transformer T2 is short-circuited.
  • HV is the input terminal of energy
  • the output side is Vin, which is opposite to the charging mode.
  • the primary side is the energy output side
  • the secondary HV is the energy input side.
  • Control mode The controller realizes the control of energy output by driving the power switches Q1—Q4 of the primary side conversion module and the power switches Q5—Q8 of the secondary side high voltage conversion module.
  • the primary-side conversion modules Q1 and Q4 are driven in the same way and both have a 50% duty cycle; Q2 and Q3 are driven in the same way and both have a 50% duty cycle.
  • the drives of Q1, Q4 and Q2, Q3 are completely opposite, as shown in Figure 6. Show.
  • Q5 and Q8 are driven in the same way, both are 50% duty cycle; Q6 and Q7 are driven the same, both are 50% duty cycle, Q5, Q8 and Q6, Q7 are driven completely opposite, as shown in Figure 7. .
  • the 50% duty cycle mentioned above is specifically implemented in order to prevent the upper and lower switch tubes of a pair of bridge arms from being turned on at the same time, and a dead time needs to be subtracted.
  • the 50% duty cycle It is collectively referred to as including dead time.
  • the control of the LV side is the same as the charging mode by controlling the duty cycle of the switching tube Q11 to stabilize the voltage and current of the LV.
  • the turn-off edge of Q11 is aligned with the turn-off edge of the switches Q5-Q8, as shown in Figure 16.
  • the controller can disconnect Q11 to achieve LV no output, and the energy is only from the HV side of the high-voltage battery to the primary side.
  • DCDC mode In the topology diagram of Figure 3, when working in DCDC mode, energy flows from the high-voltage HV side to the LV side. As shown in Figure 20, the relay K is closed, the switch Q11 remains normally on, and Q5-Q8 work in the PWM mode , The drive is shown in Figure 17, the controller realizes the stable LV voltage by adjusting the duty cycle of the switching tubes Q5-Q8.
  • FIG. 10 shows the simulation results of the respective output currents and total output currents of the first and second conversion modules on the secondary side. It can be seen from Table 2 that there is almost no deviation between the output of the secondary-side high-voltage conversion module and the output of the secondary-side rectifier module, which proves the feasibility of this control method.
  • Figure 11 shows the output current of the primary side conversion module, the bridge arm midpoint voltage of the primary side conversion module, and the secondary side first and second conversion modules (that is, the secondary side high voltage conversion module and the secondary side rectifier module) bridge arm midpoint voltage waveforms Control chart.
  • the present invention also discloses a control method of a compatible high-power dual-end output vehicle-mounted charger.
  • the charger adopts the above-mentioned compatible high-power dual-end output vehicle-mounted charger;
  • control Q5-Q8 in the secondary-side high-voltage conversion module to work in PWM mode (duty cycle adjustment mode) according to the sampled low-voltage output voltage (VoLV) and low-voltage output current (IoLV) and the reference Ref operation.
  • PWM mode duty cycle adjustment mode
  • ⁇ the timing difference between the primary side conversion module and the secondary side high voltage conversion module, and ⁇ is leading in the charging mode, and ⁇ is lagging in the inverter mode.
  • the controller controls the lead or lag of the timing difference ⁇ ; the lead of the timing difference ⁇ can increase the gain of the charger and increase the output power of the secondary side high voltage conversion module; the lag of the timing difference ⁇ can reduce the gain of the charger and reduce the secondary The output power of the side high voltage conversion module.
  • the timing difference ⁇ has left and right shifts, as follows: the secondary side high-voltage conversion module drives the primary side conversion module to drive to the right, and the secondary side high-voltage conversion module drives the primary side conversion module to lag behind. Move left.
  • the timing difference ⁇ leading can increase the gain of the charger and increase the output power of the secondary side high voltage conversion module; the timing difference ⁇ lag can reduce the gain of the charger and reduce the output power of the secondary side high voltage conversion module.
  • the fifth power switch Q5 and the eighth power switch Q8 in the secondary side high voltage conversion module are turned on, corresponding to the tenth power switch Q10 being turned on, and the sixth power switch Q6 and the seventh power switch Q7 are turned on.
  • Turning on corresponds to the turning on of the ninth power switch Q9, the switching period of the eleventh power switch Q11 is twice the switching period of the fifth power switch Q5, the turn-off edge of the eleventh power switch Q11 and the turn-off edge of the fifth power switch Q5 It is aligned with the turn-off edge of the sixth power switch Q6.
  • the eleventh power switch Q11 maintains a normally-on state.
  • the PWM Generator module that drives Q5 to Q8 and the PWM Generator module that drives Q11. This connection serves to synchronize the turn-off edges of Q11 with the turn-off edges of Q5 and Q6.
  • Q9 and Q10 are used as synchronous rectification functions, as diodes, Q9 and Q10 are turned on and correspond to Q5-Q8. Stabilize the LV output voltage by controlling the duty cycle of Q11.
  • the controller includes a collector that collects the input voltage (Vin) of the primary-side conversion module, and a collector that collects the high-voltage output voltage (VoHV) and the high-voltage output current (IoHV) common to the secondary-side high-voltage conversion module and the secondary-side rectifier module; A collector for low-voltage output voltage (VoLV) and low-voltage output current (IoLV) of the secondary-side low-voltage conversion module; in the charging mode, the controller samples and calibrates the low-voltage output voltage (VoLV) and the low-voltage output current (IoLV) respectively , And obtain the output power through Power Calculation; use the sampled and calibrated low-voltage output voltage (VoLV) to perform the difference calculation with the low-voltage output voltage reference value (VrefLV), and perform loop compensation for the difference between the two.
  • Vin input voltage
  • VoHV high-voltage output voltage
  • IoHV high-voltage output current
  • VrefLV low-
  • the obtained compensation value and the preset current loop preset value (IsetLV) are calculated to be small, and the small value is used as the current loop reference value (IrefLV), and then the low voltage output current (IoLV) after sampling and calibration is performed Difference calculation, loop compensation calculation is performed on the difference between the two to obtain the duty cycle, and then the eleventh power switch Q11 is driven by the PWM operation (PWM Generator); the high-voltage output current after sampling and calibration (IoHV ) Perform difference calculation with the high-voltage output current reference value (IrefHV), perform loop compensation on the difference between the two, and take the smaller calculation between the obtained compensation value and the preset voltage loop preset value (VsetHV), and take The small value is used as the voltage loop reference value (VrefHV), and then the difference calculation is performed with the sampled and calibrated high-voltage output current (VoHV), and the difference between the two is subjected to loop compensation, and the obtained loop compensation value is compared with the The output power calculation obtains the timing difference
  • the controller moves to the right Increase the timing difference ⁇ to increase the gain.
  • the controller Move to the left to increase the timing difference ⁇ and decrease the gain.
  • the controller includes a collector that collects the input voltage (Vin) of the primary-side conversion module, and a collector that collects the high-voltage output voltage (VoHV) and the high-voltage output current (IoHV) common to the secondary-side high-voltage conversion module and the secondary-side rectifier module; The collector of the low-voltage output voltage (VoLV) and low-voltage output current (IoLV) of the secondary side low-voltage conversion module; in the inverter mode, the controller samples and sets the low-voltage output voltage (VoLV) and the low-voltage output current (IoLV) respectively.
  • the output power is obtained by power calculation (Power Calculation); the difference calculation between the sampled and calibrated low-voltage output voltage (VoLV) and the low-voltage output voltage reference value (VrefLV) is performed, and the difference between the two is subjected to loop compensation ,
  • the obtained compensation value and the pre-set current loop preset value (IsetLV) are calculated to be small, and the small value is used as the current loop reference value (IrefLV), and then the low voltage output current after sampling and calibration (IoLV) Perform difference calculation, perform loop compensation calculation on the difference between the two to obtain the duty cycle, and then drive the eleventh power switch Q11 through the PWM operation (PWM Generator); use the sampled and calibrated input voltage (Vin ) And the voltage reference value (VrefVo) for difference calculation, loop compensation for the difference between the two, and the obtained compensation value and the output power to calculate the timing difference ⁇ , and then through the PWM operation (PWM Generator) Drive the power switches in the primary side conversion module and the secondary side high voltage conversion
  • timing difference ⁇ between the primary side conversion module and the secondary side high voltage conversion module drive that is, there is a timing difference ⁇ between the primary side conversion modules Q1, Q4 and the secondary side high voltage conversion modules Q5, Q8, the primary side conversion modules Q2, Q3 and
  • a timing difference ⁇ between the secondary high-voltage conversion modules Q6 and Q7 as shown in Figs. 8 and 12.
  • the timing difference ⁇ has left and right shifts, as follows: the secondary side high-voltage conversion module drives the primary side conversion module to drive to the right, and the secondary side high-voltage conversion module drives the primary side conversion module to lag behind. Move left. In the inverter mode, shifting ⁇ to the right can reduce the gain (that is, reducing the output power of the primary conversion module), and shifting ⁇ to the left can increase the gain (that is, increasing the output power of the primary conversion module).
  • the significance is that since the inductance Lr1, Lr2 and the capacitor C1 exist in the primary side conversion module, the two form a network that can change the equivalent reactance with the change of the switching period.
  • the mathematical expression is:
  • Z(fs) is the change of equivalent reactance of Lr1, Lr2 and C1 as fs changes.
  • fs is controlled to change Z(fs), and then the optimal matching characteristic is obtained.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Transportation (AREA)
  • Mechanical Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

本发明公开了一种兼容型大功率双端输出车载充电机及其控制方法，充电机包括原边转换模块、第一变压器T1、副边高压转换模块和副边低压转换模块、第二变压器T2和副边整流模块，第二变压器T2原边绕组W5与第一变压器T1原边绕组W1串联后接原边转换模块的输出端，第二变压器T2副边绕组W6接副边整流模块输入端，副边整流模块输出端与副边高压转换模块输出端并联，第二变压器T2原边或副边设置切换开关K；根据原边转换模块的母线电压控制切换开关K将第二变压器T2和副边整流模块投入运行或退出运行；本发明解决了大功率车载OBC后级DCDC多路并联均流的问题以及兼容单相和三相输入电压的问题；具有器件数量少、简单、容易实现等优点。

Description

一种兼容型大功率双端输出车载充电机及其控制方法 技术领域
本发明属于电源技术领域，具体涉及一种兼容型大功率双端输出车载充电机及其控制方法。
背景技术
随着社会的发展，环境污染和能源紧缺问题得到越来越多的关注，大力发展新能源汽车是解决上述两大问题的一个有效途径。随着新能源汽车技术的发展，续航里程越来越高，动力电池的容量的要求越来越高，对电池充电时间也要求越来越短，使得对车载充电机(简称：OBC)的功率急需提升。当前大功率的OBC都是采用多路并联的设计，如图1所示。在当前的设计中，由于元器件的参数都有容差，会出现不均流的问题，导致每一路的设计都要留比较的余量以及增加额外的硬件处理保证均流，造成过设计，使得成本过高。此外，当前OBC的设计组成如图2所示，由PFC和DCDC两级串联组成，大功率输出都是基于输入交流电压为三相的场景，而车载OBC往往都需要兼容单相和三相输入。在三相输入，PFC输出电压通常为800V,即DCDC的输入电压为800V；在单相输入，因为DCDC的变压器匝比是固定的，需要在单相输入时将PFC输入电压也升到800V，会增大PFC的损耗，导致效率低。
因此，如何设计一种在不增加额外的硬件成本下解决大功率的场景多路并联均流的问题，解决单相输入和三相输入不同母线电压DCDC变压器原副边匝比兼容的问题，是业界亟待解决的技术问题。
发明内容
为了解决现有技术中存在的上述缺陷，本发明提出一种兼容型大功率双端输出车载充电机及其控制方法。
本发明采用的技术方案是设计一种兼容型大功率双端输出车载充电机，包括依次连接原边转换模块、第一变压器T1、副边高压转换模块、控制器、连接第一变压器T1副边绕组的副边低压转换模块，以及第二变压器T2和副 边整流模块，其中所述第二变压器T2的原边绕组W5与第一变压器T1的原边绕组W1串联后连接原边转换模块的输出端，所述第二变压器T2的副边绕组W6连接所述副边整流模块的输入端，副边整流模块的输出端与副边高压转换模块的输出端并联，所述第二变压器T2的原边或副边设置切换开关K；控制器根据原边转换模块的母线电压控制切换开关K将第二变压器T2和副边整流模块投入运行或退出运行。
所述切换开关K可以并联在所述第二变压器T2原边绕组W5的两端。
所述切换开关K也可以并联在所述第二变压器T2副边绕组W6的两端。
所述充电机包括充电模式、逆变模式、DCDC模式；在充电模式中所述母线电压高于阈值M时，控制器控制开关K断开；在充电模式中所述母线电压不高于阈值M时，控制器控制开关K闭合；在逆变模式中，控制器控制开关K闭合；在DCDC模式中，控制器控制开关K闭合。
所述阈值M为600伏。
所述第一变压器T1原边绕组W1与第二副边绕组W2的比值和第二变压器T2原边绕组W5与副边绕组W6的比值相等；第一变压器T1原边绕组W1与二变压器T2原边绕组W5的匝数相等、线径相等；第一变压器T1第二副边绕组W2与二变压器T2副边绕组W6的匝数相等、线径相等。
所述副边高压转换模块中的功率开关采用有源器件，所述副边整流模块中的功率开关采用有无源器件。
所述副边整流模块采用桥式整流模块，包括第一二极管D1、第二二极管D2、第三二极管D2、第四二极管D4。
所述原边转换模块采用全桥结构，包括第一功率开关Q1、第二功率开关Q2、第三功率开关Q3、第四功率开关Q4；所述副边高压转换模块采用全桥结构，包括第五功率开关Q5、第六功率开关Q6、第七功率开关Q7、第八功率开关Q8；其中第五功率开关Q5和第七功率开关Q7为一对桥臂，第六功率开关Q6和第八功率开关Q8为一对桥臂，并且第五功率开关Q5和第六功率开关Q6为上桥臂，第七功率开关Q7和第八功率开关Q8为下桥臂。
所述第一变压器T1包括第二副边绕组W2、第三副边绕组W3和第四副边绕组W4，所述第二副边绕组W2连接所述副边高压转换模块；所述副边低压 转换模块包括第九功率开关Q9、第十功率开关Q10、第十一功率开关Q11；所述第九功率开关Q9的漏极连接第四副边绕组W4的同名端，所述第十功率开关Q10的漏极连接第三副边绕组W3的异名端，第四副边绕组W4的异名端与第三副边绕组W3同名端连接后串联第十一功率开关Q11和输出电感L0、之后连接副边低压转换模块的正极输出端，第九功率开关Q9和第十功率开关Q10的源极接地。
所述第一变压器T1的第二副边绕组W2通过隔直电容C2后连接所述副边高压转换模块；所述第二变压器T2的副边绕组W6直接连接所述副边整流模块。
所述第一变压器T1的原边绕组与第二变压器T2的原边绕组之间串接谐振电容C1。
所述第一变压器T1的原边绕组串接第一谐振电感Lr1。
所述第二变压器T2的原边绕组串接第二谐振电感Lr2。
所述第一变压器T1和第二变压器T2集成在同一个磁芯上。
本发明还设计了一种兼容型大功率双端输出车载充电机的控制方法，所述充电机采用上述的兼容型大功率双端输出车载充电机；驱动所述原边转换模块中功率开关的控制信号与驱动副边高压转换模块中功率开关的控制信号之间存在时序差Φ，所述控制器控制时序差Φ的超前或滞后；时序差Φ超前可增大充电机的增益，增大副边高压转换模块的输出功率；时序差Φ滞后可降低充电机的增益，减小副边高压转换模块的输出功率。
在充电模式和逆变模式中，所述副边高压转换模块中第五功率开关Q5和第八功率开关Q8导通对应第十功率开关Q10导通，第六功率开关Q6和第七功率开关Q7导通对应第九功率开关Q9导通，第十一功率开关Q11开关周期是第五功率开关Q5开关周期的2倍，第十一功率开关Q11的关断沿与第五功率开关Q5关断沿和第六功率开关Q6关断沿对齐；在DCDC模式中，第十一功率开关Q11保持常通状态。
所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边高压转换模块和副边整流模块共同的高压输出电压(VoHV)和高压输出电流 (IoHV)的采集器；采集副边低压转换模块低压输出电压(VoLV)和低压输出电流(IoLV)的采集器；在充电模式中，所述控制器对低压输出电压(VoLV)和低压输出电流(IoLV)分别进行采样定标、并通过功率运算(Power Calculation)得到输出功率；用采样定标后的低压输出电压(VoLV)同低压输出电压基准值(VrefLV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电流环预设值(IsetLV)做取小运算、将取其小值作为电流环基准值(IrefLV)，然后与采样定标后的低压输出电流(IoLV)进行差值运算，对两者的差值进行环路补偿运算得出占空比，再通过PWM运算(PWM Generator)驱动所述第十一功率开关Q11；用采样定标后的高压输出电流(IoHV)同高压输出电流基准值(IrefHV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电压环预设值(VsetHV)做取小运算、将取其小值作为电压环基准值(VrefHV)，然后与采样定标后的高压输出电流(VoHV)进行差值运算，对两者的差值进行环路补偿，用所得环路补偿值与所述输出功率运算得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边转换模块中的功率开关。
所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边高压转换模块和副边整流模块共同的高压输出电压(VoHV)和高压输出电流(IoHV)的采集器；采集副边低压转换模块低压输出电压(VoLV)和低压输出电流(IoLV)的采集器；在逆变模式中，所述控制器对低压输出电压(VoLV)和低压输出电流(IoLV)分别进行采样定标、并通过功率运算(Power Calculation)得到输出功率；用采样定标后的低压输出电压(VoLV)同低压输出电压基准值(VrefLV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电流环预设值(IsetLV)做取小运算、将取其小值作为电流环基准值(IrefLV)，然后与采样定标后的低压输出电流(IoLV)进行差值运算，对两者的差值进行环路补偿运算得出占空比，再通过PWM运算(PWM Generator)驱动所述第十一功率开关Q11；用采样定标后的输入电压(Vin)与电压基准值(VrefVo)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与所述输出功率运算得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边高压转换模块中的功率 开关。
本发明提供的技术方案的有益效果是：
本发明在不增加成本的情况下解决了大功率车载OBC后级DCDC多路并联均流的问题以及兼容单相和三相输入电压的问题；具有器件数量少、简单、容易实现等优点。
附图说明
下面结合实施例和附图对本发明进行详细说明，其中：
图1是现有技术中多路DCDC并联运行的原理框图；
图2是充电机原理框图；
图3是本发明串联一个谐振电感的电路图；
图4是本发明串联两个谐振电感的电路图；
图5是本发明第一和第二变压器合并的电路图；
图6是原边转换模块控制波形图；
图7是副边高压转换模块控制波形图；
图8是原边转换模块和副边高压转换模块控制波时序差对照图；
图9是副边高压转换模块控制波形与桥臂中点电压波形对照图；
图10是副边第一和第二转换模块各自输出电流和总输出电流波形对照图；
图11是原边转换模块输出电流、原边转换模块桥臂中点电压、副边第一和第二换模块桥臂中点电压波形对照图；
图12是原边转换模块和副边高压转换模块控制波时序差波形示意图；
图13是接单相电网时第二转换模块桥臂中点电压V_EF、第一转换模块桥臂中点电压V_CD、原边转换模块输出电压V_AB、原边转换模块输出电流Ip、充电机输出电流IoHV的波形对照图；
图14是充电模式时控制器控制原理框图；
图15是逆变模式和DCDC模式时控制器控制原理框图；
图16是充电模式中副边高、低压转换模块控制波形对照图；
图17是DCDC模式中低压转换模块控制波形对照图；
图18是充电模式能力流动方向示意图；
图19是逆变模式能力流动方向示意图；
图20是DCDC模式能力流动方向示意图。
具体实施方式
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明作进一步详细说明。应当理解，此处所描述的具体实施例仅仅用于解释本发明，并不用于限定本发明。
本发明公开了一种兼容型大功率双端输出车载充电机，包括依次连接原边转换模块、第一变压器T1、副边高压转换模块、控制器、连接第一变压器T1副边绕组的副边低压转换模块，以及第二变压器T2和副边整流模块，其中所述第二变压器T2的原边绕组W5与第一变压器T1的原边绕组W1串联后连接原边转换模块的输出端，所述第二变压器T2的副边绕组W6连接所述副边整流模块的输入端，副边整流模块的输出端与副边高压转换模块的输出端并联，所述第二变压器T2的原边或副边设置切换开关K；控制器根据原边转换模块的母线电压控制切换开关K将第二变压器T2和副边整流模块投入运行或退出运行。
参看图2示出的充电机原理框图，在较佳实施例中充电机的前端依次连接交流输入端、EMI滤波器、PFC电路构成，为DCDC电路提供直流电源。交流输入端连接外部不同输入电网，可以是三相电网，也可以是单相电网。本申请中副边高压转换模块连接高压动力电池，副边低压转换模块连接低压电池和整车用电设备。
在较佳实施例中(参看图3)，所述切换开关K可以并联在所述第二变压器T2原边绕组W5的两端。在另一些实施例中，所述切换开关K并联在所述第二变压器T2副边绕组W6的两端(未给出电路图)。
所述充电机包括充电模式、逆变模式、DCDC模式；其中充电模式(参看图18)是指从能量从交流电网传输给高压动力电池和低压侧电池和整车用电设备，逆变模式(参看图19)是指能量取自高压动力电池，逆变出交流电给到用电器同时给到低压侧电池和用电设备，DCDC模式(参看图20)是指能 量取自高压动力电池给到整车低压侧低压电池和整车用电器。在充电模式中所述母线电压高于阈值M时，控制器控制开关K断开；在充电模式中所述母线电压不高于阈值M时，控制器控制开关K闭合；在逆变模式中，控制器控制开关K闭合；在DCDC模式中，控制器控制开关K闭合。所述开关K采用双向开关或继电器中的一种。
在较佳实施例中所述阈值M为600伏。本发明应用在充电机中时，母线电压高于阈值M，代表充电机连接三相电网。母线电压不高于阈值M，代表充电机连接单相电网。
在较佳实施例中，所述第一变压器T1原边绕组W1与第二副边绕组W2的比值和第二变压器T2原边绕组W5与副边绕组W6的比值相等；第一变压器T1原边绕组W1与二变压器T2原边绕组W5的匝数相等、线径相等；第一变压器T1第二副边绕组W2与二变压器T2副边绕组W6的匝数相等、线径相等。
所述副边高压转换模块中的功率开关采用有源器件，所述副边整流模块中的功率开关采用有无源器件。
在较佳实施例中，所述副边整流模块采用桥式整流模块，包括第一二极管D1、第二二极管D2、第三二极管D2、第四二极管D4。
参看图3示出的较佳实施例，所述原边转换模块采用全桥结构，包括第一功率开关Q1、第二功率开关Q2、第三功率开关Q3、第四功率开关Q4；所述副边高压转换模块采用全桥结构，包括第五功率开关Q5、第六功率开关Q6、第七功率开关Q7、第八功率开关Q8；其中第五功率开关Q5和第七功率开关Q7为一对桥臂，第六功率开关Q6和第八功率开关Q8为一对桥臂，并且第五功率开关Q5和第六功率开关Q6为上桥臂，第七功率开关Q7和第八功率开关Q8为下桥臂。副边高压转换模块控制信号和输入电压如图9所示。所述第一功率开关Q1、第二功率开关Q2、第三功率开关Q3、第四功率开关Q4、第五功率开关Q5、第六功率开关Q6、第七功率开关Q7、第八功率开关Q8采用MOSFET、SiC MOSFET、IGBT并联二极管、GAN HEMT中的一种。
参看图3示出的较佳实施例，所述第一变压器T1包括第二副边绕组W2、第三副边绕组W3和第四副边绕组W4，所述第二副边绕组W2连接所述副边高压转换模块；所述副边低压转换模块包括第九功率开关Q9、第十功率开关 Q10、第十一功率开关Q11；所述第九功率开关Q9的漏极连接第四副边绕组W4的同名端，所述第十功率开关Q10的漏极连接第三副边绕组W3的异名端，第四副边绕组W4的异名端与第三副边绕组W3同名端连接后串联第十一功率开关Q11和输出电感L0、之后连接副边低压转换模块的正极输出端，第九功率开关Q9和第十功率开关Q10的源极接地。
在较佳实施例中，所述第一变压器T1的第二副边绕组W2通过隔直电容C2后连接所述副边高压转换模块；所述第二变压器T2的副边绕组W6直接连接所述副边整流模块。如图3、4、5所示。
在较佳实施例中，所述第一变压器T1的原边绕组与第二变压器T2的原边绕组之间串接谐振电容C1。如图3、4、5所示。
参看图3示出的实施例，所述第一变压器T1的原边绕组串接第一谐振电感Lr1。即整个原边侧只串联一个振电感。
参看图4示出的实施例，所述第二变压器T2的原边绕组串接第二谐振电感Lr2。即整个原边侧串联两个振电感。
参看图5示出的实施例，所述第一变压器T1和第二变压器T2集成在同一个磁芯上。即第一变压器T1和第二变压器T2可以分开设置，也可以合并设置。
需要指出，谐振电感设置在不同的实施例中有不同的设置方法，可以一个谐振电感，也可以是两个分立的谐振电感，还可以是集成在同一个磁芯。谐振电感可以是独立元件，也可以是变压器的漏感。
下面结合图3以本发明使用在充电机中为例，对本发明作说明。
充电模式连接三相电网：
三相输入时，母线电压为800V，即图3中的Vin＝800V，开关K断开，原边转换模块中Q1和Q3组成原边第一桥臂，桥臂中点A；Q2和Q4组成原边第二桥臂，桥臂中点B；谐振电感Lr1、变压器T1绕组W1、谐振电容C1、变压器T2绕组W5，谐振电容Lr2串联接在一起，一端接在原边第一桥臂中点A，另一端接在原边第二桥臂中点B，组成变压器T1和T2原边串联的结构。副边高压转换模块中Q5和Q7组成副边第一桥臂，桥臂中点C；Q6和Q8组成副边第二桥臂，桥臂中点D；变压器T1绕组W2与隔直电容C2串联，一端接在 副边第一桥臂中点C，另一端接在副边第二桥臂中点D，并联输出电容C4，组成副边高压转换模块输出HV1；副边整流模块中D1和D3组成副边第三桥臂，桥臂中点E；D2和D4组成副边第四桥臂，桥臂中点F；变压器T2绕组W6一端接在副边第三桥臂中点E，另一端接在副边第四桥臂中点F，并联输出电容C5组成副边整流模块输出HV2，HV1输出正端和HV2输出正端接在一起，HV1输出负端和HV2输出负端接在一起，HV1和HV2并联组成高压HV输出。按照上述连接，变压器T1绕组W2和变压器T2绕组W6组成并联连接。其中变压器T1绕组W1：W2变压器T2绕组W5：W6的匝比以及匝数，绕线线径都是相同的。
副边低压转换模块输出：变压器T1绕组W3、W4串联组成中间抽头的形式，第十功率开关Q10、D5、Lo、C3组成buck电路，输出LV电压；通过控制Q11占空比稳定LV输出电压。
控制方式：图3的拓扑结构中，控制器通过驱动原边转换模块中功率开关Q1—Q4、以及副边高压转换模块中功率开关Q5—Q8实现对HV电压和电流的控制，具体的，原边转换模块：Q1和Q4驱动一致，都是50％占空比；Q2和Q3驱动一致，都是50％占空比，Q1、Q4和Q2、Q3驱动完全相反，如图6所示；副边高压转换模块：Q5和Q8驱动一致，都是50％占空比；Q6和Q7驱动一致，都是50％占空比，Q5、Q8和Q6、Q7驱动完全相反，如图7所示；上述所说的50％占空比在具体实施的为防止一对桥臂的上下两个开关管同时导通出现短路，都需要减去一个死区时间，在这所说的50％占空比是统称包含有死区时间。
副边低压转换模块(LV)输出控制，集成变压器T1的电压由副边高压转换模块输出钳位，变压器电压翻转由Q5/Q8、Q6/Q7的驱动决定，如图9所示。副边低压转换模块同步整流Q9和Q10的导通由副边高压转换模块输出的功率开关决定。即Q5/Q8导通对应同步整流Q10导通，Q6/Q7导通对应Q9导通。在变压器电压翻转时，伴随能量的波动，所以LV输出的功率开关Q11的驱动必须在Q5/Q8或者Q6/Q7导通时完成一个开关周期,即Q11开关周期是Q5/Q6/Q7/Q8的2倍，如图16所示，Q11的关断沿和高压第一路的开关Q5-Q8关断沿对齐。通过控制Q11的占空比稳定LV输出的电压和电流。采用该控制 方式，通过控制高压转换模块输出的驱动实现了均流，降低了控制的复杂性，同时解决了磁集成中HV和LV功率自动分配的问题。
并联均流原理：两个变压器T1原副边匝比W1：W2和T2原副边匝比W5：W6相等，且变压器原边是串联的，输入电压Vin的电压加在谐振腔、变压器T1原边W1和变压器T2原边W5上，流过两个变压器原边绕组的电流是一样的,即使谐振参数Lr，谐振电容Cr参数出现偏差，流过两个变压器的电流都是一样的。两个变压器原副边匝比，匝数都是一样的，且两路输出是短接在一起的，两路的输出电压一样，则变压器T1和T2耦合到副边的电流也是一样的，即副边高压转换模块输出电流Io1和副边整流模块输出电流Io2是相等的，通过控制原边和副边高压转换模块输出的驱动控制总的输出电流，而副边整流模块输出电流自动和副边高压转换模块输出电流相等。由于变压器T1和T2的原副边绕组匝数一样，副边高压转换模块输出和副边整流模块输出的电流自动均衡，不需要额外作均流处理。此外，副边整流模块是二极管整流，二极管导通由原边电流过零决定，不存在副边高压转换模块是有源器件由于驱动的轻微差异导致的变压器偏磁的问题，在本控制方式上副边整流模块可以省略隔直电容，即C2电容在副边整流模块是不需要的。
充电模式连接单相电网：
单相输入时，母线电压降低一半，即Vin＝400V，开关K闭合。原边转换模块中，变压器T2原边绕组W5短路，输入电压加在谐振腔和变压器T1原边绕组W1上，谐振腔参数以及谐振点一样，单相和三相输入的谐振点都是如下公式1所示；
Figure PCTCN2020101135-appb-000001
由于输入电压降低了一半，同时原边转换模块的变压器也减少一半(短路T2原边绕组W5)，谐振点不变，在单相输入时，单个变压器T1增益还是一样的，则相比三相输入，单相的功率减少一半，同时原边电流也是一样的，变压器T1的匝比以及绕线线径的设计都是一样的，不会造成因为单相和三相兼容造成过设计。
逆变模式(逆变模式不分三相或单相电网)：
在图3的拓扑图中，开关K闭合，即变压器T2绕组W5短路。HV为能量的输入端，输出侧为Vin，与充电模式相反。原边侧为能量输出侧，副边HV为能量输入侧。
控制方式：控制器通过驱动原边转换模块功率开关Q1—Q4，以及副边高压转换模块功率开关Q5—Q8实现对能量输出的控制。具体地，原边转换模块Q1和Q4驱动一致，都是50％占空比；Q2和Q3驱动一致，都是50％占空比，Q1、Q4和Q2、Q3驱动完全相反，如图6所示。副边高压转换模块中Q5和Q8驱动一致，都是50％占空比；Q6和Q7驱动一致，都是50％占空比，Q5、Q8和Q6、Q7驱动完全相反，如图7所示。上述所说的50％占空比在具体实施的为防止一对桥臂的上下两个开关管同时导通出现短路，都需要减去一个死区时间，在这所说的50％占空比是统称包含有死区时间。
逆变模式中LV侧的控制和充电模式一样都是通过控制开关管Q11的占空比实现稳定LV的电压和电流。Q11的关断沿和开关管Q5-Q8关断沿对齐，如图16所示。
在逆变模式可以通过控制器断开Q11实现LV不输出，能量只从高压电池HV侧到原边侧。
DCDC模式：在图3拓扑图中，当工作在DCDC模式，能量从高压HV侧流向LV侧，如图20所示，继电器K闭合，开关管Q11保持常通状态，Q5-Q8工作在PWM模式，驱动如图17所示，控制器通过调整开关管Q5-Q8的占空比实现稳定LV电压。
具体实施举例：
运用本发明的硬件框架和控制方式；工作在三相时，Vin＝800V,高压输出300V，HV输出功率21KW，即：高压HV输出总电流70A，低压LV输出功率为0，对比副边高压转换模块输出和副边整流模块输出的均流性。
表1：仿真参数
Figure PCTCN2020101135-appb-000002
仿真结果如表2所示，副边第一和第二转换模块各自输出电流和总输出电流波形对照如图10所示。从表2中可以看出，副边高压转换模块输出和副边整流模块输出的电流几乎无偏差，证明本控制方式的可行性。图11示出了原边转换模块输出电流、原边转换模块桥臂中点电压、副边第一和第二换模块(即副边高压转换模块和副边整流模块)桥臂中点电压波形对照图。
表2：仿真结果
项目 结果
副边高压转换模块输出电流(IoHV1) 35.019A
副边整流模块输出电流(IoHV2) 35.025A
副边输出总电流(IoHV) 70.045A
图13是接单相电网时，Vin＝400V，继电器K闭合将变压器T2原边绕组W5短路状态下第二转换模块桥臂中点电压V_EF、第一转换模块桥臂中点电压V_CD、原边转换模块输出电压V_AB、原边转换模块输出电流Ip、充电机输出电流IoHV的波形对照图。从图中可以看到：(1)在单相输入时高压输出总电流相比三相输入时降低一半，即：输出功率降低一半；(2)副边整流模块输出桥臂中点E、F电压V_EF为0，说明所有的输出功率来自副边高压转换模块输出；(3)单相和三相输入时原边电流Ip有效值是一样的。
表3：单相和三相输入电流对比
  三相输入 单相输入
原边电流Ip 45.42A 45.501A
本发明还公开了一种兼容型大功率双端输出车载充电机的控制方法，所述充电机采用上述的兼容型大功率双端输出车载充电机；驱动所述原边转换模块中功率开关的控制信号与驱动副边高压转换模块中功率开关的控制信号之间存在时序差Φ，如图8和图12所示。即原边转换模块中Q1、Q4和副边高压转换模块中Q5、Q8之间存在时序差Φ，原边转换模块中Q2、Q3和边第一转换模块中Q6、Q7之间存在时序差Φ。在DCDC模式中，控制根据根据采样到的低压输出电压(VoLV)和低压输出电流(IoLV)与参考Ref运算控制副边高压转换模块中Q5-Q8工作在PWM模式(调占空比的模式)。充电模式和逆变模式时调原边转换模块与副边高压转换模块之间的时序差Φ，并且充电模式时Φ是超前，逆变模式时Φ是滞后。所述控制器控制时序差Φ的超前或滞后；时序差Φ超前可增大充电机的增益，增大副边高压转换模块的输出功率；时序差Φ滞后可降低充电机的增益，减小副边高压转换模块的输出功率。以原边转换模块驱动为参考，时序差Φ有左移和右移，具体如下：副边高压转换模块驱动超前原边转换模块驱动为右移，副边高压转换模块驱动滞后原边转换模块驱动为左移。时序差Φ超前可增大充电机的增益，增大副边高压转换模块的输出功率；时序差Φ滞后可降低充电机的增益，减小副边高压转换模块的输出功率。
在充电模式和逆变模式中，所述副边高压转换模块中第五功率开关Q5和第八功率开关Q8导通对应第十功率开关Q10导通，第六功率开关Q6和第七功率开关Q7导通对应第九功率开关Q9导通，第十一功率开关Q11开关周期是第五功率开关Q5开关周期的2倍，第十一功率开关Q11的关断沿与第五功率开关Q5关断沿和第六功率开关Q6关断沿对齐。
在DCDC模式中，第十一功率开关Q11保持常通状态。参看图14或图15，驱动Q5至Q8的PWM Generator模块和驱动Q11的PWM Generator模块之间有连接，该连接起到让Q11的关断沿与Q5和Q6关断沿对齐的同步作用。充 电模式时Q9和Q10用作同步整流功能，当做二极管用，Q9和Q10导通和Q5-Q8对应的。通过控制Q11占空比稳定LV输出电压。
参看图14示出的充电模式时控制器控制原理框图。所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边高压转换模块和副边整流模块共同的高压输出电压(VoHV)和高压输出电流(IoHV)的采集器；采集副边低压转换模块低压输出电压(VoLV)和低压输出电流(IoLV)的采集器；在充电模式中，所述控制器对低压输出电压(VoLV)和低压输出电流(IoLV)分别进行采样定标、并通过功率运算(Power Calculation)得到输出功率；用采样定标后的低压输出电压(VoLV)同低压输出电压基准值(VrefLV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电流环预设值(IsetLV)做取小运算、将取其小值作为电流环基准值(IrefLV)，然后与采样定标后的低压输出电流(IoLV)进行差值运算，对两者的差值进行环路补偿运算得出占空比，再通过PWM运算(PWM Generator)驱动所述第十一功率开关Q11；用采样定标后的高压输出电流(IoHV)同高压输出电流基准值(IrefHV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电压环预设值(VsetHV)做取小运算、将取其小值作为电压环基准值(VrefHV)，然后与采样定标后的高压输出电流(VoHV)进行差值运算，对两者的差值进行环路补偿，用所得环路补偿值与所述输出功率运算得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边转换模块中的功率开关。
当控制器采样到的副边高压转换模块电流IoHV比实际设定的IrefHV小或者采样到的副边高压转换模块侧电压VoHV比设定的电压环基准值(VrefHV)小，控制器往右移动增大时序差Φ增大增益。反之，当控制器采样到的副边高压转换模块侧电流IoHV比设定的电流基准值refHV大或者采样到的副边高压转换模块侧电压VoHV比设定的电压环基准值VrefHV大，控制器往左移动增大时序差Φ降低增益。
参看图15示出的逆变模式时控制器控制原理框图。所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边高压转换模块和副边整流模块共同的高压输出电压(VoHV)和高压输出电流(IoHV)的采集器；采集 副边低压转换模块低压输出电压(VoLV)和低压输出电流(IoLV)的采集器；在逆变模式中，所述控制器对低压输出电压(VoLV)和低压输出电流(IoLV)分别进行采样定标、并通过功率运算(Power Calculation)得到输出功率；用采样定标后的低压输出电压(VoLV)同低压输出电压基准值(VrefLV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电流环预设值(IsetLV)做取小运算、将取其小值作为电流环基准值(IrefLV)，然后与采样定标后的低压输出电流(IoLV)进行差值运算，对两者的差值进行环路补偿运算得出占空比，再通过PWM运算(PWM Generator)驱动所述第十一功率开关Q11；用采样定标后的输入电压(Vin)与电压基准值(VrefVo)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与所述输出功率运算得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边高压转换模块中的功率开关。
原边转换模块和副边高压转换模块驱动之间存在时序差Φ，即原边转换模块Q1、Q4和副边高压转换模块Q5、Q8之间存在时序差Φ，原边转换模块Q2、Q3和副边高压转换模块Q6、Q7之间存在时序差Φ，如图8和图12所示。以原边转换模块驱动为参考，时序差Φ有左移和右移，具体如下：副边高压转换模块驱动超前原边转换模块驱动为右移，副边高压转换模块驱动滞后原边转换模块驱动为左移。逆变模式Φ右移可以降低增益(即减小原边转换模块输出功率)，Φ左移可以增大增益(即增大原边转换模块输出功率)。
在控制Φ的同时，也控制原边转换模块和副边高压转换模块功率开关开关周期fs。其意义在于，由于原边转换模块中存在电感Lr1、Lr2和电容C1，两者组成一个可以随开关周期变化而等效电抗发生变化的网络，其数学表达式为：
Figure PCTCN2020101135-appb-000003
其中Z(fs)便是随着fs变化，Lr1、Lr2和C1等效电抗的变化。实际应用中，为了提高效率，避免无功能量过多，会通过控制fs，从而改变Z(fs)，进而得出最优匹配特性。
以上实施例仅为举例说明，非起限制作用。任何未脱离本申请精神与范 畴，而对其进行的等效修改或变更，均应包含于本申请的权利要求范围之中。

Claims (19)

  1. 一种兼容型大功率双端输出车载充电机，包括依次连接原边转换模块、第一变压器T1、副边高压转换模块，以及控制器，其特征在于：还包括连接第一变压器T1副边绕组的副边低压转换模块，以及第二变压器T2和副边整流模块，其中所述第二变压器T2的原边绕组W5与第一变压器T1的原边绕组W1串联后连接原边转换模块的输出端，所述第二变压器T2的副边绕组W6连接所述副边整流模块的输入端，副边整流模块的输出端与副边高压转换模块的输出端并联，所述第二变压器T2的原边或副边设置切换开关K；控制器根据原边转换模块的母线电压控制切换开关K将第二变压器T2和副边整流模块投入运行或退出运行。
  2. 如权利要求1所述的兼容型大功率双端输出车载充电机，其特征在于：所述切换开关K并联在所述第二变压器T2原边绕组W5的两端。
  3. 如权利要求1所述的兼容型大功率双端输出车载充电机，其特征在于：所述切换开关K并联在所述第二变压器T2副边绕组W6的两端。
  4. 如权利要求2或3所述的兼容型大功率双端输出车载充电机，其特征在于：所述充电机包括充电模式、逆变模式、DCDC模式；
    在充电模式中所述母线电压高于阈值M时，控制器控制开关K断开；
    在充电模式中所述母线电压不高于阈值M时，控制器控制开关K闭合；
    在逆变模式中，控制器控制开关K闭合；
    在DCDC模式中，控制器控制开关K闭合。
  5. 如权利要求4所述的适用于不同输入电网的充电机，其特征在于：所述阈值M为600伏。
  6. 如权利要求4所述的兼容型大功率双端输出车载充电机，其特征在于：所述第一变压器T1原边绕组W1与第二副边绕组W2的比值和第二变压器T2原边绕组W5与副边绕组W6的比值相等；第一变压器T1原边绕组W1与二变压器T2原边绕组W5的匝数相等、线径相等；第一变压器T1第二副边绕组W2与二变压器T2副边绕组W6的匝数相等、线径相等。
  7. 如权利要求1所述的兼容型大功率双端输出车载充电机，其特征在于：所述副边高压转换模块中的功率开关采用有源器件，所述副边整流模块 中的功率开关采用有无源器件。
  8. 如权利要求7所述的兼容型大功率双端输出车载充电机，其特征在于：所述副边整流模块采用桥式整流模块，包括第一二极管D1、第二二极管D2、第三二极管D2、第四二极管D4。
  9. 如权利要求7所述的兼容型大功率双端输出车载充电机，其特征在于：所述原边转换模块采用全桥结构，包括第一功率开关Q1、第二功率开关Q2、第三功率开关Q3、第四功率开关Q4；所述副边高压转换模块采用全桥结构，包括第五功率开关Q5、第六功率开关Q6、第七功率开关Q7、第八功率开关Q8；其中第五功率开关Q5和第七功率开关Q7为一对桥臂，第六功率开关Q6和第八功率开关Q8为一对桥臂，并且第五功率开关Q5和第六功率开关Q6为上桥臂，第七功率开关Q7和第八功率开关Q8为下桥臂。
  10. 如权利要求9所述的兼容型大功率双端输出车载充电机，其特征在于：所述第一变压器T1包括第二副边绕组W2、第三副边绕组W3和第四副边绕组W4，所述第二副边绕组W2连接所述副边高压转换模块；所述副边低压转换模块包括第九功率开关Q9、第十功率开关Q10、第十一功率开关Q11；所述第九功率开关Q9的漏极连接第四副边绕组W4的同名端，所述第十功率开关Q10的漏极连接第三副边绕组W3的异名端，第四副边绕组W4的异名端与第三副边绕组W3同名端连接后串联第十一功率开关Q11和输出电感L0、之后连接副边低压转换模块的正极输出端，第九功率开关Q9和第十功率开关Q10的源极接地。
  11. 如权利要求1所述的兼容型大功率双端输出车载充电机，其特征在于：所述第一变压器T1的第二副边绕组W2通过隔直电容C2后连接所述副边高压转换模块；所述第二变压器T2的副边绕组W6直接连接所述副边整流模块。
  12. 如权利要求1所述的兼容型大功率双端输出车载充电机，其特征在于：所述第一变压器T1的原边绕组与第二变压器T2的原边绕组之间串接谐振电容C1。
  13. 如权利要求1所述的兼容型大功率双端输出车载充电机，其特征在于：所述第一变压器T1的原边绕组串接第一谐振电感Lr1。
  14. 如权利要求11所述的兼容型大功率双端输出车载充电机，其特征在于：所述第二变压器T2的原边绕组串接第二谐振电感Lr2。
  15. 如权利要求1所述的兼容型大功率双端输出车载充电机，其特征在于：所述第一变压器T1和第二变压器T2集成在同一个磁芯上。
  16. 一种兼容型大功率双端输出车载充电机的控制方法，其特征在于：所述充电机采用权利要求1至15任一项所述的兼容型大功率双端输出车载充电机；驱动所述原边转换模块中功率开关的控制信号与驱动副边高压转换模块中功率开关的控制信号之间存在时序差Φ，所述控制器控制时序差Φ的超前或滞后；时序差Φ超前可增大充电机的增益，增大副边高压转换模块的输出功率；时序差Φ滞后可降低充电机的增益，减小副边高压转换模块的输出功率。
  17. 如权利要求16所述的兼容型大功率双端输出车载充电机，其特征在于：在充电模式和逆变模式中，所述副边高压转换模块中第五功率开关Q5和第八功率开关Q8导通对应第十功率开关Q10导通，第六功率开关Q6和第七功率开关Q7导通对应第九功率开关Q9导通，第十一功率开关Q11开关周期是第五功率开关Q5开关周期的2倍，第十一功率开关Q11的关断沿与第五功率开关Q5关断沿和第六功率开关Q6关断沿对齐；在DCDC模式中，第十一功率开关Q11保持常通状态。
  18. 如权利要求17所述的兼容型大功率双端输出车载充电机，其特征在于：所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边高压转换模块和副边整流模块共同的高压输出电压(VoHV)和高压输出电流(IoHV)的采集器；采集副边低压转换模块低压输出电压(VoLV)和低压输出电流(IoLV)的采集器；
    在充电模式中，所述控制器对低压输出电压(VoLV)和低压输出电流(IoLV)分别进行采样定标、并通过功率运算(Power Calculation)得到输出功率；用采样定标后的低压输出电压(VoLV)同低压输出电压基准值(VrefLV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电流环预设值(IsetLV)做取小运算、将取其小值作为电流环基准值(IrefLV)，然后与采样定标后的低压输出电流(IoLV)进行差值运算，对两者的差值进 行环路补偿运算得出占空比，再通过PWM运算(PWM Generator)驱动所述第十一功率开关Q11；用采样定标后的高压输出电流(IoHV)同高压输出电流基准值(IrefHV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电压环预设值(VsetHV)做取小运算、将取其小值作为电压环基准值(VrefHV)，然后与采样定标后的高压输出电流(VoHV)进行差值运算，对两者的差值进行环路补偿，用所得环路补偿值与所述输出功率运算得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边转换模块中的功率开关。
  19. 如权利要求17所述的兼容型大功率双端输出车载充电机，其特征在于：所述控制器包括采集原边转换模块输入电压(Vin)的采集器、采集副边高压转换模块和副边整流模块共同的高压输出电压(VoHV)和高压输出电流(IoHV)的采集器；采集副边低压转换模块低压输出电压(VoLV)和低压输出电流(IoLV)的采集器；
    在逆变模式中，所述控制器对低压输出电压(VoLV)和低压输出电流(IoLV)分别进行采样定标、并通过功率运算(Power Calculation)得到输出功率；用采样定标后的低压输出电压(VoLV)同低压输出电压基准值(VrefLV)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与预先设定的电流环预设值(IsetLV)做取小运算、将取其小值作为电流环基准值(IrefLV)，然后与采样定标后的低压输出电流(IoLV)进行差值运算，对两者的差值进行环路补偿运算得出占空比，再通过PWM运算(PWM Generator)驱动所述第十一功率开关Q11；用采样定标后的输入电压(Vin)与电压基准值(VrefVo)进行差值运算，对两者的差值进行环路补偿，将所得补偿值与所述输出功率运算得出所述时序差Φ、再通过PWM运算(PWM Generator)驱动所述原边转换模块和副边高压转换模块中的功率开关。
PCT/CN2020/101135 2020-05-14 2020-07-09 一种兼容型大功率双端输出车载充电机及其控制方法 WO2021227230A1 (zh)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
CN202010409356.6A CN111463878B (zh) 2020-05-14 2020-05-14 一种兼容型大功率双端输出车载充电机及其控制方法
CN202010409356.6 2020-05-14

Publications (1)

Publication Number Publication Date
WO2021227230A1 true WO2021227230A1 (zh) 2021-11-18

Family

ID=71681881

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/CN2020/101135 WO2021227230A1 (zh) 2020-05-14 2020-07-09 一种兼容型大功率双端输出车载充电机及其控制方法

Country Status (2)

Country Link
CN (1) CN111463878B (zh)
WO (1) WO2021227230A1 (zh)

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114537169A (zh) * 2022-03-28 2022-05-27 华人运通(山东)科技有限公司 大功率无线充电机及其车载端供电电路和控制方法
CN115514230A (zh) * 2022-08-31 2022-12-23 核工业西南物理研究院 一种高稳定度的大功率长脉冲高压电源及控制方法

Families Citing this family (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US11594973B2 (en) 2020-08-04 2023-02-28 Delta Electronics Inc. Multiple-port bidirectional converter and control method thereof
CN112134463B (zh) * 2020-08-31 2022-06-03 杭州中恒电气股份有限公司 全桥型llc谐振变换器及其谐振电流检测方法
CN112821483B (zh) * 2020-12-31 2024-01-05 维沃移动通信有限公司 充电器
CN113511084B (zh) * 2021-04-25 2023-06-09 深圳威迈斯新能源股份有限公司 一种输出端可进行串并联切换的车载充电机
CN113381607B (zh) * 2021-06-08 2022-09-02 哈尔滨工业大学 一种低成本高效率高变比dc/dc变换器
CN113472214B (zh) * 2021-07-30 2023-05-02 宁波吉利罗佑发动机零部件有限公司 一种车辆直流电压转换电路
CN113964923B (zh) * 2021-09-15 2024-04-19 许继电源有限公司 一种基于切档控制的大功率充电机装置及其切档控制方法
CN114221535B (zh) * 2021-12-21 2023-12-19 阳光电源股份有限公司 一种车载充电器、dcdc变换器及控制方法

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN106329940A (zh) * 2016-11-07 2017-01-11 江南大学 一种双变压器串并联结构全桥llc谐振变换器
CN106936184A (zh) * 2017-03-14 2017-07-07 深圳威迈斯电源有限公司 一种车载充电机和dcdc的集成电路
CN109808522A (zh) * 2017-11-14 2019-05-28 福特全球技术公司 用于车辆的复合双向集成充电器

Family Cites Families (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP4816908B2 (ja) * 2006-01-17 2011-11-16 サンケン電気株式会社 多出力スイッチング電源装置
CN105896998B (zh) * 2016-06-20 2018-08-24 杭州电子科技大学 一种隔离型双向有源全桥dc-dc变换器
CN107359682A (zh) * 2017-07-29 2017-11-17 深圳市国电赛思科技有限公司 一种双向充电与直流转换二合一的电源系统及其控制方法
CN108237943B (zh) * 2018-01-17 2019-05-17 深圳威迈斯新能源股份有限公司 一种双输出端口充电电路及其控制方法
CN109038736B (zh) * 2018-08-10 2019-10-18 深圳威迈斯新能源股份有限公司 一种充电电路移相控制方法
CN109167423A (zh) * 2018-09-12 2019-01-08 联合汽车电子有限公司 车载充电机与车载dcdc的电气集成装置
CN110040028A (zh) * 2019-04-15 2019-07-23 创驱（上海）新能源科技有限公司 一种车载充电机和dcdc转换装置的集成控制电路
CN110356269A (zh) * 2019-07-19 2019-10-22 宝能(广州)汽车研究院有限公司 集成dc/dc的双向车载充电机和电动汽车

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN106329940A (zh) * 2016-11-07 2017-01-11 江南大学 一种双变压器串并联结构全桥llc谐振变换器
CN106936184A (zh) * 2017-03-14 2017-07-07 深圳威迈斯电源有限公司 一种车载充电机和dcdc的集成电路
CN109808522A (zh) * 2017-11-14 2019-05-28 福特全球技术公司 用于车辆的复合双向集成充电器

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114537169A (zh) * 2022-03-28 2022-05-27 华人运通(山东)科技有限公司 大功率无线充电机及其车载端供电电路和控制方法
CN115514230A (zh) * 2022-08-31 2022-12-23 核工业西南物理研究院 一种高稳定度的大功率长脉冲高压电源及控制方法

Also Published As

Publication number Publication date
CN111463878B (zh) 2021-10-01
CN111463878A (zh) 2020-07-28

Similar Documents

Publication Publication Date Title
WO2021227230A1 (zh) 一种兼容型大功率双端输出车载充电机及其控制方法
CN108448913B (zh) 一种单级式基于交错并联无桥pfc电路和llc谐振的隔离型ac-dc变换器
WO2021227231A1 (zh) 一种适用于不同输入电网的dcdc架构及其控制方法
US11831236B2 (en) Bidirectional multiple-port power conversion system and method
EP2571154B1 (en) PV inverter with input parallel output series connected flyback converters feeding a fullbridge grid converter
CN109039121B (zh) 一种高频隔离型交直流变换电路及其控制方法
CN108988676B (zh) 一种单级式隔离型双向ac-dc变换器
CN108512256B (zh) 一种多功能车载充放电一体化系统
CN111355398A (zh) 一种集成dc/dc转换器的双向车载充电机电路
CN110061650B (zh) 单级式隔离型三相双向ac/dc变换器及控制方法
CN108900100A (zh) 一种单相高效高频隔离型整流器
CN113039712B (zh) 双向功率转换系统及控制方法
CN109742965A (zh) 一种单相交错并联三电平谐振式的高频隔离型ac-dc变换器
CN111245078A (zh) 集成dc/dc转换器的双向车载充电机电路
US11990835B2 (en) Power conversion system and control method
CN217087777U (zh) 一种宽范围谐振式软开关双向直流变换器
Xu et al. A Novel Phase-Shift Pulsewidth Modulation Method for Light-Load Bidirectional Resonant Converter
CN107171563B (zh) 紧调整输出的组合变流器
CN108306514A (zh) 一种燃料电池的dc-dc变换器
CN112202351A (zh) 一种宽范围软开关的单级式隔离型三相ac/dc整流器
TWI794329B (zh) Ｄｃ轉ａｃ功率轉換器及適合被使用在ｄｃ轉ａｃ功率轉換器之隔絕ｄｃ轉ｄｃ轉換器
CN208046459U (zh) 一种燃料电池的dc-dc变换器
WO2023029885A1 (zh) 发射端、接收端、动态无线供电系统及电动汽车
CN115765514A (zh) 一种可双向变换的三相隔离型变换器及其控制方法
CN107612340A (zh) 一种低电压应力隔离全桥变换器电路装置

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 20935448

Country of ref document: EP

Kind code of ref document: A1

NENP Non-entry into the national phase

Ref country code: DE

122 Ep: pct application non-entry in european phase

Ref document number: 20935448

Country of ref document: EP

Kind code of ref document: A1