WO2021174582A1 - 无位置传感器的永磁同步电机控制方法及汽车动力系统 - Google Patents

无位置传感器的永磁同步电机控制方法及汽车动力系统 Download PDF

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WO2021174582A1
WO2021174582A1 PCT/CN2020/079487 CN2020079487W WO2021174582A1 WO 2021174582 A1 WO2021174582 A1 WO 2021174582A1 CN 2020079487 W CN2020079487 W CN 2020079487W WO 2021174582 A1 WO2021174582 A1 WO 2021174582A1
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control
permanent magnet
magnet synchronous
current
synchronous motor
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PCT/CN2020/079487
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English (en)
French (fr)
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张驰
虞冠杰
蒋哲
乔海
陈进华
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中国科学院宁波材料技术与工程研究所
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Publication of WO2021174582A1 publication Critical patent/WO2021174582A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

Definitions

  • the application relates to a motor control method, in particular to a position sensor-less permanent magnet synchronous motor control device, a control method and an automobile power system, belonging to the technical field of motors.
  • high-performance permanent magnet synchronous motor speed control systems usually need to install mechanical sensors (such as encoders, decoders, tachogenerators) on the motor shaft to provide the rotor position and speed signals required for motor control.
  • mechanical sensors such as encoders, decoders, tachogenerators
  • the installation of mechanical sensors will bring some problems to the system, such as: increasing the system cost, increasing the size and rotation of the motor, reducing the system reliability, and limiting the application of the transmission system in harsh environments such as vibration and humidity. Wait.
  • position sensorless control technology can replace mechanical sensors. It uses easily available motor physical quantities to calculate the rotor position and speed of the motor, thereby solving the above-mentioned problems.
  • the existing position sensorless control equipment still has disadvantages such as difficult adjustment of parameters, inflexible adjustment of parameters, and inability to meet the requirements of operating performance for different permanent magnet synchronous motors and application scenarios.
  • the main purpose of this application is to provide a position sensorless permanent magnet synchronous motor control device and control method, so as to overcome the shortcomings of the prior art.
  • the embodiment of the present application provides a position sensorless permanent magnet synchronous motor control method, which includes:
  • the angle calculation device receives the back EMF signals e ⁇ and e ⁇ output by the low-pass filter device, and calculates the electrical angle of the permanent magnet synchronous motor rotor through the arctangent function
  • the angle compensation device uses the angle compensation device, according to the command speed ⁇ of the permanent magnet synchronous motor during operation and the cut-off angle frequency ⁇ c of the low-pass filter device, the lagging phase shift angle ⁇ is obtained through the arctangent algorithm, and the phase shift angle ⁇ And the electrical angle output by the angle calculation device Add together to get the estimated value ⁇ e of the electrical angle of the permanent magnet synchronous motor rotor.
  • the back-EMF observation device includes a current observation module and a current control module, and the control method includes:
  • the current observation module receives the actual ⁇ and ⁇ phase voltages and the control signal u transmitted by the current control module, and processes them to obtain the estimated values of the ⁇ and ⁇ phase currents
  • control method specifically includes: in the current observation module, processing the input actual ⁇ and ⁇ phase voltages and the control signal u through a permanent magnet synchronous motor mathematical model, thereby Obtain the estimated value of the ⁇ and ⁇ phase currents
  • the mathematical model of the permanent magnet synchronous motor is expressed as:
  • L and R are the phase inductance and phase resistance of the permanent magnet synchronous motor
  • i ⁇ , u ⁇ , e ⁇ are the current, voltage, and back EMF of the motor ⁇ phase respectively
  • i ⁇ , u ⁇ , and e ⁇ are the motor ⁇ respectively Phase current, voltage, back EMF.
  • the control method specifically includes: in the current control module, calculating the estimated values of the ⁇ and ⁇ phase currents
  • the phase current error ⁇ I s is obtained after the difference between the actual ⁇ and ⁇ phase currents I s , and the control signal u is output as the input to the current observation module to participate in the feedback control of the current observer, so that the estimated value of the phase current Converge to the actual phase current I s , that is, let ⁇ I s tend to 0, the control signal u is K*f( ⁇ I s ), where K is the control coefficient, and f( ⁇ I s ) is the control with ⁇ I s as a variable function.
  • the control method further includes: adopting adaptive switching strategy 1 for switching selection of f( ⁇ I s ), and the switching options include f 1 ( ⁇ I s ), f 2 ( ⁇ I s ),... , F i ( ⁇ I s ), ..., f n ( ⁇ I s ), where n is the number of control functions, and the range of i is [1,n], so as to obtain the best control effect for different permanent magnet synchronous motors,
  • the adaptive switching strategy 1 includes: according to different motor parameters, each motor parameter combination corresponds to a control function, and then from f 1 ( ⁇ I s ), f 2 ( ⁇ I s ), ..., f n ( ⁇ I s ) Select a control function as the application.
  • the motor parameters include, but are not limited to, stator resistance, inductance, and so on.
  • the control method further includes: adopting an adaptive switching strategy 2 to switch the value of K based on the constraint condition of the value of K and the selected control function, and the switching options include Ki1, Ki2 ,..., Kim, where the value of i ranges from 1 to n, and m is the speed segment number of the permanent magnet synchronous motor, so as to obtain the best control effect for the permanent magnet synchronous motor at different speed segments, the K is taken
  • the constraint condition of the value is greater than the maximum absolute value of the opposite potentials of ⁇ and ⁇ , so that ⁇ I s converges to 0.
  • the adaptive switching strategy 2 includes: according to the current speed range of the permanent magnet synchronous motor, each The speed section corresponds to a control coefficient, and then select a control coefficient from Ki1, Ki2,..., Kim as the application.
  • the switching option f i ( ⁇ I s ) of the control function f( ⁇ I s ) includes the switching function, saturation function, and exponential approach function shown in the following formulas (a), (b), and (c).
  • ⁇ is the current error limit.
  • the constraint condition of ⁇ is less than the rated current of the permanent magnet synchronous motor.
  • the method for obtaining ⁇ includes any one of a fixed value method, a look-up table method, a fuzzy control adaptive algorithm, a neural network adaptive algorithm, a recursive least square method, and a Kalman filter algorithm, but it is not limited to this. .
  • the cutoff angular frequency ⁇ c may be selected as the rated angular frequency of the permanent magnet synchronous motor.
  • the electrical angle of the permanent magnet synchronous motor rotor is:
  • the calculation formula of the electrical angle estimation value ⁇ e is:
  • the embodiment of the application also provides a position sensorless permanent magnet synchronous motor control device, which includes a back-EMF observation device, a low-pass filter device, an angle calculation device, and an angle compensation device.
  • the back EMF observation device includes a current observation module and a current control module.
  • the current observation module is used to receive the actual ⁇ and ⁇ phase voltages of the permanent magnet synchronous motor and the control signal u transmitted by the current control module, and process to obtain the estimated values of the ⁇ and ⁇ phase currents
  • the current control module is configured to receive the actual ⁇ and ⁇ phase current I s of the permanent magnet synchronous motor and the estimated value of the ⁇ and ⁇ phase current And processing to obtain the control signal u.
  • the low-pass filtering device is used to filter the disturbance signal from the control signal u at the cut-off angular frequency ⁇ c , and extract the continuous back EMF signals e ⁇ and e ⁇ .
  • the angle calculation device is used to receive the back EMF signals e ⁇ and e ⁇ , and calculate the electrical angle of the permanent magnet synchronous motor rotor according to the following formula (d)
  • the angle compensation device is used to obtain the lagging phase shift angle ⁇ according to the command speed ⁇ of the permanent magnet synchronous motor during operation and the cut-off angle frequency ⁇ c of the low-pass filter device, and to obtain the lagging phase shift angle ⁇ through the arctangent algorithm.
  • the phase shift angle ⁇ and the electrical angle output by the angle calculation device Add together to obtain the estimated value ⁇ e of the electrical angle of the permanent magnet synchronous motor rotor.
  • the current observation module includes a permanent magnet synchronous motor mathematical model, and the mathematical model is expressed as:
  • L and R are the phase inductance and phase resistance of the permanent magnet synchronous motor
  • i ⁇ , u ⁇ , e ⁇ are the current, voltage, and back EMF of the motor ⁇ phase respectively
  • i ⁇ , u ⁇ , and e ⁇ are the motor ⁇ respectively Phase current, voltage, back EMF.
  • the current control module is used to calculate the estimated values of the ⁇ and ⁇ phase currents
  • the phase current error ⁇ I s is obtained after the difference between the actual ⁇ and ⁇ phase currents I s , and the control signal u is input to the current observation module to participate in the feedback control of the current observer, so that the estimated value of the phase current Converge to the actual phase current I s , that is, let ⁇ I s tend to 0,
  • the control signal u is K*f( ⁇ I s ), where K is the control coefficient, and f( ⁇ I s ) is the control with ⁇ I s as a variable Function, the control function
  • the options for include the three functions shown in the following formulas (a), (b), and (c): switching function, saturation function, and exponential approach function, but are not limited to these three functions.
  • ⁇ is the current error limit, and the constraint condition of ⁇ is less than the rated current of the permanent magnet synchronous motor.
  • cut-off angular frequency ⁇ c is selected as the rated angular frequency of the permanent magnet synchronous motor.
  • the embodiment of the present application also provides an automobile power system, including a permanent magnet synchronous motor and a control system matching the permanent magnet synchronous motor, the control system including any of the aforementioned position sensorless permanent magnet synchronous motor control equipment.
  • the provided position sensorless method and system have at least the following advantages:
  • the provided permanent magnet synchronous motor control method does not need to install a position sensor, and the position and speed of the motor rotor can be obtained by collecting signals such as motor current and voltage, which not only reduces the hardware cost of the motor control equipment, but also reduces the size of the motor And moment of inertia, reducing the complexity of the mechanical structure, and also improving the environmental adaptability and reliability of the motor control equipment;
  • the provided permanent magnet synchronous motor control method effectively solves the problems of difficult adjustment of parameters for different permanent magnet synchronous motors and application scenarios, inflexible adjustment of parameters, and insufficient operating performance.
  • the provided permanent magnet synchronous motor control equipment has the characteristics of high reliability, low cost, high robustness, etc., and has broad application prospects. For example, it can be widely used in the power system of automobiles and other motor vehicles.
  • FIG. 1 is an application principle diagram of a position sensorless permanent magnet synchronous motor control device in an embodiment of the application
  • FIG. 2 is a schematic structural diagram of a permanent magnet synchronous motor control device without a position sensor in an embodiment of the application;
  • Fig. 3 is a schematic diagram of the structure of the back EMF observation unit in Fig. 2;
  • Figure 4 is a diagram of the composition of the current observation module in Figure 2;
  • Figure 5 is a diagram of the composition of the current control module in Figure 2;
  • Fig. 6 is a composition diagram of the angle compensation device in Fig. 2.
  • the following embodiments of the present application provide a position sensorless permanent magnet synchronous motor control device.
  • the position and function of the permanent magnet synchronous motor control equipment in the permanent magnet synchronous motor control equipment are shown in Figure 1. It is mainly to provide electrical angle position information for the vector control of the permanent magnet synchronous motor, so that it can carry out speed and current Closed-loop control.
  • the vector control can be FOC vector control, which generates a rotating magnetic field and performs torque control.
  • the permanent magnet synchronous motor control device of this embodiment may include four parts, namely: a back-EMF observation device, a low-pass filter device, an angle calculation device, and an angle compensation device.
  • the back-EMF observation device is mainly composed of a current observation module and a current control module. Its main function is to receive the actual ⁇ and ⁇ phase voltages and actual ⁇ and ⁇ phase currents of the permanent magnet synchronous motor, and output discontinuous back EMF information.
  • the current observation module receives the aforementioned actual ⁇ and ⁇ phase voltages and the control signal u transmitted from the current control module, processes and obtains the estimated values of the ⁇ and ⁇ phase currents, and outputs them to the current control module; and
  • the current control module receives the actual ⁇ and ⁇ phase currents and the estimated values of the ⁇ and ⁇ phase currents transmitted from the current observation module, processes and obtains the control signal u and outputs it to the current observation module.
  • the current observation module is a current observer based on the mathematical model of a permanent magnet synchronous motor.
  • the mathematical model of the motor can be expressed as:
  • L and R are the phase inductance and phase resistance of the permanent magnet synchronous motor
  • i ⁇ , u ⁇ and e ⁇ are the current, voltage and back EMF of the motor ⁇ phase respectively
  • i ⁇ , u ⁇ and e ⁇ are the motor ⁇ respectively The current, voltage and back EMF of the phase.
  • the main function of the current control module is to receive the estimated phase current output from the current observation module Compared with the actual phase current I s , the phase current error ⁇ I s is obtained by making the difference between the two inside the module.
  • the output control signal u is K*f( ⁇ I s ), and the control signal u is transmitted to the current observation module as an input. Participate in the feedback control of the current observer to estimate the phase current Converge to the actual phase current I s , that is, let ⁇ I s tend to 0, where K is the control coefficient, and the f( ⁇ I s ) is the control function with ⁇ I s as a variable.
  • the control function f( ⁇ I s ) has a composition form as shown below, which can be selected from a switching function (shown in equation (a)), a saturation function (shown in equation (b)), and an exponential approach function (shown in equation (c) ) Shows any one of), namely:
  • adaptive switching strategy 1 can be used for switching selection of f( ⁇ I s ) (adaptive switching 1 in Fig. 5), and the switching options are f 1 ( ⁇ I s ), f 2 ( ⁇ I s ), ..., f n ( ⁇ I s ), where n is the number of control functions, switch to f 2 ( ⁇ I s ) according to the motor parameters and adaptive switching strategy 1, that is, the aforementioned saturation is adopted function.
  • the control coefficient K needs to choose an appropriate value to make ⁇ I s converge to 0, and the constraint condition for its value is greater than the maximum value of the absolute value of the opposite potential of ⁇ and ⁇ .
  • the main feature of the adaptive switching strategy 1 is to select one of f 1 ( ⁇ I s ), f 2 ( ⁇ I s ),..., f n ( ⁇ I s ) after internal processing according to different motor parameters. Function as an application. Specifically, according to different motor parameters such as stator resistance and inductance, each motor parameter combination corresponds to a control function, and then from f 1 ( ⁇ I s ), f 2 ( ⁇ I s ), ..., f n ( ⁇ I s ) Choose a control function as the application.
  • the adaptive switching strategy 2 can be used to switch the value of K based on the above constraints and the selected control function (the adaptive switching in Figure 5 2)
  • the K value switching options are K21, K22,..., K2m, where m is the speed segment number of the permanent magnet synchronous motor, for example, m can be agreed to be 5.
  • the main feature of the adaptive switching strategy 2 is to select a control coefficient from Ki1, Ki2,..., Kim as an application after internal processing according to the speed segment of the current rotation speed of the permanent magnet synchronous motor. Specifically, according to the speed section where the current speed of the permanent magnet synchronous motor is located, each speed section corresponds to a control coefficient, and then a control coefficient is selected from Ki1, Ki2, ..., Kim as an application.
  • ⁇ in the saturation function is a limit of current error.
  • the control signal can be smoothed by selecting an appropriate ⁇ value.
  • the constraint condition is less than the rated current of the motor.
  • the method of obtaining it includes fixed value method and check Table method, fuzzy control adaptive algorithm, neural network adaptive algorithm, recursive least square method, Kalman filter algorithm, etc., but not limited to this.
  • a fixed value method can be used, and one-fifth of the maximum current of the system can be selected as the value of ⁇ .
  • phase current error ⁇ I s is caused by the harmonics of the motor back electromotive force
  • control signal u is composed of the back electromotive force and the disturbance signal.
  • All implementation strategies of the back-EMF observation device can be processed in devices such as DSP, MCU, CPLD, FPGA, etc., or can be implemented by hardware.
  • the main function of the low-pass filter device is to filter the high-frequency disturbance signal from the control signal u with the cut-off angular frequency ⁇ c , and extract the continuous back EMF signals e ⁇ and e ⁇ .
  • the cutoff angular frequency ⁇ c can be selected as the rated angular frequency of the motor.
  • All the implementation strategies of the low-pass filtering device can be processed in devices such as DSP, MCU, CPLD, FPGA, etc., or can be implemented by hardware.
  • the main function of the angle calculation device is to receive the ⁇ -phase and ⁇ opposite potential information output by the low-pass filter device, and obtain the electrical angle of the permanent magnet synchronous motor rotor through the arctangent function calculation
  • the calculation method used is as follows:
  • All implementation strategies of the angle calculation device can be processed in devices such as DSP, MCU, CPLD, FPGA, etc., or can be implemented by hardware.
  • the angle compensation device is shown in Figure 6. Since the low-pass filter device uses low-pass filter to obtain the back EMF, the phase delay is introduced, so according to the operating command speed ⁇ and the low-pass filter device cut-off angle frequency ⁇ c , through anyway The tangent algorithm obtains the lagging phase shift angle ⁇ , and finally compares it with the electrical angle calculated by the angle calculation device Add together to obtain the estimated value ⁇ e of the rotor electrical angle of the permanent magnet synchronous motor that is closer to the true value.
  • All implementation strategies of the angle compensation device can also be processed in devices such as DSP, MCU, CPLD, FPGA, etc., or can be implemented by hardware.
  • the position sensorless permanent magnet synchronous motor control device of this embodiment has the characteristics of high reliability, low cost, high robustness, etc., and has a wide range of applications in wind power, hydropower, new energy vehicles, underwater propulsion, and aviation. Application prospects.
  • this embodiment also provides an automobile power system, including a permanent magnet synchronous motor and a control system matching the permanent magnet synchronous motor, and the control system includes any of the aforementioned position sensorless permanent magnet synchronous motors. Motor control equipment.

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Abstract

本申请公开了一种无位置传感器的永磁同步电机控制方法及汽车动力系统。所述控制方法包括：以反电势观测装置接收永磁同步电机的实际α、β相电压和实际α、β相电流，并输出控制信号u；利用低通滤波装置以截止角频率ωc从前述u中滤除扰动信号，并提取出连续反电势信号eα和eβ；以角度计算装置接收eα和eβ，并计算得到电机转子的电角度aa；利用角度补偿装置，根据电机运行时的指令速度ω和ωc，计算获得滞后的相移角△θ，再与 aa相加，得到电机转子的电角度估计值θe。本申请提供的控制设备具有高可靠、低成本、高鲁棒性等特点，而相应的控制方法可以针对不同的永磁同步电机和应用场景而方便灵活的调参数，使电机运行性能可以更好的适应实际需求。

Description

无位置传感器的永磁同步电机控制方法及汽车动力系统 技术领域
本申请涉及一种电机控制方法，特别是一种无位置传感器的永磁同步电机控制设备、控制方法及汽车动力系统，属于电机技术领域。
背景技术
目前，高性能永磁同步电机调速系统通常需要在电机轴上安装机械传感器(如编码器、解码器、测速发电机),以提供电机控制所需要的转子位置和速度信号。但是，安装机械传感器会给系统带来一些问题，例如：增加了系统成本，增大了电机的尺寸和转动量，降低了系统可靠性，限制了传动系统在振动、潮湿等恶劣环境下的应用等。
无位置传感器控制技术作为一种新的电机控制技术可以代替机械传感器，其利用容易获得的电机物理量来计算电机的转子位置和转速，从而解决上述问题。但现有的无位置传感器控制设备针对不同的永磁同步电机和应用场景还存在不易调参数、调参不灵活、运行性能不能达到要求等缺陷。
有鉴于此，业界亟待发展出一种针对不同的电机和应用场景具有灵活调参数能力的无位置传感器的永磁同步电机控制方式。
发明内容
本申请的主要目的在于提供一种无位置传感器的永磁同步电机控制设备及控制方法，从而克服现有技术的不足。
为了达到前述发明目的，本申请采用了以下方案：
本申请实施例提供了一种无位置传感器的永磁同步电机控制方法，其包括：
以反电势观测装置接收永磁同步电机的实际α、β相电压和实际α、β相电流，并输出控制信号u，所述控制信号u包括反电势信号和扰动信号；
利用低通滤波装置，以截止角频率ω c从所述控制信号u中滤除扰动信号，并提取出连续的反电势信号e α和e β；
以角度计算装置接收低通滤波装置输出的反电势信号e α和e β，通过反正切函数计算得到永磁同步电机转子的电角度
Figure PCTCN2020079487-appb-000001
利用角度补偿装置，根据永磁同步电机运行时的指令速度ω和低通滤波装置的截止角频率ω c，通过反正切算法获得滞后的相移角△θ，并将所述相移角△θ与由角度计算装置输出的电角度
Figure PCTCN2020079487-appb-000002
相加，得到永磁同步电机转子的电角度估计值θ e。
进一步的，所述反电势观测装置包括电流观测模块和电流控制模块，并且所述控制方法包括：
以电流观测模块接收所述实际α、β相电压和由电流控制模块传输的控制信号u，并处理获得α、β相电流的估计值
Figure PCTCN2020079487-appb-000003
以电流控制模块接收所述实际α、β相电流I s和所述α、β相电流的估计值
Figure PCTCN2020079487-appb-000004
并处理获得控制信号u。
在一些实施方式中，所述的控制方法具体包括：在所述电流观测模块内，通过永磁同步电机数学模型对输入的所述实际α、β相电压和所述控制信号u进行处理，从而获得所述α、β相电流的估算值
Figure PCTCN2020079487-appb-000005
所述永磁同步电机数学模型表示为：
Figure PCTCN2020079487-appb-000006
其中L、R分别为永磁同步电机的相电感、相电阻，i α、u α、e α分别为电机α相的电流、电压、反电势，i β、u β、e β分别为电机β相的电流、电压、反电势。
在一些实施方式中，所述的控制方法具体包括：在所述电流控制模块内，将所述α、β相电流的估计值
Figure PCTCN2020079487-appb-000007
与实际α、β相电流I s做差后得到相电流误差ΔI s，且将控制信号u作为输入量输往电流观测模块从而参与电流观测器的反馈控制，使所述相电流估算值
Figure PCTCN2020079487-appb-000008
收敛于所述实际相电流I s，即令ΔI s趋向于0，所述控制信号u为K*f(ΔI s)，其中K为控制系数，f(ΔI s)为以ΔI s作为变量的控制函数。
在一些实施方式中，所述的控制方法还包括：采用自适应切换策略1进行f(ΔI s)的切换选择，且切换选择项包括f 1(ΔI s)、f 2(ΔI s)、…、f i(ΔI s)、…、f n(ΔI s)，其中n为控制函数的种类数，i的范围为[1,n]，从而获得针对不同永磁同步电机的最佳控制效果，所述自适应切换策略1 包括：根据不同的电机参数，以每个电机参数组合对应一个控制函数，然后从f 1(ΔI s)、f 2(ΔI s)、…、f n(ΔI s)中选择一个控制函数作为应用。所述电机参数包括但不限于定子电阻、电感等。
在一些实施方式中，所述的控制方法还包括：采用自适应切换策略2在K取值的约束条件和所选控制函数的基础上进行K值的切换选择，且切换选择项包括Ki1、Ki2、…、Kim，其中i的取值范围为1到n，m为永磁同步电机的速度段分段数，从而获得针对永磁同步电机不同速度段下的最佳控制效果，所述K取值的约束条件是大于α、β相反电势绝对值的最大值，以使ΔI s收敛于0，所述自适应切换策略2包括：根据永磁同步电机当前转速所处的速度段，以每个速度段对应一个控制系数，然后从Ki1、Ki2、…、Kim中选择一个控制系数作为应用。
进一步的，所述控制函数f(ΔI s)的切换选择项f i(ΔI s)包括下式(a)、(b)、(c)所示开关函数、饱和函数、指数趋近函数这3种函数，但不限于这3种函数，
Figure PCTCN2020079487-appb-000009
式(b)中ε为电流误差界限。
进一步的，所述ε的约束条件为小于永磁同步电机的额定电流。
进一步的，所述ε的获得方法包括定值法、查表法、模糊控制自适应算法、神经网络自适应算法、递推最小二乘法、卡尔曼滤波算法中的任一种，但不限于此。
在一些实施方式中，所述截止角频率ω c可以选择为永磁同步电机的额定角频率。
进一步的，在所述角度计算装置中，所述永磁同步电机转子的电角度
Figure PCTCN2020079487-appb-000010
的计算公式为：
Figure PCTCN2020079487-appb-000011
进一步的，在所述角度补偿装置中，所述电角度估计值θ e的计算公式为：
Figure PCTCN2020079487-appb-000012
本申请实施例还提供了一种无位置传感器的永磁同步电机控制设备，其包括反电势观测装置、低通滤波装置、角度计算装置和角度补偿装置。
进一步的，所述反电势观测装置包括电流观测模块和电流控制模块。
进一步的，所述电流观测模块用于接收永磁同步电机的实际α、β相电压和由电流控制模块传输的控制信号u，并处理获得α、β相电流的估计值
Figure PCTCN2020079487-appb-000013
进一步的，所述电流控制模块用于接收永磁同步电机的实际α、β相电流I s和所述α、β相电流的估计值
Figure PCTCN2020079487-appb-000014
并处理获得控制信号u。
进一步的，所述低通滤波装置用于以截止角频率ω c从所述控制信号u中滤除扰动信号，并提取出连续的反电势信号e α和e β。
进一步的，所述角度计算装置用于接收所述反电势信号e α和e β，并依据下式(d)计算得到永磁同步电机转子的电角度
Figure PCTCN2020079487-appb-000015
Figure PCTCN2020079487-appb-000016
进一步的，所述角度补偿装置用于根据永磁同步电机运行时的指令速度ω和低通滤波装置的截止角频率ω c，并通过反正切算法获得滞后的相移角△θ，再将所述相移角△θ与由角度计算装置输出的电角度
Figure PCTCN2020079487-appb-000017
相加，从而得到永磁同步电机转子的电角度估计值θ e。
进一步的，所述电流观测模块包含永磁同步电机数学模型，所述数学模型表示为：
Figure PCTCN2020079487-appb-000018
其中L、R分别为永磁同步电机的相电感、相电阻，i α、u α、e α分别为电机α相的电流、电压、反电势，i β、u β、e β分别为电机β相的电流、电压、反电势。
进一步的，所述电流控制模块用于将所述α、β相电流的估计值
Figure PCTCN2020079487-appb-000019
与实际α、β相电流I s做差后得到相电流误差ΔI s，并将控制信号u作为输入量输往电流观测模块从而参与电流观测器的 反馈控制，使所述相电流估算值
Figure PCTCN2020079487-appb-000020
收敛于所述实际相电流I s，即令ΔI s趋向于0，所述控制信号u为K*f(ΔI s)，其中K为控制系数，f(ΔI s)为以ΔI s作为变量的控制函数，所述控制函数
Figure PCTCN2020079487-appb-000021
的选择项包含下式(a)、(b)、(c)所示的开关函数、饱和函数、指数趋近函数这3种函数，但不限于这3种函数，
Figure PCTCN2020079487-appb-000022
式(b)中ε为电流误差界限，且ε的约束条件为小于永磁同步电机的额定电流。
进一步的，所述截止角频率ω c选择为永磁同步电机的额定角频率。
进一步的，所述角度补偿装置中采用的反正切算法表示为：
Figure PCTCN2020079487-appb-000023
本申请实施例还提供了一种汽车动力系统，包括永磁同步电机以及与所述永磁同步电机匹配的控制系统，所述控制系统包括前述的任一种无位置传感器的永磁同步电机控制设备。
较之现有技术，本申请至少具有如下优点：提供的无位置传感器的方法及系统至少具有如下优点：
(1)提供的永磁同步电机控制方法无需安装位置传感器，通过采集电机电流、电压等信号就可以获知电机转子的位置和速度，不仅降低了电机控制设备的硬件成本，减小了电机的尺寸和转动惯量，减少了机械结构的复杂度，还可提高电机控制设备的环境适应性和可靠性；
(2)提供的永磁同步电机控制方法获取永磁同步电机转子位置信息时不会受到恶劣环境的影响，提高了系统的环境适应性；
(3)提供的永磁同步电机控制方法有效解决了针对不同的永磁同步电机和应用场景存在不易调参数，调参不灵活，运行性能不能达到要求等问题。
(4)提供的永磁同步电机控制设备具有高可靠、低成本、高鲁棒性等特点，应用前景广阔，例如可以广泛应用于汽车等机动车的动力系统中。
附图说明
为了更清楚地说明本申请实施例或现有技术中的技术方案，下面将对实施例或现有技术描述中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本发明中记载的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图获得其他的附图。
图1为本申请实施例中一种无位置传感器的永磁同步电机控制设备的应用原理图；
图2为本申请实施例中一种无位置传感器的永磁同步电机控制设备的结构示意图；
图3为图2中反电势观测单元的结构示意图；
图4为图2中电流观测模块的组成图；
图5为图2中电流控制模块的组成图；
图6为图2中角度补偿装置的组成图。
具体实施方式
为使本申请的目的、技术方案和优点更加清楚，下面结合附图对本申请的具体实施方式进行详细说明。这些优选实施方式的示例在附图中进行了例示。附图中所示和根据附图描述的本申请的实施方式仅仅是示例性的，并且本申请并不限于这些实施方式。
在此，还需要说明的是，为了避免因不必要的细节而模糊了本申请，在附图中仅仅示出了与根据本申请的方案密切相关的结构和/或处理步骤，而省略了与本申请关系不大的其他细节。
本申请的如下实施例提供了一种无位置传感器的永磁同步电机控制设备。该永磁同步电机控制设备在永磁同步电机控制设备中所处的位置和功能如图1所示，主要是为永磁同步电机的矢量控制提供电角度位置信息，使其能进行速度和电流的闭环控制。其中的矢量控制可以是FOC矢量控制，产生旋转磁场，并进行力矩控制。
进一步的，请参阅图2所示，本实施例的永磁同步电机控制设备可以包括四个部分，即：反电势观测装置、低通滤波装置、角度计算装置和角度补偿装置。
参阅图3所示，其中的反电势观测装置主要由一个电流观测模块和一个电流控制模块组成。其主要作用是接收永磁同步电机的实际α、β相电压和实际α、β相电流，输出不连续的反电势信息。
进一步的，所述电流观测模块通过接收前述实际α、β相电压和从电流控制模块传来的控制信号u，处理获得α、β相电流的估计值，并将其输出给电流控制模块；而所述电流控制模块通过接收实际α、β相电流和从电流观测模块传来的α、β相电流的估计值，处理获得控制信号u并把其输出给电流观测模块。
更进一步的，参阅图4所示，所述电流观测模块是一个基于永磁同步电机数学模型的电流观测器，该电机数学模型可以表示为：
Figure PCTCN2020079487-appb-000024
其中L、R分别为永磁同步电机的相电感、相电阻，i α、u α和e α分别为电机α相的电流、电压和反电势，i β、u β和e β分别为电机β相的电流、电压和反电势。向所述电流观测模块输入永磁同步电机实际α、β相电压和电流控制模块输出的控制信号u之后，其内部经过该电机数学模型的处理，会输出永磁同步电机α、β相电流的估算值
Figure PCTCN2020079487-appb-000025
参阅图5所示，所述电流控制模块的主要作用是接收电流观测模块输出的相电流估算值
Figure PCTCN2020079487-appb-000026
和实际相电流I s，在模块内部通过将两者做差后得到相电流误差ΔI s，其输出控制信号u为K*f(ΔI s)，控制信号u作为输入量传给电流观测模块，参与电流观测器的反馈控制，使相电流估算值
Figure PCTCN2020079487-appb-000027
收敛于实际相电流I s，即让ΔI s趋向于0，其中K为控制系数，所述f(ΔI s)为以ΔI s作为变量的控制函数。
所述控制函数f(ΔI s)的构成形式如下所示，其可以选自开关函数(式(a)所示)、饱和函数(式(b)所示)和指数趋近函数(式(c)所示)中的任一者，即：
Figure PCTCN2020079487-appb-000028
为了获得针对不同永磁同步电机的最佳控制效果，可以采用自适应切换策略1进行f(ΔI s)的切换选择(图5中的自适应切换1)，其切换选择项有f 1(ΔI s)、f 2(ΔI s)、…、f n(ΔI s)，其中 n为控制函数的种类数，根据电机参数和自适应切换策略1切换到f 2(ΔI s)，即采用前述饱和函数。控制系数K需要选择合适的值才能使得ΔI s收敛于0，其取值的约束条件是大于α、β相反电势绝对值的最大值。
进一步的，所述自适应切换策略1的主要特点是根据不同的电机参数经过内部处理后从f 1(ΔI s)、f 2(ΔI s)、…，f n(ΔI s)中选择一个控制函数作为应用。具体的可以根据不同的定子电阻、电感等电机参数，以每个电机参数组合对应一个控制函数，然后从f 1(ΔI s)、f 2(ΔI s)、…、f n(ΔI s)中选择一个控制函数作为应用。
为了获得针对永磁同步电机不同速度段下的最佳控制效果，可以采用自适应切换策略2在上述约束条件和所选控制函数的基础上进行K值的切换选择(图5中的自适应切换2)，K值的切换选择项有K21、K22、…、K2m，m为永磁同步电机的速度段分段数，例如可以约定m为5。
所述自适应切换策略2的主要特点是根据永磁同步电机当前转速所处的速度段经过内部处理后从Ki1、Ki2、…、Kim中选择一个控制系数作为应用。具体的，可以根据永磁同步电机当前转速所处的速度段，以每个速度段对应一个控制系数，然后从Ki1、Ki2、…、Kim中选择一个控制系数作为应用。
进一步的，所述饱和函数中的ε是一个电流误差的界限，通过选择合适的ε值可以使控制信号变得平滑，其约束条件为小于电机的额定电流，其获得方法包括定值法、查表法、模糊控制自适应算法、神经网络自适应算法、递推最小二乘法、卡尔曼滤波算法等，且不限于此。例如可以采用定值法，并选取系统最大电流的五分之一作为ε的值。
另外，所述相电流误差ΔI s是由电机反电动势的谐波引起的，所述控制信号u由反电势和扰动信号组成。
所述反电势观测装置的所有实现策略可以在DSP、MCU、CPLD、FPGA等器件中进行处理，也可以由硬件进行实现。
所述低通滤波装置的主要作用是以截止角频率ω c从控制信号u中过滤掉高频的扰动信号，提取出连续的反电势信号e α和e β。所述截止角频率ω c可以选择为电机额定角频率。
所述低通滤波装置的所有实现策略可以在DSP、MCU、CPLD、FPGA等器件中进行处理，也可以由硬件进行实现。
所述角度计算装置主要作用是接收低通滤波装置输出的α相和β相反电势信息，通过反正切函数计算得到永磁同步电机转子的电角度
Figure PCTCN2020079487-appb-000029
其中采用的计算方式如下：
Figure PCTCN2020079487-appb-000030
所述角度计算装置的所有实现策略可以在DSP、MCU、CPLD、FPGA等器件中进行处理，也可以由硬件进行实现。
所述角度补偿装置如图6所示，由于低通滤波装置采用低通滤波获取反电动势时引入了相位延迟，所以根据运行时的指令速度ω和低通滤波装置截止角频率ω c，通过反正切算法获得滞后的相移角△θ，最终与角度计算装置计算得到的电角度
Figure PCTCN2020079487-appb-000031
相加，得到更接近真实值的永磁同步电机转子电角度估计值θ e。
所述角度补偿装置的所有实现策略也可以在DSP、MCU、CPLD、FPGA等器件中进行处理，也可以由硬件进行实现。
本实施例的无位置传感器的永磁同步电机控制设备具有高可靠、低成本、高鲁棒性等特点，在风力发电、水力发电、新能源汽车、水下推进和航空等应用方面具有广大的应用前景。
相应的，本实施例还提供了一种汽车动力系统，包括永磁同步电机以及与所述永磁同步电机匹配的控制系统，所述控制系统包括前述的任一种无位置传感器的永磁同步电机控制设备。
此外，需要说明的是，在本说明书中，“包括”、“包含”或者其任何其他变体意在涵盖非排他性的包含，从而使得包括一系列要素的过程、方法、物品或者设备不仅包括那些要素，而且还包括没有明确列出的其他要素，或者是还包括为这种过程、方法、物品或者设备所固有的要素。
应当理解，虽然本说明书按照实施方式加以描述，但并非每个实施方式仅包含一个独立的技术方案，说明书的这种叙述方式仅仅是为清楚起见，本领域技术人员应当将说明书作为一个整体，各实施例中的技术方案也可以经适当组合，形成本领域技术人员可以理解的其他实施方式。

Claims (16)

  1. 一种无位置传感器的永磁同步电机控制方法，其特征在于包括：
    以反电势观测装置接收永磁同步电机的实际α、β相电压和实际α、β相电流，并输出控制信号u，所述控制信号u包括反电势信号和扰动信号；
    利用低通滤波装置，以截止角频率ω c从所述控制信号u中滤除扰动信号，并提取出连续的反电势信号e α和e β；
    以角度计算装置接收低通滤波装置输出的反电势信号e α和e β，通过反正切函数计算得到永磁同步电机转子的电角度
    Figure PCTCN2020079487-appb-100001
    利用角度补偿装置，根据永磁同步电机运行时的指令速度ω和低通滤波装置的截止角频率ω c，通过反正切算法获得滞后的相移角△θ，并将所述相移角△θ与由角度计算装置输出的电角度
    Figure PCTCN2020079487-appb-100002
    相加，得到永磁同步电机转子的电角度估计值θ e。
  2. 根据权利要求1所述的控制方法，其特征在于：所述反电势观测装置包括电流观测模块和电流控制模块，并且所述控制方法包括：
    以电流观测模块接收所述实际α、β相电压和由电流控制模块传输的控制信号u，并处理获得α、β相电流的估计值
    Figure PCTCN2020079487-appb-100003
    以电流控制模块接收所述实际α、β相电流I s和所述α、β相电流的估计值
    Figure PCTCN2020079487-appb-100004
    并处理获得控制信号u。
  3. 根据权利要求2所述的控制方法，其特征在于具体包括：在所述电流观测模块内，通过永磁同步电机数学模型对输入的所述实际α、β相电压和所述控制信号u进行处理，从而获得所述α、β相电流的估算值
    Figure PCTCN2020079487-appb-100005
    所述永磁同步电机数学模型表示为：
    Figure PCTCN2020079487-appb-100006
    其中L、R分别为永磁同步电机的相电感、相电阻，i α、u α、e α分别为电机α相的电流、电压、反电势，i β、u β、e β分别为电机β相的电流、电压、反电势。
  4. 根据权利要求2所述的控制方法，其特征在于具体包括：在所述电流控制模块内，将所述α、β相电流的估计值
    Figure PCTCN2020079487-appb-100007
    与实际α、β相电流I s做差后得到相电流误差ΔI s，且将控制信号u作为输入量输往电流观测模块从而参与电流观测器的反馈控制，使所述相电流估算值
    Figure PCTCN2020079487-appb-100008
    收敛于所述实际相电流I s，即令ΔI s趋向于0，所述控制信号u为K*f(ΔI s)，其中K为控制系数，f(ΔI s)为以ΔI s作为变量的控制函数。
  5. 根据权利要求4所述的控制方法，其特征在于：
    采用自适应切换策略1进行f(ΔI s)的切换选择，且切换选择项包括f 1(ΔI s)、f 2(ΔI s)、…、f i(ΔI s)、…、f n(ΔI s)，其中n为控制函数的种类数，i的范围为[1,n]，从而获得针对不同永磁同步电机的最佳控制效果，所述自适应切换策略1包括：根据不同的电机参数，所述电机参数包括定子电阻或电感，以每个电机参数组合对应一个控制函数，然后从f 1(ΔI s)、f 2(ΔI s)、…、f n(ΔI s)中选择一个控制函数作为应用；
    采用自适应切换策略2在K取值的约束条件和所选控制函数的基础上进行K值的切换选择，且切换选择项包括Ki1、Ki2、…、Kim，其中i的取值范围为1到n，m为永磁同步电机的速度段分段数，从而获得针对永磁同步电机不同速度段下的最佳控制效果，所述K取值的约束条件是大于α、β相反电势绝对值的最大值，以使ΔI s收敛于0，所述自适应切换策略2包括：根据永磁同步电机当前转速所处的速度段，以每个速度段对应一个控制系数，然后从Ki1、Ki2、…、Kim中选择一个控制系数作为应用。
  6. 根据权利要求5所述的控制方法，其特征在于：所述控制函数f(ΔI s)的切换选择项f i(ΔI s)包括下式(a)、(b)、(c)所示开关函数、饱和函数、指数趋近函数；
    Figure PCTCN2020079487-appb-100009
    Figure PCTCN2020079487-appb-100010
    式(b)中ε为电流误差界限。
  7. 根据权利要求6所述的控制方法，其特征在于：所述ε的约束条件为小于永磁同步电机的额定电流，并且所述ε的获得方法包括定值法、查表法、模糊控制自适应算法、神经网络自适应算法、递推最小二乘法、卡尔曼滤波算法中的任一种。
  8. 根据权利要求1所述的控制方法，其特征在于：所述截止角频率ω c选择为永磁同步电机的额定角频率。
  9. 根据权利要求1所述的控制方法，其特征在于，在所述角度计算装置中，所述永磁同步电机转子的电角度
    Figure PCTCN2020079487-appb-100011
    的计算公式为：
    Figure PCTCN2020079487-appb-100012
  10. 根据权利要求1所述的控制方法，其特征在于：在所述角度补偿装置中，所述永磁同步电机转子的电角度估计值θ e的计算公式为：
    Figure PCTCN2020079487-appb-100013
    Figure PCTCN2020079487-appb-100014
  11. 一种无位置传感器的永磁同步电机控制设备，其特征在于包括反电势观测装置、低通滤波装置、角度计算装置和角度补偿装置，所述反电势观测装置包括电流观测模块和电流控制模块；
    所述电流观测模块用于接收永磁同步电机的实际α、β相电压和由电流控制模块传输的控制信号u，并处理获得α、β相电流的估计值
    Figure PCTCN2020079487-appb-100015
    所述电流控制模块用于接收永磁同步电机的实际α、β相电流I s和所述α、β相电流的估计值
    Figure PCTCN2020079487-appb-100016
    并处理获得控制信号u；
    所述低通滤波装置用于以截止角频率ω c从所述控制信号u中滤除扰动信号，并提取出连续的反电势信号e α和e β；
    所述角度计算装置用于接收所述反电势信号e α和e β，并依据下式(d)计算得到永磁同步电机转子的电角度
    Figure PCTCN2020079487-appb-100017
    Figure PCTCN2020079487-appb-100018
    所述角度补偿装置用于根据永磁同步电机运行时的指令速度ω和低通滤波装置的截止角频率ω c，并通过反正切算法获得滞后的相移角△θ，再将所述相移角△θ与由角度计算装置输出的电角度
    Figure PCTCN2020079487-appb-100019
    相加，从而得到永磁同步电机转子的电角度估计值θ e。
  12. 根据权利要求11所述的控制设备，其特征在于：所述电流观测模块包含永磁同步电机数学模型，所述数学模型表示为：
    Figure PCTCN2020079487-appb-100020
    其中L、R分别为永磁同步电机的相电感、相电阻，i α、u α和e α分别为电机α相的电流、电压和反电势，i β、u β和e β分别为电机β相的电流、电压和反电势。
  13. 根据权利要求12所述的控制设备，其特征在：所述电流控制模块用于将所述α、β相电流的估计值
    Figure PCTCN2020079487-appb-100021
    与实际α、β相电流I s做差后得到相电流误差ΔI s，并将控制信号u作为输入量输往电流观测模块从而参与电流观测器的反馈控制，使所述相电流估算值
    Figure PCTCN2020079487-appb-100022
    收敛于所述实际相电流I s，即令ΔI s趋向于0，所述控制信号u为K*f(ΔI s)，其中K为控制系数，f(ΔI s)为以ΔI s作为变量的控制函数，所述控制函数f(ΔI s)的选择项包含下式(a)、(b)、(c)所示的开关函数、饱和函数、指数趋近函数；
    Figure PCTCN2020079487-appb-100023
    Figure PCTCN2020079487-appb-100024
    式(b)中ε为电流误差界限，且ε的约束条件为小于永磁同步电机的额定电流。
  14. 根据权利要求11所述的控制设备，其特征在于：所述截止角频率ω c选择为永磁同步电机的额定角频率。
  15. 根据权利要求11所述的控制设备，其特征在于：所述角度补偿装置中采用的反正切算法表示为：
    Figure PCTCN2020079487-appb-100025
    Figure PCTCN2020079487-appb-100026
  16. 一种汽车动力系统，包括永磁同步电机以及与所述永磁同步电机匹配的控制系统，其特征在于：所述控制系统包括权利要求11-15中任一项所述的无位置传感器的永磁同步电机控制设备。
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