WO2019063333A1 - Dispositif de commande de lampe ayant un convertisseur en dcm - Google Patents

Dispositif de commande de lampe ayant un convertisseur en dcm Download PDF

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Publication number
WO2019063333A1
WO2019063333A1 PCT/EP2018/075050 EP2018075050W WO2019063333A1 WO 2019063333 A1 WO2019063333 A1 WO 2019063333A1 EP 2018075050 W EP2018075050 W EP 2018075050W WO 2019063333 A1 WO2019063333 A1 WO 2019063333A1
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WIPO (PCT)
Prior art keywords
switch
current
time
operating device
control circuit
Prior art date
Application number
PCT/EP2018/075050
Other languages
German (de)
English (en)
Inventor
Ludwig Erasmus DE CLERCQ
Harald Netzer
Frank Lochmann
Clemens KUCERA
Original Assignee
Tridonic Gmbh & Co Kg
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from DE202018104924.0U external-priority patent/DE202018104924U1/de
Application filed by Tridonic Gmbh & Co Kg filed Critical Tridonic Gmbh & Co Kg
Priority to EP18778838.5A priority Critical patent/EP3673714A1/fr
Publication of WO2019063333A1 publication Critical patent/WO2019063333A1/fr

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Classifications

    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
    • H05B45/10—Controlling the intensity of the light
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
    • H05B45/30—Driver circuits
    • H05B45/37—Converter circuits
    • H05B45/3725—Switched mode power supply [SMPS]
    • H05B45/375—Switched mode power supply [SMPS] using buck topology
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0025—Arrangements for modifying reference values, feedback values or error values in the control loop of a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0064—Magnetic structures combining different functions, e.g. storage, filtering or transformation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/157—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators with digital control
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
    • H05B45/30—Driver circuits
    • H05B45/37—Converter circuits
    • H05B45/3725—Switched mode power supply [SMPS]
    • H05B45/38—Switched mode power supply [SMPS] using boost topology
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B45/00—Circuit arrangements for operating light-emitting diodes [LED]
    • H05B45/30—Driver circuits
    • H05B45/37—Converter circuits
    • H05B45/3725—Switched mode power supply [SMPS]
    • H05B45/385—Switched mode power supply [SMPS] using flyback topology
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B47/00—Circuit arrangements for operating light sources in general, i.e. where the type of light source is not relevant
    • H05B47/10—Controlling the light source
    • H05B47/165—Controlling the light source following a pre-assigned programmed sequence; Logic control [LC]
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates to a lamp operating device, which is designed in particular for the dimmable operation of light sources, such as LEDs.
  • LEDs also mean organic LEDs (OLEDs).
  • the invention relates to dimmable control gear for lighting devices that use an active clocked converter.
  • a control circuit controls a switch of the clocked converter, such that in the switched (conductive) state of the switch, an energy storage element (such as an inductance) is charged, which energy storage element in the off state of the switch (non-conductive state of the switch) again discharges via the light path or preferably a further energy storage device (eg, capacitor) loads, which in turn feeds the LED track with a possibly rippled with DC voltage.
  • an energy storage element such as an inductance
  • the current drop through the inductor is limited to a value greater than zero, and the switch is again rendered conductive before the current through the inductor has dropped to zero.
  • the restart threshold or the corresponding time switch-on time
  • the reconnection time extended until finally the state is reached that the current through the inductance actually drops to zero may before the switch is again switched conductive and the current thus rises again.
  • This mode of operation (restarting on reaching the zero level) is typically referred to as “critical mode” (BCM, also abbreviated to “limit operating mode”).
  • a positive zero crossing of the current through the inductance is a current which at the same time shows a positive gradient at a time when the current has the value zero, ie a first derivative of the course of the current at the time of the zero crossing greater than zero (positive) is.
  • the time ranges of the positive zero crossings can be detected by measurement (for example of the current through the inductance), or can be predicted on the basis of the resonant frequency known from the dimensioning of the components.
  • the light output can not be changed continuously by continuously changing the reconnection time, but only in increments ("valley switching operation") between a or more positive zero crossings of the current through the inductance, which can lead to jumps in the light output, which can also be perceived optically as flickering during a dimming ramp.
  • the invention addresses this problem and provides a technique which, at dimming in the leaky mode and transitioning from the critical mode to the spanning mode, at least reduces the problem of jumps in the output power of the converter, even if reconnecting only at the times the positive zero crossings of the current through the inductance takes place. Repeated jumping of the converter between two operating points, wherein the two operating points are characterized by the respective discretely spaced reclosing times of the switch, in a steady state operation of the converter is to be prevented.
  • the embodiment should be such that an ASIC is not necessarily required as a control device, but that a microcontroller can also initiate the corresponding activation of the switch.
  • a first aspect of the invention relates to an operating device for the dimmable operation of lighting devices, in particular one or more LED (s). It has a control circuit which has a clocked converter circuit with an energy storage element, in particular at least one inductance, and at least one switch, which is clocked starting from the control circuit.
  • the converter circuit can be, for example, a buck converter or a boost converter.
  • the control circuit is designed to operate the converter circuit selectively, in particular by a signal specifying the power, by driving the switch, at least in the critical mode or in the mode with a gaping current.
  • a first dimming range therefore, there is an operation in the latching mode
  • another separate dimming range there is an operation in the critical mode.
  • These two modes may be adjacent to each other, however, a hybrid mode may also be provided between them, in which the gaping or critical mode is temporally multiplexed. (Optionally, the critical mode can also be connected to a continuous mode).
  • the control circuit sets the switch-on instant of the switch in discrete increments into one of the time ranges after the first zero crossing, in which the current during the discharge of the energy storage element performs a rising zero crossing.
  • the control circuit is designed to regulate, with an incremental change in the reconnection time, a feedback variable influencing the luminous flux power by direct or indirect change in the switch-on duration of the switch.
  • the control circuit can be designed to
  • the control circuit can be designed to carry out an incremental change of the reclosing time, if the control with the control variable "switch-on time" has predetermined minimum or maximum values of the duty cycle or of a variable influencing them, so to speak, this regulation by means of the change of the switch-on to their Limits are encountered. Accordingly, the control circuit may be designed so that when the switch-on time is maximum and at the same time a set long dead time is not sufficient to achieve a desired average current, make an incremental reduction of the dead time by selecting a time earlier Wiedererschaltzeithuis.
  • the on-time is maximum when, for example, a maximum allowable peak current value I pea k is achieved by the inductor current II (coil current II).
  • An incremental extension of the reconnection time is to be understood as a time shift of the reconnection time at a later time later by a predetermined period of time.
  • the predetermined period of time may in particular comprise one or more periods of the resonance oscillation.
  • the decrease in the reconnection timing is accordingly a time shift of the reconnection timing to a time earlier by a predetermined period of time.
  • the control circuit may be configured to specify the change in the switch-on period of the switch indirectly by specifying a switch-off threshold for the current through the switch or directly by specifying the switch-on period.
  • the control circuit may be configured to, in the event of a transition from the critical mode to the latching current mode, suddenly increase the value for the switch-on time duration of the switch.
  • the converter circuit can be, for example, a boost, buck, buck / boost or flyback converter.
  • control circuit configured to perform in the latching operation, a change in the reconnection time to a calculated different reclosing time different from a current reconnection time only if the new reclosing time of the switch from the current reconnection time by a time greater than deviates predetermined value.
  • the switch-on time can be changed from period to period t pe period, that is to say very quickly.
  • the turn-off because it is generated for example by a digital / analog converter and supplied to a comparator, not be changed arbitrarily fast.
  • the turn-off threshold can be set to achieve the ideal value for Ipea k, fluctuations in the average load current can occur because the turn-off threshold can not be varied as rapidly as it is for the reclosing time.
  • this hysteresis is a frequent change (engl .: toggle) of the reconnection time.
  • a stability of the average load current is improved, so that, for example, a light output of the lamp can now be flicker-free.
  • the predetermined value is a period of time that is longer than 50% of a period of a resonant oscillation of the current through the energy storage element during a dead time of the clocked converter.
  • the predetermined value is a period of time greater than 55%, preferably a period of time greater than 62.5% of the period of a resonant oscillation of the current through the energy storage element during a dead time of the clocked converter.
  • Other values for the predetermined value between 0.5 and 1.0 are also possible.
  • Another Aspect of the invention relates to a method for dimmable operation of lamps, in particular one or more LED (s), using a control circuit having a clocked converter circuit with an energy storage element and at least one switch, which is driven by the control circuit.
  • the converter circuit is operated in at least a partial area of the total dimming range in the mode with a gaping current.
  • the reclosing time is placed in discrete increments in the range of a rising zero crossing of the current through the storage element.
  • a switch-off threshold for the current through the switch or a switch-on time of the switch predefined for the dimming value to be set is set, and a rising zero crossing is set as the actual switch-on time closest to the theoretical, calculated switch-on time which results from the switch-off threshold or the turn-on time period and the dimming value to be set, and - the switch-off threshold or the switch-on duration changed depending on the deviation of the actual switch-on time from the theoretical switch-on time.
  • the predefined switch-off threshold for the current through the switch or a switch-on period of the switch can depend not only on the dimming value to be set, but on at least one further parameter, such as the voltage across the lighting means.
  • the invention also relates to a control circuit, for example. ASIC or microcontroller, which is designed for such a method.
  • a further aspect of the invention relates to a circuit for detecting the zero crossing of the current through the inductance of a buck converter, in particular a buck converter in an operating device of the type described above, having one with the potential at a connection point of the diode of the Buck converter and a switch of the Buck converter connected diode circuit which generates a preferably digital detection signal whose preferably logic level changes at a zero crossing of the current through the inductance.
  • This circuit may include a software or hardware block that prevents the level of the detection signal from changing after a first zero crossing at further zero crossings.
  • This circuit may comprise a control circuit, preferably a microcontroller, to which the detection signal is supplied.
  • a defined level change of the detection signal can trigger a starting of a counter in the microcontroller.
  • the detection signal starts the dead time Tdead.
  • Another aspect of the invention relates to a circuit for detecting the lower reversal point of the current through the inductance of a buck converter, comprising an auxiliary winding magnetically coupled to the inductance of the buck converter which generates a signal which switches a transistor producing a detection signal when the current through the inductor reaches its lower reversal point.
  • the detection signal thus triggers a storage of the counter readings of the lower reversal points (temporal position of the "valleys").
  • Yet another aspect of the invention relates to an operating device for lighting, comprising a clocked by a switch converter (clocked converter), in particular Buck converter, comprising a control circuit, the switch after a predetermined time t on or upon reaching a switch-off of the rising Switch current again nonconductive switches,
  • a switch converter locked converter
  • Buck converter comprising a control circuit
  • control circuit is supplied with a feedback signal which reproduces the voltage via the light source supplied by the operating device, the control circuit correcting the set time period t on or the switch-off threshold depending on the feedback signal.
  • the switch-off threshold is set as a function of a set average output current I aV g_nom.
  • the set peak current I pea k may consist of a part which is adaptively calculated on the basis of the selected reclosing time, and a corrective part, which is given for example by a regulator (I regulator), in order to achieve the average output current Lvg j om to be set ,
  • the turn-off threshold is preferably set lower at lower voltage across the lighting means than at higher voltage across the lighting means.
  • a correction value (further correction value) is thus taken into account, which is determined on the basis of the voltage across the lighting means. This ensures that the set average output current is really reached, for example independently of one Slope of a current increase via the inductance.
  • the correction value which takes into account the voltage across the lighting means, prevents higher peak currents resulting from given turn-off current delays at the same turn-off threshold at lower output voltage than at high output voltage. This is because a steepness of a current increase through the switch also depends on a voltage across the lamps.
  • time duration t on can be set independently of a voltage across the lighting means. In this case, a value for a switch-off delay can be taken into account, in particular subtracted.
  • Fig. 1 shows a schematic view of a known buck converter for
  • Fig. 2 shows a current waveform as a function of switch state
  • FIG. 3 shows the application of the so-called. Valley switching in a selected
  • Fig. 4 shows a block diagram for carrying out the invention.
  • Fig. 5 shows waveforms in the practice of the invention.
  • Fig. 6 shows an implementation of the invention with two possible zero-crosses
  • FIGS. 7 to 10 show signal and current waveforms as well as a schematic circuit in FIG.
  • Fig. 12 shows, associated with the time course of the predetermined load current after
  • Fig. 13 shows, associated with the time course of the predetermined load current after
  • Fig. 11 calculated values for a dead time tdead_dom and set values for the dead time tdead after a further improved design with a hysteresis.
  • Fig. 14 shows further time courses of the predetermined average load current I ; avg nom for predefined dimming curves.
  • FIG. 15 shows, for the time courses of the predetermined average load current according to FIG.
  • Dead time tdead with a hysteresis for time-increasing and time-decreasing values of the given load current in a section.
  • FIG. 1 schematically shows a known buck converter 1 for operating a light path 6 as an example of a typical load of the Buck converter 1.
  • a DC operating voltage VLED for a light path 6 is provided.
  • the voltage VLED increases by charging the capacitor, while it decreases in the off state of the switch 5.
  • the human eye perceives only the time average with appropriate high-frequency control of the switch 5.
  • the switch 5 is controlled by a control circuit 2 via a signal input (control input 3).
  • the control circuit may be an ASIC or preferably a microcontroller.
  • Fig. 2 shows corresponding signal, - voltage and current waveforms.
  • the signal HS is the level at the control input 3 of the switch 5. During a time t on , this switch 5 is turned on , while it is switched non-conducting in the periods and tdead.
  • the diode 7 in Fig. 1 is conductive.
  • the lowermost curve in FIG. 2 is the current course through the inductance 4, which is denoted by II.
  • this current increases, while in the turn-off period f it decreases until it makes a first (falling) zero crossing. Due to resonance effects, a vibration pattern then occurs during the time period tdead.
  • a reconnection in this current gap according to the invention is always carried out to a temporal range (time) in which the current II through the inductor 4 performs a rising zero crossing.
  • the on-time t on can be output by the control circuit 2 timed.
  • the on-time t on can be specified by the control circuit 2 by the switch 5 is switched non-conductive as soon as the rising switch current reaches a predetermined upper cut-off threshold I pea k.
  • reconnecting only in discrete increments to the positive zero-crossing ranges of the current means that jumps in the light output can occur in intermittent operation, or at the transition from the critical mode to the mode with intermittent operation, since just no continuous Change of the reconnection time should be made and thus the supply of the light path 6 could not be continuous.
  • the luminous flux power is to be reduced in intermittent operation, a chronologically higher reclosing time for the switch 5 must be triggered abruptly.
  • the thus threatening power jump is now compensated for by maintaining a time- on- control when the switch-off period is held fixed.
  • the on-time t on not so held in lopsided operation, but designed adaptive and maintain a t on- time control.
  • this has the advantage, in terms of control technology, that a continuous on-time regulation is maintained throughout, ie both in intermittent operation and in critical mode or possibly also in continuous operation mode.
  • the dead time tdead is reduced by a jump of the zero crossing increments, and at the same time the t on- time is suddenly reduced to an estimated operating point to avoid light power jumps by then continuing the timing at this estimated operating point.
  • the switch-on time t on of the switch 5 can result either directly from a time- on-time regulation, but also indirectly, namely in particular by predefining a switch-off threshold I pea k for the switch current.
  • the new operating point can thus be a new switch-on time or a new switch-off threshold.
  • a jump in the dead time increment that is to say the selected positive zero crossing during the time tdead, can also be triggered when the time control or the shutdown threshold regulation encounters an upper or lower predetermined limit value for the t on time or the switch off threshold I pea k.
  • the control circuit 2 After specifying a change, in particular a jump for the setpoint value for the average LED current, the control circuit 2 first calculates how long the idle time tdead would have to be in the latching mode in order to achieve the average time current at a given switch-off threshold I pe ak_nom , This calculated dead time tdeadjom will normally not fall on a zero crossing of the current with positive gradients, so that then on the one hand the nearest zero crossing with positive gradient is selected, but at the same time the difference between the calculated dead time tdead j om and the nearest zero crossing is determined. Because of this known difference can then be set according to the new operating point of the cut-off threshold I pea k or WZeitregelung accordingly, whereupon the WZeitregelung is continued at this operating point.
  • the new shutdown threshold calculation circuit tdead 2 may not use the difference between tdead j om and the selected tdead, but may use the absolute value of the selected tdead based on the selected reclosing time.
  • the given switch-off threshold I pe ak_nom also depends on the desired value of the mean output current Iavgjom (average current Lvgjom). With this predetermined switch-off threshold I pe ak_nom, the dead time tdeadjom is then calculated. In order to prevent unnecessary jumping between different zero-crossings ("valleys"), which may possibly be visible as a jump in the luminous power, always taking into account a new average current Lvg j om the previous value of the zero crossing is taken into account to jump between to prevent two different zero crossings with positive gradients by a kind of hysteresis control.
  • the conversion of a set average current value Lvg j om to a switch-off threshold Ipeak or t on- time can be multidimensional, such that the selection of the switch-off threshold I pea k (or the direct t on- time default) takes place taking into account further parameters (via the Average current value addition). These further parameters can be, for example: the LED voltage V LED , since this has an influence on how sensitively the average value I aV g of the current reacts to the change in the switch-off threshold I pea k or the t on time.
  • the determination of the switch-off threshold I pea k or the t on- time dependent on the dimming value to be achieved determines the frequency of the occurrence of the switch-on.
  • the switch-off threshold I pea k is set so that as far as possible a constant frequency of the occurrence of the switch-on results, ie if possible the dead time tdead is kept constant.
  • FIG. 3 shows that the valley switching according to the invention, ie the incremental jump between different increments at positive zero crossings of the current through the energy storage element (inductance 4), is preferably only in an upper dimming range, then at 100% nominal output of the illuminant path 6 is performed.
  • the dead time tdead in the lopsided operation is so large that the decaying reverberation of the current II through the inductance 4 is no longer present or no longer plays a role.
  • the reclosing time t on the switch 5 can be adjusted continuously.
  • the dead time tdead can be changed continuously for a change in the mean output current I aV g.
  • the dead time tdead If the dead time tdead is very large, the switching frequency of the switch 5 becomes correspondingly low. Losses due to the switching operations are correspondingly low. At a low dimming level, therefore, the dead time tdead can also be changed continuously accordingly.
  • Fig. 4 shows a schematic block diagram for carrying out the invention.
  • a block A designates a calculation block for calculating the nominal dead time tdead nom and the nominal shutdown threshold I pe ak_nom.
  • Block A in FIG. 4 calculates the nominal values for the dead time tdead nom and the switch-off threshold I pe ak_nom (or the switch-on time t on _nom) on the basis of a function or an adjustment table.
  • Block B then serves to implement the dead time control on the basis of these calculated nominal values.
  • a dead time compensation unit "tdead compensator” which receives as input information the selected valley from the block “select nearest neighbor” as well as the time mean value of the current through the LED path Lvg j om
  • the output variable of the current regulator is changed by the dead time compensator tdeadcompensator for shifting the operating point, resulting in a new Switch-off threshold for the switch current "new I pea k” results.
  • a buck converter 1 Shown in FIG. 6 is a buck converter 1 according to the present invention, the power path of which comprises a buck switch M1, a buck diode D1, an output filter capacitor C1 fed by the converter, which may be any low-pass filter, and the buck - Inductance Ll-A and a measuring resistor (shunt) Rl for measuring the buck current.
  • Diode-based circuit 8
  • a first detection circuit 8 and a second detection circuit 9 which may be present alternatively or simultaneously, and which are each adapted to generate a signal representing the time at which the inductor Ström II through the buck Inductance Ll-A crosses the zero line, which is referred to in English as 'zero crossing ZX'.
  • This circuit has two diodes D20, D21, two ohmic resistors R20, R21 and a Zener diode Z20.
  • This circuit is used to generate a signal which detects the beginning of the dead time tdead, ie the first drop of the inductor current II to zero.
  • the current II decreases from a maximum value ⁇ to zero.
  • the voltage at the midpoint (measured from the cathode of the diode D1 to ground potential) is at the logic low level, which means that the signal ZCD_l_filtered, which can be supplied to a pin of a microcontroller or ASIC, for example, during this time is also pulled low because the diode D20 is conductive.
  • the control circuit 2 outputs a ZCD_Filter_l_out signal at a logic high (high), which in turn pulls the signal ZCD l filtered high via the diode D21 has the advantage that thereby re-starting the dead time counter is avoided, ie restarting at further rising edges of the signal ZCD_l_filtered be prevented or hidden.
  • This circuit is used to determine possible reclosing times for the switch M1 (in the event that the reclosing times are not preliminarily stored due to known component dimensions).
  • the voltage across inductor L1-A is positive when the slope of current II is positive and the voltage is negative when the slope of the current is negative. Due to the 90 ° phase shift between current and voltage at an inductance, a voltage maximum occurs at the positive going zero crossing of the current, or a voltage minimum at the negative going zero crossing of the current. A reversal of the voltage takes place in each case at the maximum and minimum of the current through the inductance. In order to achieve minimum switching losses, so therefore the switch Ml should be turned on again at the positive going zero crossing of the current II. A reversal of the voltage across the inductance Ll-A occurs at a current minimum before this zero-crossing with a positive gradient.
  • the detected time in this approach according to the invention is half a half wave of the oscillation of the inductor current II after its first zero crossing before the optimal time for reconnection.
  • an auxiliary winding Ll-B is provided, which is coupled to the actual inductance Ll-A in the power path of the converter. If the voltage above LI is positive, diode D30 will turn off and switch Q30 will be on. The signal ZCD_1 which can be supplied to a pin of a control unit, in particular the control circuit 2, is thus logically low. If the voltage across LI is negative, diode D30 will conduct and switch Q30 will be off. Thus, the signal ZCD_1 is pulled up to the potential high (high) via the resistor R30, which corresponds to the potential of a supplied low-voltage power supply VDD.
  • the falling edges of the signal ZCD_1 represent the times at which the voltage across the inductance changes from negative to positive, which means that the current II has a minimum, followed by a positive zero crossing with a certain time delay of the inductance current takes place.
  • the current counter value of the dead time counter is stored in a memory "valley array”.
  • the rule algorithm that calculates the target dead time now selects one of the values in this filed valley array.
  • the control algorithm (block B) then selects, based on tdead nom, the valley which is closest to tdead nom. This tdead_vaiiey is then set ("New Tdead" in FIG. 4).
  • the calculated dead time will differ from the available Valley values, such that a new turn-off threshold I pe ak_comp for the peak current is calculated (or a new tone time is calculated) to provide the desired average current Lvgjom with the selected dead time tdead_vaiiey to reach.
  • measurement paths are provided, which can be defined as follows:
  • R40, R41 and C40 are used for LED voltage measurement.
  • R50, C50 These are used for averaging the measuring voltage across the measuring resistor Rl, in order to determine the average current, which can then be fed via an AD converter of the control circuit.
  • R51 This signal is compared by means of a comparator with a set threshold. When the threshold is reached, the switch Ml is turned off. So this is one possible implementation for a threshold shutdown.
  • turn-off delays are caused for example by a so-called propagation delay of the comparator and the gate driver, as well as finite edge slopes and parasitic capacitances.
  • the actually flowing peak current and the thus resulting time average is thus falsified, this deviation depending on the steepness of the current dl / dt.
  • typical turn-off delays may be in the range of several hundred nanoseconds, which may, for example, cause the peak current at 1.2 amps to be turned off instead of a desired turn-off value of 1 amp at a steep current increase dl / dt of the switch current.
  • the steepness of the current increase through the primary-side switch 5 depends inter alia on the voltage V LED across the load, for example an LED load.
  • V LED voltage across the load
  • the current increase is very steep, resulting in a large overshoot of the peak current, conversely, a high LED voltage V LED leads to a less steep increase in current when the primary side is on Switch of the clocked converter, which in turn leads to a lower overshoot of the peak current.
  • the switch-off threshold value is now set as a function of the detected output-side voltage, in particular a measured LED voltage.
  • the threshold value I pea k is reduced relatively little (for example, from 1 ampere to 0.95 amperes), while it is significantly reduced at a lower detected LED voltage V LED , for example, from a nominal value of 1 amp to 0.9 amperes, so that in fact results in the desired peak current of 1 ampere in both cases due to the off-delay.
  • the requirements for the accuracy of keeping the peak current are particularly large when the peak current is calculated, for example, as described above.
  • FIG. 7 shows how a different peak current can actually be set at two different operating points (slope of the current through the buck inductance when the switch 5 is conductive).
  • a switch-off threshold value of 1A depending on the operating point, a real peak current of 1.2A or 1.1A can be set, for example, even if the switch-off delay (which is dependent on the component) is constant.
  • the threshold values are now varied according to the operating point in order to compensate for the switch-off delay.
  • the shutdown threshold is set to 0.9A and in the second case to 0.8A. Due to the switch-off delay, in both scenarios (with a different slope of the current through the inductance 4), the desired (for example, pre-calculated) peak current of 1A is obtained, and the switch-off delay is thus compensated in this regard.
  • An example is calculated peak current I pea k_com P , which is calculated, for example, from a desired dimming value (current average) and a selected dead time, said calculated peak current I pea k_com P by the output signal of a controller, such as a Integral controller, deltalpk is easily corrected. The sum of both paths then gives a desired peak current of, for example, 1A. Depending on the operating point, a correction value I p k_overshoot is now deducted from this calculated desired target value. With small LED voltages with steep current increase (large dl / dt) is corrected with a relatively high value (for example, the value I p k_overshoot be 0.2A). At high LED voltages V LED with flat current increase (low dl / dt) is corrected accordingly less strong (then, for example, the value of I p k_overshoot be 0.1A).
  • the digital peak current value is converted in a DA converter DAC into an analog threshold value.
  • the threshold value is passed to a comparator, which compares the threshold value with the actual voltage drop at, for example, a shunt, the voltage drop the shunt reflects the current through the buck inductor.
  • the comparator switches its output, which leads to switching off the Buck switch 5.
  • the factor "I pe ak_overshoot" is determined as a function of the measured LED voltage V LED , for example according to an analytical function or a stored table which reproduces the function, for example, according to FIG. 10.
  • Information regarding the slope of the current increase by the switch 5. the use of at least one of the value of the inductance 4, the input voltage Vbus or the output voltage V LED and any combinations thereof.
  • the relationship between the correction value I pea k_overshoot and the selected parameter is linear in an example of FIG. 10 (linear dependence of the voltage across the LED segment V LED ). However, this dependence can not be linear.
  • the correction factor I pea k_overshoot is calculated continuously or at regular intervals.
  • the buck current increases by a value I p kov, so that, with a constant switch-off delay, the overshoot value I p k_ov to be corrected can be calculated at any time according to the following formula:
  • L denotes the value of the inductance 4 of the clocked converter.
  • 11 shows a time characteristic of a predetermined load current I aV g_nom.
  • the abscissa represents the time t for a time segment of a dimming process of a light path 6, for example a slow dimming for 4 seconds for a transition from a dimming level (dimming value) 100% to a dimming level of 1% percent.
  • Fig. 11 only a portion of the dimming process is shown in the drawing. For the complete dimming process, the value reached on the y-axis at the end of the dimming process has fallen to exactly 1% of the initial value.
  • the target brightness value in FIG. 11 is plotted in the form of a nominal average load current Lvg j om.
  • the dimming process for the light path 6 is characterized by a desired falling curve 10 of the nominal average load current Lvg j om, the clocked converter 1 provides at its output as a nominal value Lvg j om possible actual value of the average load current Lvg.
  • FIG. 12 shows values for the dead time tdead_do m and set values for the dead time tdead assigned to the time profile 10 of the predetermined load current I aV g_nom according to FIG. 11.
  • the time t for the time interval of the dimming process according to FIG. 11 is plotted on the abscissa of FIG.
  • the calculated dead time tdead j om shows a gradient 11 rising in the opposite direction to the falling curve 10 of the nominal average load current Iavg j om.
  • the clocked converter 1 generates the sinking load current I aV g with an increase in the calculated dead time tdeadjom. Since a restart of the switch 5 is to take place only at discretely spaced time points, an increase of the actually set dead time tdead takes place only in discrete steps. This leads to the stepped rising curve 12 of the dead time tdead shown in FIG. 12 over the time t for the time segment of the dimming process of the light path 6 in FIG. 11.
  • FIG. 12 shows that always the, the calculated (nominal) dead time tdead j om nearest zero crossing of the inductor current II is selected, and thus determines the current value for tdead.
  • the threshold lies at 50% of a period of the oscillation process of the clocked converter 1. In terms of time, this corresponds exactly to half the distance between two "valleys", ie two negative half-waves of the inductance current II or half a period of the resonance oscillation.
  • FIG. 12 also shows that in the case of three transitions 12.1 of the total of four transitions of the curve 12 shown, the dead time tdead enters a next higher value of the dead time tdead toggle occurs between two adjacent positive-gradient zero crossings.
  • This short-term change between adjacent discrete reconnection times is based on non-ideal determination of the peak current I pea k in conjunction with a frequent change of the positive zero crossing of the inductance current II closest to the calculated reclosing time. Also, the detection of the valleys is subject to fluctuations. The detected possible reclosing times fluctuate. This results in short-term fluctuations in the average load current Lvg and thus ILED, which can be perceptible, for example, as flickering of the light output of the illuminant section 6.
  • a change in the reconnection time to a calculated new reconnection time other than a current reclosing time is performed only if the new reconnection time of the switch deviates from the current reconnection time by a time greater than a predetermined value, in particular greater than 50% of the period of the resonance oscillation a frequent short-term change of the reconnection time is prevented by this hysteresis.
  • a stability of the average load current is improved, so that, for example, a light output of the light source can be flicker-free.
  • FIG. 13 shows values for a dead time tdead_d om and set values for the dead time tdead, assigned to the time characteristic of the predetermined load current Lvg j om according to FIG. 11, according to a further improved embodiment with a hysteresis.
  • the predetermined value is set to 10/16 and 62.5%, respectively, in the embodiment of FIG. This means that for the dimming process in the direction of a lower light output or a lower average load current I aV g, one of the calculated dead time tdeadjom remains adjacent smaller dead time tdead until the calculated dead time tdead j om is greater than 10/16 of the period the resonance oscillation deviates from the currently selected and set dead time tdead.
  • Fig. 13 shows in the illustrated step-shaped curve 13 of the dead time tdead reached by means of the introduced hysteresis suppression of choiritigen alternation at the transitions 12.1 of the course 12 of the dead time tdead to a next higher value of the dead time tdead in Fig. 12.
  • an improved since flicker-free light output of the light-emitting means path 6 is achieved by an increased temporal stability of the average load current I aV g.
  • the hysteresis introduced in the previously discussed embodiment of the clocked converter 1 further causes different operating points of the clocked converter 1 to result for a specific dimming value of the load current I aV g_nom.
  • the different operating points are each characterized by a set dead time tdead and a set peak current I pea k. This will be further explained with reference to FIG. 16.
  • FIG. 14 shows two further time courses of the predetermined average load current Ia V g_nom for the clocked converter 1 over the time t plotted in the direction of the abscissa.
  • the predetermined average load current is given in representative digital values.
  • the first curve 14 represents a dimming process in the direction of a lower average load current I aV g.
  • the second curve 15 represents a dimming process in the direction of a higher average load current I aV g.
  • the first curve 14 of the predetermined average load current I aV g_nom and the second curve 15 of the predetermined average load current I aV g_nom intersect at an intersection point 16.
  • Intersection point 16 is characterized by an assigned value of the predetermined average load current I aV g_nom which is both part of the curve 14 as well as the course 15 of the dimming operations of the average load current I aV g.
  • FIG. 15 shows values for the dead time tdead_dom and set values for the dead time tdead with a hysteresis for the time profiles 14, 15 of the predetermined load current according to FIG. 14.
  • the first curve 16 of the calculated average dead time tdeadjom represents the calculated dead time d nom for the second curve 15 of the predetermined average load current Iavg j om for a dimming operation in the direction of a higher average load current I aV g.
  • the curve 17 is the actually set dead time tdead associated with the first curve 16 of the calculated average dead time d nom.
  • the dead time tdead decreases with increasing time t.
  • the dimming value increases towards a higher light output of the light-emitting means path 6 due to an increasing average load current I aV g.
  • the second curve 18 of the calculated average dead time tdead j om represents the calculated dead time for the first curve 14 of the predetermined average load current Iavg j om for a dimming operation in the direction of a reduced average load current I aV g. This reduces the dimming value towards a weaker light output of the illuminant path 6 due to a decreasing average load current I aV g.
  • the curve 19 is the actually set dead time tdead to the second curve 18 of the calculated average dead time tdead j om corresponding to a dimming operation towards smaller dimming values.
  • a change of the selected zero crossing of the inductance current II as a reclosing time of the switch 5 is made by the control circuit 2, if the time interval between the calculated theoretical reconnection time and the currently selected reclose time greater than the product of hysteresis factor k and the period of the resonant oscillation of the clocked converter 1 is.
  • the period of the resonant oscillation of the clocked converter 1 corresponds to the time interval between two in-phase current values of the inductor current II during the resonant oscillation during the dead time tdead.
  • the time interval of two in-phase current values of the inductor current II during resonant oscillation corresponds to a height of one stage of the curves 17, 19 of the set dead time tdead in the ordinate direction of Fig. 15.
  • the height 20 in Fig. 15 is counter values , according to a normalized time.
  • a change of the selected reconnection time takes place in the exemplary embodiment selected in FIG. 15 with a hysteresis factor k of approximately 0.625. This can be seen from the fact that the change does not take place at half, ie 0.5 or 50% of the step height 20, which would correspond to the selection of the zero crossing with a positive gradient of the inductance current II temporally closest to a calculated switch-on time, but after approximately 0.625 or 62.5% corresponding to the first distance 21 at the curve 16 for decreasing dead time tdead or the distance 22 of the curve 18 for increasing dead time tdead.
  • 16 shows calculated values for the dead time tdead_d om and set values for the dead time tdead with a hysteresis for time-increasing and time-decreasing values of the predetermined load current in a section.
  • a predetermined average load current I aV g with an average current of 100 mA is provided by the clocked converter 1. This can be done on the one hand in a first operating point, which is characterized by a dead time tdead with a duration of four periods of resonance oscillation. There is thus a reconnection of the switch 5 of the clocked converter 1 at a zero crossing of the inductor current Ibuck with a positive gradient of the course of the coil current II in the fourth period of the resonant oscillation after turning off the switch 5. Turning off the switch 5 takes place in the first operating point upon reaching a peak current I pea k with a current of 300 mA.
  • a second operating point is characterized by a dead time tdead with a duration of five periods of the resonance oscillation. There is thus a reconnection of the switch 5 of the clocked converter 1 at a zero crossing of the inductor current II with a positive gradient of the course of the inductor current II in the fifth period of the resonant oscillation after turning off the switch 5. Turning off the switch 5 takes place in the second operating point upon reaching a peak current I pea k with a current of 330 mA.
  • FIG. 16 shows a value 26 of the dead time tdead of the rising dimming curve ""
  • the control circuit 2 sets the dead time with a hysteresis in FIG. 16.
  • the hysteresis may be 62.5% of the period of the resonant oscillation of the clocked converter 1.
  • the corresponding value 26 of the dead time tdead can alternatively be achieved via a falling dimming curve of the average load current I av g and a correspondingly increasing curve 23 of the dead time tdead_nom. This results in a step-increasing course 24 of the dead time tdead.
  • the control circuit 2 can determine the switch-off threshold for the peak current I pea k corresponding to the different values for the dead time tdeadi 28, tdead2 29 to I pea ki or I pe ak2 to the same setpoint I av g _nom with the current actual value of the load current I. reach aV g.
  • hysteresis factors each having different values for increasing and decreasing dimming characteristics of the load current I av g can be selected.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Circuit Arrangement For Electric Light Sources In General (AREA)
  • Dc-Dc Converters (AREA)

Abstract

L'invention concerne un dispositif de commande permettant de commander de manière modulable des moyens d'éclairage, en particulier une ou plusieurs DEL. Le dispositif de commande comporte un circuit de commande et un circuit convertisseur pourvu d'un élément accumulateur d'énergie et d'au moins un commutateur qui est commandé par le circuit de commande ; le circuit de commande est conçu pour commander, par actionnement du commutateur, le circuit convertisseur au choix au moins en mode critique ou en mode avec fonctionnement en courant intempestif. En fonctionnement intempestif, l'instant de reconnexion est défini par incréments discrets dans la zone d'un passage à zéro croissant du courant traversant l'élément accumulateur. Le circuit de commande est adapté pour effectuer, en cas de variation incrémentielle de l'instant de reconnexion, un réglage d'un paramètre ayant une influence sur la puissance des moyens d'éclairage par modification directe ou indirecte de la durée de connexion du commutateur.
PCT/EP2018/075050 2017-09-29 2018-09-17 Dispositif de commande de lampe ayant un convertisseur en dcm WO2019063333A1 (fr)

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ATGM218/2017U AT16163U1 (de) 2017-09-29 2017-09-29 Lampenbetriebsgerät
DE102017221786.3A DE102017221786A1 (de) 2017-09-29 2017-12-04 Lampenbetriebsgerät mit Konverter im DCM
DE102017221786.3 2017-12-04
DE202018104924.0U DE202018104924U1 (de) 2017-09-29 2018-08-28 Lampenbetriebsgerät mit Konverter in DCM
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EP3758204A1 (fr) 2019-06-28 2020-12-30 Infineon Technologies Austria AG Procédé de commande d'un commutateur électronique dans un circuit convertisseur de puissance et circuit convertisseur de puissance
DE102020103921B4 (de) 2020-02-14 2021-12-30 Vossloh-Schwabe Deutschland Gmbh Betriebsvorrichtung und Verfahren zum Betreiben einer Leuchtmittelanordnung

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US20150244272A1 (en) * 2014-02-26 2015-08-27 Infineon Technologies Austria Ag Valley to valley switching in quasi-resonant mode for driver
US20150303796A1 (en) * 2014-04-17 2015-10-22 Cirrus Logic, Inc. Systems and methods for valley switching in a switching power converter
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