WO2018157114A1 - Filtre à ondes acoustiques à élimination améliorée - Google Patents

Filtre à ondes acoustiques à élimination améliorée Download PDF

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Publication number
WO2018157114A1
WO2018157114A1 PCT/US2018/019899 US2018019899W WO2018157114A1 WO 2018157114 A1 WO2018157114 A1 WO 2018157114A1 US 2018019899 W US2018019899 W US 2018019899W WO 2018157114 A1 WO2018157114 A1 WO 2018157114A1
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WO
WIPO (PCT)
Prior art keywords
capacitive
interdigitated
fingers
acoustic
resonator
Prior art date
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PCT/US2018/019899
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English (en)
Inventor
Kurt F. Raihn
Gregory L. Hey-Shipton
Original Assignee
Resonant Inc.
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from US15/443,740 external-priority patent/US10305447B2/en
Application filed by Resonant Inc. filed Critical Resonant Inc.
Priority to CN201880012977.1A priority Critical patent/CN110506391B/zh
Priority to DE112018001025.2T priority patent/DE112018001025T5/de
Publication of WO2018157114A1 publication Critical patent/WO2018157114A1/fr

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/64—Filters using surface acoustic waves
    • H03H9/6423—Means for obtaining a particular transfer characteristic
    • H03H9/6433—Coupled resonator filters
    • H03H9/6483—Ladder SAW filters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/54—Filters comprising resonators of piezoelectric or electrostrictive material
    • H03H9/542—Filters comprising resonators of piezoelectric or electrostrictive material including passive elements
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/54—Filters comprising resonators of piezoelectric or electrostrictive material
    • H03H9/56—Monolithic crystal filters
    • H03H9/564—Monolithic crystal filters implemented with thin-film techniques
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/54—Filters comprising resonators of piezoelectric or electrostrictive material
    • H03H9/56—Monolithic crystal filters
    • H03H9/566—Electric coupling means therefor
    • H03H9/568—Electric coupling means therefor consisting of a ladder configuration

Definitions

  • the present inventions generally relate to microwave filters, and more
  • Electrical filters have long been used in the processing of electrical signals.
  • such electrical filters are used to select desired electrical signal frequencies from an input signal by passing the desired signal frequencies, while blocking or attenuating other undesirable electrical signal frequencies.
  • Filters may be classified in some general categories that include low-pass filters, high-pass filters, band-pass filters, and band-stop filters, indicative of the type of frequencies that are selectively passed by the filter.
  • filters can be classified by type, such as Butterworth, Chebyshev, Inverse Chebyshev, and Elliptic, indicative of the type of bandshape frequency response (frequency cutoff characteristics) the filter provides relative to the ideal frequency response.
  • radio frequency (RF) filters are conventionally used in telecommunications systems where information is transmitted over wireless links, for example, in the RF front-ends of base stations, mobile devices, satellite transceivers and ground receivers, Internet of Things (IOT) devices, laptop computers and tablets, fixed point links, and other similar systems, to filter out or block RF signals in all but one or more predefined bands.
  • IOT Internet of Things
  • the frequency range from approximately 500-3,500 MHz.
  • band 2 ⁇ 1 800-1 900 MHz
  • Band 4 -1 700-21 00 MHz
  • Band 5 -800-900 MHz
  • Band 1 3 Band 1 3
  • Band 1 7 Band 1 7
  • Performance enhancements to the RF filters in a telecommunications system can have broad impact to system performance.
  • Improvements realized can come in several ways: cell size, battery life, data rates, network capacity, cost, security, reliability, etc. These improvements can be realized at many levels of the wireless system both separately and in combination, for example, at the RF module, RF transceivers, mobile or fixed sub-system, or network levels.
  • RF filters are generally built using two circuit building blocks: a plurality of resonators, which store energy very efficiently at a resonant frequency (which may be a fundamental resonant frequency fo or any one of a variety of higher order resonant frequencies fi-fn); and couplings, which couple electromagnetic energy between the resonators to form multiple reflection zeros providing a broader spectral response.
  • a resonant frequency which may be a fundamental resonant frequency fo or any one of a variety of higher order resonant frequencies fi-fn
  • couplings which couple electromagnetic energy between the resonators to form multiple reflection zeros providing a broader spectral response.
  • a four-resonator filter may include four reflection zeros. The strength of a given coupling is determined by its reactance (i.e., inductance and/or capacitance).
  • the relative strengths of the couplings determine the filter shape, and the topology of the couplings determines whether the filter performs a band-pass or a band-stop function.
  • the resonant frequency fo is largely determined by the inductance and capacitance of the respective resonator.
  • the frequency at which the filter is active is determined by the resonant frequencies of the resonators that make up the filter.
  • Each resonator must have very low internal resistance to enable the response of the filter to be sharp and highly selective for the reasons discussed above. This requirement for low resistance tends to drive the size and cost of the resonators for a given technology.
  • the duplexer a specialized kind of filter is a key component in the front-end of mobile devices.
  • Modern mobile communications devices transmit and receive at the same time (using LTE, WCDMA or CDMA) and use the same antenna.
  • the duplexer separates the transmit signal, which can be up to 0.5 Watt power, from the receive signal, which can be as low as a pico-Watt.
  • the transmit and receive signals are modulated on carriers at different frequencies allowing the duplexer to select them, even so the duplexer must provide the frequency selection, isolation and low insertion loss in a very small size often only about two millimeters square.
  • the front-end receive filter preferably takes the form of a sharply defined bandpass filter to eliminate various adverse effects resulting from strong interfering signals at frequencies near the desired received signal frequency. Because of the location of the front-end receiver filter at the antenna input, the insertion loss must be very low so as to not degrade the noise figure. In most filter technologies, achieving a low insertion loss requires a corresponding compromise in filter steepness or selectivity.
  • acoustic resonator technology such as surface acoustic wave (SAW), bulk acoustic wave (BAW), and film bulk acoustic resonator (FBAR) technologies.
  • SAW surface acoustic wave
  • BAW bulk acoustic wave
  • FBAR film bulk acoustic resonator
  • the equivalent circuit of an acoustic resonator has two resonances closely spaced in frequency called the "resonance" frequency and the "anti-resonance” frequency (see K.S. Van Dyke, Piezo- Electric Resonator and its Equivalent Network Proc. IRE, Vol. 16, 1928, pp.742-764).
  • Such acoustic resonator filters have the advantages of low insertion loss (on the order of 1 dB at the center frequency), compact size, and low cost compared to equivalent inductor/capacitor resonators. For this reason, acoustic resonator implementations are often used for RF filtering applications in the front-end receive filter of mobile devices.
  • Acoustic resonators are typically arranged in a ladder topology (alternating series and shunt resonators) in order to create band pass filters.
  • Acoustic ladder filters have been very successful for handset applications, with more than a billion units currently sold each year.
  • the recent trend in wireless technology towards multifunctional devices and a more crowded electromagnetic spectrum requires filters for ever more bands with sharper line shapes, while simultaneously demanding reduction in the size, cost, and power consumption.
  • a typical acoustic resonator has a plurality of interdigitized fingers (e.g., 80-100 fingers) that reflect acoustic waves back and forth between the fingers.
  • the frequency band over which the acoustic reflections between the fingers add in phase to create the resonance may be referred to as the "Bragg Band.”
  • a discontinuity feature in the frequency response occurs at the upper edge of the Bragg Band, i.e., the highest frequency at which the acoustic reflections add in phase.
  • This Bragg resonance can distort the high side of the passband of the bandpass filter, resulting in excessive loss at these frequencies.
  • the performance of the filter may be compromised if this discontinuity feature falls within the passband, it is important to ensure that the discontinuity feature falls well outside of the passband of the filter.
  • an acoustic filter comprises a piezoelectric layer, and an acoustic resonator structure monolithically disposed on the piezoelectric layer, and a lumped capacitive structure monolithically disposed on the piezoelectric layer and being electrically coupled (e.g., in parallel) to the acoustic resonator structure.
  • the piezoelectric layer may be, e.g., a piezoelectric substrate or may be thin-film piezoelectric monolithically disposed on a non-piezoelectric substrate.
  • the acoustic resonator structure comprises an arrangement of planar interdigitated resonator fingers
  • the lumped capacitive structure comprises an arrangement of planar interdigitated capacitive fingers.
  • the interdigitated capacitive fingers and interdigitated resonator fingers have orientations that are parallel, oblique, or orthogonal to each other.
  • the acoustic filter further may further comprise a metalized signal plane monolithically disposed on the piezoelectric layer, and a metalized ground plane monolithically disposed on the piezoelectric layer.
  • each of the acoustic resonator structure and the capacitive resonator structure is electrically coupled between the signal plane and the ground plane.
  • the lumped capacitive structure may be least partially nested, and perhaps fully nested, within at least one of the signal plane and the ground plane.
  • the acoustic filter may further comprise a metalized input signal plane portion monolithically disposed on the piezoelectric layer, and a metalized output signal plane portion monolithically disposed on the piezoelectric layer.
  • each of the acoustic resonator structure and the capacitive resonator structure is electrically coupled between the input signal plane portion and the output signal plane portion.
  • the lumped capacitive structure may be least partially nested, and perhaps fully nested, within at least one of the input signal plane portion and the output signal plane portion.
  • each of at least one, and perhaps all, of the interdigitated capacitive fingers has an edge that is entirely continuous. In one embodiment, the entirely continuous edge of each of the
  • interdigitated capacitive finger(s) has one or more undulations (e.g., a single undulation or a plurality of undulations).
  • the interdigitated capacitive fingers and interdigitated resonator fingers have orientations that are parallel to each other, but no finite portion of an entire edge along the length of each of at least one, and perhaps all, of the interdigitated capacitive fingers is parallel to any finite portions of entire edges along the lengths of the interdigitated resonator fingers.
  • the entire edge of each of the interdigitated capacitive finger(s) has one or more undulations (e.g., a single undulation or a plurality of undulations).
  • the entire edge of each of the interdigitated capacitive finger(s) has at least one sawtooth (e.g., a single sawtooth or a plurality of sawteeth).
  • each of at least one, and perhaps all, of the interdigitated capacitive fingers comprises a plurality of interdigitated capacitive sub-fingers.
  • the plurality of interdigitated capacitive sub-fingers has an orientation that is orthogonal to the orientation of the interdigitated capacitive finger(s).
  • each of the sub-fingers has an edge that forms at least one undulation.
  • each of the sub-fingers has an edge that forms at least one sawtooth.
  • no finite portion of an edge along the length of each of at least one of the interdigitated capacitive sub-fingers is parallel to any finite portions of edges along the lengths of the interdigitated resonator fingers.
  • each of at least one of the interdigitated capacitive fingers has a length/width ratio of less than two, and preferably less than one.
  • each of the interdigitated capacitive finger(s) forms a single undulation.
  • each of the interdigitated capacitive finger(s) forms a single sawtooth.
  • FIG. 1 is a block diagram of a wireless telecommunications system
  • FIG. 2 is a schematic diagram of a conventional microwave acoustic filter arranged in an Nth order ladder topology
  • Fig. 3 is a schematic diagram illustrating the transformation of an acoustic resonator of the acoustic filter of Fig. 2 into an equivalent modified Butterworth-Van Dyke (MBVD) model;
  • Fig. 4 is a schematic diagram illustrating the MBVD equivalent circuit of the conventional acoustic filter of Fig. 2;
  • FIG. 5 is a schematic diagram illustrating an enhanced microwave acoustic filter constructed in accordance with one embodiment of the present inventions
  • Fig. 6 is a schematic diagram illustrating the MBVD equivalent circuit of the enhanced acoustic filter of Fig. 5;
  • Fig. 7 is a frequency response plot comparing the passbands of the
  • Fig. 8 is a frequency response plot comparing the upper edges of the passbands of the conventional acoustic filter of Fig. 3 and the enhanced acoustic filter of Fig. 5;
  • Fig. 9 is another frequency response plot comparing the upper edges of the passbands of the conventional acoustic filter of Fig. 3 and the enhanced acoustic filter of Fig. 5;
  • Fig. 10 is a frequency response plot comparing the out-of-band rejection of the conventional acoustic filter of Fig. 3 and the enhanced acoustic filter of Fig. 5;
  • Fig. 11a is a plan view of an actual in-line acoustic resonator with additional capacitive elements fabricated for use in the enhanced acoustic filter of Fig. 5;
  • Fig. 11 b is a plan view of a portion of the actual acoustic resonator of Fig. 11 ;
  • Fig. 12 is a plan view of an actual in-shunt acoustic resonator with an additional capacitive element fabricated for use in the enhanced acoustic filter of Fig. 5;
  • Fig. 13a is a schematic diagram of a conventional single section band-pass acoustic filter circuit
  • Fig. 13b is a schematic diagram illustrating the MBVD equivalent circuit of the conventional acoustic filter circuit of Fig. 13a;
  • Fig. 13c is a frequency response plot of the passband of the conventional acoustic filter of Fig. 13b;
  • Figs. 14a-14c are frequency response plots of passbands of varying
  • Fig. 15 is a frequency response plot comparing the passbands of Figs. 14a- 14c;
  • Fig. 16a is a schematic diagram of an enhanced single section band-pass acoustic filter circuit, wherein a capacitive element is added in parallel with the in-shunt resonator;
  • Fig. 16b is a schematic diagram illustrating the MBVD equivalent circuit of the enhanced acoustic filter circuit of Fig. 16a;
  • Fig. 16c is a frequency response plot of the passband of the enhanced acoustic filter of Fig. 16b;
  • Fig. 17a is a schematic diagram of an enhanced single section band-pass acoustic filter circuit, wherein a capacitive element is added in parallel with the in-line resonator;
  • Fig. 17b is a schematic diagram illustrating the MBVD equivalent circuit of the enhanced acoustic filter circuit of Fig. 17a;
  • Fig. 17c is a frequency response plot of the passband of the enhanced acoustic filter of Fig. 17b;
  • Fig. 18 is a frequency response plot comparing an enhanced acoustic resonator with a conventional acoustic resonator
  • Fig. 19a is a plan view of one embodiment of a lumped capacitive structure that can be used in the acoustic filter of Fig. 5, particularly showing the lumped capacitive structure oriented orthogonal to the lumped resonant structure;
  • Fig. 19b is a plan view of the embodiment of the lumped capacitive structure of Fig. 19a, particularly showing the lumped capacitive structure oriented parallel to the lumped resonant structure;
  • Fig. 19c is a plan view of the embodiment of the lumped capacitive structure of Fig. 19a, particularly showing the lumped capacitive structure oriented oblique to the lumped resonant structure;
  • Fig. 20a is a plan view of another embodiment of a lumped capacitive structure that can be used in the acoustic filter of Fig. 5, particularly showing the lumped capacitive structure oriented orthogonal to the lumped resonant structure;
  • Fig. 20b is a plan view of the embodiment of the lumped capacitive structure of Fig. 20a, particularly showing the lumped capacitive structure oriented parallel to the lumped resonant structure;
  • Fig. 20c is a plan view of the embodiment of the lumped capacitive structure of Fig. 20a, particularly showing the lumped capacitive structure oriented oblique to the lumped resonant structure;
  • Fig. 21a is a plan view of still another embodiment of a lumped capacitive structure that can be used in the acoustic filter of Fig. 5, particularly showing the lumped capacitive structure oriented orthogonal to the lumped resonant structure;
  • Fig. 21 b is a plan view of the embodiment of the lumped capacitive structure of Fig. 21a, particularly showing the lumped capacitive structure oriented parallel to the lumped resonant structure;
  • Fig. 21c is a plan view of the embodiment of the lumped capacitive structure of Fig. 21a, particularly showing the lumped capacitive structure oriented oblique to the lumped resonant structure;
  • Fig. 22a is a plan view of yet another embodiment of a lumped capacitive structure that can be used in the acoustic filter of Fig. 5, particularly showing the lumped capacitive structure oriented orthogonal to the lumped resonant structure;
  • Fig. 22b is a plan view of the embodiment of the lumped capacitive structure of Fig. 22a, particularly showing the lumped capacitive structure oriented parallel to the lumped resonant structure; and
  • Fig. 22c is a plan view of the embodiment of the lumped capacitive structure of Fig. 22a, particularly showing the lumped capacitive structure oriented oblique to the lumped resonant structure.
  • the present disclosure describes a design technique that enhances the rejection on one or both sides of a passband of an acoustic wave (AW) microwave filter, such as, e.g., a surface acoustic wave (SAW), bulk acoustic wave (BAW), film bulk acoustic resonator (FBAR), or microelectromechanical system (MEMS) filter.
  • AW acoustic wave
  • SAW surface acoustic wave
  • BAW bulk acoustic wave
  • FBAR film bulk acoustic resonator
  • MEMS microelectromechanical system
  • Narrow band filters/duplexers can be designed, which increases the number of designs possible for a given piezoelectric material.
  • the acoustic microwave filter may operate at microwave frequencies in the range of 300 MHz to 300 GHz, but is most applicable at frequencies in the range of 300 MHz to 10 GHz, and most particularly, at frequencies in the range of 500 MHz to 3.5 GHz.
  • the AW microwave filter described herein exhibits a frequency response with a single passband, which is particularly useful in telecommunication system duplexers where a passband with a closely spaced stopband is required.
  • a telecommunications system 10 for use in the front-end of devices that transmit or receive information over wireless links, such as, e.g., base stations, mobile devices, satellite transceivers and ground receivers, Internet of Things (IOT) devices, laptop computers and tablets, fixed point links, and other similar systems.
  • the telecommunications system 10 may include a transceiver 12 capable of transmitting and receiving wireless signals, and a controller/processor 14 capable of controlling the functions of the transceiver 12.
  • the transceiver 1 2 generally comprises a broadband antenna 16, a duplexer 18 having a transmit filter 24 and a receive filter 26, a
  • transmitter 20 coupled to the antenna 16 via the transmit filter 24 of the duplexer 18, and a receiver 22 coupled to the antenna 16 via the receive filter 26 of the duplexer 18.
  • the transmitter 20 includes an upconverter 28 configured for converting a baseband signal provided by the controller/processor 14 to a radio frequency (RF) signal, a variable gain amplifier (VGA) 30 configured for amplifying the RF signal, a bandpass filter 32 configured for outputting the RF signal at an operating frequency selected by the controller/processor 14, and a power amplifier 34 configured for amplifying the filtered RF signal, which is then provided to the antenna 16 via the transmit filter 24 of the duplexer 18.
  • RF radio frequency
  • VGA variable gain amplifier
  • the receiver 22 includes a notch or stopband filter 36 configured for rejecting transmit signal interference from the RF signal input from the antenna 16 via the receiver filter 26, a low noise amplifier (LNA) 38 configured for amplifying the RF signal from the stop band filter 36 with a relatively low noise, a tunable bandpass filter 40 configured for outputting the amplified RF signal at a frequency selected by the controller/processor 14, and a downconverter 42 configured for downconverting the RF signal to a baseband signal that is provided to the controller/processor 14.
  • the function of rejecting transmit signal interference performed by the stop-band filter 36 can instead be performed by the duplexer 18.
  • the power amplifier 34 of the transmitter 20 can be designed to reduce the transmit signal interference.
  • Fig. 1 the block diagram illustrated in Fig. 1 is functional in a nature, and that several functions can be performed by one electronic component or one function can be performed by several electronic components.
  • the functions performed by the up converter 28, VGA 30, bandpass filter 40, downconverter 42, and controller/processor 14 are oftentimes performed by a single transceiver chip.
  • the function of the bandpass filter 32 can be into the power amplifier 34 and the transmit filter 24 of the duplexer 18.
  • the exemplary design techniques described herein are used to design acoustic microwave filters for the front-end of the telecommunications system 10, and in particular, the transmit filter 24 of the duplexer 18, although the same techniques can be used to design acoustic microwave filters for the receive filter 26 of the duplexer 18 and for other RF filters. Furthermore, although the exemplary design techniques are described herein for use in the front-end of the telecommunications system 10, such exemplary design techniques can be used in other systems, such as radio and electronic and information warfare systems. [0062] Referring now to Fig. 2, one embodiment of a conventional band pass filter 100 will be described.
  • the filter 100 comprises a voltage source V, a source resistance S, a load resistance L, five series (or in-line) acoustic resonators Zsi-Zss, and four parallel (or in-shunt) acoustic resonators ZPI-ZP4.
  • each of the acoustic resonators Z may be described by a modified Butterworth-Van Dyke (MBVD) model 1 10.
  • MBVD models 1 10 may also describe SAW resonators, which may be fabricated by disposing interdigital transducers (IDTs) on a piezoelectric substrate, such as crystalline Quartz, Lithium Niobate
  • Each MBVD model 1 10 includes a motional capacitance Cm, a static capacitance Co, a motional inductance L m , and a resistance R.
  • capacitance Cm and motional inductance L m may result from the interactions of electrical and acoustical behavior, and thus, may be referred to as the motional arm of the MBVD model.
  • the static capacitance Co may result from the capacitance of the structure, and thus, may be referred to as the static (non-motional) capacitance of the MBVD model.
  • the resistance R may result from the electrical resistance of the acoustic resonator.
  • each of the acoustic resonators Z of the conventional filter 100 can be replaced with the MBVD model 1 10 illustrated in Fig. 3.
  • the in-band and out-of-band rejection of the conventional band-pass filter 100 can be significantly improved by adding at least one capacitive element in parallel with at least one of the acoustic resonators 1 10. For example, as illustrated in Fig.
  • an embodiment of an enhanced band-pass filter 200 with improved in-band and out-of-band rejection is similar to the conventional band-pass filter 100, with the exception that the enhanced band-pass filter 200 comprises a plurality of additional capacitive elements 120 (Csi-Css and CPI -CP4), each of which is in parallel with a respective one of the acoustic resonators (Zsi-Zss and Zpi-Zp 4 ).
  • Each of the capacitive elements 120 may, e.g., have a capacitance in the range of 0.5pF-2.0pF, specifically, in the range of 0.8pF-1 .5pF, and more specifically, in the range of 0.9pF- 1 .1 pF.
  • each of the acoustic resonators Z of the enhanced filter 200 can be replaced with the MBVD model 1 10 illustrated in Fig. 3.
  • a simulated frequency response of the enhanced band- pass filter 200 where the values Csi, Cs2, Cs3, Cs4, and Css are set to 0.4 pF, and the values of CPI , CP2, CP3, and CP4 are set to 0.0 pF, can be compared to the simulated frequency response of the conventional band-pass filter 100 in terms of insertion loss ⁇ S21 ⁇ 2 .
  • the enhanced frequency response has been aligned to the conventional acoustic filter at the upper -3dB insertion loss point for comparison of the slopes at the upper sides of the passbands.
  • the lower edge of the passband of the enhanced band-pass filter 200 is sharper than the lower edge of the nominal passband of the conventional band-pass filter 100.
  • the enhanced filter 200 has improved rejection over that of the conventional filter 100 at the upper edge of the passband.
  • the band-pass filter 200 has a substantially improved out-of-band rejection compared to that of the band-pass filter 100.
  • the capacitive elements 120 can be easily incorporated into an already existing conventional filter structure to create the band-pass filter 200. For example, as shown in Figs.
  • a portion of the filter 200a comprises a piezoelectric layer 252, and a metalized signal plane 254, acoustic resonator structure 258a, and a distinct lumped capacitive structure 260a all monolithically disposed on the piezoelectric layer 252.
  • the piezoelectric layer 252 may be, e.g., a piezoelectric substrate or may be monolithically disposed on a non-piezoelectric substrate, e.g., as a thin-film
  • the signal plane 254 comprises an input signal plane portion 254a and an output signal plane portion 254b.
  • the acoustic resonator structure 258a which corresponds to one of the in-line resonators Zs in Fig. 5, is electrically coupled between the input signal plane portion 254a and the output signal plane portion 254b, and in the illustrated embodiment, is directly connected to the input signal plane portion 254a and output signal plane portion 254b.
  • the acoustic resonator structure 258a comprises an interdigitated transducer (IDT) 262 formed by a plurality of interdigitated resonator fingers 266, for generating the acoustic waves, and an optional reflector 264 for reflecting the acoustic waves back into the IDT 262.
  • the lumped capacitive structure 260 is shown coupled to both ends of the acoustic resonator structure 258a, and in particular, is directly electrically coupled between the input signal plane portion 254a and output signal plane portion 254b, and in the illustrated embodiment, is directly connected to the input signal plane portion 254a and output signal plane portion 254b.
  • the lumped capacitive structure 260a comprises a plurality of interdigitated capacitive fingers 268.
  • the interdigitated capacitive fingers 268 are orthogonal to the interdigitated resonator fingers 266 to avoid excitation of acoustic waves.
  • another portion of the filter 200b comprises the piezoelectric layer 252 and the signal plane 254, a metallized ground plane 256, an acoustic resonator structure 258b, and a distinct lumped capacitive structure 260b all monolithically disposed on the piezoelectric layer 252.
  • the acoustic resonator structure 258b which corresponds to one of the in-shunt resonators Zp in Fig. 5, is electrically coupled between the signal plane 254 and the ground plane 256.
  • the acoustic resonator structure 258b comprises an IDT 262 formed by a plurality of interdigitated resonator fingers 266, for generating the acoustic waves, and an optional reflector 264 for reflecting the acoustic waves back into the IDT 262.
  • the lumped capacitive structure 260b is shown coupled to both ends of the acoustic resonator structure 258b, and in particular, is directly electrically coupled between the signal plane 254 and ground plane 256, and in the illustrated embodiment is directly connected to the signal plane 254 and ground plane 256.
  • the lumped capacitive structure 260b comprises a plurality of interdigitated capacitive fingers 268 that are orthogonal to the interdigitated resonator fingers 266 to avoid excitation of acoustic waves.
  • the interdigitated capacitive fingers 268 are at least partially nested within one or both of the signal plane 254 and ground plane 256 in order to utilize the limited space on the piezoelectric layer 252. In this manner, the lumped capacitive structure 260b can be more easily incorporated into an already existing filter layout. In the illustrated embodiment, the interdigitated capacitive fingers 268 are fully nested within the ground plane 256.
  • the interdigitated capacitive fingers 268 of the lumped capacitive structure 260a may be at least partially nested, and perhaps fully nested, within one or both of the input signal plane portion 254a and output signal plane portion 254b illustrated in Fig. 11 b.
  • a conventional single section band-pass filter circuit 300 may have a single acoustic resonator pair 302 consisting of a series (or in-line) acoustic resonator Zs and a parallel (or in-shunt) acoustic resonator Zp (Fig. 13a).
  • acoustic resonator pair 302 consisting of a series (or in-line) acoustic resonator Zs and a parallel (or in-shunt) acoustic resonator Zp (Fig. 13a).
  • Four of such acoustic resonator pairs can be found in the conventional band-pass filter 100 or the enhanced band-pass filter 200.
  • the acoustic resonators pairs may be identified in the filters 100, 200 as the resonators ZSI/ZPI , ZS2/ZP2, ZS3/ZP3, and Zs4/Zp 4 , or as the resonators ZPI/ZS2, ZP2/ZS3, ZP3/ZS4, and Zp 4 /Zss.
  • each of the acoustic resonators Z of the filter circuit 300 can be replaced with the BVD model 100' (i.e., the MBVD model 1 10 illustrated in Fig. 3 without the resistance R) to create an equivalent filter circuit, and modeled to create a passband having the profile represented by the
  • the resonance and anti-resonance frequencies of the series resonator Zs be respectively designated as ⁇ 3 ⁇ 43 ⁇ 4 and ⁇ 3 ⁇ 4 s
  • the resonance and anti-resonance frequencies of each of the shunt resonator Zpbe respectively designated as ⁇ 3 ⁇ 4- p and ⁇ 3 ⁇ 4 p .
  • ⁇ /? and UJA may be the respective resonance and anti-resonance frequencies for any given acoustic resonator, and gamma ⁇ may depend on a material's property, which may be further defined by:
  • the Quality factor (Q) for an acoustic resonator may be an important figure of merit in acoustic filter design, relating to the loss of the element within the filter.
  • Q of a circuit element represents the ratio of the energy stored per cycle to the energy dissipated per cycle.
  • the Q factor models the real loss in each acoustic resonator, and generally more than one Q factor may be required to describe the loss in an acoustic resonator.
  • Q factors may be defined as follows for the filter examples.
  • Circuit designers may typically characterize SAW resonators by resonant frequency OUR, static capacitance Co, gamma ⁇ , and Quality factor QL m .
  • QL m may be about 1000 for SAW resonators, and about 3000 for BAW resonators.
  • Typical ⁇ values may range from about 12 to about 18 for 42- degree X Y cut LiTaO3.
  • the transmission zeroes and reflection zeroes of the equivalent filter circuit of Fig. 13c can be computed as follows.
  • the transmission zero at the lower edge of the passband is effectively the resonance created by the circuit L m i and Cmi (i.e., the resonance of the acoustic resonator Zp) and is given by: .
  • This resonance creates an effective short circuit to the return path and no power is transmitted from the input to the output of the filter.
  • One reflection zero located in the passband is effectively the resonance created by the circuit Lmi, Cm-i, and Coi (i.e., the anti-resonance of the acoustic resonator Zp) and is given by:
  • Fb This resonance creates an effective open circuit to the return path, allowing power to be transmitted from the input to the output of the filter.
  • the other reflection zero located in the passband is effectively the resonance created by the circuit Lm2 and Cm2 (i.e., the resonance of the acoustic resonator Zs) and is given by:
  • a capacitance Csh is added in parallel with the shunt acoustic resonator Zp of the original filter circuit 300 (Fig. 16a), and the shunt acoustic resonator Zp is replaced with the BVD model 1 1 0' to create a new filter circuit 300' (Fig. 16b), which yields an
  • the transmission zero located at the lower edge of the passband is effectively the resonance created by the circuit L m i and Cmi (i.e., the resonance of the acoustic resonator Zp).
  • This transmission zero therefore, remains unchanged with the addition of the capacitance Csh, and is thus, located at the frequency F a given by equation [5] above.
  • One reflection zero located in the passband is effectively the resonance created by the circuit Lmi and Cmi (i.e., the resonance of the acoustic resonator Zp).
  • the relationship between the reflection zero Fb in equation [6] and the reflection zero Fb in equation [9] can be determined by assigning values to the lumped elements and
  • the transmission zero F a causes the reflection zero Fb to move down in frequency to Fb . Because the filter match is affected (degraded), the transmission zero F a can be moved higher to return the filter match to its original response, which also narrows the filter bandwidth. The resulting filter has a steeper skirt on the lower side of the passband.
  • a capacitance Cse is added in parallel with the in-line acoustic resonator Zs of the original filter circuit 300 (Fig. 17a), and the in-line acoustic resonator Zs is replaced with the MBVD model 1 10 illustrated in Fig. 3 to create a new filter circuit 300" (Fig. 17b), which yields an
  • the reflection zero located in the passband is effectively the resonance created by Lm2 and Cm2 (i.e., the resonance of the acoustic resonator Zs).
  • This reflection zero therefore, remains unchanged with the addition of the capacitance Cse, and is thus, located at the frequency F c given by equation [7] above.
  • the transmission zero located at the right edge of the passband is effectively the resonance created by the circuit Lm2, Cm2, and C02 (i.e., the anti-resonance of the acoustic resonator Zs), and is given by:
  • the relationship between the transmission zero Fd in equation [8] and the transmission zero Fd in equation [10] can be determined by assigning values to the lumped elements k
  • the result of adding a capacitance Cse in parallel with the in-line resonator Zs does not affect the location of the reflection zero Fc, but causes the transmission zero Fd to move down in frequency to FJ.
  • the filter match is not greatly affected, and the frequency response of the resulting filter is narrower and also steeper on the high side of the passband.
  • adding capacitance in parallel to shunt resonators of an acoustic filter narrows and steepens the lower edge of the passband, while adding capacitance in parallel to in-line resonators of an acoustic filter narrows and steepens the upper edge of the passband. It follows that adding capacitance in parallel to both the shunt and inline resonators of an acoustic filter narrows and steepens both edges of the passband.
  • narrower filters can be realized with piezoelectric materials that are normally used for wider bandwidth filters.
  • the passband insertion loss increases and the filter skirts become steeper.
  • the benefits of increasing the steepness of the passband can be realized by moving the entire filter up or down in frequency to maximize customer specification from band edge to rejection frequencies.
  • Adding capacitance in parallel to shunt resonators of the acoustic filter also effectively moves the upper Bragg Band resonance further from the passband.
  • the frequency response (in terms of the real impedance) of a conventional shunt resonator without an added parallel capacitance and the frequency response (in terms of real impedance) of an enhanced shunt resonator with an added parallel capacitance of 1 .0 pF can be compared.
  • the resonances of both the conventional acoustic resonator and the enhanced acoustic resonator are the same and are shown at marker ml (1 .898 GHz).
  • the upper Bragg Band frequencies of both the conventional acoustic resonator and the enhanced acoustic resonator are the same and are shown at markers m4 and m5 (2.028 GHz).
  • the anti-resonance of the conventional acoustic resonator is shown at marker m3 (1 .964 GHz), while the anti- resonance of the enhanced acoustic resonator is shown at marker m2 (1 .964 GHz).
  • the upper Bragg band frequency at marker m4 is higher than the anti-resonant frequency at marker m3 by 64 MHz
  • the upper Bragg band frequency at marker m5 is higher than the anti-resonant frequency at marker m2 by 82 MHz.
  • the lumped capacitive structures 260 avoid coupling the acoustic waves originating from the lumped resonator structures 258. Furthermore, it is desirable that the lumped capacitive structures 260 be capable of handling high power without incurring damage. To this end, several novel designs of lumped capacitive structures 260 that avoid or minimize coupling acoustic waves originating from the interdigitated resonator fingers 266 and/or handle high power without incurring damage, will now be described.
  • the following described lumped capacitive structures 260 are monolithically disposed on the piezoelectric layer 252 (which may, e.g., be a piezoelectric substrate or a thin-film piezoelectric monolithically disposed on a non- piezoelectric substrate) and are electrically coupled (e.g., in parallel) to the acoustic resonator structure 258.
  • the piezoelectric layer 252 which may, e.g., be a piezoelectric substrate or a thin-film piezoelectric monolithically disposed on a non- piezoelectric substrate
  • electrically coupled e.g., in parallel
  • the following lumped capacitive structures 260 are described as being electrically coupled between the metalized input signal plane portion 254a and the metalized output signal plane portion 254b, similar to the lumped capacitive structure 260a illustrated in Fig. 11 b, although the following lumped capacitive structures 260 may be alternatively electrically coupled between the metalized signal plane 254 and the metalized ground plane 256, similar to the lumped capacitive structure 260b illustrated in Fig. 12.
  • the following described lumped capacitive structures 260 may be partially nested within, fully nested within, or not nested at all within, the metalized input signal plane portion 254a, metalized output signal plane portion 254b, metalized input signal plane 254, or metalized ground plane 256.
  • one embodiment of a lumped capacitive structure 260(1 ) comprises an arrangement of planar interdigitated capacitive fingers 268a.
  • the orientation of the interdigitated capacitive fingers 268a may be orthogonal to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a (Fig. 19a), parallel to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a (Fig. 19b), or oblique to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a (Fig. 19c).
  • Each of at least one of the interdigitated capacitive fingers 268a (in the illustrated embodiment, all of the interdigitated capacitive fingers 268a) has an edge 270 that is entirely continuous (i.e., there are no discontinuities, such as sharp corners, along the entire edge).
  • the edge 270 of each interdigitated capacitive finger 268a takes the form of a single undulation.
  • edges 270 of the interdigitated capacitive fingers 268a Because there are no discontinuities in the edges 270 of the interdigitated capacitive fingers 268a, electrical power will not be concentrated or focused on any portions of the edges 270.
  • At least one of the interdigitated capacitive fingers 268a (in the illustrated embodiment, all of the interdigitated capacitive fingers 268a) has a length(l)-to-width(w) ratio of less than two, and preferably, a length-to-width ratio of less than one.
  • edge 270 along the length of each of at least one of the interdigitated capacitive fingers 268a is parallel to any finite portions of edges 282 (in this case, the straight edges 282) along the lengths of the interdigitated resonator fingers 266. This remain true regardless of the orientation of the interdigitated capacitive fingers 268a relative to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a, as can be seen in Figs. 19a-19c.
  • a lumped capacitive structure 260(2) comprises an arrangement of planar interdigitated capacitive fingers 268b.
  • the orientation of the interdigitated capacitive fingers 268b may be orthogonal to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a (Fig. 20a), parallel to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a (Fig. 20b), or orthogonal to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a (Fig. 20c).
  • interdigitated capacitive fingers 268b has a length(l)-to-width(w) ratio of less than two, and preferably, a length-to-width ratio of less than one.
  • w length-to-width ratio
  • each of at least one of the interdigitated capacitive fingers 268b (in the illustrated embodiment, all of the interdigitated capacitive fingers 268b) has straight edges 272a, 272b that takes the form of a single sawtooth instead of a single undulation.
  • no finite portion of the edge 272a, 272b along the length of each of at least one of the interdigitated capacitive fingers 268b is parallel to any finite portions of edges 282 (in this case, the straight edges 282) along the length of the interdigitated resonator fingers 266.
  • edges 282 in this case, the straight edges 282
  • the straight edges 272a, 272b of each of the interdigitated capacitive fingers 268b are oriented at a 45-degree angle relative to the orientation of the straight edges 282 of the interdigitated resonator fingers 266.
  • the edge 272a of each of the interdigitated capacitive fingers 268b is orthogonal to the straight edges 272 of each of the interdigitated capacitive fingers 268b, whereas the edge 272b of each of the interdigitated capacitive fingers 268b is parallel to the straight edges 272 of each of the interdigitated capacitive fingers 268b.
  • a lumped capacitive structure 260(3) comprises an arrangement of planar interdigitated capacitive fingers 268c.
  • the orientation of the interdigitated capacitive fingers 268c may be orthogonal to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a (Fig. 21a), parallel to the orientation of the
  • each of the interdigitated capacitive fingers 268c has an edge that forms a plurality of undulations, instead of a single undulation.
  • each of the interdigitated capacitive fingers 268c comprises a plurality of interdigitated capacitive sub-fingers 274, thereby increasing the effective lengths, and thus the static capacitances, of the interdigitated capacitive fingers 268c.
  • all of the interdigitated capacitive sub-fingers 274 of each interdigitated capacitive finger 268c have an orientation that is orthogonal to the orientation of the respective interdigitated capacitive finger 268c.
  • interdigitated capacitive sub-fingers 274 of each least one of the interdigitated capacitive fingers 268c has an edge 276 that is entirely continuous (i.e., there are no
  • yet another embodiment of a lumped capacitive structure 260(4) comprises an arrangement of planar interdigitated capacitive fingers 268d.
  • the orientation of the interdigitated capacitive fingers 268d may be orthogonal to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a (Fig. 22a), parallel to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a (Fig. 22b), or oblique to the orientation of the interdigitated resonator fingers 266 of the lumped resonator structure 258a (Fig. 22c).
  • each of the interdigitated capacitive fingers 268d comprises a plurality of interdigitated capacitive sub-fingers 278, thereby increasing the effective lengths, and thus the static capacitances, of the interdigitated capacitive fingers 268d.
  • all of the interdigitated capacitive sub-fingers 278 of each interdigitated capacitive finger 268d have an orientation that is orthogonal to the orientation of the respective interdigitated capacitive finger 268d.
  • each interdigitated capacitive sub-finger 278 has two straight edges 280a, 280b that forms a sawtooth.
  • no finite portion of the straight edges 280a, 280b along the length of each of at least one of the interdigitated capacitive sub-fingers 278 is parallel to any finite portions of edges 282 (in this case, the straight edges 282) along the length of the interdigitated resonator fingers 266.
  • the straight edges 280a, 280b of each of the interdigitated capacitive sub-fingers 278 are oriented at a 45-degree angle relative to the orientation of the straight edges 282 of the interdigitated resonator fingers 266.
  • the edge 280a of each of the interdigitated capacitive sub-fingers 278 is orthogonal to the straight edges 282 along the length of the interdigitated resonator fingers 266, whereas the edge 280b of each of the interdigitated capacitive sub-fingers 278 is parallel to the straight edges 282 along the length of the interdigitated resonator fingers 266.

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  • Physics & Mathematics (AREA)
  • Acoustics & Sound (AREA)
  • Chemical & Material Sciences (AREA)
  • Crystallography & Structural Chemistry (AREA)
  • Piezo-Electric Or Mechanical Vibrators, Or Delay Or Filter Circuits (AREA)
  • Surface Acoustic Wave Elements And Circuit Networks Thereof (AREA)

Abstract

L'invention concerne un filtre acoustique comprenant une couche piézoélectrique; une structure de résonateur acoustique disposée de façon monolithique sur la couche piézoélectrique, la structure de résonateur acoustique comprenant un agencement de doigts interdigités planaires; et une structure capacitive localisée disposée de façon monolithique sur la couche piézoélectrique et couplée électriquement à la structure de résonateur acoustique, la structure capacitive localisée comprenant un agencement de doigts capacitifs interdigités planaires, chacun desdits au moins un des doigts capacitifs interdigités présentant un bord totalement continu.
PCT/US2018/019899 2017-02-27 2018-02-27 Filtre à ondes acoustiques à élimination améliorée WO2018157114A1 (fr)

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CN201880012977.1A CN110506391B (zh) 2017-02-27 2018-02-27 增强抑制性的声波滤波器
DE112018001025.2T DE112018001025T5 (de) 2017-02-27 2018-02-27 Akustischer Wellenfilter mit verstärkter Abweisung

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