WO2011103841A2 - 一种微带天线 - Google Patents

一种微带天线 Download PDF

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Publication number
WO2011103841A2
WO2011103841A2 PCT/CN2011/073006 CN2011073006W WO2011103841A2 WO 2011103841 A2 WO2011103841 A2 WO 2011103841A2 CN 2011073006 W CN2011073006 W CN 2011073006W WO 2011103841 A2 WO2011103841 A2 WO 2011103841A2
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WO
WIPO (PCT)
Prior art keywords
dielectric plate
microstrip
microstrip antenna
dielectric
coupling window
Prior art date
Application number
PCT/CN2011/073006
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English (en)
French (fr)
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WO2011103841A3 (zh
Inventor
冯祖建
唐振宇
廖星
郭智力
Original Assignee
华为技术有限公司
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Publication date
Application filed by 华为技术有限公司 filed Critical 华为技术有限公司
Priority to EP11746883A priority Critical patent/EP2482383A4/en
Priority to PCT/CN2011/073006 priority patent/WO2011103841A2/zh
Priority to CN2011800002497A priority patent/CN102959801A/zh
Publication of WO2011103841A2 publication Critical patent/WO2011103841A2/zh
Publication of WO2011103841A3 publication Critical patent/WO2011103841A3/zh

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Classifications

    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/0407—Substantially flat resonant element parallel to ground plane, e.g. patch antenna
    • H01Q9/045—Substantially flat resonant element parallel to ground plane, e.g. patch antenna with particular feeding means
    • H01Q9/0457—Substantially flat resonant element parallel to ground plane, e.g. patch antenna with particular feeding means electromagnetically coupled to the feed line
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/30—Arrangements for providing operation on different wavebands
    • H01Q5/378—Combination of fed elements with parasitic elements
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/0407—Substantially flat resonant element parallel to ground plane, e.g. patch antenna
    • H01Q9/0414—Substantially flat resonant element parallel to ground plane, e.g. patch antenna in a stacked or folded configuration

Definitions

  • the embodiments of the present invention relate to the field of mobile communications technologies, and in particular, to a microstrip antenna. Background technique
  • microstrip antennas have been widely used in microwave and millimeter wave fields due to their low cost and ease of processing.
  • the microstrip antenna is small in thickness and easy to integrate.
  • the technology to expand the antenna bandwidth has been widely used, such as: L-probe feed, parasitic patch, U-slot loading, caliber coupling and other technologies.
  • the aperture-coupled multi-layer microstrip antenna has the widest application and the best comprehensive performance, and has the advantages of high bandwidth, low cross polarization and small size.
  • a cross-sectional view of a conventional microstrip antenna is generally composed of a dielectric plate 1a, a dielectric plate 2a, and a dielectric plate 3a disposed in parallel, and a microstrip patch 1 is disposed at the center of the upper surface of the dielectric plate 1a.
  • a microstrip patch 21a is disposed at the center of the upper surface of the dielectric plate 2a
  • a ground layer 31a is disposed on the upper surface of the dielectric plate 3a
  • a coupling window 32a is defined at the center of the ground layer 31a
  • a center conductor is disposed on the lower surface of the dielectric plate 3a.
  • the microstrip antenna of such a structure is fed by a microstrip line composed of a ground layer 31a, a dielectric plate 3a, and a center conductor 33a. Since the coupling window 32a will have back radiation and spurious radiation of the microstrip antenna, it will affect the ratio of the forward radiant energy and the back radiant energy of the antenna, that is, affect the front to back ratio of the antenna. As shown in FIG. 2, another microstrip antenna of the prior art generally adds an aluminum honeycomb reflective plate 4a at a medium wavelength of 1/4 below the antenna dielectric plate 3a to counteract the back radiation, the dielectric plate 3a and the reflective plate 4a. Fill the foam 5a between.
  • Embodiments of the present invention provide a microstrip antenna to solve the problem that the existing microstrip antenna has a thick thickness and is not easy to integrate the transmitting and receiving circuits of the antenna.
  • An embodiment of the present invention provides a microstrip antenna, including: a four-layer dielectric plate disposed in parallel; a first microstrip patch is disposed at a center of a surface of the first dielectric plate; and a center of the surface of the second dielectric plate is disposed at a center a second microstrip patch; a surface of the third dielectric plate is provided with a first ground layer, a coupling window is defined at a center of the first ground layer, and a center conductor is disposed at a center of the lower surface of the third dielectric plate; a second ground layer is disposed on a lower surface of the four-layer dielectric plate;
  • the third dielectric plate and the fourth dielectric plate are asymmetric media such that an electric field above the central conductor is greater than an electric field below.
  • the third dielectric plate and the fourth dielectric plate are asymmetric media, so that the electric field strength in the upper half of the strip line is much larger than the electric field strength in the lower half, and the energy is increased.
  • the coupling efficiency ensures the bandwidth of the antenna and reduces the back radiation of the microstrip antenna, thereby improving the front-to-back ratio of the microstrip antenna.
  • the overall thickness of the microstrip antenna is greatly reduced, which is advantageous for the integration of the transmitting and receiving circuits of the antenna.
  • 1 is a cross-sectional view of a conventional aperture coupled multilayer microstrip antenna
  • FIG. 2 is a cross-sectional view of another conventional aperture-coupled multilayer microstrip antenna
  • FIG. 3 is a cross-sectional view of a microstrip antenna according to an embodiment of the present invention.
  • FIG. 4 is a cross-sectional view of a microstrip antenna according to still another embodiment of the present invention.
  • Figure 5 is a perspective view of the microstrip antenna shown in Figure 4.
  • FIG. 6 is a schematic diagram of return loss obtained by HFSS simulation of a microstrip antenna provided by the present invention
  • FIG. 7 is a far-field pattern obtained by HFSS simulation of a microstrip antenna provided by the present invention
  • FIG. 8 is a microstrip antenna provided by the present invention. Schematic diagram of the in-band gain obtained by HFSS simulation. detailed description
  • the microstrip antenna includes: a four-layer dielectric plate disposed in parallel;
  • the first microstrip patch 11 is disposed at the center of the upper surface of the first dielectric sheet 1; the second microstrip patch 21 is disposed at the center of the upper surface of the second dielectric panel 2; and the upper surface of the third dielectric panel 3
  • a first ground layer 31 is disposed, a coupling window 32 is defined at a center of the first ground layer 31, a center conductor 33 is disposed at a center of a lower surface of the third dielectric plate 3, and a second ground layer is disposed on a lower surface of the fourth dielectric plate 4. 41;
  • the third dielectric plate 3 and the fourth dielectric plate 4 are asymmetric media such that the electric field above the center conductor 33 is greater than the lower electric field.
  • the third dielectric plate 3 and the fourth dielectric plate 4 are asymmetric media, the first ground layer 31, the third dielectric plate 3, the center conductor 33, the fourth dielectric plate 4, and the second ground layer 41 are common. Forming an asymmetric dielectric strip line.
  • the entire microstrip antenna is composed of a first dielectric plate 1, a second dielectric plate 2, a third dielectric plate 3, and a fourth dielectric plate 4.
  • a double-sided copper-clad dielectric plate can be used, the copper foil is etched by photolithography to remove the unnecessary copper foil, the desired pattern is retained, and the entire microstrip is obtained through a lamination process. antenna.
  • the center of the upper surface of the first dielectric plate 1 is etched by photolithography to retain the first microstrip patch 1 1 , and the copper foil on the lower surface of the first dielectric plate 1 is completely etched away; the second dielectric plate 2
  • the second microstrip patch 21 is retained at the center of the upper surface, and the copper foil on the lower surface of the second dielectric panel 2 is also completely etched away;
  • the copper foil on the upper surface of the third dielectric panel 3 is retained as an asymmetric dielectric stripline a first ground layer 31, the first ground layer 31 is etched with a coupling window 32, and the center conductor 33 is retained at the center of the lower surface of the third dielectric sheet 3,
  • the remaining portion of the lower surface of the third dielectric plate 3 is completely etched away; the upper surface of the fourth dielectric plate 4 is completely etched away, and the copper foil on the lower surface of the fourth dielectric plate 4 remains as an asymmetric dielectric strip line.
  • the second ground layer 41 is used to retain the first microstrip patch 1 1
  • the first ground layer 31, the third dielectric plate 3, the center conductor 33, the fourth dielectric plate 4 and the second ground layer 41 together form an asymmetric dielectric strip line: the third layer medium above the center conductor 33
  • the plate 3 and the lower fourth dielectric plate 4 are different and asymmetrical.
  • the difference in the asymmetry may be:
  • the thickness of the third dielectric plate 3 and the fourth dielectric plate 4 are different, or the dielectric constants are different, or the thickness and the dielectric constant are different.
  • the dielectric constant of the third dielectric plate 3 needs to be larger than that of the fourth dielectric plate 4, and the thickness of the third dielectric plate 3 needs to be smaller than that of the fourth dielectric plate 4.
  • the thickness of the third dielectric plate 3 and the fourth dielectric plate 4 in the asymmetric dielectric strip line are different, or the materials have different dielectric constants, or the thicknesses of the third dielectric plate 3 and the fourth dielectric plate 4 And materials are different.
  • the electric field strength above the center conductor 33 will be much greater than the underlying electric field strength, and the energy is mainly concentrated in the region of the third dielectric sheet 3 between the center conductor 33 and the first ground layer 31.
  • the energy exchange in the vicinity of the coupling window 32 has little effect on the electric field strength of the fourth dielectric plate 4, and the entire energy can be efficiently transmitted to the first microstrip patch 1 1 and the second microstrip patch 21, therefore,
  • the symmetrical strip line can ensure the effective transmission of energy and reduce the back radiation and stray radiation of energy.
  • the presence of the second ground plane 41 acts as a shielding energy, preventing the energy from being directed away from the radiation, ensuring that most of the energy is radiated from the front side of the microstrip antenna.
  • the third dielectric plate and the fourth dielectric plate are asymmetric media, so that the electric field intensity in the upper half of the strip line is much larger than the electric field strength in the lower half, thereby improving energy coupling.
  • the efficiency ensures the bandwidth of the antenna and reduces the back radiation of the microstrip antenna, thereby improving the front-to-back ratio of the microstrip antenna.
  • the overall thickness of the microstrip antenna is greatly reduced, which is advantageous for the integration of the antenna's transmitting and receiving circuits.
  • the microstrip antenna includes: a four-layer dielectric plate disposed in parallel;
  • the first microstrip patch 11 is disposed at the center of the upper surface of the first dielectric sheet 1;
  • a second microstrip patch 21 is disposed at the center of the surface;
  • a first ground layer 31 is disposed on the upper surface of the third dielectric panel 3, and a coupling window 32 is defined at a center of the first ground layer 31, and the third dielectric panel 3 is located at the center of the lower surface is provided with a center conductor 33;
  • a lower surface of the fourth layer of dielectric plate 4 is provided with a second ground layer 41;
  • the third dielectric plate 3 and the fourth dielectric plate 4 are asymmetric media such that the electric field above the center conductor 33 is greater than the lower electric field.
  • the first ground layer 31, the third dielectric plate 3, the center conductor 33, the fourth dielectric plate 4, and the second ground layer 41 constitute an asymmetric dielectric strip line, and the asymmetric dielectric strip line mainly refers to the center conductor.
  • the third dielectric plate 3 above the 33 and the fourth dielectric plate 4 below are asymmetric media, specifically the thickness of the third dielectric plate 3 and the fourth dielectric plate 4 are different, or the dielectric constant is different, or the thickness And dielectric constant are different.
  • the dielectric constant of the third dielectric plate 3 is greater than that of the fourth dielectric plate 4, and the third dielectric plate 3 and the thickness are smaller than the fourth dielectric plate 4.
  • the first microstrip patch 11 can overlap the center of the second microstrip patch 21 to ensure maximum energy transfer to the top of the microstrip antenna.
  • the first microstrip patch 11 and the second microstrip patch 21 may each be set to a square shape, the side length of the first microstrip patch 11 is slightly larger than 1/2 medium wavelength, and the side length of the second microstrip patch 21 is slightly smaller than 1/2 medium wavelength, because the first microstrip patch 11 and the second microstrip patch 21 respectively have a resonance point, and the first microstrip patch 11 has a side length slightly larger than 1/2 medium wavelength,
  • the length of the second microstrip patch 21 is slightly less than 1 /2 medium wavelength to ensure better energy transfer between the two resonance points, which is beneficial to expand the bandwidth of the microstrip antenna.
  • the first ground layer 31 is a ground shared by the second microstrip patch 21 and the asymmetric dielectric strip line.
  • the coupling window 32 is formed on the first ground layer 31.
  • the coupling window 21 is an elongated slit. As seen in the perspective view, the coupling window 32 is located directly below the second microstrip patch 21, perpendicular to the center conductor 33.
  • the coupling window 32 and the center conductor 33 are etched at the center of the upper and lower surfaces of the third dielectric plate 3, thereby ensuring the alignment accuracy of the coupling window 32 and the center conductor 33.
  • the length of the coupling window 32 can be set to be less than 1 / 2 dielectric wavelength to make the coupling window 32 non-resonant State. This is because the coupling window 32 is a long slit, and the slit in the resonant state also radiates electromagnetic waves, which may cause the microstrip antenna to radiate back, thereby mismatching the working energy of the entire antenna, and affecting the normal working state of the microstrip antenna. . Therefore, the coupling window 32 needs to be in a non-resonant state.
  • the center conductor 33 is located directly below the coupling window 32, and the length of the center conductor 33 beyond the edge portion of the coupling window 32 is less than 1/4 of the medium wavelength, and the length of the edge portion of the coupling window 32 is not more than 1/4 of the medium.
  • the wavelength, the width of the center conductor 33 beyond the edge portion of the coupling window 32 and the portion not exceeding the edge portion of the coupling window 32 is greater than the width of the other portions of the center conductor 33.
  • the open condition of the open section is that the width is more tolerant under the condition that the length is less than 1/4 of the medium wavelength.
  • a section of the conductor in front of the coupling window 32 serves to transform the impedance of the antenna to a high impedance for matching.
  • the impedance transform segment has a length of about 1/4 of the dielectric wavelength, and the width is wider than the original width of the center conductor 33, transforming the smaller antenna impedance to match the impedance of the asymmetric dielectric stripline.
  • the optimal bandwidth of the microstrip antenna can be achieved by fine-tuning the length and width of the reactance regulation section and the impedance transformation section.
  • a non-resonant cavity may be disposed in the third dielectric plate 3 and the fourth dielectric plate 4 around the coupling window 32, the non-resonant cavity connecting the first ground layer 31 and the second ground layer 41.
  • the non-resonant cavity may be composed of at least four metal posts 5 that connect the first ground layer 31 and the second ground layer 41.
  • the non-resonant cavity may be composed of four metal pillars 5, and the distance between two adjacent metal pillars 5 in the non-resonant cavity is less than 1/2 medium wavelength to prevent the vicinity of the center conductor 33 from being generated. Additional resonance leads to energy mismatch.
  • Transverse electric and magnetic fields are transmitted in the asymmetric medium strip line (Transverse Electric and
  • Magnetic Field; TEM Magnetic Field; TEM Wave or quasi-TEM wave, which is determined by the metal boundary conditions and dielectric boundary conditions of the transmission line (consisting of the center conductor 33, the first ground layer 31 and the second ground layer 41), if the third layer medium
  • the dielectric constants of the plate 3 and the fourth dielectric plate 4 are different, then the division of the two media
  • the longitudinal electric field component or the magnetic field component is generated at the boundary, but the longitudinal component is weak, so the quasi-TEM wave propagates in the asymmetric dielectric strip line; if the third dielectric plate 3 and the fourth dielectric plate 4 are only thick The same and the same dielectric constant, then the TEM wave propagates in the asymmetric medium stripline.
  • the first ground layer 31 and the second ground layer 41 in the asymmetric dielectric strip line together form a loop of surface current with the center conductor 33.
  • the direction of the surface current on the first ground layer 31 and the second ground layer 41 is opposite to the direction on the center conductor 33, and the sum of the magnitudes of the surface currents of the first ground layer 31 and the second ground layer 41 is on the center conductor 33.
  • the size is equal.
  • the electric field strength is different in the two dielectric plates, resulting in the first ground layer 31.
  • the magnitude of the surface current on the second ground layer 41 is also different.
  • the dielectric constant of the third dielectric plate 3 is greater than the dielectric constant of the fourth dielectric plate 4, or the thickness of the third dielectric plate 3 is smaller than the thickness of the fourth dielectric plate 4. It is also possible that the dielectric constant of the third dielectric plate 3 is greater than the dielectric constant of the fourth dielectric plate 4 and the thickness is smaller than the thickness of the fourth dielectric plate 4. Therefore, the surface current of the first ground layer 31 is much larger than that of the second ground layer 41.
  • the coupling window 32 When the surface current on the first ground layer 31 passes through the coupling window 32, it is shunted in front of the coupling window 32 and merges behind the coupling window 32. This change in surface current direction forms a non-TEM field near the coupling window 32.
  • the coupling window 32 is located directly above the center conductor 33 for maximum non-TEM field strength. According to the boundary condition of the asymmetric stripline, under the action of this non-TEM field, an induced current is generated on the second ground layer 41, corresponding to a mirror image non-TEM field, and the mirror non-TEM field is opposite to the original non-TEM field.
  • the non-TEM field acts as a counteracting effect.
  • the length of the coupling window 32 can be reduced to be as small as possible in the resonant state to reduce the intensity of the induced current.
  • changing the dielectric filling mode in the third dielectric plate 3 and the fourth dielectric plate 4 in the asymmetric strip line can also serve to reduce the induced current.
  • the energy also in a resonant state, radiates energy externally, because the resonant frequency of the second microstrip patch 21 and the first microstrip patch 11 is different but close, so that the bandwidth of the microstrip antenna is improved.
  • the non-resonant cavity composed of the metal pillars 5 can destroy the boundary conditions of parallel two-line TEM wave transmission, suppress the generation of parallel two-line TEM waves, and help to reduce the interference of each antenna element in the array antenna, which is beneficial to reduce the active Interference in the integrated antenna to the active devices behind it.
  • the relative bandwidth of the antenna is an important indicator to measure the performance of the antenna.
  • the relative bandwidth of the microstrip antenna is the ratio of the frequency range of the electromagnetic wave radiated by the microstrip antenna to the center frequency of the electromagnetic wave radiated by the microstrip antenna.
  • the return loss of the antenna should also be considered.
  • the return loss refers to the ratio of the electromagnetic wave radiated by the microstrip antenna to the reflected electromagnetic wave.
  • the abscissa is the frequency of the electromagnetic wave radiated by the microstrip antenna
  • the ordinate is the electromagnetic wave radiated by the microstrip antenna. Return loss.
  • the relative bandwidth of the microstrip antenna can be calculated according to the frequency range and the center frequency of the electromagnetic wave of the microstrip antenna. In FIG. 6, the corresponding return loss can be found according to the frequency range corresponding to the partial relative bandwidth of the microstrip antenna.
  • the microstrip antenna provided by the present invention has a return loss of less than -20 dB in a relative bandwidth of 10%, and a return loss of less than -18 dB in a relative bandwidth of 12%. Therefore, the microstrip antenna provided by the embodiment of the present invention has a return loss that satisfies most application scenarios.
  • FIG. 7 is a far-field pattern obtained by HFSS simulation of the microstrip antenna provided by the present invention.
  • Theta and phi are spherical coordinates, and FIG. 7 is taken as two electromagnetic wave energy radiated by the microstrip antenna.
  • a single microstrip antenna can be used as an array unit, and the second ground layer can be increased by expanding the number of array units, so that energy is difficult to be diffracted behind the microstrip antenna, that is, the back radiation can be reduced, thereby improving the microstrip.
  • Antenna front to back ratio can be used as an array unit, and the second ground layer can be increased by expanding the number of array units, so that energy is difficult to be diffracted behind the microstrip antenna, that is, the back radiation can be reduced, thereby improving the microstrip.
  • the front-to-back ratio of the microstrip antenna array can usually be increased to 30 dB, and the size of the microstrip antenna array is only 54mm 54mm, if the number of array units of the microstrip antenna is further expanded, the front-to-back ratio of the microstrip antenna can be increased to 50 dB or more.
  • FIG. 8 is a schematic diagram of the in-band gain obtained by the HFSS simulation of the microstrip antenna provided by the present invention.
  • the abscissa is the frequency of the electromagnetic wave radiated by the microstrip antenna
  • the ordinate is the gain of the electromagnetic wave radiated by the microstrip antenna
  • Gain is another important indicator of antenna performance, reflecting the energy bunching ability of the antenna as an energy transceiver.
  • the average value of the gain of the radiated electromagnetic wave of the microstrip antenna provided by the embodiment of the present invention is about 7.33 dB, and the average value is around each frequency point in the frequency band of the microstrip antenna.
  • the amplitude of the fluctuation, that is, the in-band flatness is around 0.1 dB. It can be seen that the microstrip antenna provided by the present invention has relatively close electromagnetic wave gains at various frequency points in the frequency band.

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Description

一种微带天线
技术领域
本发明实施例涉及移动通信技术领域， 特别涉及一种微带天线。 背景技术
随着射频技术的发展， 微带天线由于成本低， 加工制作容易， 在微波、 毫米波领域得到了大量应用。 微带天线厚度小、 易于集成。 随着天线技术的 不断发展， 拓展天线带宽的技术得到了广泛应用， 例如： L探针馈电、 寄生贴 片、 U型槽加载、 口径耦合等技术。 其中， 口径耦合多层微带天线应用最广， 综合性能最好， 具有高带宽、 低交叉极化、 尺寸小的优点。
现有的微带天线的截面图通常如图 1所示， 由平行设置的介质板 1 a、介质 板 2a和介质板 3a构成， 介质板 1 a上表面中心处设置有一个微带贴片 1 1a, 介 质板 2a上表面中心处设置有一个微带贴片 21 a, 介质板 3a上表面设有一接地 层 31a, 接地层 31a中心处开设一耦合窗口 32a, 介质板 3a下表面设有一中心 导体 33a。 这种结构的微带天线， 由接地层 31a、 介质板 3a和中心导体 33a构 成的微带线进行馈电。由于耦合窗口 32a会出现背向辐射和微带天线的杂散辐 射， 因此会影响天线的正向辐射能量和背向辐射能量的比值， 即影响天线的 前后比。 如图 2所示， 现有的另一种微带天线通常在距天线介质板 3a下方 1/4 介质波长处增加铝蜂窝材质的反射板 4a以抵消背向辐射， 介质板 3a和反射板 4a之间填充泡沫 5a。
然而， 现有的微带天线厚度较厚， 不易于天线的发射和接收电路的集成。 发明内容
本发明实施例提供一种微带天线， 以解决现有的微带天线厚度较厚， 不 易于天线的发射和接收电路的集成的问题。 本发明实施例提供一种微带天线， 包括： 平行设置的四层介质板； 第一 层介质板上表面中心处设有第一微带贴片； 第二层介质板上表面中心处设有 第二微带贴片； 第三层介质板上表面设有第一接地层， 所述第一接地层中心 处开设耦合窗口， 所述第三层介质板下表面中心处设有中心导体； 第四层介 质板下表面设有第二接地层；
所述第三层介质板与所述第四层介质板为非对称介质， 以使所述中心导 体上方电场大于下方电场。
本发明实施例提供的微带天线， 第三层介质板与所述第四层介质板为非 对称介质， 使得带状线内上半部电场强度远远大于下半部电场强度， 提高了 能量耦合效率， 保证了天线的带宽， 降低了微带天线的背向辐射， 从而提高 了微带天线的前后比。 大大降低了微带天线的整体厚度， 有利于天线的发射 和接收电路的集成。 附图说明
为了更清楚地说明本发明实施例或现有技术中的技术方案， 下面将对实 施例或现有技术描述中所需要使用的附图作简单地介绍， 显而易见地， 下面 描述中的附图仅仅是本发明的一些实施例， 对于本领域普通技术人员来讲， 在不付出创造性劳动的前提下， 还可以根据这些附图获得其他的附图。
图 1为现有的一种口径耦合多层微带天线的截面图；
图 2为现有的另一种口径耦合多层微带天线截面图；
图 3为本发明一个实施例提供的一种微带天线的截面图；
图 4为本发明又一个实施例提供的一种微带天线的截面图；
图 5为图 4所示微带天线的透视图；
图 6为本发明提供的微带天线由 HFSS仿真得到的回波损耗示意图； 图 7为本发明提供的微带天线由 HFSS仿真得到的远场方向图； 图 8为本发明提供的微带天线由 HFSS仿真得到的带内增益示意图。 具体实施方式
下面将结合本发明实施例中的附图， 对本发明实施例中的技术方案进行 清楚、 完整地描述， 显然， 所描述的实施例仅仅是本发明一部分实施例， 而 不是全部的实施例。 基于本发明中的实施例， 本领域普通技术人员在没有做 出创造性劳动前提下所获得的所有其他实施例， 都属于本发明保护的范围。
图 3为本发明一个实施例提供的一种微带天线的截面图， 如图 3所示， 该 微带天线包括： 平行设置的四层介质板；
其中，第一层介质板 1上表面中心处设有第一微带贴片 11 ; 第二层介质板 2上表面中心处设有第二微带贴片 21 ; 第三层介质板 3上表面设有第一接地层 31 , 第一接地层 31中心处开设耦合窗口 32, 第三层介质板 3下表面中心处设 有中心导体 33; 第四层介质板 4下表面设有第二接地层 41；
第三层介质板 3与第四层介质板 4为非对称介质， 以使中心导体 33上方电 场大于下方电场。
由于第三层介质板 3与第四层介质板 4为非对称介质， 则第一接地层 31、 第三层介质板 3、 中心导体 33、 第四层介质板 4和第二接地层 41共同构成非对 称介质带状线。
具体的， 整个微带天线由第一层介质板 1、 第二层介质板 2、 第三层介质 板 3和第四层介质板 4这四层介质板组成。 作为形成微带天线的一种可行的工 艺流程： 可以采用双面覆铜介质板， 铜箔经过光刻腐蚀后去除不需要的铜箔， 保留需要的图案， 再通过层压工艺得到整个微带天线。
其中，第一层介质板 1上表面的中心处通过光刻腐蚀后保留第一微带贴片 1 1 , 第一层介质板 1下表面的铜箔完全刻蚀掉； 第二层介质板 2上表面中心处 保留第二微带贴片 21 ,第二层介质板 2下表面的铜箔亦完全刻蚀掉； 第三层介 质板 3上表面的铜箔保留，作为非对称介质带状线的第一接地层 31 ,第一接地 层 31上蚀刻耦合窗口 32, 第三层介质板 3下表面的中心处保留中心导体 33, 第三层介质板 3下表面的其余部分完全刻蚀掉； 第四层介质板 4上表面完全刻 蚀掉， 第四层介质板 4下表面的铜箔保留，作为非对称介质带状线的第二接地 层 41。
所谓第一接地层 31、 第三层介质板 3、 中心导体 33、 第四层介质板 4和第 二接地层 41共同构成非对称介质带状线是指： 中心导体 33上方的第三层介质 板 3和下方的第四层介质板 4是不相同、 不对称的。 这种不相同、 不对称具体 可以是： 第三层介质板 3和第四层介质板 4的厚度不同， 或者是介电常数不同， 或者厚度和介电常数均不同。 在本发明实施例中， 第三层介质板 3的介电常数 需要大于第四层介质板 4, 第三层介质板 3的厚度需要小于第四层介质板 4。
非对称介质带状线中第三层介质板 3与第四层介质板 4的厚度不同， 或者 材质不同而造成介电常数不同， 或者第三层介质板 3与第四层介质板 4的厚度 和材质都不同。 采用非对称介质带状线， 中心导体 33上方的电场强度将远远 大于下方的电场强度， 能量主要集中在中心导体 33和第一接地层 31之间的第 三层介质板 3的区域。 这样在耦合窗口 32附近的能量交换对第四层介质板 4的 电场强度影响很小， 整个能量能有效地传输给第一微带贴片 1 1和第二微带贴 片 21 , 因此， 非对称带状线更能保证能量的有效传输， 降低能量的背向辐射 和杂散辐射。 同时， 第二接地层 41的存在起到了屏蔽能量的作用， 阻止能量 背向辐射， 保证绝大部分能量从微带天线的正面辐射出去。
本实施例提供的微带天线， 第三层介质板与所述第四层介质板为非对称 介质， 使得带状线内上半部电场强度远远大于下半部电场强度， 提高了能量 耦合效率， 保证了天线的带宽， 降低了微带天线的背向辐射， 从而提高了微 带天线的前后比。 大大降低了微带天线的整体厚度， 有利于天线的发射和接 收电路的集成。
图 4为本发明又一个实施例提供的一种微带天线的截面图， 如图 4所示， 该微带天线包括： 平行设置的四层介质板；
第一层介质板 1上表面中心处设有第一微带贴片 11； 第二层介质板 2上表 面中心处设有第二微带贴片 21； 第三层介质板 3上表面设有第一接地层 31 , 所述第一接地层 31中心处开设耦合窗口 32 , 所述第三层介质板 3下表面中心 处设有中心导体 33; 第四层介质板 4下表面设有第二接地层 41；
第三层介质板 3与第四层介质板 4为非对称介质， 以使中心导体 33上方电 场大于下方电场。
其中， 第一接地层 31、 第三层介质板 3、 中心导体 33、 第四层介质板 4和 第二接地层 41构成非对称介质带状线， 非对称介质带状线主要是指中心导体 33上方的第三层介质板 3和下方的第四层介质板 4是非对称介质， 具体可以是 第三层介质板 3和第四层介质板 4的厚度不同， 或者介电常数不同， 或者厚度 和介电常数都不同。 本发明实施例中， 第三层介质板 3的介电常数大于第四层 介质板 4, 第三层介质板 3和厚度小于第四层介质板 4。
第一微带贴片 11可与第二微带贴片 21的中心重叠， 以保证向微带天线的 上方传递的能量最大。 可以将第一微带贴片 11和第二微带贴片 21均设置为正 方形， 第一微带贴片 11边长略大于 1/2介质波长， 第二微带贴片 21边长略小于 1/2介质波长，这是由于第一微带贴片 11处和第二微带贴片 21处分别有一个谐 振点， 第一微带贴片 11边长略大于 1/2介质波长， 第二微带贴片 21边长略小于 1 /2介质波长以保证两个谐振点之间的能量较好地传输，有利于拓展微带天线 的带宽。
第一接地层 31是第二微带贴片 21和非对称介质带状线共用的地， 耦合窗 口 32开设在第一接地层 31上， 该耦合窗口 21为长条形的缝隙， 从图 5所示的 透视图上看，耦合窗口 32位于第二微带贴片 21的正下方，与中心导体 33垂直。 耦合窗口 32和中心导体 33蚀刻在第三层介质板 3的上下表面中心处， 从而保 证耦合窗口 32和中心导体 33的对位精度。 由于末端开路的中心导体 33上的驻 波电压呈余弦分布， 在距离开路端约 1/4介质波长处， 电压幅度最大， 耦合窗 口 32处于这个位置可以实现能量的最大耦合， 从而保证能量的有效传递。 耦 合窗口 32的长度可设置成小于 1 /2介质波长， 以使耦合窗口 32处于非谐振状 态。 这是由于耦合窗口 32为长条形缝隙， 处于谐振状态的缝隙也会辐射电磁 波， 从而会导致微带天线背向辐射， 进而使整个天线的工作能量失配， 影响 微带天线的正常工作状态。 因此， 耦合窗口 32需要处于非谐振状态。
如图 5所示， 中心导体 33位于耦合窗口 32的正下方， 中心导体 33中超出 耦合窗口 32边缘部分的长度小于 1/4介质波长，未超出耦合窗口 32边缘部分的 长度等于 1 /4介质波长，中心导体 33中超出耦合窗口 32边缘部分以及未超出耦 合窗口 32边缘部分的宽度均大于所述中心导体 33其他部分的宽度。 这是由于 能量耦合与天线辐射的存在，使得耦合窗口 32附近的传输线阻抗发生了变化， 需要对微带天线进行阻抗匹配， 在中心导体 33末端开路段引入的容性电抗以 抵消耦合窗口 32产生的感性电抗。开路段呈容性的条件是在长度小于 1/4介质 波长的条件下， 宽度越宽容性越强。 中心导体 33上， 耦合窗口 32前面的一段 导体起着把天线阻抗变换到高阻抗实现匹配的作用。 阻抗变换段长度约为 1/4 介质波长， 宽度比中心导体 33本来的宽度宽， 把较小的天线阻抗变换到和非 对称介质带状线阻抗一致。通过微调电抗调节段和阻抗变换段的长度和宽度， 可以实现微带天线的最佳带宽。
为了增加微带天线的带宽，可以在耦合窗口 32周围的第三介质板 3和第四 介质板 4中设置非谐振腔， 该非谐振腔连接第一接地层 31和第二接地层 41。 该非谐振腔可以由至少四个金属柱 5构成， 金属柱 5连接第一接地层 31和第二 接地层 41。
作为一种较佳的实施方式， 非谐振腔可以由四个金属柱 5构成， 非谐振腔 中相邻两个金属柱 5的距离均小于 1/2介质波长， 以防止中心导体 33附近区域 产生额外的谐振而导致能量失配。
非对称介质带状线中传输的是横向电磁场 （Transverse Electric and
Magnetic Field; TEM ) 波或准 TEM波， 这是由传输线 （由中心导体 33, 第 一接地层 31和第二接地层 41构成） 的金属边界条件和介质边界条件决定的， 如果第三层介质板 3和第四层介质板 4的介电常数不同， 那么在两种介质的分 界处就会产生纵向电场分量或磁场分量， 但纵向分量很弱， 因此非对称介质 带状线中传播的是准 TEM波； 如果第三层介质板 3和第四层介质板 4只是厚度 不相同而介电常数相同， 那么非对称介质带状线中传播的就是 TEM波。
非对称介质带状线中的第一接地层 31和第二接地层 41共同与中心导体 33构成表面电流的回路。 表面电流在第一接地层 31和第二接地层 41上的方向 与在中心导体 33上的方向相反， 而第一接地层 31和第二接地层 41上表面电流 大小的总和与中心导体 33上的大小相等。 在非对称介质带状线中由于第三层 介质板 3和第四层介质板 4的厚度和介电常数不同， 使得电场强度在这两层介 质板中的大小不同， 导致第一接地层 31上的表面电流大小和第二接地层 41也 不同。 由于本发明实施例中， 第三层介质板 3的介电常数大于所述第四层介质 板 4的介电常数，或者第三层介质板 3的厚度小于第四层介质板 4的厚度，还有 可能是第三层介质板 3的介电常数大于所述第四层介质板 4的介电常数并且厚 度小于第四层介质板 4的厚度。 因此，第一接地层 31的表面电流远远大于第二 接地层 41。
当第一接地层 31上的表面电流经过耦合窗口 32的时候， 在耦合窗口 32前 面分流， 在耦合窗口 32后方合流。 这种表面电流方向的改变在耦合窗口 32附 近形成了一个非 TEM场。 耦合窗口 32位于中心导体 33的正上方可以得到最大 的非 TEM场强度。根据非对称带状线的边界条件，在这个非 TEM场的作用下， 第二接地层 41上产生感应电流， 对应一个镜像非 TEM场， 镜像非 TEM场与原 非 TEM场方向相反， 对原非 TEM场起到抵消作用。
需要说明的是， 耦合窗口 32是否处于谐振状态对感应电流影响较大， 因 此， 可以减小耦合窗口 32的长度， 使其尽量小于处于谐振状态的长度以降低 感应电流的强度。 另外， 改变非对称带状线中第三层介质板 3和第四层介质板 4中的介质填充方式也可以起到了降低感应电流的作用。
耦合窗口 32附近形成的非 TEM场的磁场部分恰好和第二微带贴片 21辐 射时第二层介质板 2中形成的横向磁场一致， 因此，耦合窗口 32能够激励第二 微带贴片 21的辐射， 同样第二微带贴片 21需要位于耦合窗口 32的正上方才能 激励起最高效的天线辐射 , 第一微带贴片 11通过耦合第二微带贴片 21的能 量， 也处于谐振状态， 对外辐射能量， 由于激励第二微带贴片 21和第一微带 贴片 1 1谐振频率不同但接近， 从而使得微带天线的带宽得以提高。
耦合窗口 32附近由于电磁场模式快速变化， 会在非对称带状线中激励起 许多不需要的传输模式， 最主要的就是平行双线 TEM波。 由于平行双线 TEM 波的存在， 会使天线辐射效率下降， 并带来杂散辐射， 降低天线交叉极化性 能、 侧向与背向辐射抑制能力。 通过金属柱 5组成的非谐振腔， 可以破坏平行 双线 TEM波传输的边界条件， 抑制平行双线 TEM波的产生， 有利于减小阵列 天线中各天线单元的干扰， 有利于减小有源集成天线中对背后各有源器件的 干扰。
天线的相对带宽是衡量天线性能的一个重要指标， 微带天线的相对带宽 为微带天线辐射的电磁波的频率范围与微带天线辐射的电磁波的中心频率的 比值。 而在保证天线相对带宽的前提下， 还应考虑天线的回波损耗， 回波损 耗是指微带天线辐射出去的电磁波与反射回来的电磁波的比值， 回波损耗可 以用来衡量天线的电磁波辐射效率。 图 6所示为本发明提供的微带天线由 HFSS仿真得到的回波损耗的部分示意图， 图 6中， 横坐标为微带天线辐射的 电磁波的频率， 纵坐标是微带天线辐射的电磁波的回波损耗。 可以根据本发 明提供微带天线的电磁波的频率范围和中心频率计算出该微带天线的相对带 宽， 在图 6中， 可以根据微带天线的部分相对带宽对应的频率范围找到对应的 回波损耗纵坐标， 可以得出， 本发明提供的微带天线在 10%的相对带宽内回 波损耗小于 -20dB, 在 12%的相对带宽内回波损耗小于 -18dB。 因此， 本发明 实施例提供的微带天线 , 其回波损耗满足绝大多数应用场景。
图 7所示为本发明提供的微带天线由 HFSS仿真得到的远场方向图， 图 7 中， Theta和 phi均为球坐标， 图 7中取的是微带天线辐射的电磁波能量的两个 正交截面， 而同一个截面内有两个极化分量， 共四条曲线， 根据坐标刻度， 从能量大的主极化的最大值和能量小的交叉极化的最大值之间的差值， 或者 从同一条主极化曲线的前向 （0度）和后向 （180度） 的比值均可以得出， 本 发明提供的微带天线的交叉极化在 -27.2dB左右， 微带天线的前后比为 16.5dB。 由于第二接地层的大小在一定程度上影响天线的前后比， 而通常增 大第二接地层能够减少背向辐射。 因此， 可以将单个微带天线作为阵列单元， 通过扩充阵列单元的数量来增大第二接地层， 从而使能量难以绕射到微带天 线的背后， 即能够减少背向辐射， 从而提高微带天线的前后比。 例如： 采用 4 X 4的微带天线阵列， 即将 4行乘以 4列个微带天线拼接起来，该微带天线阵列 的前后比通常可以提高到 30dB , 而该微带天线阵列的大小仅为 54mm 54mm, 如果进一步扩充微带天线的阵列单元数量， 能够实现将微带天线的 前后比高到 50dB以上。
图 8为本发明提供的微带天线由 HFSS仿真得到的带内增益示意图， 图 8 中， 横坐标为微带天线辐射的电磁波的频率， 纵坐标为微带天线辐射的电 磁波的增益， 其中， 增益是天线性能的又一个重要指标， 体现了天线作为能 量收发器的能量聚束能力。 从图 8可以计算得出， 本发明实施例提供的微带 天线，其辐射的电磁波的增益平均值为 7.33dB左右， 而在该微带天线的频带 范围内的各个频点上， 围绕平均值的波动幅度， 即带内平坦度在 0.1 dB左右。 可以看出， 本发明提供的微带天线， 在频带范围内的各个频点上， 其辐射的 电磁波增益较为接近。
最后应说明的是： 以上实施例仅用以说明本发明的技术方案， 而非对其 限制； 尽管参照前述实施例对本发明进行了详细的说明， 本领域的普通技术 人员应当理解： 其依然可以对前述各实施例所记载的技术方案进行修改， 或 者对其中部分技术特征进行等同替换； 而这些修改或者替换， 并不使相应技 术方案的本质脱离本发明各实施例技术方案的精神和范围。

Claims

权 利 要求
1、 一种微带天线， 其特征在于， 包括： 平行设置的四层介质板； 第一层 介质板上表面中心处设有第一微带贴片； 第二层介质板上表面中心处设有第 二微带贴片； 第三层介质板上表面设有第一接地层， 所述第一接地层中心处 开设耦合窗口， 所述第三层介质板下表面中心处设有中心导体； 第四层介质 板下表面设有第二接地层；
所述第三层介质板与所述第四层介质板为非对称介质， 以使所述中心导 体上方电场大于下方电场。
2、 根据权利要求 1所述的微带天线， 其特征在于， 所述第三层介质板的 介电常数大于所述第四层介质板的介电常数， 和 /或所述第三层介质板的厚度 小于所述第四层介质板的厚度。
3、根据权利要求 1或 2所述的微带天线， 其特征在于， 所述耦合窗口周围 的所述第三介质板和所述第四介质板中设有非谐振腔， 所述非谐振腔连接所 述第一接地层和所述第二接地层。
4、 根据权利要求 3所述的微带天线， 其特征在于， 所述非谐振腔由至少 四个金属柱构成， 所述金属柱连接所述第一接地层和所述第二接地层。
5、 根据权利要求 4所述的微带天线， 其特征在于， 所述非谐振腔由四个 所述金属柱构成， 所述非谐振腔中相邻两个所述金属柱的距离均小于 1 /2介质 波长。
6、 根据权利要求 1所述的微带天线， 其特征在于， 所述耦合窗口为长条 形缝隙， 所述耦合窗口的长度小于 1/2介质波长。
7、 根据权利要求 6所述的微带天线， 其特征在于， 所述中心导体中超出 所述耦合窗口边缘部分的长度小于 1/4介质波长， 未超出所述耦合窗口边缘部 分的长度等于 1 /4介质波长， 所述中心导体中超出所述耦合窗口边缘部分以及 未超出所述耦合窗口边缘部分的宽度均大于所述中心导体其他部分的宽度。
8、 根据权利要求 1所述的微带天线， 其特征在于， 所述第一微带贴片与 所述第二微带贴片的中心重叠。
9、根据权利要求 1或 8所述的微带天线， 其特征在于， 所述第一微带贴 片的边长略大于 1/2介质波长， 所述第二啟带贴片的边长略小于 1/2介质波 长。
PCT/CN2011/073006 2011-04-19 2011-04-19 一种微带天线 WO2011103841A2 (zh)

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