WO2009152704A1 - 一种信道估计方法、装置和系统 - Google Patents

一种信道估计方法、装置和系统 Download PDF

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Publication number
WO2009152704A1
WO2009152704A1 PCT/CN2009/071246 CN2009071246W WO2009152704A1 WO 2009152704 A1 WO2009152704 A1 WO 2009152704A1 CN 2009071246 W CN2009071246 W CN 2009071246W WO 2009152704 A1 WO2009152704 A1 WO 2009152704A1
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WIPO (PCT)
Prior art keywords
channel
interpolation
subcarriers
unit
error signal
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PCT/CN2009/071246
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English (en)
French (fr)
Inventor
方李明
卫东
陈子欢
辛德瑞那⋅拉斐尔
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华为技术有限公司
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Publication date
Priority claimed from CN200810067958.7A external-priority patent/CN101610103B/zh
Application filed by 华为技术有限公司 filed Critical 华为技术有限公司
Priority to EP09765344A priority Critical patent/EP2282409A4/en
Publication of WO2009152704A1 publication Critical patent/WO2009152704A1/zh
Priority to US12/971,389 priority patent/US8582688B2/en

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/06—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station
    • H04B7/0613—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission
    • H04B7/0615—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission of weighted versions of same signal
    • H04B7/0619—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission of weighted versions of same signal using feedback from receiving side
    • H04B7/0621—Feedback content
    • H04B7/0623—Auxiliary parameters, e.g. power control [PCB] or not acknowledged commands [NACK], used as feedback information
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B3/00—Line transmission systems
    • H04B3/02—Details
    • H04B3/46—Monitoring; Testing
    • H04B3/487—Testing crosstalk effects
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0204—Channel estimation of multiple channels
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/022—Channel estimation of frequency response
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0224—Channel estimation using sounding signals
    • H04L25/0228—Channel estimation using sounding signals with direct estimation from sounding signals
    • H04L25/023—Channel estimation using sounding signals with direct estimation from sounding signals with extension to other symbols
    • H04L25/0232—Channel estimation using sounding signals with direct estimation from sounding signals with extension to other symbols by interpolation between sounding signals
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03343—Arrangements at the transmitter end
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00—Arrangements affording multiple use of the transmission path
    • H04L5/0001—Arrangements for dividing the transmission path
    • H04L5/0003—Two-dimensional division
    • H04L5/0005—Time-frequency
    • H04L5/0007—Time-frequency the frequencies being orthogonal, e.g. OFDM(A), DMT
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04M—TELEPHONIC COMMUNICATION
    • H04M11/00—Telephonic communication systems specially adapted for combination with other electrical systems
    • H04M11/06—Simultaneous speech and data transmission, e.g. telegraphic transmission over the same conductors
    • H04M11/062—Simultaneous speech and data transmission, e.g. telegraphic transmission over the same conductors using different frequency bands for speech and other data

Definitions

  • the present invention relates to the field of communications technologies, and in particular, to a channel estimation method, apparatus, and system. Background technique
  • XDSL digital subscriber line
  • UDP Unshielded Twist Pair
  • xDSL with passband transmission uses frequency division multiplexing technology to coexist xDSL with traditional telephony service (POTS) on the same pair of twisted pairs, where xDSL occupies the high frequency band and POTS occupies the baseband below 4 kHz.
  • POTS traditional telephony service
  • the POTS signal and the xDSL signal are separated by a splitter, and the xDSL transmitted by the passband uses discrete multi-tone modulation (DMT).
  • DMT discrete multi-tone modulation
  • vector DSL vectored-DSL
  • FIG. 1 A schematic structural diagram of synchronously transmitting signals for an existing DSLAM; as shown in FIG. 2, a schematic structural diagram of a synchronous DSLAM receiving signal.
  • the shared channel H shown in Figures 1 and 2 can be represented in the form of a matrix in the frequency domain:
  • H (1 ⁇ ⁇ N, 1 ⁇ ⁇ N) represents the crosstalk channel transfer function of the pair pair ' ' on the kth tone
  • H 1 ⁇ ⁇ N) indicates the line pair '' at the kth tone
  • N is the line number, which is the number of users.
  • ⁇ is an NxN channel transmission matrix.
  • X be an Nxl channel input vector
  • Y is a Nxl channel output vector
  • N is an Nxl channel noise vector.
  • the central office performs the joint reception processing of the signal, and introduces a crosstalk canceller W at the receiving end, and the received signal is:
  • crosstalk canceller enables WH to be a diagonal matrix, crosstalk can be eliminated.
  • the joint transmission processing of the signal at the CO end introduces a precoder P at the CO end, and the signal received by the receiving end is:
  • crosstalk precoder When the crosstalk precoder enables HP to be a diagonal matrix, crosstalk can be eliminated.
  • the methods for calculating the crosstalk canceller and the crosstalk precoder include first-order approximation and the like, all of which need to be calculated according to the channel transmission matrix. Therefore, in order to eliminate crosstalk, the crosstalk channel must be estimated first to obtain a channel transmission matrix.
  • the crosstalk channel is estimated at the CO end, which is easier to implement.
  • the downlink crosstalk channel still needs to be estimated at the CO end, so the terminal needs to cooperate with the feedback error signal to the CO end.
  • the CO end estimates the crosstalk channel by using the error signal fed back by the terminal, and then calculates the precoder based on the obtained crosstalk channel. Since the feedback channel uses the uplink channel for feedback of the error signal, it occupies a certain amount of uplink channel capacity. Convergence of channel estimation The speed depends on the capacity of the feedback channel and the data of the feedback error signal.
  • Embodiments of the present invention provide a channel estimation method, apparatus, and system, which speed up convergence of channel estimation by reducing data of an error signal fed back by a terminal.
  • An embodiment of the present invention provides a method for channel estimation, where the method includes:
  • the error signal is an error signal of at least two subcarriers in a downlink frequency band
  • An embodiment of the present invention further provides an apparatus for channel estimation, including:
  • a receiving unit configured to receive an error signal fed back by the terminal, where the error signal is an error signal of at least two subcarriers in a downlink frequency band;
  • a first channel acquiring unit configured to obtain, according to an error signal received by the receiving unit, a channel of the at least two subcarriers
  • a second channel acquiring unit configured to perform interpolation on channels of the at least two subcarriers to obtain channels of the remaining subcarriers in the downlink frequency band.
  • the embodiment of the present invention further provides a digital subscriber line system, including a terminal and an access device, where the terminal is configured to feed back an error signal to the access device, where the error signal is an error of at least two subcarriers in a downlink frequency band.
  • a digital subscriber line system including a terminal and an access device, where the terminal is configured to feed back an error signal to the access device, where the error signal is an error of at least two subcarriers in a downlink frequency band.
  • the access device is configured to receive an error signal fed back by the terminal, obtain a channel of the at least two subcarriers according to the received error signal, and perform interpolation on a channel of the at least two subcarriers to obtain the remaining subbands in the downlink frequency band.
  • the channel of the carrier is configured to receive an error signal fed back by the terminal, obtain a channel of the at least two subcarriers according to the received error signal, and perform interpolation on a channel of the at least two subcarriers to obtain the remaining subbands in the downlink frequency band.
  • the channel estimation method, apparatus and system provided by the embodiments of the present invention accelerate the convergence rate of the channel estimation by reducing the data of the error signal fed back by the terminal.
  • DRAWINGS 1 is a schematic structural diagram of a conventional DSLAM synchronous transmission signal
  • FIG. 2 is a schematic structural diagram of a synchronous receiving signal of an existing DSLAM
  • FIG. 3 is a flowchart of a channel estimation method according to an embodiment of the present invention.
  • FIG. 5 is a schematic diagram of channel amplitude, phase, and error obtained by interpolation method 2 according to an embodiment of the present invention
  • FIG. 6 is a schematic diagram of a channel estimation apparatus according to an embodiment of the present invention.
  • FIG. 7 is a schematic structural diagram of a DSLAM according to an embodiment of the present invention.
  • the embodiment of the present invention performs channel estimation by reducing data of an error signal fed back by the terminal.
  • the channel estimation method provided by the embodiment of the present invention includes the following steps: S301: Receive an error signal fed back by a terminal, where the error signal is an error signal of at least two subcarriers in a downlink frequency band;
  • S302 Obtain a channel of the at least two subcarriers according to the error signal.
  • S303 Interpolate channels of the at least two subcarriers to obtain channels of the remaining subcarriers in the downlink frequency band.
  • S301 The error signal fed back by the receiving terminal, where the error signal is an error signal of at least two subcarriers in the downlink frequency band.
  • the downlink frequency band is divided into a plurality of sub-carriers, and the channel characteristics of the adjacent sub-carriers (tones) are relatively close.
  • the channels of each tone can be estimated without performing channel estimation, and at least part of the tone, that is, at least Two tone channels are estimated, where the tone channel Includes amplitude and phase information.
  • VDSL2 very high speed Digital Subscriber Line 2
  • the downlink frequency band DS1 is / ⁇
  • DS2 is y ⁇ / 4
  • the subcarrier spacing is ⁇ /.
  • a tone for the feedback error signal is selected, wherein the error signal for feedback can be selected by the terminal, or the error signal for feedback can be selected by the transmitting end.
  • S302 Obtain a channel of the at least two subcarriers according to an error signal fed back by the terminal in S301.
  • the obtaining the subcarrier channel may use a method based on the orthogonal sequence to calculate the channel, which is of course not limited to the method.
  • the following is an example of a method based on an orthogonal sequence.
  • the real and imaginary parts of the feedback error signal are respectively modulated by two orthogonal sequences
  • t n (l) Re ⁇ M. ⁇ 5 n (l)+ _ Im ⁇ M. ⁇ 5*2 n (l)
  • the channel can be found by the real or imaginary part of the error signal, if the feedback is the real part of the error signal:
  • the normalized channel can also be calculated.
  • S303 Interpolate the channels of the at least two subcarriers obtained in S302 to obtain channels of the remaining subcarriers in the downlink frequency band.
  • the interpolation function can be used to calculate the channel of the remaining tone.
  • the interpolation function may use an interpolation function such as a linear interpolation function or a Gaussian interpolation function, but is not limited to the above two interpolation functions. There are two ways to do interpolation:
  • Interpolation method 2 Interpolating the amplitude and phase information of the channels of the at least two subcarriers obtained in step S302, respectively, to obtain the amplitude and phase of the remaining tones in the downlink frequency band, thereby obtaining channels of all tone. If you use a linear interpolation function, the magnitude is:
  • the phase is:
  • H ⁇ i) H()
  • exp(7'0()) , i k,, ''.k 2 ,k 3 ,'', k 4
  • the interpolation function used is a linear interpolation function.
  • the crosstalk channel is measured data.
  • the downlink frequency band DS1 is 276 kHz to 3.75 MHz
  • the DS2 is 5.2 MHz to 8.5 MHz
  • the subcarrier spacing is 4.3125 kHz.
  • the down tone is the 64th tone to the 870th tone, and the 1205th tone to the 1970th tone.
  • the channel amplitude represents the amplitude signal of the crosstalk channel.
  • the dotted line is the amplitude of the actually measured channel, the broken line is the crosstalk channel amplitude calculated by the embodiment of the present invention;
  • the channel phase represents the phase information of the crosstalk channel, which is the phase of the actually measured channel, and is the invention The crosstalk channel amplitude calculated by the embodiment. D
  • Fig. 4 It can be seen from Fig. 4 that the channel amplitude and phase calculated by the embodiment of the present invention are substantially the same as the actually measured channel amplitude and phase.
  • the error curve between the actual channel and the channel calculated by the embodiment of the present invention is also shown in Fig. 4, and it can be seen that the relative error is below 2e-3.
  • the relative errors of all subcarriers are averaged to give an average relative error of 4.2757e-004.
  • the channel amplitude represents the amplitude information of the crosstalk channel, which is the amplitude of the measured channel, and is the crosstalk channel amplitude calculated by the embodiment of the present invention
  • the channel phase represents the phase information of the crosstalk channel, which is the phase of the measured channel, which is The crosstalk channel amplitude calculated by the embodiment of the present invention.
  • the channel amplitude and phase calculated using the embodiment of the present invention are substantially the same as the actually measured channel amplitude and phase.
  • the error curve between the actual channel and the channel calculated by the embodiment of the present invention is also shown in the figure. It can also be seen that the relative error is below 7e-4.
  • the relative error of all subcarriers is averaged to give an average relative error of 5.4587e-005.
  • the channel of the embodiment of the present invention can be used to estimate a crosstalk channel, and can also be used to estimate a direct channel.
  • the embodiment of the present invention further provides an apparatus for channel estimation, where the apparatus includes: a receiving unit, configured to receive an error signal fed back by a terminal, where the error signal is at least two subcarriers in a downlink frequency band. Error signal; a first channel acquisition unit, configured to receive according to the receiving unit Obtaining an error signal, obtaining a channel of the at least two subcarriers; a second channel acquiring unit, configured to perform interpolation on a channel of the at least two subcarriers to obtain a channel of the remaining subcarriers in the downlink frequency band.
  • the second channel acquiring unit includes: an interpolation unit and an obtaining unit. And an interpolating unit, configured to perform interpolation on a channel of the at least two subcarriers obtained by the first channel acquiring unit, and an acquiring unit, configured to obtain a channel of the remaining subcarriers according to an interpolation result of the interpolating unit.
  • the interpolation unit includes a first interpolation unit and a second interpolation unit.
  • a first interpolation unit configured to perform interpolation on a magnitude of a channel of the at least two subcarriers obtained by the first channel acquiring unit
  • a second interpolation unit configured to perform, on the at least two subcarriers obtained by the first channel acquiring unit
  • the phase of the channel is interpolated.
  • the obtaining unit is configured to obtain a channel of the remaining subcarriers in the downlink frequency band according to the amplitude value obtained by the first interpolation unit and the phase value obtained by the second interpolation unit.
  • the interpolation method may use an interpolation function such as a linear interpolation function or a Gaussian interpolation function, but is not limited to the above two interpolation functions.
  • the device for channel estimation can be integrated in a Digital Subscriber Liner Access Multiplexer (DSLAM).
  • DSLAM Digital Subscriber Liner Access Multiplexer
  • an embodiment of the present invention further provides a digital subscriber line system, where the system includes a terminal and an access device.
  • the terminal is configured to feed back an error signal to the access device, where the error signal is an error signal of at least two subcarriers in a downlink frequency band;
  • the access device is configured to receive an error signal fed back by the terminal, obtain a channel of the at least two subcarriers according to the received error signal, and perform interpolation on the channels of the at least two subcarriers to obtain the remaining subcarriers in the downlink frequency band. channel.
  • the access device includes: a receiving unit, configured to receive an error signal fed back by the terminal; a first channel acquiring unit, configured to obtain, according to an error signal received by the receiving unit, a channel of the at least two subcarriers; And an obtaining unit, configured to perform interpolation on channels of the at least two subcarriers to obtain channels of the remaining subcarriers in the downlink frequency band.
  • the second channel acquiring unit includes: an interpolation unit and an obtaining unit.
  • An interpolating unit configured to perform interpolation on a channel of the at least two subcarriers obtained by the first channel acquiring unit, and an acquiring unit, configured to obtain, according to an interpolation result of the interpolating unit, a channel of the remaining subcarriers.
  • the interpolation unit includes a first interpolation unit and a second interpolation unit.
  • a first interpolation unit configured to perform interpolation on a magnitude of a channel of the at least two subcarriers obtained by the first channel acquiring unit
  • a second interpolation unit configured to perform, on the at least two subcarriers obtained by the first channel acquiring unit
  • the phase of the channel is interpolated.
  • the acquiring unit is configured to obtain a channel of the remaining subcarriers according to the amplitude value obtained by the first interpolation unit and the phase value obtained by the second interpolation unit.
  • the interpolation method may use an interpolation function such as a linear interpolation function or a Gaussian interpolation function, but is not limited to the above two interpolation functions.
  • the access device may be a Digital Subscriber Line Access Multiplexer (DSLAM).
  • DSLAM Digital Subscriber Line Access Multiplexer
  • the above-mentioned storage medium may be a read only memory, a magnetic disk or an optical disk or the like.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
  • Telephonic Communication Services (AREA)

Description

一种信道估计方法、 装置和系统
本申请要求于 2008 年 6 月 20 日提交中国专利局、 申请号为 200810067958. 7 , 发明名称为 "一种信道估计方法、 装置和系统" 的中国专 利申请的优先权， 其全部内容通过引用结合在本申请中。
技术领域
本发明涉及通信技术领域， 尤其涉及一种信道估计方法、 装置和系统。 背景技术
不同种类的数字用户线路 ( X Digital Subscriber Line, XDSL )技术是一 种在电话双绞线， 即无屏蔽双绞线（Unshielded Twist Pair, UTP )传输的高速 数据传输技术， 除了基带传输的数字用户线路（Digital Subscriber Line, DSL ) 外， 通带传输的 xDSL利用频分复用技术使得 xDSL与传统电话业务 ( POTS ) 共存于同一对双绞线上， 其中 xDSL占据高频段， POTS占用 4KHz以下基带 部分， POTS信号与 xDSL信号通过分离器分离， 通带传输的 xDSL釆用离散 多音频调制 （DMT )。
随着 xDSL技术使用频带的提高， 串扰（ Crosstalk )尤其是高频段的串扰 问题表现得日益突出，由于 xDSL上下行信道釆用频分复用，近端串扰 ( NEXT ) 对系统的性能产生的危害不太大； 但远端串扰（FEXT )会严重影响线路的传 输性能； 当一捆电缆内有多路用户都要求开通 xDSL业务时，会因为远端串扰 ( FEXT )使一些线路速率低、 性能不稳定、 甚至不能开通等， 最终导致 DSL 接入复用器（DSLAM ) 的出线率比较低。
目前， 向量 DSL ( vectored - DSL )技术主要是利用在 DSLAM端进行联 合收发的可能性， 使用信号处理的方法来抵消 FEXT 的干扰， 最终使每一路 信号中不存在 FEXT干扰， 如图 1所示， 为现有 DSLAM同步发送信号的结 构示意图； 如图 2所示， 为现有 DSLAM同步接收信号的结构示意图。
将图 1、 图 2所示的共享信道 H在频率域可以表示为矩阵形式：
Figure imgf000004_0001
其中， H (1≤ ≤N,1≤ ≤N)表示线对 对线对' '在第 k个 tone上的串扰信道传 递函数， H 1≤ ≤N)表示线对''在第 k个 tone上的直接信道传递函数， N为线对 数， 即用户数。 那么 Η是一个 NxN的信道传输矩阵。 为方便起见， 在后面的 表述中忽略上标1^。 又分别设 X是一个 Nxl的信道输入向量， Y是一个 Nxl的 信道输出向量， N是一个 Nxl的信道噪声向量。 最终， 把信道传输方程表达为 如下形式：
Υ=ΗΧ+Ν
对于上行， 在局端 （Central Office, CO)做信号的联合接收处理， 在接 收端引入一个串扰抵消器 W, 接收到的信号为：
Y^WY-WHX + WN
若串扰消除器能使得 WH为一对角矩阵， 串扰可以得到消除。
对于下行，在 CO端做信号的联合发送处理， 在 CO端引入一个预编码器 P, 那么接收端接收到的信号为：
Ϋ-ΗΡΧ + Ν
当串扰预编码器能使得 HP为一对角矩阵， 串扰可以得到消除。
计算串扰消除器和串扰预编码器的方法包括一阶逼近等方法， 这些方法 都需要根据信道传输矩阵来进行计算。 因此为了消除串扰， 必须先对串扰信 道进行估计， 获得信道传输矩阵。
对于上行， 串扰信道在 CO端进行估计， 较容易实现。 对于下行， 由于只 在 CO端进行联合处理， 下行串扰信道还是需要在 CO端进行估计， 因此需要 终端配合反馈误差信号至 CO端。 CO端利用终端反馈的误差信号估计出串扰 信道， 然后根据得到的串扰信道计算出预编码器。 由于反馈信道利用上行信 道进行误差信号的反馈， 因此占用了一定的上行信道容量。 信道估计的收敛 速度依赖于反馈信道的容量和反馈的误差信号的数据。
发明内容
本发明实施例提供了一种信道估计方法、 装置及系统， 通过减少终端反 馈的误差信号的数据来加快信道估计的收敛速度。
本发明实施例提供了一种信道估计的方法， 该方法包括：
接收终端反馈的误差信号， 所述误差信号是下行频带中的至少两个子载 波的误差信号；
根据所述误差信号， 获得所述至少两个子载波的信道；
对所述至少两个子载波的信道进行插值， 获得下行频带中其余子载波的 信道。
本发明实施例还提供了一种信道估计的装置， 包括：
接收单元， 用于接收终端反馈的误差信号， 所述误差信号是下行频带中 的至少两个子载波的误差信号；
第一信道获取单元， 用于根据接收单元所接收的误差信号， 获得所述至 少两个子载波的信道；
第二信道获取单元， 用于对所述至少两个子载波的信道进行插值， 获得 下行频带中其余子载波的信道。
本发明实施例还提供了一种数字用户线系统， 包括终端和接入设备， 所述终端， 用于向接入设备反馈误差信号， 所述误差信号是下行频带中 的至少两个子载波的误差信号；
所述接入设备， 用于接收终端反馈的误差信号， 根据所接收的误差信号， 获得所述至少两个子载波的信道； 对所述至少两个子载波的信道进行插值， 获得下行频带中其余子载波的信道。
本发明实施例提供的信道估计方法、 装置和系统， 通过减少终端反馈的 误差信号的数据， 加快信道估计的收敛速度。
附图说明 图 1为现有 DSLAM同步发送信号的结构示意图；
图 2为现有 DSLAM同步接收信号的结构示意图；
图 3为本发明实施例提供的信道估计方法流程图；
图 4为本发明实施例提供的用插值方法一得到的信道幅度、 相位以及误 差曲线图；
图 5为本发明实施例提供的用插值方法二得到的信道幅度、 相位以及误 差曲线图；
图 6为本发明实施例提供的信道估计装置示意图；
图 7为本发明实施例提供的 DSLAM结构示意图。
具体实施方式
为了加快信道估计的收敛速度， 可以通过增大反馈信道的信道容量实现， 也可以通过减少终端 CPE反馈的误差信号的数据实现。 增大反馈信道的信道 容量会带来较大的开销， 降低数据传输效率， 带来一定的损失。 本发明实施 例通过降低终端反馈的误差信号的数据， 来进行信道估计。
如图 3所示， 本发明实施例提供的信道估计方法， 包括下列步骤： S301 : 接收终端反馈的误差信号， 所述误差信号是下行频带中的至少两 个子载波的误差信号；
S302: 根据所述误差信号， 获得所述至少两个子载波的信道；
S303 : 对所述至少两个子载波的信道进行插值， 获得下行频带中其余子 载波的信道。
下面结合附图对本发明实施例提供的信道估计方法进行详细描述： S301 : 接收终端反馈的误差信号， 所述误差信号是下行频带中的至少两 个子载波的误差信号。
下行频带被分成多个子载波， 邻近子载波（tone ) 的信道特性比较接近， 利用这一特性， 可以在进行信道估计时不对每个 tone的信道都进行估计， 而 对其中的一部分 tone, 即至少两个 tone的信道进行估计， 其中， tone的信道 包括幅度和相位信息。
例如， 对于 VDSL2 ( Very High Speed Digital Subscriber Line 2 , 第二代甚 高速数字用户环路）， 假设下行频带 DS1为/ ^Λ, DS2为 y〜 /4, 子载波间 隔为 Δ/。下行 tone为第 个 tone〜第 k2个 tone,以及第 个 tone〜第 k4个 tone。
首先， 选择用于反馈误差信号的 tone, 其中， 可以由终端来选择用于反 馈的误差信号， 也可以由发送端来选择用于反馈的误差信号。 这里选择每隔 M-1个 tone (即每 M个 tone )反馈误差信号进行信道估计， 即第 ^ +Mx(i-l) 个 tone ( = !,···
Figure imgf000007_0001
表示向下取整。
S302: 根据 S301中终端反馈的误差信号， 获得所述至少两个子载波的信 道。
其中， 获取子载波信道可以釆用基于正交序列的方法来计算信道， 当然 不局限于该方法。 下面以基于正交序列的方法为例进行说明。
反馈的误差信号的实部和虚部分别由两个正交序列调制， 为
tn(l) = Re{M。}5 n(l)+ _ Im{M。}5*2n(l)
不同线路的正交序列两两正交， 实部与虚部的正交序列也两两正交， 即 c- =1
Figure imgf000007_0002
(1,2
4叚设归一化的信道 „m = ^ L = am +jbm , (m≠n,m = 1,2,... ,N),得出判决误差
Figure imgf000007_0003
通过误差信号的实部或者虚部就可以求出信道， 如果反馈的是误差信号 的实部：
N
Rc{en (λ)} = ^ [am Re{w。 ( ) - bm Im{w。 )S2m ( )]
m≠n
那么根据误差信号的实部， 计算用户 i对用户 n的信道实部为 bm lm{ Q }S2m ( ) Ξΐ λ)
Figure imgf000008_0001
O J
得出归一化的信道的实部为
Figure imgf000008_0002
同理可以求出用户 i对用户 n的归一化信道的虚部为
L
^Re{e„(^)}« 52; (^)
b. = 从而得到归一化的信道。
同样的， 如果反馈的是误差信号的虚部， 那么也可以计算出归一化的信 道。
S303: 对 S302中所获得的至少两个子载波的信道进行插值， 获得下行频 带中其余子载波的信道。
具体的， 可以釆用内插函数计算出其余 tone的信道。 内插函数可以釆用 线性内插函数、 高斯内插函数等内插函数， 但不局限于上述两种内插函数。 进行插值有两种方法：
插值方法一、 对 S302中获得的至少两个子载波的信道进行插值， 从而获 得所有 tone的信道。 如果釆用线性内插函数， 得出： H[t + M x(i - I) + m] =
Figure imgf000009_0001
M-m k
H[k3+Mx(i-l) + m] H[k3 +Mx(i-l)] +― H[k3+Mxi], i + 1
M M M
式中， 》? = 1，··.，Μ_1。
插值方法二、 对步骤 S302中获得的至少两个子载波的信道的幅度和相位 信息分别进行插值， 得到下行频带中其余 tone的幅度和相位， 从而获得所有 tone的信道。 如果釆用线性内插函数， 得出幅度为：
M-m k2—k、
H[kl + Mx(i-l) + m H[^ +Mx(i- 1)1 +― H[^ +Mxi] , i = l + 1
M M M
Figure imgf000009_0002
式中， (表示第 i个 tone的信道幅度，
相位为：
Α/ί γη ^ ίΥΙ ^ — k
[k{ + Mx(i-\) + m] Φ[^, + χ(ζ-1)] +— Φ[^, +Μχί] + 1
Μ Μ Μ
, Μ— m ' "
Φ| 3 + χ (ζ' - 1) + w] = + Μχ(ί-ϊ)] + ηιΦ[^ +Μχί] , i = 1,···, + 1
Μ Μ
式中， 表示第 i个 tone的信道相位， 》? = 1，···，Μ_1。
最后， 根据计算出的信道幅度和相位值， 得出所有 tone的信道。 H{i) = H()| exp(7'0()) , i = k、,''.k2,k3,''、k4
为 300m, 耦合长度也为 300m的串扰信道进行仿真。 釆用的内插函数是线性 内插函数。 串扰信道是实测数据。 下行频带 DS1为 276kHz〜 3.75MHz, DS2 为 5.2MHz〜 8.5MHz,子载波间隔为 4.3125kHz。下行 tone为第 64个 tone〜 第 870个 tone, 以及第 1205个 tone〜第 1970个 tone。
图 4是以 M=2的情况进行仿真， 实际信道和利用插值方法一估算得出的 信道幅度、 相位以及误差曲线图。 图 4 中， 信道幅度表示串扰信道的幅度信 息， 点划线 是实际测量的信道的幅度，虚线 是釆用本发明实施例 计算出的串扰信道幅度； 信道相位表示串扰信道的相位信息， 是实际 测量的信道的相位， 是釆用本发明实施例计算出的串扰信道幅度。 从图
4中可以看出，釆用本发明实施例计算得出的信道幅度和相位与实际测量的信 道幅度和相位基本一样。 在图 4 中还给出了实际信道与釆用本发明实施例计 算得出的信道之间的误差曲线， 同样可以看出相对误差在 2e-3 以下。 将所有 子载波的相对误差取平均得出平均相对误差为 4.2757e-004。
图 5是以 M=2的情况进行仿真， 实际信道和利用插值方法二估算得出的 信道幅度、 相位以及误差曲线图。 图 5 中， 信道幅度表示串扰信道的幅度信 息， 是实测信道的幅度， 是釆用本发明实施例计算出的串扰信道 幅度； 信道相位表示串扰信道的相位信息， 是实测信道的相位， 是釆用本发明实施例计算出的串扰信道幅度。 从图 5 中可以看出， 釆用本发 明实施例计算得出的信道幅度和相位与实际测量的信道幅度和相位基本一 样。 在图中还给出了实际信道与釆用本发明实施例计算得出的信道之间的误 差曲线， 同样可以看出相对误差在 7e-4以下。 将所有子载波的相对误差取平 均得出平均相对误差为 5.4587e-005。
从上面的结果可以看出， 在每 2个 tone反馈误差信号的情况下， 带来的误 差都是较小的， 但其反馈的数据量却减少为原来的 1/2 , 大大降低了反馈数据 量， 艮大程度地提高信道估计算法的收敛速度。
以 M = 6的情况进行仿真， 实际信道和利用方法一插值估算得到的平均相 对误差为 0.0050。 实际信道和利用插值方法二估算得到的平均相对误差为 5.3223e-004。 这时， 反馈的误差数据量减少为原来的 1/6。
本发明实施例的信道可以用来估计串扰信道， 也可以用来估计直接信道。 如图 6所示， 本发明实施例还提供了一种信道估计的装置， 该装置包括： 接收单元， 用于接收终端反馈的误差信号， 所述误差信号是下行频带中 的至少两个子载波的误差信号； 第一信道获取单元， 用于根据接收单元所接 收的误差信号， 获得所述至少两个子载波的信道； 第二信道获取单元， 用于 对所述至少两个子载波的信道进行插值， 获得下行频带中其余子载波的信道。
具体实施方式如方法实施例中所述， 这里不再详细描述。
进一步的， 第二信道获取单元包括： 插值单元和获取单元。 插值单元， 用于对所述第一信道获取单元获得的至少两个子载波的信道进行插值； 获取 单元， 用于根据插值单元的插值结果得到其余子载波的信道。
进一步的， 插值单元包括第一插值单元和第二插值单元。 第一插值单元， 用于对所述第一信道获取单元获得的至少两个子载波的信道的幅度进行插 值； 第二插值单元， 用于对所述第一信道获取单元获得的至少两个子载波的 信道的相位进行插值。 所述获取单元， 用于根据第一插值单元得到的幅度值 和第二插值单元得到的相位值得到下行频带中其余子载波的信道。
其中， 所述的插值方法可以釆用线性内插函数、 高斯内插函数等内插函 数， 但不局限于上述两种内插函数。
所述的信道估计的装置可以集成在数字用户线路接入复用器 （ Digital Subscriber Liner Access Multiplexer, DSLAM ) 中。
如图 7所示， 本发明实施例还提供了一种数字用户线系统， 该系统包括终 端和接入设备。
所述终端， 用于向接入设备反馈误差信号， 所述误差信号是下行频带中 的至少两个子载波的误差信号；
所述接入设备， 用于接收终端反馈的误差信号， 根据所接收的误差信号， 获得至少两个子载波的信道； 对所述至少两个子载波的信道进行插值， 获得 下行频带中其余子载波的信道。
所述接入设备， 包括： 接收单元， 用于接收终端反馈的误差信号； 第一 信道获取单元， 用于根据接收单元所接收的误差信号， 获得所述至少两个子 载波的信道； 第二信道获取单元， 用于对所述至少两个子载波的信道进行插 值， 获得下行频带中其余子载波的信道。 进一步的， 第二信道获取单元包括： 插值单元和获取单元。 插值单元， 用于对所述第一信道获取单元获得的至少两个子载波的信道进行插值； 获取 单元， 用于根据插值单元的插值结果得到所述其余子载波的信道。
进一步的， 插值单元包括第一插值单元和第二插值单元。 第一插值单元， 用于对所述第一信道获取单元获得的至少两个子载波的信道的幅度进行插 值； 第二插值单元， 用于对所述第一信道获取单元获得的至少两个子载波的 信道的相位进行插值。 所述获取单元， 用于根据第一插值单元得到的幅度值 和第二插值单元得到的相位值得到所述其余子载波的信道。
其中， 所述的插值方法可以釆用线性内插函数、 高斯内插函数等内插函 数， 但不局限于上述两种内插函数。
所述的接入设备可以是数字用户线接入复用器 （DSLAM , Digital Subscriber Line Access Multiplexer ) 。
本领域普通技术人员可以理解实现上述方法实施例中的全部或部分步骤 是可以通过程序来指令相关的硬件完成， 所述的程序可以存储于一种计算机 可读存储介质中。
上述提到的存储介质可以是只读存储器， 磁盘或光盘等。
最后应说明的是： 以上实施例仅用以说明本发明的技术方案， 而非对其 限制； 尽管参照前述实施例对本发明进行了详细的说明， 本领域的普通技术 人员应当理解： 其依然可以对前述各实施例所记载的技术方案进行修改， 或 者对其中部分技术特征进行等同替换； 而这些修改或者替换， 并不使相应技 术方案的本质脱离本发明各实施例技术方案的精神和范围。

Claims

权利 要 求 书
1、 一种信道估计方法， 其特征在于， 包括：
接收终端反馈的误差信号， 所述误差信号是下行频带中的至少两个子载波 的误差信号；
根据所述误差信号， 获得所述至少两个子载波的信道；
对所述至少两个子载波的信道进行插值， 获得下行频带中其余子载波的信 道。
2、 如权利要求 1所述的信道估计方法， 其特征在于， 所述对至少两个子载 波的信道进行插值具体包括： 分别对至少两个子载波的信道幅度和相位进行插 值。
3、 如权利要求 1或 2所述的信道估计方法， 其特征在于， 所述的插值包括 用线性内插函数或高斯内插函数进行插值。
4、 如权利要求 1或 2所述的信道估计方法， 其特征在于， 所述获得所述至 少两个子载波的信道釆用基于正交序列的方法来获取所述子载波的信道。
5、 一种信道估计的装置， 该装置包括：
接收单元， 用于接收终端反馈的误差信号， 所述误差信号是下行频带中的 至少两个子载波的误差信号；
第一信道获取单元， 用于根据接收单元所接收的误差信号， 获得所述至少 两个子载波的信道；
第二信道获取单元， 用于对所述至少两个子载波的信道进行插值， 获得下 行频带中其余子载波的信道。
6、 如权利要求 5所述的信道估计的装置， 其特征在于， 所述第二信道获取 单元包括插值单元和获取单元；
所述插值单元， 用于对所述第一信道获取单元获得的至少两个子载波的信 道进行插值；
所述获取单元， 用于根据插值单元的插值结果得到下行频带中其余子载波 的信道。
7、 如权利要求 6所述的信道估计的装置， 其特征在于， 所述插值单元包括 第一插值单元和第二插值单元；
所述第一插值单元， 用于对所述第一信道获取单元获得的至少两个子载波 的信道的幅度进行插值；
所述第二插值单元， 用于对所述第一信道获取单元获得的至少两个子载波 的信道的相位进行插值。
8、 如权利要求 7所述的信道估计的装置， 其特征在于， 所述获取单元， 用 于根据第一插值单元得到的幅度值和第二插值单元得到的相位值获得下行频带 中其余子载波的信道。
9、 如权利要求 5至 8任一项所述的信道估计的装置， 其特征在于， 所述装 置集成在数字用户线^ 入复用器中。
10、 一种数字用户线系统， 其特征在于， 该系统包括终端和接入设备， 所述终端， 用于向接入设备反馈误差信号， 所述误差信号是下行频带中的 至少两个子载波的误差信号；
所述接入设备， 用于接收终端反馈的误差信号， 根据所接收的误差信号， 获得所述至少两个子载波的信道； 对所述至少两个子载波的信道进行插值， 获 得下行频带中其余子载波的信道。
11、 如权利要求 10所述的数字用户线系统， 其特征在于， 所述接入设备包 括：
接收单元， 用于接收终端反馈的误差信号；
第一信道获取单元， 用于根据所述接收单元接收的误差信号， 获得所述至 少两个子载波的信道；
第二信道获取单元， 用于对所述至少两个子载波的信道进行插值， 获得下 行频带中其余子载波的信道。
12、 如权利要求 11所述的数字用户线系统， 其特征在于， 所述第二信道获 取单元包括插值单元和获取单元；
所述插值单元， 用于对所述第一信道获取单元获得的至少两个子载波的信 道进行插值；
所述获取单元， 用于根据插值单元的插值结果得到下行频带中其余子载波 的信道。
13、 如权利要求 12所述的数字用户线系统， 其特征在于， 所述插值单元包 括第一插值单元和第二插值单元；
所述第一插值单元， 用于对所述第一信道获取单元获得的至少两个子载波 的信道的幅度进行插值；
所述第二插值单元， 用于对所述第一信道获取单元获得的至少两个子载波 的信道的相位进行插值。
14、 如权利要求 13所述的数字用户线系统， 其特征在于， 所述获取单元， 用于根据第一插值单元得到的幅度值和第二插值单元得到的相位值获得下行频 带中其余子载波的信道。
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US8582688B2 (en) 2013-11-12

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