Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2647—Arrangements specific to the receiver only
  • H04L27/2655—Synchronisation arrangements
  • H04L27/2656—Frame synchronisation, e.g. packet synchronisation, time division duplex [TDD] switching point detection or subframe synchronisation
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/69—Spread spectrum techniques
  • H04B1/707—Spread spectrum techniques using direct sequence modulation
  • H04B1/7073—Synchronisation aspects
  • H04B1/7075—Synchronisation aspects with code phase acquisition
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2647—Arrangements specific to the receiver only
  • H04L27/2655—Synchronisation arrangements
  • H04L27/2668—Details of algorithms
  • H04L27/2681—Details of algorithms characterised by constraints
  • H04L27/2688—Resistance to perturbation, e.g. noise, interference or fading
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2602—Signal structure
  • H04L27/2605—Symbol extensions, e.g. Zero Tail, Unique Word [UW]
  • H04L27/2607—Cyclic extensions
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2602—Signal structure
  • H04L27/261—Details of reference signals
  • H04L27/2613—Structure of the reference signals

Definitions

• Time domain synchronous OFDM is one of the fundamental physical schemes of the DMB-T specification for terrestrial broadcasting in China, in which a multiplex frame structure is used as shown in FIG. 2.
• the frame structure is important for time synchronization in the DMB-T receiver.
• a signal frame is the basic element of the multiplex frame structure. As shown in FIG. 3, a signal frame consists of two time-domain signal parts, namely frame header and frame body. The frame header and frame body have the same baseband symbol data rate (7.56Msym/sec). In the signal frame header, a PN sequence is sent for purposes of synchronization and channel estimation. At the same time, the PN header also serves as a guard time interval for the following OFDM frame body in the place of the conventional cyclic prefix (CP).
• CP cyclic prefix
• the PN header with length of LPN includes three parts: a complete period of the PN sequence with length of N PN symbols, a length L pre PN preamble and a length L post PN post-amble.
• Three types of signal frame options are defined in the specification with different parameter combinations.
• type 2 the same segment of the PN sequence is used in every frame, which makes PN phase synchronization easier than in types 1 and 3, where different PN sequences are sent in different signal frames.
• the present invention concerns PN phase recovery for types 1 and 3.
• the PN sequences used in signal frames will change from frame to frame. Knowledge of PN sequences used in each frame is necessary for the synchronization of sampling frequency and sampling time phase, and for channel estimation. Also, it is necessary to maintain PN sequence phase synchronization throughout the reception procedure. As described herein, the object of PN phase recovery (PPR) is to recognize the index/of the ⁇ th signal frame given the received data sampled from several signal frame headers.
• the receiver may not necessarily always operate in an AWGN channel.
• the radio channel may experience severe fading as well as strong interference.
• the OFDM frame body part may unfortunately interfere with the frame header part.
• the peak of the correlation is not always easy to determine correctly due to possible strong noise and interference. Because of the PN phase change from frame to frame, traditional averaging or accumulation of the correlation result within multiple successive signal frames to suppress the noise and interference becomes impossible.
• sampling frequency error What makes the measurement of the time difference between the two peaks more difficult is sampling frequency error. There may be large sampling frequency error if a low-cost crystal is adopted. In digital processing receivers, the above correlation will be done using samples of the received signal. Large sampling frequency error will add more difficulties to PN phase recovery, because large sampling frequency error will cause uncertainty in the determination of the frame header from the analysis based on the sampled data.
• the PN phase needs to be captured as fast as possible.
• the receiver has little knowledge about the radio channel. It is therefore a challenge to capture the PN phase quickly and robustly, even in a severe wireless environment.
• PN Phase Recovery (PPR) methods and apparatus are described to acquire PN sequence phase synchronization in a system such as DMB-T, where the time offset of the positions of the basic PN sequence in successive signal frames is estimated robustly.
• An accurate decision of the signal frame index is made based on multiple time offsets measured in successive signal frames through a voting mechanism with little calculation complexity. In this manner, a DMB-T receiver can be made more robust and can be rapidly synchronized with the transmitter in PN sequence phase, even in an environment with very low signal to noise ratio or in the presence of large sample frequency errors.
• FIG. 1 is a diagram of a DMB-T receiver in which the present invention can be used.
• FIG. 2 is a diagram showing the multiplex frame structure for DMB-T.
• FIG. 3 is a diagram showing the structure of a signal frame.
• FIG. 4 is a diagram showing the PN phase offset for the signal frames in a super frame.
• FIG. 5 is a block diagram of a time offset estimator.
• FIG. 6 is a diagram showing a constellation for obtaining information about the signal frame index.
• FIG. 7 is a block diagram of a PN phase recovery module.
• FIG. 1 The general structure of a DTV receiver is shown in FIG. 1.
• a signal is received by an RF module (not shown) and sampled in an ADC 105.
• the sampled signal 107 is applied to a digital front end 109 that performs synchronization in response to information 110 from a synchronization block 111.
• the synchronization block performs calculations to enable, for example, PN phase recovery I l ia, carrier offset recovery 111b, symbol offset recovery 111c, and sampling frequency recovery 11 Id.
• An output signal 113 of the digital front end 109 is applied to the synchronization block 111 and to a channel estimation and equalization block 115, which receives information from and provides information to the synchronization block 111 via lines 117 and 119, respectively.
• An output signal 121 of the channel estimation and equalization block is applied to a decoder 123, which decodes and outputs received information 125.
• the present invention is concerned particularly with PN phase recovery (11 Ia).
• Table 1 lists the parameters of the three types.
• Table 1 The parameters of the three types of the signal frame header
• Type 2 the same segment of the PN sequence is used in every frame, which makes PN phase synchronization easier than in types 1 and 3, where different PN sequences are sent in different signal frames.
• the present invention concerns PN phase recovery for types 1 and 3.
• Types 1 and 3 are similar; type 1 (PN420) will be taken as an example for illustration.
• L PN 420.
• the PN sequence used is derived from an m-sequence and has a special initial phase in an LFSR PN generator for each signal frame of a super frame.
• the PN sequence in 0-th signal frame, PN(OJ) can be generated in an LFSR with the initial phase of "10110000" (binary number).
• the different PN sequences used in F signal frames have some internal relations.
• a basic PN sequence P 0 (i) may be defined which satisfies
• N the period of the m-sequence.
• N 255.
• All the PN sequences used in different signal frames can be regarded as a derivative of the basic PN sequence P 0 (i) . From analysis of the initial phases of the LFSR for the signal frames in a super frame as tabulated in the DMB-T specification, the PN sequence with length of 420 used in the/-th signal frame can be generated as
• O(f) is a variable PN phase offset for the/ ⁇ th signal frame.
• O(f) can be calculated using the following formula:
• O(f) is shown in FIG. 4.
• the PN sequences used in signal frames will change from frame to frame. Knowledge of PN sequences used in each frame is necessary for the synchronization of sampling frequency and sampling time phase, and for channel estimation. Also, it is necessary to maintain PN sequence phase synchronization throughout the reception procedure. As described herein, the object of PN phase recovery (PPR) is to recognize the index/of the/-th signal frame given the received data sampled from several signal frame headers.
• the received signal around the header of the/-th signal frame header is given as: [ ⁇ P 0 (i ⁇ (t -iT s )]® ⁇ (t-[82 + O(f)])®g(t) + n(t)
• g(t) is the impulse response of the combined equivalent channel, which includes the effect of the SRRC pulse shaping filters at the transmitter and receiver and the effect of the radio channel.
• the impulse response g(t) is given by:
• n(t) is AWGN noise.
• the receiver calculates the correlation between received data samples and the local basic PN sequence P 0 (i) .
• the correlation is
• R(t) R 0 (t) ® g(t) ®8 (t -[Z2 + O(f)]T s -fl' F O + M.t) ( 6 )
• R 0 (t) V R 0 (k) ⁇ (t - kT s ) .
• t l ⁇ %2 + O(J) + fl. F ) T s + ⁇ ( 7 ) where ⁇ o is a constant time reference point.
• t 2 (82 + O(f + 1) + (/ + ⁇ )L F ) T s + A 0 .
• the distance between these two peaks is a function of the index of the frame, enabling the index of the signal frame to be determined.
• the receiver may not necessarily always operate in an AWGN channel.
• the radio channel may experience severe fading as well as strong interference.
• the OFDM frame body part may unfortunately interfere with the frame header part.
• the peak of the correlation is not always easy to determine correctly due to possible strong noise and interference. Because of the PN phase change from frame to frame, traditional averaging or accumulation of the correlation result within multiple successive signal frames to suppress the noise and interference becomes impossible.
• sampling frequency error There may be large sampling frequency error if a low-cost crystal is adopted.
• the above correlation R(t) will be done using samples of the received signal.
• the accumulated sampling time error in one signal frame is Lp*df*T s .
• Large sampling frequency error will add more difficulty to PN phase recovery, because large sampling frequency error will cause uncertainty in the determination of the frame header from the analysis based on the sampled data.
• the PN phase needs to be captured as fast as possible.
• the receiver has little knowledge about the radio channel. It is therefore a challenge to capture the PN phase quickly and robustly, even in a severe wireless environment.
• PN phase recovery is achieved using two main modules.
• One module performs time offset estimation (TOE) of the basic PN sequence in two successive signal frames.
• the other module is a decision module that decides the signal frame index.
• Measurement of the time offset of the basic PN sequence is done by finding the correlation of the header part of two successive signal frames, represented as follows:
• RR 1 (t) R 0 (t) ® R 0 (t) ® g(t) ®g(t) ® ⁇ (t -[0(f + l) - 0(f)]T -L F T ) + u(t) ( 1 O )
• R 0 (t) ® R 0 (t) is almost a wave of impulse shape and u(t) is a noise term.
• This value t / indicates the distance between the same PN segment in the two successive signal frames, which may indicate the index of the signal frame.
• a block diagram of a time offset estimator (TOE) 500 for performing time offset estimation is shown in FIG. 5.
• Received signal samples 501 are applied to a first correlation block 503, a buffer 505, and a second correlation block 507.
• a generator 502 for generating a basic PN sequence Po is coupled to the correlation block 503 such that a correlation is performed between the received signal samples 501 and the basic PN sequence Po.
• the results of this correlation operation determine a peak correlaton time ti and determine which portion of the received signal samples stored in the buffer 505 will be used in a subsequent correlation operation.
• the correlation block 503 is coupled to the buffer 505 in order to perform this selection.
• the selected portion of the received signal samples is delayed by one signal frame by a delay element 509.
• the correlation block 507 then performs a further correlation, this time between received signal samples of the current frame and the selected portion of the received signal samples of the preceding frame.
• the results of this correlation operation determine a peak correlaton time t 2 .
• a time offset calculation block 511 then calculates the time offset t f as the difference t 2 - ti.
• the time offset estimator implements a sequence of steps that is repeated in every frame as summarized below.
• step (d) Sort the elements in set ⁇ , get Ji 1 , i 2 ,...., ⁇ L ⁇
• minimum distance detection is not necessarily the optimum detection method. Moreover, there is considerable calculation burden entailed by minimum distance detection due to too many hypotheses.
• T f+1 ,T f is illustrated as an asterisk, and the number beside it is the index of the signal frame.
• the pair is (-1 ,2). That is to say, the time offset between signal frame 0 and signal frame 1 is -1 and the time offset between signal frame 1 and signal frame 2 is 2.
• S (2) ⁇ /(2) - 1,/ 2 (2) - 1 ⁇ .
• FIG. 7 The block diagram of a PN phase recovery (PPR) module is shown in FIG. 7.
• Successive time offset estimates from the time offset estimator 500 are applied to successive delay elements 701, 703, 705, etc., to 7Ox.
• a corresponding number of slicers 711, 713, 715, etc., to 71x are provided.
• Each sheer receives a different pair of time offset estimates delayed by one signal frame and maps the pair to the constellation of FIG. 6.
• Outputs of the slicers are applied to a voting machine 721 to produce a final time offset estimate/.

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• 2007
  • 2007-11-30 CN CNA2007101962210A patent/CN101453554A/zh active Pending
• 2008
  • 2008-11-26 EP EP08853927A patent/EP2225864A2/en not_active Withdrawn
  • 2008-11-26 CN CN200880117823A patent/CN101874392A/zh active Pending
  • 2008-11-26 WO PCT/IB2008/054961 patent/WO2009069083A2/en active Application Filing

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