WO2006107838A1 - Systems, methods, and apparatus for highband time warping - Google Patents

Systems, methods, and apparatus for highband time warping Download PDF

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Publication number
WO2006107838A1
WO2006107838A1 PCT/US2006/012232 US2006012232W WO2006107838A1 WO 2006107838 A1 WO2006107838 A1 WO 2006107838A1 US 2006012232 W US2006012232 W US 2006012232W WO 2006107838 A1 WO2006107838 A1 WO 2006107838A1
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Prior art keywords
signal
highband
narrowband
excitation signal
speech
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PCT/US2006/012232
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French (fr)
Inventor
Koen Bernard Vos
Ananthapadmanabhan A. Kandhadai
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Qualcomm Incorporated
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Application filed by Qualcomm Incorporated filed Critical Qualcomm Incorporated
Priority to BRPI0607691A priority Critical patent/BRPI0607691B1/en
Priority to JP2008504479A priority patent/JP5203930B2/en
Priority to MX2007012187A priority patent/MX2007012187A/en
Priority to CA2603231A priority patent/CA2603231C/en
Priority to AU2006232362A priority patent/AU2006232362B2/en
Priority to EP06740356A priority patent/EP1864283B1/en
Priority to CN200680018212.6A priority patent/CN101185126B/en
Publication of WO2006107838A1 publication Critical patent/WO2006107838A1/en
Priority to IL186405A priority patent/IL186405A/en
Priority to NO20075512A priority patent/NO20075512L/en

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Classifications

    • G—PHYSICS
    • G10—MUSICAL INSTRUMENTS; ACOUSTICS
    • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
    • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
    • G10L19/02—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis using spectral analysis, e.g. transform vocoders or subband vocoders
    • G10L19/0204—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis using spectral analysis, e.g. transform vocoders or subband vocoders using subband decomposition
    • G10L19/0208—Subband vocoders
    • G—PHYSICS
    • G10—MUSICAL INSTRUMENTS; ACOUSTICS
    • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
    • G10L21/00—Speech or voice signal processing techniques to produce another audible or non-audible signal, e.g. visual or tactile, in order to modify its quality or its intelligibility
    • G10L21/02—Speech enhancement, e.g. noise reduction or echo cancellation
    • G10L21/0208—Noise filtering
    • G—PHYSICS
    • G10—MUSICAL INSTRUMENTS; ACOUSTICS
    • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
    • G10L21/00—Speech or voice signal processing techniques to produce another audible or non-audible signal, e.g. visual or tactile, in order to modify its quality or its intelligibility
    • G10L21/02—Speech enhancement, e.g. noise reduction or echo cancellation
    • G10L21/038—Speech enhancement, e.g. noise reduction or echo cancellation using band spreading techniques
    • G10L21/0388—Details of processing therefor
    • G—PHYSICS
    • G10—MUSICAL INSTRUMENTS; ACOUSTICS
    • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
    • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
    • G10L19/02—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis using spectral analysis, e.g. transform vocoders or subband vocoders
    • G10L19/032—Quantisation or dequantisation of spectral components
    • G10L19/038—Vector quantisation, e.g. TwinVQ audio
    • G—PHYSICS
    • G10—MUSICAL INSTRUMENTS; ACOUSTICS
    • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
    • G10L19/00—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis
    • G10L19/04—Speech or audio signals analysis-synthesis techniques for redundancy reduction, e.g. in vocoders; Coding or decoding of speech or audio signals, using source filter models or psychoacoustic analysis using predictive techniques
    • G10L19/16—Vocoder architecture
    • G10L19/18—Vocoders using multiple modes
    • G10L19/24—Variable rate codecs, e.g. for generating different qualities using a scalable representation such as hierarchical encoding or layered encoding
    • G—PHYSICS
    • G10—MUSICAL INSTRUMENTS; ACOUSTICS
    • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
    • G10L21/00—Speech or voice signal processing techniques to produce another audible or non-audible signal, e.g. visual or tactile, in order to modify its quality or its intelligibility
    • G10L21/02—Speech enhancement, e.g. noise reduction or echo cancellation
    • G10L21/0208—Noise filtering
    • G10L21/0216—Noise filtering characterised by the method used for estimating noise
    • G10L21/0232—Processing in the frequency domain
    • G—PHYSICS
    • G10—MUSICAL INSTRUMENTS; ACOUSTICS
    • G10L—SPEECH ANALYSIS TECHNIQUES OR SPEECH SYNTHESIS; SPEECH RECOGNITION; SPEECH OR VOICE PROCESSING TECHNIQUES; SPEECH OR AUDIO CODING OR DECODING
    • G10L21/00—Speech or voice signal processing techniques to produce another audible or non-audible signal, e.g. visual or tactile, in order to modify its quality or its intelligibility
    • G10L21/02—Speech enhancement, e.g. noise reduction or echo cancellation
    • G10L21/038—Speech enhancement, e.g. noise reduction or echo cancellation using band spreading techniques

Definitions

  • This invention relates to signal processing.
  • PSTN public switched telephone network
  • New networks for voice communications such as cellular telephony and voice over IP (Internet Protocol, VoIP) may not have the same bandwidth limits, and it may be desirable to transmit and receive voice communications that include a wideband frequency range over such networks. For example, it may be desirable to support an audio frequency range that extends down to 50 Hz and/or up to 7 or 8 kHz. It may also be desirable to support other applications, such as high-quality audio or audio/video conferencing, that may have audio speech content in ranges outside the traditional PSTN limits.
  • Extension of the range supported by a speech coder into higher frequencies may improve intelligibility. For example, the information that differentiates fricatives such as V and T is largely in the high frequencies. Highband extension may also improve other qualities of speech, such as presence. For example, even a voiced vowel may have spectral energy far above the PSTN limit.
  • One approach to wideband speech coding involves scaling a narrowband speech coding technique (e.g., one configured to encode the range of 0-4 kHz) to cover the wideband spectrum. For example, a speech signal may be sampled at a higher rate to include components at high frequencies, and a narrowband coding technique may be reconfigured to use more filter coefficients to represent this wideband signal.
  • a narrowband speech coding technique e.g., one configured to encode the range of 0-4 kHz
  • Narrowband coding techniques such as CELP (codebook excited linear prediction) are computationally intensive, however, and a wideband CELP coder may consume too many processing cycles to be practical for many mobile and other embedded applications. Encoding the entire spectrum of a wideband signal to a desired quality using such a technique may also lead to an unacceptably large increase in bandwidth. Moreover, transcoding of such an encoded signal would be required before even its narrowband portion could be transmitted into and/or decoded by a system that only supports narrowband coding.
  • CELP codebook excited linear prediction
  • Another approach to wideband speech coding involves extrapolating the highband spectral envelope from the encoded narrowband spectral envelope. While such an approach may be implemented without any increase in bandwidth and without a need for transcoding, the coarse spectral envelope or formant structure of the highband portion of a speech signal generally cannot be predicted accurately from the spectral envelope of the narrowband portion.
  • wideband speech coding such that at least the narrowband portion of the encoded signal may be sent through a narrowband channel (such as a PSTN channel) without transcoding or other significant modification.
  • Efficiency of the wideband coding extension may also be desirable, for example, to avoid a significant reduction in the number of users that may be serviced in applications such as wireless cellular telephony and broadcasting over wired and wireless channels.
  • a method of signal processing including encoding a low- frequency portion of a speech signal into at least an encoded lowband excitation signal and a plurality of lowband filter parameters; generating a highband excitation signal based on the encoded lowband excitation signal.
  • the method also includes encoding, according to at least the highband excitation signal, a high-frequency portion of the speech signal into at least a plurality of highband filter parameters.
  • the encoded lowband excitation signal describes a signal that is warped in time, with respect to the speech signal, according to a time-varying time warping.
  • the method includes applying, based on information relating to the time warping, a plurality of different time shifts to a corresponding plurality of successive portions in time of the high-frequency portion.
  • an apparatus in another embodiment, includes a lowband speech encoder configured to encode a low-frequency portion of a speech signal into at least an encoded lowband excitation signal and a plurality of lowband filter parameters; and a highband speech encoder configured to generate a highband excitation signal based on the encoded lowband excitation signal.
  • the highband encoder is configured to encode a high-frequency portion of the speech signal into at least a plurality of highband filter parameters according to at least the highband excitation signal.
  • the narrowband speech encoder is configured to output a regularization data signal describing a time-varying time warping, with respect to the speech signal, that is included in the encoded narrowband excitation signal.
  • the apparatus includes a delay line configured to apply a plurality of different time shifts to a corresponding plurality of successive portions in time of the high-frequency portion, wherein the plurality of different time shifts are based on the regularization data signal.
  • an apparatus in another embodiment, includes means for encoding a low- frequency portion of a speech signal into at least an encoded lowband excitation signal and a plurality of lowband filter parameters; means for generating a highband excitation signal based on the encoded lowband excitation signal; and means for encoding a high- frequency portion of the speech signal into at least a plurality of highband filter parameters according to at least the highband excitation signal.
  • the encoded narrowband excitation signal describes a signal that is warped in time, with respect to the speech signal, according to a time-varying time warping.
  • the apparatus includes means for applying, based on information relating to the time warping, a plurality of different time shifts to a corresponding plurality of successive portions in time of the high-frequency portion.
  • FIGURE Ia shows a block diagram of a wideband speech encoder AlOO according to an embodiment.
  • FIGURE Ib shows a block diagram of an implementation A102 of wideband speech encoder AlOO.
  • FIGURE 2a shows a block diagram of a wideband speech decoder BlOO according to an embodiment.
  • FIGURE 2b shows a block diagram of an implementation B 102 of wideband speech encoder BlOO.
  • FIGURE 3a shows a block diagram of an implementation Al 12 of filter bank AIlO.
  • FIGURE 3b shows a block diagram of an implementation B 122 of filter bank B120.
  • FIGURE 4a shows bandwidth coverage of the low and high bands for one example of filter bank Al 10.
  • FIGURE 4b shows bandwidth coverage of the low and high bands for another example of filter bank Al 10.
  • FIGURE 4c shows a block diagram of an implementation Al 14 of filter bank A112.
  • FIGURE 4d shows a block diagram of an implementation B 124 of filter bank B 122.
  • FIGURE 5a shows an example of a plot of frequency vs. log amplitude for a speech signal.
  • FIGURE 5b shows a block diagram of a basic linear prediction coding system.
  • FIGURE 6 shows a block diagram of an implementation A122 of narrowband encoder A 120.
  • FIGURE 7 shows a block diagram of an implementation Bl 12 of narrowband decoder BIlO.
  • FIGURE 8a shows an example of a plot of frequency vs. log amplitude for a residual signal for voiced speech.
  • FIGURE 8b shows an example of a plot of time vs. log amplitude for a residual signal for voiced speech.
  • FIGURE 9 shows a block diagram of a basic linear prediction coding system that also performs long-term prediction.
  • FIGURE 10 shows a block diagram of an implementation A202 of highband encoder A200.
  • FIGURE 11 shows a block diagram of an implementation A302 of highband excitation generator A300.
  • FIGURE 12 shows a block diagram of an implementation A402 of spectrum extender A400.
  • FIGURE 12a shows plots of signal spectra at various points in one example of a spectral extension operation.
  • FIGURE 12b shows plots of signal spectra at various points in another example of a spectral extension operation.
  • FIGURE 13 shows a block diagram of an implementation A304 of highband excitation generator A302.
  • FIGURE 14 shows a block diagram of an implementation A306 of highband excitation generator A302.
  • FIGURE 15 shows a flowchart for an envelope calculation task TlOO.
  • FIGURE 16 shows a block diagram of an implementation 492 of combiner 490.
  • FIGURE 17 illustrates an approach to calculating a measure of periodicity of highband signal S30.
  • FIGURE 18 shows a block diagram of an implementation A312 of highband excitation generator A302.
  • FIGURE 19 shows a block diagram of an implementation A314 of highband excitation generator A302.
  • FIGURE 20 shows a block diagram of an implementation A316 of highband excitation ⁇ ge v nerator A302.
  • FIGURE 21 shows a flowchart for a gain calculation task T200.
  • FIGURE 22 shows a flowchart for an implementation T210 of gain calculation task T200.
  • FIGURE 23a shows a diagram of a windowing function.
  • FIGURE 23b shows an application of a windowing function as shown in FIGURE 23a to subframes of a speech signal.
  • FIGURE 24 shows a block diagram for an implementation B202 of highband decoder B200.
  • FIGURE 25 shows a block diagram of an implementation ADlO of wideband speech encoder AlOO.
  • FIGURE 26a shows a schematic diagram of an implementation D122 of delay line D120.
  • FIGURE 26b shows a schematic diagram of an implementation D 124 of delay line D120.
  • FIGURE 27 shows a schematic diagram of an implementation D 130 of delay line D 120.
  • FIGURE 28 shows a block diagram of an implementation AD 12 of wideband speech encoder ADlO.
  • FIGURE 29 shows a flowchart of a method of signal processing MDlOO according to an embodiment.
  • FIGURE 30 shows a flowchart for a method MlOO according to an embodiment.
  • FIGURE 31a shows a flowchart for a method M200 according to an embodiment.
  • FIGURE 31b shows a flowchart for an implementation M210 of method M200.
  • FIGURE 32 shows a flowchart for a method M300 according to an embodiment.
  • Embodiments as described herein include systems, methods, and apparatus that may be configured to provide an extension to a narrowband speech coder to support transmission and/or storage of wideband speech signals at a bandwidth increase of only about 800 to 1000 bps (bits per second).
  • Potential advantages of such implementations include embedded coding to support compatibility with narrowband systems, relatively easy allocation and reallocation of bits between the narrowband and highband coding channels, avoiding a computationally intensive wideband synthesis operation, and maintaining a low sampling rate for signals to be processed by computationally intensive waveform coding routines.
  • the term “calculating” is used herein to indicate any of its ordinary meanings, such as computing, generating, and selecting from a list of values. Where the term “comprising” is used in the present description and claims, it does not exclude other elements or operations.
  • the term “A is based on B” is used to indicate any of its ordinary meanings, including the cases (i) "A is equal to B” and (ii) "A is based on at least B.”
  • Internet Protocol includes version 4, as described in IETF (Internet Engineering Task Force) RFC (Request for Comments) 791, and subsequent versions such as version 6.
  • FIGURE Ia shows a block diagram of a wideband speech encoder AlOO according to an embodiment.
  • Filter bank AIlO is configured to filter a wideband speech signal SlO to produce a narrowband signal S20 and a highband signal S30.
  • Narrowband encoder A 120 is configured to encode narrowband signal S20 to produce narrowband (NB) filter parameters S40 and a narrowband residual signal S50.
  • narrowband encoder A 120 is typically configured to produce narrowband filter parameters S40 and encoded narrowband excitation signal S50 as codebook indices or in another quantized form.
  • Highband encoder A200 is configured to encode highband signal S30 according to information in encoded narrowband excitation signal S50 to produce highband coding parameters S60.
  • highband encoder A200 is typically configured to produce highband coding parameters S60 as codebook indices or in another quantized form.
  • wideband speech encoder AlOO is configured to encode wideband speech signal SlO at a rate of about 8.55 kbps (kilobits per second), with about 7.55 kbps being used for narrowband filter parameters S40 and encoded narrowband excitation signal S50, and about 1 kbps being used for highband coding parameters S60.
  • FIGURE Ib shows a block diagram of an implementation A102 of wideband speech encoder AlOO that includes a multiplexer A130 configured to combine narrowband filter parameters S40, encoded narrowband excitation signal S50, and highband filter parameters S60 into a multiplexed signal S70.
  • An apparatus including encoder A102 may also include circuitry configured to transmit multiplexed signal S70 into a transmission channel such as a wired, optical, or wireless channel. Such an apparatus may also be configured to perform one or more channel encoding operations on the signal, such as error correction encoding (e.g., rate- compatible convolutional encoding) and/or error detection encoding (e.g., cyclic redundancy encoding), and/or one or more layers of network protocol encoding (e.g., Ethernet, TCP/IP, cdma2000).
  • error correction encoding e.g., rate- compatible convolutional encoding
  • error detection encoding e.g., cyclic redundancy encoding
  • layers of network protocol encoding e.g., Ethernet, TCP/IP, cdma2000.
  • multiplexer A130 may be configured to embed the encoded narrowband signal (including narrowband filter parameters S40 and encoded narrowband excitation signal S50) as a separable substream of multiplexed signal S70, such that the encoded narrowband signal may be recovered and decoded independently of another portion of multiplexed signal S70 such as a highband and/or lowband signal.
  • multiplexed signal S70 may be arranged such that the encoded narrowband signal may be recovered by stripping away the highband filter parameters S60.
  • One potential advantage of such a feature is to avoid the need for transcoding the encoded wideband signal before passing it to a system that supports decoding of the narrowband signal but does not support decoding of the highband portion.
  • FIGURE 2a is a block diagram of a wideband speech decoder BlOO according to an embodiment.
  • Narrowband decoder BIlO is configured to decode narrowband filter parameters S40 and encoded narrowband excitation signal S50 to produce a narrowband signal S90.
  • Highband decoder B200 is configured to decode highband coding parameters S60 according to a narrowband excitation signal S80, based on encoded narrowband excitation signal S50, to produce a highband signal SlOO.
  • narrowband decoder BIlO is configured to provide narrowband excitation signal S 80 to highband decoder B200.
  • Filter bank B 120 is configured to combine narrowband signal S90 and highband signal SlOO to produce a wideband speech signal SIlO.
  • FIGURE 2b is a block diagram of an implementation B 102 of wideband speech decoder BlOO that includes a demultiplexer B 130 configured to produce encoded signals S40, S50, and S60 from multiplexed signal S70.
  • An apparatus including decoder B 102 may include circuitry configured to receive multiplexed signal S70 from a transmission channel such as a wired, optical, or wireless channel.
  • Such an apparatus may also be configured to perform one or more channel decoding operations on the signal, such as error correction decoding (e.g., rate-compatible convolutional decoding) and/or error detection decoding (e.g., cyclic redundancy decoding), and/or one or more layers of network protocol decoding (e.g., Ethernet, TCP/IP, cdma2000).
  • error correction decoding e.g., rate-compatible convolutional decoding
  • error detection decoding e.g., cyclic redundancy decoding
  • layers of network protocol decoding e.g., Ethernet, TCP/IP, cdma2000.
  • Filter bank AIlO is configured to filter an input signal according to a split- band scheme to produce a low-frequency subband and a high-frequency subband. Depending on the design criteria for the particular application, the output subbands may have equal or unequal bandwidths and may be overlapping or nonoverlapping. A configuration of filter bank Al 10 that produces more than two subbands is also
  • such a filter bank may be configured to produce one or more lowband signals that include components in a frequency range below that of narrowband signal S20 (such as the range of 50-300 Hz). It is also possible for such a filter bank to be configured to produce one or more additional highband signals that include components in a frequency range above that of highband signal S30 (such as a range of 14-20, 16-20, or 16-32 kHz). In such case, wideband speech encoder AlOO may be implemented to encode this signal or signals separately, and multiplexer A130 may be configured to include the additional encoded signal or signals in multiplexed signal S70 (e.g., as a separable portion).
  • FIGURE 3a shows a block diagram of an implementation Al 12 of filter bank AIlO that is configured to produce two subband signals having reduced sampling rates.
  • Filter bank AIlO is arranged to receive a wideband speech signal SlO having a high- frequency (or highband) portion and a low-frequency (or lowband) portion.
  • Filter bank Al 12 includes a lowband processing path configured to receive wideband speech signal SlO and to produce narrowband speech signal S20, and a highband processing path configured to receive wideband speech signal SlO and to produce highband speech signal S30.
  • Lowpass filter 110 filters wideband speech signal SlO to pass a selected low-frequency subband
  • highpass filter 130 filters wideband speech signal SlO to pass a selected high-frequency subband.
  • Downsampler 120 reduces the sampling rate of the lowpass signal according to a desired decimation factor (e.g., by removing samples of the signal and/or replacing samples with average values), and downsampler 140 likewise reduces the sampling rate of the highpass signal according to another desired decimation factor.
  • a desired decimation factor e.g., by removing samples of the signal and/or replacing samples with average values
  • FIGURE 3b shows a block diagram of a corresponding implementation B 122 of filter bank B 120.
  • Upsampler 150 increases the sampling rate of narrowband signal S90 (e.g., by zero-stuffing and/or by duplicating samples), and lowpass filter 160 filters the upsampled signal to pass only a lowband portion (e.g., to prevent aliasing).
  • upsampler 170 increases the sampling rate of highband signal SlOO and highpass filter 180 filters the upsampled signal to pass only a highband portion. The two passband signals are then summed to form wideband speech signal SIlO.
  • filter bank B 120 is configured to produce a weighted sum of the two passband signals according to one or more weights received and/or calculated by highband decoder B200.
  • a configuration of filter bank B 120 that combines more than two passband signals is also contemplated.
  • Each of the filters 110, 130, 160, 180 may be implemented as a finite-impulse- response (FIR) filter or as an infinite-impulse-response (IIR) filter.
  • the frequency responses of encoder filters 110 and 130 may have symmetric or dissimilarly shaped transition regions between stopband and passband.
  • the frequency responses of decoder filters 160 and 180 may have symmetric or dissimilarly shaped transition regions between stopband and passband. It may be desirable but is not strictly necessary for lowpass filter 110 to have the same response as lowpass filter 160, and for highpass filter 130 to have the same response as highpass filter 180.
  • the two filter pairs 110, 130 and 160, 180 are quadrature mirror filter (QMF) banks, with filter pair 110, 130 having the same coefficients as filter pair 160, 180.
  • QMF quadrature mirror filter
  • lowpass filter 110 has a passband that includes the limited PSTN range of 300-3400 Hz (e.g., the band from 0 to 4 kHz).
  • FIGURES 4a and 4b show relative bandwidths of wideband speech signal SlO, narrowband signal S20, and highband signal S30 in two different implementational examples.
  • wideband speech signal SlO has a sampling rate of 16 kHz (representing frequency components within the range of 0 to 8 kHz)
  • narrowband signal S20 has a sampling rate of 8 kHz (representing frequency components within the range of 0 to 4 kHz).
  • a highband signal S30 as shown in this example may be obtained using a highpass filter 130 with a passband of 4-8 kHz. In such a case, it may be desirable to reduce the sampling rate to 8 kHz by downsampling the filtered signal by a factor of two. Such an operation, which may be expected to significantly reduce the computational complexity of further processing operations on the signal, will move the passband energy down to the range of 0 to 4 kHz without loss of information.
  • the upper and lower subbands have an appreciable overlap, such that the region of 3.5 to 4 kHz is described by both subband signals.
  • a highband signal S30 as in this example may be obtained using a highpass filter 130 with a passband of 3.5-7 kHz. In such a case, it may be desirable to reduce the sampling rate to 7 kHz by downsampling the filtered signal by a factor of 16/7. Such an operation, which may be expected to significantly reduce the computational complexity of further processing operations on the signal, will move the passband energy down to the range of 0 to 3.5 kHz without loss of information.
  • one or more of the transducers i.e., the microphone and the earpiece or loudspeaker
  • the portion of wideband speech signal SlO between 7 and 8 kHz is not included in the encoded signal.
  • Other particular examples of highpass filter 130 have passbands of 3.5- 7.5 kHz and 3.5-8 kHz.
  • providing an overlap between subbands as in the example of FIGURE 4b allows for the use of a lowpass and/or a highpass filter having a smooth rolloff over the overlapped region.
  • Such filters are typically easier to design, less computationally complex, and/or introduce less delay than filters with sharper or "brick-wall" responses.
  • Filters having sharp transition regions tend to have higher sidelobes (which may cause aliasing) than filters of similar order that have smooth rolloff s. Filters having sharp transition regions may also have long impulse responses which may cause ringing artifacts.
  • allowing for a smooth rolloff over the overlapped region may enable the use of a filter or filters whose poles are farther away from the unit circle, which may be important to ensure a stable fixed-point implementation.
  • Overlapping of subbands allows a smooth blending of lowband and highband that may lead to fewer audible artifacts, reduced aliasing, and/or a less noticeable transition from one band to the other.
  • the coding efficiency of narrowband encoder A120 may drop with increasing frequency.
  • coding quality of the narrowband coder may be reduced at low bit rates, especially in the presence of background noise.
  • providing an overlap of the subbands may increase the quality of reproduced frequency components in the overlapped region.
  • overlapping of subbands allows a smooth blending of lowband and highband that may lead to fewer audible artifacts, reduced aliasing, and/or a less noticeable transition from one band to the other.
  • Such a feature may be especially desirable for an implementation in which narrowband encoder A120 and highband encoder A200 operate according to different coding methodologies.
  • different coding techniques may produce signals that sound quite different.
  • a coder that encodes a spectral envelope in the form of codebook indices may produce a signal having a different sound than a coder that encodes the amplitude spectrum instead.
  • a time-domain coder (e.g., a pulse-code-modulation or PCM coder) may produce a signal having a different sound than a frequency-domain coder.
  • a coder that encodes a signal with a representation of the spectral envelope and the corresponding residual signal may produce a signal having a different sound than a coder that encodes a signal with only a representation of the spectral envelope.
  • a coder that encodes a signal as a representation of its waveform may produce an output having a different sound than that from a sinusoidal coder. In such cases, using filters having sharp transition regions to define nonoverlapping subbands may lead to an abrupt and perceptually noticeable transition between the subbands in the synthesized wideband signal.
  • QMF filter banks having complementary overlapping frequency responses are often used in subband techniques, such filters are unsuitable for at least some of the wideband coding implementations described herein.
  • a QMF filter bank at the encoder is configured to create a significant degree of aliasing that is canceled in the corresponding QMF filter bank at the decoder. Such an arrangement may not be appropriate for an application in which the signal incurs a significant amount of distortion between the filter banks, as the distortion may reduce the effectiveness of the alias cancellation property.
  • applications described herein include coding implementations configured to operate at very low bit rates.
  • the decoded signal is likely to appear significantly distorted as compared to the original signal, such that use of QMF filter banks may lead to uncanceled aliasing.
  • Applications that use QMF filter banks typically have higher bit rates (e.g., over 12 kbps for AMR, and 64 kbps for G.722).
  • a coder may be configured to produce a synthesized signal that is perceptually similar to the original signal but which actually differs significantly from the original signal.
  • a coder that derives the highband excitation from the narrowband residual as described herein may produce such a signal, as the actual highband residual may be completely absent from the decoded signal.
  • Use of QMF filter banks in such applications may lead to a significant degree of distortion caused by uncanceled aliasing.
  • the amount of distortion caused by QMF aliasing may be reduced if the affected subband is narrow, as the effect of the aliasing is limited to a bandwidth equal to the width of the subband.
  • each subband includes about half of the wideband bandwidth
  • distortion caused by uncanceled aliasing could affect a significant part of the signal.
  • the quality of the signal may also be affected by the location of the frequency band over which the uncanceled aliasing occurs. For example, distortion created near the center of a wideband speech signal (e.g., between 3 and 4 kHz) may be much more objectionable than distortion that occurs near an edge of the signal (e.g., above 6 IdEIz).
  • the lowband and highband paths of filter banks Al 10 and B 120 may be configured to have spectra that are completely unrelated apart from the overlapping of the two subbands.
  • the overlap of the two subbands as the distance from the point at which the frequency response of the highband filter drops to -20 dB up to the point at which the frequency response of the lowband filter drops to -20 dB.
  • this overlap ranges from around 200 Hz to around 1 kHz.
  • the range of about 400 to about 600 Hz may represent a desirable tradeoff between coding efficiency and perceptual smoothness.
  • the overlap is around 500 Hz.
  • FIGURE 4c shows a block diagram of an implementation Al 14 of filter bank Al 12 that performs a functional equivalent of highpass filtering and downsampling operations using a series of interpolation, resampling, decimation, and other operations.
  • Such an implementation may be easier to design and/or may allow reuse of functional blocks of logic and/or code.
  • the same functional block may be used to perform the operations of decimation to 14 kHz and decimation to 7 kHz as shown in FIGURE 4c.
  • the spectral reversal operation may be implemented by multiplying the signal with the function e Jn ⁇ or the sequence (—1)", whose values alternate between +1 and -1.
  • the spectral shaping operation may be implemented as a lowpass filter configured to shape the signal to obtain a desired overall filter response.
  • highband excitation generator A300 as described herein may be configured to produce a highband excitation signal S 120 that also has a spectrally reversed form.
  • FIGURE 4d shows a block diagram of an implementation B 124 of filter bank B 122 that performs a functional equivalent of upsampling and highpass filtering operations using a series of interpolation, resampling, and other operations.
  • Filter bank B 124 includes a spectral reversal operation in the highband that reverses a similar operation as performed, for example, in a filter bank of the encoder such as filter bank Al 14.
  • filter bank B124 also includes notch filters in the lowband and highband that attenuate a component of the signal at 7100 Hz, although such filters are optional and need not be included.
  • Narrowband encoder A120 is implemented according to a source-filter model that encodes the input speech signal as (A) a set of parameters that describe a filter and (B) an excitation signal that drives the described filter to produce a synthesized reproduction of the input speech signal.
  • FIGURE 5a shows an example of a spectral envelope of a speech signal. The peaks that characterize this spectral envelope represent resonances of the vocal tract and are called formants. Most speech coders encode at least this coarse spectral structure as a set of parameters such as filter coefficients.
  • FIGURE 5b shows an example of a basic source-filter arrangement as applied to coding of the spectral envelope of narrowband signal S20.
  • An analysis module calculates a set of parameters that characterize a filter corresponding to the speech sound over a period of time (typically 20 msec).
  • a whitening filter also called an analysis or prediction error filter
  • the resulting whitened signal (also called a residual) has less energy and thus less variance and is easier to encode than the original speech signal. Errors resulting from coding of the residual signal may also be spread more evenly over the spectrum.
  • the filter parameters and residual are typically quantized for efficient transmission over the channel.
  • a synthesis filter configured according to the filter parameters is excited by a signal based on the residual to produce a synthesized version of the original speech sound.
  • the synthesis filter is typically configured to have a transfer function that is the inverse of the transfer function of the whitening filter.
  • FIGURE 6 shows a block diagram of a basic implementation A122 of narrowband encoder A120.
  • a linear prediction coding (LPC) analysis module 210 encodes the spectral envelope of narrowband signal S20 as a set of linear prediction (LP) coefficients (e.g., coefficients of an all-pole filter 1/A(z)).
  • the analysis module typically processes the input signal as a series of nonoverlapping frames, with a new set of coefficients being calculated for each frame.
  • the frame period is generally a period over which the signal may be expected to be locally stationary; one common example is 20 milliseconds (equivalent to 160 samples at a sampling rate of 8 kHz).
  • LPC analysis module 210 is configured to calculate a set of ten LP filter coefficients to characterize the formant structure of each 20-millisecond frame. It is also possible to implement the analysis module to process the input signal as a series of overlapping frames.
  • the analysis module may be configured to analyze the samples of each frame directly, or the samples may be weighted first according to a windowing function (for example, a Hamming window). The analysis may also be performed over a window that is larger than the frame, such as a 30-msec window. This window may be symmetric (e.g. 5-20-5, such that it includes the 5 milliseconds immediately before and after the 20-millisecond frame) or asymmetric (e.g. 10-20, such that it includes the last 10 milliseconds of the preceding frame).
  • An LPC analysis module is typically configured to calculate the LP filter coefficients using a Levinson-Durbin recursion or the Leroux-Gueguen algorithm. In another implementation, the analysis module may be configured to calculate a set of cepstral coefficients for each frame instead of a set of LP filter coefficients.
  • the output rate of encoder A120 may be reduced significantly, with relatively little effect on reproduction quality, by quantizing the filter parameters.
  • Linear prediction filter coefficients are difficult to quantize efficiently and are usually mapped into another representation, such as line spectral pairs (LSPs) or line spectral frequencies (LSFs), for quantization and/or entropy encoding.
  • LSPs line spectral pairs
  • LSFs line spectral frequencies
  • LP filter coefficient-to-LSF transform 220 transforms the set of LP filter coefficients into a corresponding set of LSFs.
  • LP filter coefficients include parcor coefficients; log-area-ratio values; immittance spectral pairs (ISPs); and immittance spectral frequencies (ISFs), which are used in the GSM (Global System for Mobile Communications) AMR-WB (Adaptive Multirate- Wideband) codec.
  • ISPs immittance spectral pairs
  • ISFs immittance spectral frequencies
  • GSM Global System for Mobile Communications
  • AMR-WB Adaptive Multirate- Wideband
  • Quantizer 230 is configured to quantize the set of narrowband LSFs (or other coefficient representation), and narrowband encoder A122 is configured to output the result of this quantization as the narrowband filter parameters S40.
  • Such a quantizer typically includes a vector quantizer that encodes the input vector as an index to a corresponding vector entry in a table or codebook.
  • narrowband encoder A 122 also generates a residual signal by passing narrowband signal S20 through a whitening filter 260 (also called an analysis or prediction error filter) that is configured according to the set of filter coefficients.
  • whitening filter 260 is implemented as a FIR filter, although IIR implementations may also be used.
  • This residual signal will typically contain perceptually important information of the speech frame, such as long- term structure relating to pitch, that is not represented in narrowband filter parameters S40.
  • Quantizer 270 is configured to calculate a quantized representation of this residual signal for output as encoded narrowband excitation signal S50.
  • Such a quantizer typically includes a vector quantizer that encodes the input vector as an index to a corresponding vector entry in a table or codebook.
  • a quantizer may be configured to send one or more parameters from which the vector may be generated dynamically at the decoder, rather than retrieved from storage, as in a sparse codebook method.
  • Such a method is used in coding schemes such as algebraic CELP (codebook excitation linear prediction) and codecs such as 3GPP2 (Third Generation Partnership 2) EVRC (Enhanced Variable Rate Codec).
  • narrowband encoder A120 It is desirable for narrowband encoder A120 to generate the encoded narrowband excitation signal according to the same filter parameter values that will be available to the corresponding narrowband decoder. In this manner, the resulting encoded narrowband excitation signal may already account to some extent for nonidealities in those parameter values, such as quantization error. Accordingly, it is desirable to configure the whitening filter using the same coefficient values that will be available at the decoder.
  • inverse quantizer 240 dequantizes narrowband coding parameters S40
  • LSF-to-LP filter coefficient transform 250 maps the resulting values back to a corresponding set of LP filter coefficients, and this set of coefficients is used to configure whitening filter 260 to generate the residual signal that is quantized by quantizer 270.
  • narrowband encoder A120 Some implementations of narrowband encoder A120 are configured to calculate encoded narrowband excitation signal S50 by identifying one among a set of codebook vectors that best matches the residual signal. It is noted, however, that narrowband encoder A120 may also be implemented to calculate a quantized representation of the residual signal without actually generating the residual signal. For example, narrowband encoder A120 may be configured to use a number of codebook vectors to generate corresponding synthesized signals (e.g., according to a current set of filter parameters), and to select the codebook vector associated with the generated signal that best matches the original narrowband signal S20 in a perceptually weighted domain.
  • FIGURE 7 shows a block diagram of an implementation B 112 of narrowband decoder BIlO.
  • Inverse quantizer 310 dequantizes narrowband filter parameters S40 (in this case, to a set of LSFs), and LSF-to-LP filter coefficient transform 320 transforms the LSFs into a set of filter coefficients (for example, as described above with reference to inverse quantizer 240 and transform 250 of narrowband encoder A122).
  • Inverse quantizer 340 dequantizes narrowband residual signal S40 to produce a narrowband excitation signal S80.
  • narrowband synthesis filter 330 synthesizes narrowband signal S90.
  • narrowband synthesis filter 330 is configured to spectrally shape narrowband excitation signal S 80 according to the dequantized filter coefficients to produce narrowband signal S90.
  • Narrowband decoder Bl 12 also provides narrowband excitation signal S 80 to highband encoder A200, which uses it to derive the highband excitation signal S 120 as described herein.
  • narrowband decoder BIlO may be configured to provide additional information to highband decoder B200 that relates to the narrowband signal, such as spectral tilt, pitch gain and lag, and speech mode.
  • the system of narrowband encoder A122 and narrowband decoder Bl 12 is a basic example of an analysis-by-synthesis speech codec.
  • Codebook excitation linear prediction (CELP) coding is one popular family of analysis-by-synthesis coding, and implementations of such coders may perform waveform encoding of the residual, including such operations as selection of entries from fixed and adaptive codebooks, error minimization operations, and/or perceptual weighting operations.
  • Other implementations of analysis-by-synthesis coding include mixed excitation linear prediction (MELP), algebraic CELP (ACELP), relaxation CELP (RCELP), regular pulse excitation (RPE), multi-pulse CELP (MPE), and vector-sum excited linear prediction (VSELP) coding.
  • MELP mixed excitation linear prediction
  • ACELP algebraic CELP
  • RPE regular pulse excitation
  • MPE multi-pulse CELP
  • VSELP vector-sum excited linear prediction
  • MBE multi-band excitation
  • PWI prototype waveform interpolation
  • ETSI European Telecommunications Standards Institute
  • GSM 06.10 GSM full rate codec
  • RELP residual excited linear prediction
  • GSM enhanced full rate codec ETSI-GSM 06.60
  • ITU International Telecommunication Union
  • IS-641 IS-136
  • GSM-AMR GSM adaptive multirate
  • 4GVTM Full- Generation VocoderTM codec
  • Narrowband encoder A 120 and corresponding decoder Bl 10 may be implemented according to any of these technologies, or any other speech coding technology (whether known or to be developed) that represents a speech signal as (A) a set of parameters that describe a filter and (B) an excitation signal used to drive the described filter to reproduce the speech signal.
  • FIGURE 8a shows a spectral plot of one example of a residual signal, as may be produced by a whitening filter, for a voiced signal such as a vowel.
  • the periodic structure visible in this example is related to pitch, and different voiced sounds spoken by the same speaker may have different formant structures but similar pitch structures.
  • FIGURE 8b shows a time-domain plot of an example of such a residual signal that shows a sequence of pitch pulses in time.
  • Coding efficiency and/or speech quality may be increased by using one or more parameter values to encode characteristics of the pitch structure.
  • One important characteristic of the pitch structure is the frequency of the first harmonic (also called the fundamental frequency), which is typically in the range of 60 to 400 Hz. This characteristic is typically encoded as the inverse of the fundamental frequency, also called the pitch lag.
  • the pitch lag indicates the number of samples in one pitch period and may be encoded as one or more codebook indices. Speech signals from male speakers tend to have larger pitch lags than speech signals from female speakers.
  • Periodicity indicates the strength of the harmonic structure or, in other words, the degree to which the signal is harmonic or nonharmonic.
  • Two typical indicators of periodicity are zero crossings and normalized autocorrelation functions (NACFs).
  • Periodicity may also be indicated by the pitch gain, which is commonly encoded as a codebook gain (e.g., a quantized adaptive codebook gain).
  • Narrowband encoder Al 20 may include one or more modules configured to encode the long-term harmonic structure of narrowband signal S20.
  • one typical CELP paradigm that may be used includes an open-loop LPC analysis module, which encodes the short-term characteristics or coarse spectral envelope, followed by a closed-loop long-term prediction analysis stage, which encodes the fine pitch or harmonic structure.
  • the short-term characteristics are encoded as filter coefficients, and the long-term characteristics are encoded as values for parameters such as pitch lag and pitch gain.
  • narrowband encoder A120 may be configured to output encoded narrowband excitation signal S50 in a form that includes one or more codebook indices (e.g., a fixed codebook index and an adaptive codebook index) and corresponding gain values. Calculation of this quantized representation of the narrowband residual signal (e.g., by quantizer 270) may include selecting such indices and calculating such values. Encoding of the pitch structure may also include interpolation of a pitch prototype waveform, which operation may include calculating a difference between successive pitch pulses. Modeling of the long-term structure may be disabled for frames corresponding to unvoiced speech, which is typically noise-like and unstructured.
  • codebook indices e.g., a fixed codebook index and an adaptive codebook index
  • Calculation of this quantized representation of the narrowband residual signal may include selecting such indices and calculating such values.
  • Encoding of the pitch structure may also include interpolation of a pitch prototype waveform, which operation may include calculating a difference between successive pitch pulses.
  • An implementation of narrowband decoder BIlO may be configured to output narrowband excitation signal S 80 to highband decoder B200 after the long-term structure (pitch or harmonic structure) has been restored.
  • a decoder may be configured to output narrowband excitation signal S 80 as a dequantized version of encoded narrowband excitation signal S50.
  • narrowband decoder BIlO it is also possible to implement narrowband decoder BIlO such that highband decoder B200 performs dequantization of encoded narrowband excitation signal S50 to obtain narrowband excitation signal S80.
  • highband encoder A200 may be configured to receive the narrowband excitation signal as produced by the short-term analysis or whitening filter.
  • narrowband encoder A120 may be configured to output the narrowband excitation signal to highband encoder A200 before encoding the long-term structure. It is desirable, however, for highband encoder A200 to receive from the narrowband channel the same coding information that will be received by highband decoder B200, such that the coding parameters produced by highband encoder A200 may already account to some extent for nonidealities in that information.
  • highband encoder A200 may reconstruct narrowband excitation signal S 80 from the same parametrized and/or quantized encoded narrowband excitation signal S50 to be output by wideband speech encoder AlOO.
  • One potential advantage of this approach is more accurate calculation of the highband gain factors S60b described below.
  • narrowband encoder A120 may produce parameter values that relate to other characteristics of narrowband signal S20. These values, which may be suitably quantized for output by wideband speech encoder AlOO, may be included among the narrowband filter parameters S40 or outputted separately. Highband encoder A200 may also be configured to calculate highband coding parameters S60 according to one or more of these additional parameters (e.g., after dequantization). At wideband speech decoder BlOO, highband decoder B200 may be configured to receive the parameter values via narrowband decoder BIlO (e.g., after dequantization). Alternatively, highband decoder B200 may be configured to receive (and possibly to dequantize) the parameter values directly.
  • narrowband encoder A 120 produces values for spectral tilt and speech mode parameters for each frame.
  • Spectral tilt relates to the shape of the spectral envelope over the passband and is typically represented by the quantized first reflection coefficient.
  • the spectral energy decreases with increasing frequency, such that the first reflection coefficient is negative and may approach — 1.
  • Most unvoiced sounds have a spectrum that is either flat, such that the first reflection coefficient is close to zero, or has more energy at high frequencies, such that the first reflection coefficient is positive and may approach +1.
  • Speech mode indicates whether the current frame represents voiced or unvoiced speech.
  • This parameter may have a binary value based on one or more measures of periodicity (e.g., zero crossings, NACFs, pitch gain) and/or voice activity for the frame, such as a relation between such a measure and a threshold value.
  • the speech mode parameter has one or more other states to indicate modes such as silence or background noise, or a transition between silence and voiced speech.
  • Highband encoder A200 is configured to encode highband signal S30 according to a source-filter model, with the excitation for this filter being based on the encoded narrowband excitation signal.
  • FIGURE 10 shows a block diagram of an implementation A202 of highband encoder A200 that is configured to produce a stream of highband coding parameters S60 including highband filter parameters S60a and highband gain factors S60b.
  • Highband excitation generator A300 derives a highband excitation signal S 120 from encoded narrowband excitation signal S50.
  • Analysis module A210 produces a set of parameter values that characterize the spectral envelope of highband signal S30.
  • analysis module A210 is configured to perform LPC analysis to produce a set of LP filter coefficients for each frame of highband signal S30.
  • Linear prediction filter coefficient-to-LSF transform 410 transforms the set of LP filter coefficients into a corresponding set of LSFs.
  • analysis module A210 and/or transform 410 may be configured to use other coefficient sets (e.g., cepstral coefficients) and/or coefficient representations (e.g., ISPs).
  • Quantizer 420 is configured to quantize the set of highband LSFs (or other coefficient representation, such as ISPs), and highband encoder A202 is configured to output the result of this quantization as the highband filter parameters S60a.
  • a quantizer typically includes a vector quantizer that encodes the input vector as an index to a corresponding vector entry in a table or codebook.
  • Highband encoder A202 also includes a synthesis filter A220 configured to produce a synthesized highband signal S 130 according to highband excitation signal S 120 and the encoded spectral envelope (e.g., the set of LP filter coefficients) produced by analysis module A210.
  • Synthesis filter A220 is typically implemented as an BDR. filter, although FIR implementations may also be used.
  • synthesis filter A220 is implemented as a sixth-order linear autoregressive filter.
  • Highband gain factor calculator A230 calculates one or more differences between the levels of the original highband signal S30 and synthesized highband signal S 130 to specify a gain envelope for the frame.
  • Quantizer 430 which may be implemented as a vector quantizer that encodes the input vector as an index to a corresponding vector entry in a table or codebook, quantizes the value or values specifying the gain envelope, and highband encoder A202 is configured to output the result of this quantization as highband gain factors S60b.
  • synthesis filter A220 is arranged to receive the filter coefficients from analysis module A210.
  • highband encoder A202 includes an inverse quantizer and inverse transform configured to decode the filter coefficients from highband filter parameters S60a, and in this case synthesis filter A220 is arranged to receive the decoded filter coefficients instead. Such an alternative arrangement may support more accurate calculation of the gain envelope by highband gain calculator A230.
  • analysis module A210 and highband gain calculator A230 output a set of six LSFs and a set of five gain values per frame, respectively, such that a wideband extension of the narrowband signal S20 may be achieved with only eleven additional values per frame.
  • the ear tends to be less sensitive to frequency errors at high frequencies, such that highband coding at a low LPC order may produce a signal having a comparable perceptual quality to narrowband coding at a higher LPC order.
  • a typical implementation of highband encoder A200 may be configured to output 8 to 12 bits per frame for high-quality reconstruction of the spectral envelope and another 8 to 12 bits per frame for high-quality reconstruction of the temporal envelope.
  • analysis module A210 outputs a set of eight LSFs per frame.
  • highband encoder A200 are configured to produce highband excitation signal S 120 by generating a random noise signal having highband frequency components and amplitude-modulating the noise signal according to the time- domain envelope of narrowband signal S20, narrowband excitation signal S80, or highband signal S30. While such a noise-based method may produce adequate results for unvoiced sounds, however, it may not be desirable for voiced sounds, whose residuals are usually harmonic and consequently have some periodic structure.
  • Highband excitation generator A300 is configured to generate highband excitation signal S 120 by extending the spectrum of narrowband excitation signal S 80 into the highband frequency range.
  • FIGURE 11 shows a block diagram of an implementation A302 of highband excitation generator A300.
  • Inverse quantizer 450 is configured to dequantize encoded narrowband excitation signal S50 to produce narrowband excitation signal S80.
  • Spectrum extender A400 is configured to produce a harmonically extended signal S 160 based on narrowband excitation signal S80.
  • Combiner 470 is configured to combine a random noise signal generated by noise generator 480 and a time-domain envelope calculated by envelope calculator 460 to produce a modulated noise signal S 170.
  • Combiner 490 is configured to mix harmonically extended signal S60 and modulated noise signal S 170 to produce highband excitation signal S 120.
  • spectrum extender A400 is configured to perform a spectral folding operation (also called mirroring) on narrowband excitation signal S 80 to produce harmonically extended signal S 160. Spectral folding may be performed by zero-stuffing excitation signal S80 and then applying a highpass filter to retain the alias.
  • spectrum extender A400 is configured to produce harmonically extended signal S 160 by spectrally translating narrowband excitation signal S 80 into the highband (e.g., via upsampling followed by multiplication with a constant-frequency cosine signal).
  • Spectral folding and translation methods may produce spectrally extended signals whose harmonic structure is discontinuous with the original harmonic structure of narrowband excitation signal S 80 in phase and/or frequency. For example, such methods may produce signals having peaks that are not generally located at multiples of the fundamental frequency, which may cause tinny-sounding artifacts in the reconstructed speech signal. These methods also tend to produce high-frequency harmonics that have unnaturally strong tonal characteristics.
  • a PSTN signal may be sampled at 8 kHz but bandlimited to no more than 3400 Hz, the upper spectrum of narrowband excitation signal S 80 may contain little or no energy, such that an extended signal generated according to a spectral folding or spectral translation operation may have a spectral hole above 3400 Hz.
  • harmonically extended signal S 160 include identifying one or more fundamental frequencies of narrowband excitation signal S 80 and generating harmonic tones according to that information.
  • the harmonic structure of an excitation signal may be characterized by the fundamental frequency together with amplitude and phase information.
  • Another implementation of highband excitation generator A300 generates a harmonically extended signal S 160 based on the fundamental frequency and amplitude (as indicated, for example, by the pitch lag and pitch gain). Unless the harmonically extended signal is phase-coherent with narrowband excitation signal S 80, however, the quality of the resulting decoded speech may not be acceptable.
  • a nonlinear function may be used to create a highband excitation signal that is phase-coherent with the narrowband excitation and preserves the harmonic structure without phase discontinuity.
  • a nonlinear function may also provide an increased noise level between high-frequency harmonics, which tends to sound more natural than the tonal high-frequency harmonics produced by methods such as spectral folding and spectral translation.
  • Typical memoryless nonlinear functions that may be applied by various implementations of spectrum extender A400 include the absolute value function (also called fullwave rectification), halfwave rectification, squaring, cubing, and clipping. Other implementations of spectrum extender A400 may be configured to apply a nonlinear function having memory.
  • FIG. 12 is a block diagram of an implementation A402 of spectrum extender A400 that is configured to apply a nonlinear function to extend the spectrum of narrowband excitation signal S 80.
  • Upsampler 510 is configured to upsample narrowband excitation signal S80. It may be desirable to upsample the signal sufficiently to minimize aliasing upon application of the nonlinear function. In one particular example, upsampler 510 upsamples the signal by a factor of eight. Upsampler 510 may be configured to perform the upsampling operation by zero-stuffing the input signal and lowpass filtering the result.
  • Nonlinear function calculator 520 is configured to apply a nonlinear function to the upsampled signal.
  • Nonlinear function calculator 520 may also be configured to perform an amplitude warping of the upsampled or spectrally extended signal.
  • Downsampler 530 is configured to downsample the spectrally extended result of applying the nonlinear function. It may be desirable for downsampler 530 to perform a bandpass filtering operation to select a desired frequency band of the spectrally extended signal before reducing the sampling rate (for example, to reduce or avoid aliasing or corruption by an unwanted image). It may also be desirable for downsampler 530 to reduce the sampling rate in more than one stage.
  • FIGURE 12a is a diagram that shows the signal spectra at various points in one example of a spectral extension operation, where the frequency scale is the same across the various plots. Plot (a) shows the spectrum of one example of narrowband excitation signal S80.
  • Plot (b) shows the spectrum after signal S80 has been upsampled by a factor of eight.
  • Plot (c) shows an example of the extended spectrum after application of a nonlinear function.
  • Plot (d) shows the spectrum after lowpass filtering. In this example, the passband extends to the upper frequency limit of highband signal S30 (e.g., 7 kHz or 8 kHz).
  • Plot (e) shows the spectrum after a first stage of downsampling, in which the sampling rate is reduced by a factor of four to obtain a wideband signal.
  • Plot (f) shows the spectrum after a highpass filtering operation to select the highband portion of the extended signal
  • plot (g) shows the spectrum after a second stage of downsampling, in which the sampling rate is reduced by a factor of two.
  • downsampler 530 performs the highpass filtering and second stage of downsampling by passing the wideband signal through highpass filter 130 and downsampler 140 of filter bank Al 12 (or other structures or routines having the same response) to produce a spectrally extended signal having the frequency range and sampling rate of highband signal S30.
  • downsampling of the highpass signal shown in plot (f) causes a reversal of its spectrum.
  • downsampler 530 is also configured to perform a spectral flipping operation on the signal.
  • Plot (h) shows a result of applying the spectral flipping operation, which may be performed by multiplying the signal with the function e Jn ⁇ or the sequence (-l) n , whose values alternate between +1 and -1. Such an operation is equivalent to shifting the digital spectrum of the signal in the frequency domain by a distance of ⁇ , It is noted that the same result may also be obtained by applying the downsampling and spectral flipping operations in a different order.
  • the operations of upsampling and/or downsampling may also be configured to include resampling to obtain a spectrally extended signal having the sampling rate of highband signal S30 (e.g., 7 kHz).
  • filter banks Al 10 and B 120 may be implemented such that one or both of the narrowband and highband signals S20, S30 has a spectrally reversed form at the output of filter bank Al 10, is encoded and decoded in the spectrally reversed form, and is spectrally reversed again at filter bank B 120 before being output in wideband speech signal SIlO.
  • a spectral flipping operation as shown in FIGURE 12a would not be necessary, as it would be desirable for highband excitation signal S 120 to have a spectrally reversed form as well.
  • FIGURE 12b is a diagram that shows the signal spectra at various points in another example of a spectral extension operation, where the frequency scale is the same across the various plots.
  • Plot (a) shows the spectrum of one example of narrowband excitation signal S80.
  • Plot (b) shows the spectrum after signal S 80 has been upsampled by a factor of two.
  • Plot (c) shows an example of the extended spectrum after application of a nonlinear function. In this case, aliasing that may occur in the higher frequencies is accepted.
  • Plot (d) shows the spectrum after a spectral reversal operation.
  • Plot (e) shows the spectrum after a single stage of downsampling, in which the sampling rate is reduced by a factor of two to obtain the desired spectrally extended signal.
  • the signal is in spectrally reversed form and may be used in an implementation of highband encoder A200 which processed highband signal S30 in such a form.
  • Spectral extender A402 includes a spectral flattener 540 configured to perform a whitening operation on the downsampled signal.
  • Spectral flattener 540 may be configured to perform a fixed whitening operation or to perform an adaptive whitening operation.
  • spectral flattener 540 includes an LPC analysis module configured to calculate a set of four filter coefficients from the downsampled signal and a fourth-order analysis filter configured to whiten the signal according to those coefficients.
  • Other implementations of spectrum extender A400 include configurations in which spectral flattener 540 operates on the spectrally extended signal before downsampler 530.
  • Highband excitation generator A300 may be implemented to output harmonically extended signal S 160 as highband excitation signal S 120. In some cases, however, using only a harmonically extended signal as the highband excitation may result in audible artifacts.
  • the harmonic structure of speech is generally less pronounced in the highband than in the low band, and using too much harmonic structure in the highband excitation signal can result in a buzzy sound. This artifact may be especially noticeable in speech signals from female speakers.
  • Embodiments include implementations of highband excitation generator A300 that are configured to mix harmonically extended signal S 160 with a noise signal.
  • highband excitation generator A302 includes a noise generator 480 that is configured to produce a random noise signal.
  • noise generator 480 is configured to produce a unit-variance white pseudorandom noise signal, although in other implementations the noise signal need not be white and may have a power density that varies with frequency. It may be desirable for noise generator 480 to be configured to output the noise signal as a deterministic function such that its state may be duplicated at the decoder.
  • noise generator 480 may be configured to output the noise signal as a deterministic function of information coded earlier within the same frame, such as the narrowband filter parameters S40 and/or encoded narrowband excitation signal S50.
  • the random noise signal produced by noise generator 480 may be amplitude-modulated to have a time- domain envelope that approximates the energy distribution over time of narrowband signal S20, highband signal S30, narrowband excitation signal S 80, or harmonically extended signal S160.
  • highband excitation generator A302 includes a combiner 470 configured to amplitude-modulate the noise signal produced by noise generator 480 according to a time-domain envelope calculated by envelope calculator 460.
  • combiner 470 may be implemented as a multiplier arranged to scale the output of noise generator 480 according to the time-domain envelope calculated by envelope calculator 460 to produce modulated noise signal S 170.
  • envelope calculator 460 is arranged to calculate the envelope of harmonically extended signal S 160.
  • envelope calculator 460 is arranged to calculate the envelope of narrowband excitation signal S80. Further implementations of highband excitation generator A302 may be otherwise configured to add noise to harmonically extended signal S 160 according to locations of the narrowband pitch pulses in time.
  • Envelope calculator 460 may be configured to perform an envelope calculation as a task that includes a series of subtasks.
  • FIGURE 15 shows a flowchart of an example TlOO of such a task.
  • Subtask Tl 10 calculates the square of each sample of the frame of the signal whose envelope is to be modeled (for example, narrowband excitation signal S 80 or harmonically extended signal S 160) to produce a sequence of squared values.
  • Subtask T120 performs a smoothing operation on the sequence of squared values.
  • subtask T 120 applies a first-order IIR lowpass filter to the sequence according to the expression
  • x is the filter input
  • y is the filter output
  • n is a time-domain index
  • a is a smoothing coefficient having a value between 0.5 and 1.
  • the value of the smoothing coefficient a may be fixed or, in an alternative implementation, may be adaptive according to an indication of noise in the input signal, such that a is closer to 1 in the absence of noise and closer to 0.5 in the presence of noise.
  • Subtask T130 applies a square root function to each sample of the smoothed sequence to produce the time- domain envelope.
  • envelope calculator 460 may be configured to perform the various subtasks of task TlOO in serial and/or parallel fashion.
  • subtask TIlO may be preceded by a bandpass operation configured to select a desired frequency portion of the signal whose envelope is to be modeled, such as the range of 3-4 kHz.
  • Combiner 490 is configured to mix harmonically extended signal S 160 and modulated noise signal S 170 to produce highband excitation signal S 120.
  • Implementations of combiner 490 may be configured, for example, to calculate highband excitation signal S 120 as a sum of harmonically extended signal S 160 and modulated noise signal S 170.
  • Such an implementation of combiner 490 may be configured to calculate highband excitation signal S 120 as a weighted sum by applying a weighting factor to harmonically extended signal S 160 and/or to modulated noise signal S 170 before the summation.
  • Each such weighting factor may be calculated according to one or more criteria and may be a fixed value or, alternatively, an adaptive value that is calculated on a frame-by-frame or subframe-by-subframe basis.
  • FIGURE 16 shows a block diagram of an implementation 492 of combiner 490 that is configured to calculate highband excitation signal S 120 as a weighted sum of harmonically extended signal S 160 and modulated noise signal S 170.
  • Combiner 492 is configured to weight harmonically extended signal S 160 according to harmonic weighting factor S 180, to weight modulated noise signal S 170 according to noise weighting factor S 190, and to output highband excitation signal S 120 as a sum of the weighted signals.
  • combiner 492 includes a weighting factor calculator 550 that is configured to calculate harmonic weighting factor S180 and noise weighting factor S 190.
  • Weighting factor calculator 550 may be configured to calculate weighting factors S 180 and S 190 according to a desired ratio of harmonic content to noise content in highband excitation signal S 120. For example, it may be desirable for combiner 492 to produce highband excitation signal S 120 to have a ratio of harmonic energy to noise energy similar to that of highband signal S30. In some implementations of weighting factor calculator 550, weighting factors S 180, S 190 are calculated according to one or more parameters relating to a periodicity of narrowband signal S20 or of the narrowband residual signal, such as pitch gain and/or speech mode.
  • weighting factor calculator 550 may be configured to assign a value to harmonic weighting factor S 180 that is proportional to the pitch gain, for example, and/or to assign a higher value to noise weighting factor S 190 for unvoiced speech signals than for voiced speech signals.
  • weighting factor calculator 550 is configured to calculate values for harmonic weighting factor S 180 and/or noise weighting factor S 190 according to a measure of periodicity of highband signal S30.
  • weighting factor calculator 550 calculates harmonic weighting factor S 180 as the maximum value of the autocorrelation coefficient of highband signal S30 for the current frame or subframe, where the autocorrelation is performed over a search range that includes a delay of one pitch lag and does not include a delay of zero samples.
  • FIGURE 17 shows an example of such a search range of length n samples that is centered about a delay of one pitch lag and has a width not greater than one pitch lag.
  • FIGURE 17 also shows an example of another approach in which weighting factor calculator 550 calculates a measure of periodicity of highband signal S30 in several stages.
  • the current frame is divided into a number of subframes, and the delay for which the autocorrelation coefficient is maximum is identified separately for each subframe.
  • the autocorrelation is performed over a search range that includes a delay of one pitch lag and does not include a delay of zero samples.
  • a delayed frame is constructed by applying the corresponding identified delay to each subframe, concatenating the resulting subframes to construct an optimally delayed frame, and calculating harmonic weighting factor S 180 as the correlation coefficient between the original frame and the optimally delayed frame.
  • weighting factor calculator 550 calculates harmonic weighting factor S 180 as an average of the maximum autocorrelation coefficients obtained in the first stage for each subframe. Implementations of weighting factor calculator 550 may also be configured to scale the correlation coefficient, and/or to combine it with another value, to calculate the value for harmonic weighting factor S 180.
  • weighting factor calculator 550 may be configured to calculate a measure of periodicity of highband signal S30 only in cases where a presence of periodicity in the frame is otherwise indicated.
  • weighting factor calculator 550 may be configured to calculate a measure of periodicity of highband signal S30 according to a relation between another indicator of periodicity of the current frame, such as pitch gain, and a threshold value.
  • weighting factor calculator 550 is configured to perform an autocorrelation operation on highband signal S30 only if the frame's pitch gain (e.g., the adaptive codebook gain of the narrowband residual) has a value of more than 0.5 (alternatively, at least 0.5).
  • weighting factor calculator 550 is configured to perform an autocorrelation operation on highband signal S30 only for frames having particular states of speech mode (e.g., only for voiced signals). In such cases, weighting factor calculator 550 may be configured to assign a default weighting factor for frames having other states of speech mode and/or lesser values of pitch gain.
  • Embodiments include further implementations of weighting factor calculator 550 that are configured to calculate weighting factors according to characteristics other than or in addition to periodicity. For example, such an implementation may be configured to assign a higher value to noise gain factor S 190 for speech signals having a large pitch lag than for speech signals having a small pitch lag.
  • weighting factor calculator 550 is configured to determine a measure of harmonicity of wideband speech signal SlO, or of highband signal S30, according to a measure of the energy of the signal at multiples of the fundamental frequency relative to the energy of the signal at other frequency components.
  • Some implementations of wideband speech encoder AlOO are configured to output an indication of periodicity or harmonicity (e.g. a one-bit flag indicating whether the frame is harmonic or nonharmonic) based on the pitch gain and/or another measure of periodicity or harmonicity as described herein.
  • an indication of periodicity or harmonicity e.g. a one-bit flag indicating whether the frame is harmonic or nonharmonic
  • a corresponding wideband speech decoder BlOO uses this indication to configure an operation such as weighting factor calculation.
  • such an indication is used at the encoder and/or decoder in calculating a value for a speech mode parameter.
  • weighting factor calculator 550 may be configured to calculate a value for harmonic weighting factor S 180 or for noise weighting factor S 190 (or to receive such a value from storage or another element of highband encoder A200) and to derive a value for the other weighting factor according to an expression such as
  • weighting factor calculator 550 may be configured to select, according to a value of a periodicity measure for the current frame or subframe, a corresponding one among a plurality of pairs of weighting factors S 180, S 190, where the pairs are precalculated to satisfy a constant-energy ratio such as expression (2).
  • expression (2) For an implementation of weighting factor calculator 550 in which expression (2) is observed, typical values for harmonic weighting factor S 180 range from about 0.7 to about 1.0, and typical values for noise weighting factor S 190 range from about 0.1 to about 0.7.
  • Other implementations of weighting factor calculator 550 may be configured to operate according to a version of expression (2) that is modified according to a desired baseline weighting between harmonically extended signal S 160 and modulated noise signal S 170.
  • Artifacts may occur in a synthesized speech signal when a sparse codebook (one whose entries are mostly zero values) has been used to calculate the quantized representation of the residual.
  • Codebook sparseness occurs especially when the narrowband signal is encoded at a low bit rate. Artifacts caused by codebook sparseness are typically quasi-periodic in time and occur mostly above 3 kHz. Because the human ear has better time resolution at higher frequencies, these artifacts may be more noticeable in the highband.
  • Embodiments include implementations of highband excitation generator A300 that are configured to perform anti-sparseness filtering.
  • FIGURE 18 shows a block diagram of an implementation A312 of highband excitation generator A302 that includes an anti-sparseness filter 600 arranged to filter the dequantized narrowband excitation signal produced by inverse quantizer 450.
  • FIGURE 19 shows a block diagram of an implementation A314 of highband excitation generator A302 that includes an anti-sparseness filter 600 arranged to filter the spectrally extended signal produced by spectrum extender A400.
  • FIGURE 20 shows a block diagram of an implementation A316 of highband excitation generator A302 that includes an anti- sparseness filter 600 arranged to filter the output of combiner 490 to produce highband excitation signal S 120.
  • an anti- sparseness filter 600 arranged to filter the output of combiner 490 to produce highband excitation signal S 120.
  • implementations of highband excitation generator A300 that combine the features of any of implementations A304 and A306 with the features of any of implementations A312, A314, and A316 are contemplated and hereby expressly disclosed.
  • Anti-sparseness filter 600 may also be arranged within spectrum extender A400: for example, after any of the elements 510, 520, 530, and 540 in spectrum extender A402. It is expressly noted that anti-sparseness filter 600 may also be used with implementations of spectrum extender A400 that perform spectral folding, spectral translation, or harmonic extension.
  • Anti-sparseness filter 600 may be configured to alter the phase of its input signal. For example, it may be desirable for anti-sparseness filter 600 to be configured and arranged such that the phase of highband excitation signal S 120 is randomized, or otherwise more evenly distributed, over time. It may also be desirable for the response of anti-sparseness filter 600 to be spectrally flat, such that the magnitude spectrum of the filtered signal is not appreciably changed. In one example, anti-sparseness filter 600 is implemented as an all-pass filter having a transfer function according to the following expression:
  • H ⁇ z -°- 1 + Z ⁇ 0 - 6 + Z ⁇ 6 6 . (3). 1 -0.7Z "4 l + 0.6z- 6
  • One effect of such a filter may be to spread out the energy of the input signal so that it is no longer concentrated in only a few samples.
  • Unvoiced signals are characterized by a low pitch gain (e.g. quantized narrowband adaptive codebook gain) and a spectral tilt (e.g. quantized first reflection coefficient) that is close to zero or positive, indicating a spectral envelope that is flat or tilted upward with increasing frequency.
  • a low pitch gain e.g. quantized narrowband adaptive codebook gain
  • a spectral tilt e.g. quantized first reflection coefficient
  • Typical implementations of anti-sparseness filter 600 are configured to filter unvoiced sounds (e.g., as indicated by the value of the spectral tilt), to filter voiced sounds when the pitch gain is below a threshold value (alternatively, not greater than the threshold value), and otherwise to pass the signal without alteration.
  • anti-sparseness filter 600 include two or more filters that are configured to have different maximum phase modification angles (e.g., up to 180 degrees).
  • anti-sparseness filter 600 may be configured to select among these component filters according to a value of the pitch gain (e.g., the quantized adaptive codebook or LTP gain), such that a greater maximum phase modification angle is used for frames having lower pitch gain values.
  • An implementation of anti- sparseness filter 600 may also include different component filters that are configured to modify the phase over more or less of the frequency spectrum, such that a filter configured to modify the phase over a wider frequency range of the input signal is used for frames having lower pitch gain values.
  • highband encoder A200 may be configured to characterize highband signal S30 by specifying a temporal or gain envelope.
  • highband encoder A202 includes a highband gain factor calculator A230 that is configured and arranged to calculate one or more gain factors according to a relation between highband signal S30 and synthesized highband signal S 130, such as a difference or ratio between the energies of the two signals over a frame or some portion thereof.
  • highband gain calculator A230 may be likewise configured but arranged instead to calculate the gain envelope according to such a time-varying relation between highband signal S30 and narrowband excitation signal S80 or highband excitation signal S120.
  • highband encoder A202 is configured to output a quantized index of eight to twelve bits that specifies five gain factors for each frame.
  • Highband gain factor calculator A230 may be configured to perform gain factor calculation as a task that includes one or more series of subtasks.
  • FIGURE 21 shows a flowchart of an example T200 of such a task that calculates a gain value for a corresponding subframe according to the relative energies of highband signal S30 and synthesized highband signal S 130.
  • Tasks 220a and 220b calculate the energies of the corresponding subframes of the respective signals.
  • tasks 220a and 220b may be configured to calculate the energy as a sum of the squares of the samples of the respective subframe.
  • Task T230 calculates a gain factor for the subframe as the square root of the ratio of those energies.
  • task T230 calculates the gain factor as the square root of the ratio of the energy of highband signal S30 to the energy of synthesized highband signal S 130 over the subframe.
  • FIGtORE 22 shows a flowchart of such an implementation T210 of gain factor calculation task T200.
  • Task T215a applies a windowing function to highband signal S30, and task T215b applies the same windowing function to synthesized highband signal S 130.
  • Implementations 222a and 222b of tasks 220a and 220b calculate the energies of the respective windows, and task T230 calculates a gain factor for the subframe as the square root of the ratio of the energies.
  • highband gain factor calculator A230 is configured to apply a trapezoidal windowing function as shown in FIGURE 23 a, in which the window overlaps each of the two adjacent subframes by one millisecond.
  • FIGURE 23b shows an application of this windowing function to each of the five subframes of a 20- millisecond frame.
  • highband gain factor calculator A230 may be configured to apply windowing functions having different overlap periods and/or different window shapes (e.g., rectangular, Hamming) that may be symmetrical or asymmetrical. It is also possible for an implementation of highband gain factor calculator A230 to be configured to apply different windowing functions to different subframes within a frame and/or for a frame to include subframes of different lengths.
  • windowing functions having different overlap periods and/or different window shapes (e.g., rectangular, Hamming) that may be symmetrical or asymmetrical. It is also possible for an implementation of highband gain factor calculator A230 to be configured to apply different windowing functions to different subframes within a frame and/or for a frame to include subframes of different lengths.
  • each frame has 140 samples. If such a frame is divided into five subframes of equal length, each subframe will have 28 samples, and the window as shown in FIGURE 23a will be 42 samples wide. For a highband signal sampled at 8 kHz, each frame has 160 samples. If such frame is divided into five subframes of equal length, each subframe will have 32 samples, and the window as shown in FIGURE 23a will be 48 samples wide. In other implementations, subframes of any width may be used, and it is even possible for an implementation of highband gain calculator A230 to be configured to produce a different gain factor for each sample of a frame.
  • FIG. 24 shows a block diagram of an implementation B202 of highband decoder B200.
  • Highband decoder B202 includes a highband excitation generator B300 that is configured to produce highband excitation signal S 120 based on narrowband excitation signal S80.
  • highband excitation generator B300 may be implemented according to any of the implementations of highband excitation generator A300 as described herein. Typically it is desirable to implement highband excitation generator B300 to have the same response as the highband excitation generator of the highband encoder of the particular coding system.
  • narrowband decoder BIlO will typically perform dequantization of encoded narrowband excitation signal S50, however, in most cases highband excitation generator B300 may be implemented to receive narrowband excitation signal S80 from narrowband decoder BIlO and need not include an inverse quantizer configured to dequantize encoded narrowband excitation signal S50. It is also possible for narrowband decoder BIlO to be implemented to include an instance of anti-sparseness filter 600 arranged to filter the dequantized narrowband excitation signal before it is input to a narrowband synthesis filter such as filter 330.
  • Inverse quantizer 560 is configured to dequantize highband filter parameters S60a (in this example, to a set of LSFs), and LSF-to-LP filter coefficient transform 570 is configured to transform the LSFs into a set of filter coefficients (for example, as described above with reference to inverse quantizer 240 and transform 250 of narrowband encoder A122).
  • different coefficient sets e.g., cepstral coefficients
  • coefficient representations e.g., ISPs
  • Highband synthesis filter B200 is configured to produce a synthesized highband signal according to highband excitation signal S 120 and the set of filter coefficients.
  • Hi ghband decoder B202 also includes an inverse quantizer 580 configured to dequantize highband gain factors S60b, and a gain control element 590 (e.g., a multiplier or amplifier) configured and arranged to apply the dequantized gain factors to the synthesized highband signal to produce highband signal SlOO.
  • a gain control element 590 e.g., a multiplier or amplifier
  • gain control element 590 may include logic configured to apply the gain factors to the respective subframes, possibly according to a windowing function that may be the same or a different windowing function as applied by a gain calculator (e.g., highband gain calculator A230) of the corresponding highband encoder.
  • gain control element 590 is similarly configured but is arranged instead to apply the dequantized gain factors to narrowband excitation signal S80 or to highband excitation signal S 120.
  • highband excitation generators A300 and B300 of such an implementation may be configured such that the state of the noise generator is a deterministic function of information already coded within the same frame (e.g., narrowband filter parameters S40 or a portion thereof and/or encoded narrowband excitation signal S50 or a portion thereof).
  • One or more of the quantizers of the elements described herein may be configured to perform classified vector quantization.
  • a quantizer may be configured to select one of a set of codebooks based on information that has already been coded within the same frame in the narrowband channel and/or in the highband channel.
  • Such a technique typically provides increased coding efficiency at the expense of additional codebook storage.
  • the residual signal may contain a sequence of roughly periodic pulses or spikes over time.
  • Such structure which is typically related to pitch, is especially likely to occur in voiced speech signals.
  • Calculation of a quantized representation of the narrowband residual signal may include encoding of this pitch structure according to a model of long-term periodicity as represented by, for example, one or more codebooks.
  • the pitch structure of an actual residual signal may not match the periodicity model exactly.
  • the residual signal may include small jitters in the regularity of the locations of the pitch pulses, such that the distances between successive pitch pulses in a frame are not exactly equal and the structure is not quite regular. These irregularities tend to reduce coding efficiency.
  • narrowband encoder A120 are configured to perform a regularization of the pitch structure by applying an adaptive time warping to the residual before or during quantization, or by otherwise including an adaptive time warping in the encoded excitation signal.
  • an encoder may be configured to select or otherwise calculate a degree of warping in time (e.g., according to one or more perceptual weighting and/or error minimization criteria) such that the resulting excitation signal optimally fits the model of long-term periodicity.
  • Regularization of pitch structure is performed by a subset of CELP encoders called Relaxation Code Excited Linear Prediction (RCELP) encoders.
  • RELP Relaxation Code Excited Linear Prediction
  • An RCELP encoder is typically configured to perform the time warping as an adaptive time shift. This time shift may be a delay ranging from a few milliseconds negative to a few milliseconds positive, and it is usually varied smoothly to avoid audible discontinuities.
  • such an encoder is configured to apply the regularization in a piecewise fashion, wherein each frame or subframe is warped by a corresponding fixed time shift.
  • the encoder is configured to apply the regularization as a continuous warping function, such that a frame or subframe is warped according to a pitch contour (also called a pitch trajectory).
  • the encoder is configured to include a time warping in the encoded excitation signal by applying the shift to a perceptually weighted input signal that is used to calculate the encoded excitation signal.
  • the encoder calculates an encoded excitation signal that is regularized and quantized, and the decoder dequantizes the encoded excitation signal to obtain an excitation signal that is used to synthesize the decoded speech signal.
  • the decoded output signal thus exhibits the same varying delay that was included in the encoded excitation signal by the regularization. Typically, no information specifying the regularization amounts is transmitted to the decoder.
  • Regularization tends to make the residual signal easier to encode, which improves the coding gain from the long-term predictor and thus boosts overall coding efficiency, generally without generating artifacts. It may be desirable to perform regularization only on frames that are voiced. For example, narrowband encoder A124 may be configured to shift only those frames or subframes having a long-term structure, such as voiced signals. It may even be desirable to perform regularization only on subframes that include pitch pulse energy.
  • RCELP coding are described in U.S. Pats. Nos. 5,704,003 (Kleijn et al.) and 6,879,955 (Rao) and in U.S. Pat. Appl. Publ.
  • RCELP coders include the Enhanced Variable Rate Codec (EVRC), as described in Telecommunications Industry Association (TIA) IS-127, and the Third Generation Partnership Project 2 (3GPP2) Selectable Mode Vocoder (SMV).
  • EVRC Enhanced Variable Rate Codec
  • TIA Telecommunications Industry Association
  • 3GPP2 Third Generation Partnership Project 2
  • SMV Selectable Mode Vocoder
  • a misalignment in time between the warped highband excitation signal and the original highband speech signal may cause several problems.
  • the warped highband excitation signal may no longer provide a suitable source excitation for a synthesis filter that is configured according to the filter parameters extracted from the original highband speech signal.
  • the synthesized highband signal may contain audible artifacts that reduce the perceived quality of the decoded wideband speech signal.
  • the misalignment in time may also cause inefficiencies in gain envelope encoding. As mentioned above, a correlation is likely to exist between the temporal envelopes of narrowband excitation signal S80 and highband signal S30.
  • Embodiments include methods of wideband speech encoding that perform time warping of a highband speech signal according to a time warping included in a corresponding encoded narrowband excitation signal. Potential advantages of such methods include improving the quality of a decoded wideband speech signal and/or improving the efficiency of coding a highband gain envelope.
  • FIGURE 25 shows a block diagram of an implementation ADlO of wideband speech encoder AlOO.
  • Encoder ADlO includes an implementation A124 of narrowband encoder A120 that is configured to perform regularization during calculation of the encoded narrowband excitation signal S50.
  • narrowband encoder A124 may be configured according to one or more of the RCELP implementations discussed above.
  • Narrowband encoder Al 24 is also configured to output a regularization data signal SDlO that specifies the degree of time warping applied.
  • regularization data signal SDlO may include a series of values indicating each time shift amount as an integer or non-integer value in terms of samples, milliseconds, or some other time increment.
  • regularization information signal SDlO may include a corresponding description of the modification, such as a set of function parameters.
  • narrowband encoder A124 is configured to divide a frame into three subframes and to calculate a fixed time shift for each subframe, such that regularization data signal SDlO indicates three time shift amounts for each regularized frame of the encoded narrowband signal.
  • Wideband speech encoder ADlO includes a delay line D 120 configured to advance or retard portions of highband speech signal S30, according to delay amounts indicated by an input signal, to produce time-warped highband speech signal S30a.
  • delay line D 120 is configured to time warp highband speech signal S30 according to the warping indicated by regularization data signal SDlO. In such manner, the same amount of time warping that was included in encoded narrowband excitation signal S50 is also applied to the corresponding portion of highband speech signal S30 before analysis.
  • delay line D 120 is arranged as part of the highband encoder.
  • highband encoder A200 may be configured to perform spectral analysis (e.g., LPC analysis) of the unwarped highband speech signal S30 and to perform time warping of highband speech signal S30 before calculation of highband gain parameters S60b.
  • spectral analysis e.g., LPC analysis
  • Such an encoder may include, for example, an implementation of delay line D 120 arranged to perform the time warping.
  • highband filter parameters S60a based on the analysis of unwarped signal S30 may describe a spectral envelope that is misaligned in time with highband excitation signal S 120.
  • Delay line D 120 may be configured according to any combination of logic elements and storage elements suitable for applying the desired time warping operations to highband speech signal S30.
  • delay line D 120 may be configured to read highband speech signal S30 from a buffer according to the desired time shifts.
  • FIGURE 26a shows a schematic diagram of such an implementation D 122 of delay line D120 that includes a shift register SRl.
  • Shift register SRl is a buffer of some length m that is configured to receive and store the m most recent samples of highband speech signal S30.
  • the value m is equal to at least the sum of the maximum positive (or "advance") and negative (or "retard”) time shifts to be supported.
  • Delay line D 122 is configured to output the time-warped highband signal S30a from an offset location OL of shift register SRl.
  • the position of offset location OL varies about a reference position (zero time shift) according to the current time shift as indicated by, for example, regularization data signal SDlO.
  • Delay line D 122 may be configured to support equal advance and retard limits or, alternatively, one limit larger than the other such that a greater shift may be performed in one direction than in the other.
  • FIGURE 26a shows a particular example that supports a larger positive than negative time shift.
  • Delay line D 122 may be configured to output one or more samples at a time (depending on an output bus width, for example).
  • a regularization time shift having a magnitude of more than a few milliseconds may cause audible artifacts in the decoded signal.
  • the magnitude of a regularization time shift as performed by a narrowband encoder A124 will not exceed a few milliseconds, such that the time shifts indicated by regularization data signal SDlO will be limited.
  • delay line D 122 it may be desired in such cases for delay line D 122 to be configured to impose a maximum limit on time shifts in the positive and/or negative direction (for example, to observe a tighter limit than that imposed by the narrowband encoder).
  • FIGURE 26b shows a schematic diagram of an implementation D 124 of delay line D122 that includes a shift window SW.
  • the position of offset location OL is limited by the shift window SW.
  • FIGURE 26b shows a case in which the buffer length m is greater than the width of shift window SW, delay line D 124 may also be implemented such that the width of shift window SW is equal to m.
  • delay line D 120 is configured to write highband speech signal S30 to a buffer according to the desired time shifts.
  • FIGURE 27 shows a schematic diagram of such an implementation D 130 of delay line D 120 that includes two shift registers SR2 and SR3 configured to receive and store highband speech signal S30.
  • Delay line D 130 is configured to write a frame or subframe from shift register SR2 to shift register SR3 according to a time shift as indicated by, for example, regularization data signal SDlO.
  • Shift register SR3 is configured as a FIFO buffer arranged to output time-warped highband signal S30.
  • shift register SR2 includes a frame buffer portion FBI and a delay buffer portion DB
  • shift register SR3 includes a frame buffer portion FB2, an advance buffer portion AB, and a retard buffer portion RB.
  • the lengths of advance buffer AB and retard buffer RB may be equal, or one may be larger than the other, such that a greater shift in one direction is supported than in the other.
  • Delay buffer DB and retard buffer portion RB may be configured to have the same length.
  • delay buffer DB may be shorter than retard buffer RB to account for a time interval required to transfer samples from frame buffer FB 1 to shift register SR3, which may include other processing operations such as warping of the samples before storage to shift register SR3.
  • frame buffer FBI is configured to have a length equal to that of one frame of highband signal S30.
  • frame buffer FBI is configured to have a length equal to that of one subframe of highband signal S30.
  • delay line D 130 may be configured to include logic to apply the same (e.g., an average) delay to all subframes of a frame to be shifted.
  • Delay line D130 may also include logic to average values from frame buffer FBI with values to be overwritten in retard buffer RB or advance buffer AB.
  • shift register SR3 may be configured to receive values of highband signal S30 only via frame buffer FBI, and in such case delay line D 130 may include logic to interpolate across gaps between successive frames or subframes written to shift register SR3.
  • delay line D130 may be configured to perform a warping operation on samples from frame buffer FBI before writing them to shift register SR3 (e.g., according to a function described by regularization data signal SDlO).
  • FIGURE 28 shows a block diagram of an implementation AD12 of wideband speech encoder ADlO that includes a delay value mapper Dl 10.
  • Delay value mapper Dl 10 is configured to map the warping indicated by regularization data signal SDlO into mapped delay values SDlOa.
  • Delay line D 120 is arranged to produce time-warped highband speech signal S30a according to the warping indicated by mapped delay values SDlOa.
  • the time shift applied by the narrowband encoder may be expected to evolve smoothly over time.
  • delay value mapper Dl 10 is configured to calculate an average of the subframe delay values for each frame, and delay line D120 is configured to apply the calculated average to a corresponding frame of highband signal S30.
  • delay value mapper DIlO may be configured to round the value to an integer number of samples before outputting it to delay line D 120.
  • Narrowband encoder A124 may be configured to include a regularization time shift of a non-integer number of samples in the encoded narrowband excitation signal.
  • delay value mapper DIlO it may be desirable for delay value mapper DIlO to be configured to round the narrowband time shift to an integer number of samples and for delay line D120 to apply the rounded time shift to highband speech signal S30.
  • delay value mapper DIlO may be configured to adjust time shift amounts indicated in regularization data signal SDlO to account for a difference between the sampling rates of narrowband speech signal S20 (or narrowband excitation signal S80) and highband speech signal S30.
  • delay value mapper Dl 10 may be configured to scale the time shift amounts according to a ratio of the sampling rates. In one particular example as mentioned above, narrowband speech signal S20 is sampled at 8 kHz, and highband speech signal S30 is sampled at 7 kHz. In this case, delay value mapper Dl 10 is configured to multiply each shift amount by 7/8. Implementations of delay value mapper DIlO may also be configured to perform such a scaling operation together with an integer-rounding and/or a time shift averaging operation as described herein.
  • delay line D 120 is configured to otherwise modify the time scale of a frame or other sequence of samples (e.g., by compressing one portion and expanding another portion).
  • narrowband encoder A124 may be configured to perform the regularization according to a function such as a pitch contour or trajectory.
  • regularization data signal SDlO may include a corresponding description of the function, such as a set of parameters
  • delay line D 120 may include logic configured to warp frames or subframes of highband speech signal S30 according to the function.
  • delay value mapper Dl 10 is configured to average, scale, and/or round the function before it is applied to highband speech signal S30 by delay line D 120.
  • delay value mapper DIlO may be configured to calculate one or more delay values according to the function, each delay value indicating a number of samples, which are then applied by delay line D 120 to time warp one or more corresponding frames or subframes of highband speech signal S30.
  • FIG. 29 shows a flowchart for a method MDlOO of time warping a highband speech signal according to a time warping included in a corresponding encoded narrowband excitation signal.
  • Task TDlOO processes a wideband speech signal to obtain a narrowband speech signal and a highband speech signal.
  • task TDlOO may be configured to filter the wideband speech signal using a filter bank having lowpass and highpass filters, such as an implementation of filter bank AIlO.
  • Task TD200 encodes the narrowband speech signal into at least a encoded narrowband excitation signal and a plurality of narrowband filter parameters.
  • the encoded narrowband excitation signal and/or filter parameters may be quantized, and the encoded narrowband speech signal may also include other parameters such as a speech mode parameter.
  • Task TD200 also includes a time warping in the encoded narrowband excitation signal.
  • Task TD300 generates a highband excitation signal based on a narrowband excitation signal.
  • the narrowband excitation signal is based on the encoded narrowband excitation signal.
  • task TD400 encodes the highband speech signal into at least a plurality of highband filter parameters.
  • task TD400 may be configured to encode the highband speech signal into a plurality of quantized LSFs.
  • Task TD500 applies a time shift to the highband speech signal that is based on information relating to a time warping included in the encoded narrowband excitation signal.
  • Task TD400 may be configured to perform a spectral analysis (such as an LPC analysis) on the highband speech signal, and/or to calculate a gain envelope of the highband speech signal.
  • task TD500 may be configured to apply the time shift to the highband speech signal prior to the analysis and/or the gain envelope calculation.
  • wideband speech encoder AlOO are configured to reverse a time warping of highband excitation signal S 120 caused by a time warping included in the encoded narrowband excitation signal.
  • highband excitation generator A300 may be implemented to include an implementation of delay line D120 that is configured to receive regularization data signal SDlO or mapped delay values SDlOa, and to apply a corresponding reverse time shift to narrowband excitation signal S 80, and/or to a subsequent signal based on it such as harmonically extended signal S 160 or highband excitation signal S 120.
  • Further wideband speech encoder implementations may be configured to encode narrowband speech signal S20 and highband speech signal S30 independently from one another, such that highband speech signal S30 is encoded as a representation of a highband spectral envelope and a highband excitation signal.
  • Such an implementation may be configured to perform time warping of the highband residual signal, or to otherwise include a time warping in an encoded highband excitation signal, according to information relating to a time warping included in the encoded narrowband excitation signal.
  • the highband encoder may include an implementation of delay line D120 and/or delay value mapper DIlO as described herein that are configured to apply a time warping to the highband residual signal. Potential advantages of such an operation include more efficient encoding of the highband residual signal and a better match between the synthesized narrowband and highband speech signals.
  • embodiments as described herein include implementations that may be used to perform embedded coding, supporting compatibility with narrowband systems and avoiding a need for transcoding.
  • Support for highband coding may also serve to differentiate on a cost basis between chips, chipsets, devices, and/or networks having wideband support with backward compatibility, and those having narrowband support only.
  • Support for highband coding as described herein may also be used in conjunction with a technique for supporting lowband coding, and a system, method, or apparatus according to such an embodiment may support coding of frequency components from, for example, about 50 or 100 Hz up to about 7 or 8 kHz.
  • highband support may improve intelligibility, especially regarding differentiation of fricatives. Although such differentiation may usually be derived by a human listener from the particular context, highband support may serve as an enabling feature in speech recognition and other machine interpretation applications, such as systems for automated voice menu navigation and/or automatic call processing.
  • An apparatus may be embedded into a portable device for wireless communications such as a cellular telephone or personal digital assistant (PDA).
  • a portable device for wireless communications
  • such an apparatus may be included in another communications device such as a VoIP handset, a personal computer configured to support VoIP communications, or a network device configured to route telephonic or VoIP communications.
  • an apparatus according to an embodiment may be implemented in a chip or chipset for a communications device.
  • such a device may also include such features as analog-to-digital and/or digital-to-analog conversion of a speech signal, circuitry for performing amplification and/or other signal processing operations on a speech signal, and/or radio- frequency circuitry for transmission and/or reception of the coded speech signal.
  • embodiments may include and/or be used with any one or more of the other features disclosed in the U.S. Provisional Pat. Appls. Nos. 60/667,901 and 60/673,965 of which this application claims benefit.
  • Such features include removal of high-energy bursts of short duration that occur in the highband and are substantially absent from the narrowband.
  • Such features include fixed or adaptive smoothing of coefficient representations such as highband LSFs.
  • Such features include fixed or adaptive shaping of noise associated with quantization of coefficient representations such as LSFs.
  • Such features also include fixed or adaptive smoothing of a gain envelope, and adaptive attenuation of a gain envelope.
  • an embodiment may be implemented in part or in whole as a hard-wired circuit, as a circuit configuration fabricated into an application-specific integrated circuit, or as a firmware program loaded into non- volatile storage or a software program loaded from or into a data storage medium as machine-readable code, such code being instructions executable by an array of logic elements such as a microprocessor or other digital signal processing unit.
  • the data storage medium may be an array of storage elements such as semiconductor memory (which may include without limitation dynamic or static RAM (random-access memory), ROM (read-only memory), and/or flash RAM), or ferroelectric, magnetoresistive, ovonic, polymeric, or phase-change memory; or a disk medium such as a magnetic or optical disk.
  • semiconductor memory which may include without limitation dynamic or static RAM (random-access memory), ROM (read-only memory), and/or flash RAM), or ferroelectric, magnetoresistive, ovonic, polymeric, or phase-change memory
  • a disk medium such as a magnetic or optical disk.
  • the term "software” should be understood to include source code, assembly language code, machine code, binary code, firmware, macrocode, microcode, any one or more sets or sequences of instructions executable by an array of logic elements, and any combination of such examples.
  • highband excitation generators A300 and B300, highband encoder AlOO, highband decoder B200, wideband speech encoder AlOO, and wideband speech decoder BlOO may be implemented as electronic and/or optical devices residing, for example, on the same chip or among two or more chips in a chipset, although other arrangements without such limitation are also contemplated.
  • One or more elements of such an apparatus may be implemented in whole or in part as one or more sets of instructions arranged to execute on one or more fixed or programmable arrays of logic elements (e.g., transistors, gates) such as microprocessors, embedded processors, IP cores, digital signal processors, FPGAs (field-programmable gate arrays), ASSPs (application-specific standard products), and ASICs (application-specific integrated circuits). It is also possible for one or more such elements to have structure in common (e.g., a processor used to execute portions of code corresponding to different elements at different times, a set of instructions executed to perform tasks corresponding to different elements at different times, or an arrangement of electronic and/or optical devices performing operations for different elements at different times). Moreover, it is possible for one or more such elements to be used to perform tasks or execute other sets of instructions that are not directly related to an operation of the apparatus, such as a task relating to another operation of a device or system in which the apparatus is embedded.
  • logic elements e.g., transistors,
  • FIGURE 30 shows a flowchart of a method MlOO, according to an embodiment, of encoding a highband portion of a speech signal having a narrowband portion and the highband portion.
  • Task XlOO calculates a set of filter parameters that characterize a spectral envelope of the highband portion.
  • Task X200 calculates a spectrally extended signal by applying a nonlinear function to a signal derived from the narrowband portion.
  • Task X300 generates a synthesized highband signal according to (A) the set of filter parameters and (B) a highband excitation signal based on the spectrally extended signal.
  • Task X400 calculates a gain envelope based on a relation between (C) energy of the highband portion and (D) energy of a signal derived from the narrowband portion.
  • FIGURE 31a shows a flowchart of a method M200 of generating a highband excitation signal according to an embodiment.
  • Task YlOO calculates a harmonically extended signal by applying a nonlinear function to a narrowband excitation signal , derived from a narrowband portion of a speech signal.
  • Task Y200 mixes the harmonically extended signal with a modulated noise signal to generate a highband excitation signal.
  • FIGURE 31b shows a flowchart of a method M210 of generating a highband excitation signal according to another embodiment including tasks Y300 and Y400.
  • Task Y300 calculates a time-domain envelope according to energy over time of one among the narrowband excitation signal and the harmonically extended signal.
  • Task Y400 modulates a noise signal according to the time-domain envelope to produce the modulated noise signal.
  • FIGURE 32 shows a flowchart of a method M300 according to an embodiment, of decoding a highband portion of a speech signal having a narrowband portion and the highband portion.
  • Task ZlOO receives a set of filter parameters that characterize a spectral envelope of the highband portion and a set of gain factors that characterize a temporal envelope of the highband portion.
  • Task Z200 calculates a spectrally extended signal by applying a nonlinear function to a signal derived from the narrowband portion.
  • Task Z300 generates a synthesized highband signal according to (A) the set of filter parameters and (B) a highband excitation signal based on the spectrally extended signal.
  • Task Z400 modulates a gain envelope of the synthesized highband signal based on the set of gain factors.
  • task Z400 may be configured to modulate the gain envelope of the synthesized highband signal by applying the set of gain factors to an excitation signal derived from the narrowband portion, to the spectrally extended signal, to the highband excitation signal, or to the synthesized highband signal.
  • Embodiments also include additional methods of speech coding, encoding, and decoding as are expressly disclosed herein, e.g., by descriptions of structural embodiments configured to perform such methods.
  • Each of these methods may also be tangibly embodied (for example, in one or more data storage media as listed above) as one or more sets of instructions readable and/or executable by a machine including an array of logic elements (e.g., a processor, microprocessor, microcontroller, or other finite state machine).
  • logic elements e.g., a processor, microprocessor, microcontroller, or other finite state machine.

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Abstract

In one embodiment, a method of signal processing includes encoding a low-frequency portion of a speech signal into at least an encoded narrowband excitation signal and a plurality of narrowband filter parameters; and generating a highband excitation signal based on a narrowband excitation signal. The narrowband excitation signal is based on the encoded narrowband excitation signal. The method also includes encoding a high-frequency portion of the speech signal into at least a plurality of highband filter parameters according to at least the highband excitation signal. The encoded narrowband excitation signal includes a time warping, and the method includes applying a time shift to the high-frequency portion based on information related to the time warping.

Description

SYSTEMS, METHODS, AND APPARATUS FOR HIGHBAND TIME
WARPING
RELATED APPLICATIONS
[0001] This application claims benefit of U.S. Provisional Pat. Appl. No. 60/667,901, entitled "CODING THE HIGH-FREQUENCY BAND OF WIDEBAND SPEECH," filed April 1, 2005. This application also claims benefit of U.S. Provisional Pat. Appl. No. 60/673,965, entitled "PARAMETER CODING IN A HIGH-BAND SPEECH CODER," filed April 22, 2005.
FIELD OF THE INVENTION
[0002] This invention relates to signal processing.
BACKGROUND
[0003] Voice communications over the public switched telephone network (PSTN) have traditionally been limited in bandwidth to the frequency range of 300-3400 kHz. New networks for voice communications, such as cellular telephony and voice over IP (Internet Protocol, VoIP), may not have the same bandwidth limits, and it may be desirable to transmit and receive voice communications that include a wideband frequency range over such networks. For example, it may be desirable to support an audio frequency range that extends down to 50 Hz and/or up to 7 or 8 kHz. It may also be desirable to support other applications, such as high-quality audio or audio/video conferencing, that may have audio speech content in ranges outside the traditional PSTN limits.
[0004] Extension of the range supported by a speech coder into higher frequencies may improve intelligibility. For example, the information that differentiates fricatives such as V and T is largely in the high frequencies. Highband extension may also improve other qualities of speech, such as presence. For example, even a voiced vowel may have spectral energy far above the PSTN limit. [0005] One approach to wideband speech coding involves scaling a narrowband speech coding technique (e.g., one configured to encode the range of 0-4 kHz) to cover the wideband spectrum. For example, a speech signal may be sampled at a higher rate to include components at high frequencies, and a narrowband coding technique may be reconfigured to use more filter coefficients to represent this wideband signal. Narrowband coding techniques such as CELP (codebook excited linear prediction) are computationally intensive, however, and a wideband CELP coder may consume too many processing cycles to be practical for many mobile and other embedded applications. Encoding the entire spectrum of a wideband signal to a desired quality using such a technique may also lead to an unacceptably large increase in bandwidth. Moreover, transcoding of such an encoded signal would be required before even its narrowband portion could be transmitted into and/or decoded by a system that only supports narrowband coding.
[0006] Another approach to wideband speech coding involves extrapolating the highband spectral envelope from the encoded narrowband spectral envelope. While such an approach may be implemented without any increase in bandwidth and without a need for transcoding, the coarse spectral envelope or formant structure of the highband portion of a speech signal generally cannot be predicted accurately from the spectral envelope of the narrowband portion.
[0007] It may be desirable to implement wideband speech coding such that at least the narrowband portion of the encoded signal may be sent through a narrowband channel (such as a PSTN channel) without transcoding or other significant modification. Efficiency of the wideband coding extension may also be desirable, for example, to avoid a significant reduction in the number of users that may be serviced in applications such as wireless cellular telephony and broadcasting over wired and wireless channels.
SUMMARY
[0008] In one embodiment, a method of signal processing including encoding a low- frequency portion of a speech signal into at least an encoded lowband excitation signal and a plurality of lowband filter parameters; generating a highband excitation signal based on the encoded lowband excitation signal. The method also includes encoding, according to at least the highband excitation signal, a high-frequency portion of the speech signal into at least a plurality of highband filter parameters. In this method, the encoded lowband excitation signal describes a signal that is warped in time, with respect to the speech signal, according to a time-varying time warping. The method includes applying, based on information relating to the time warping, a plurality of different time shifts to a corresponding plurality of successive portions in time of the high-frequency portion.
[0009] In another embodiment, an apparatus includes a lowband speech encoder configured to encode a low-frequency portion of a speech signal into at least an encoded lowband excitation signal and a plurality of lowband filter parameters; and a highband speech encoder configured to generate a highband excitation signal based on the encoded lowband excitation signal. In this apparatus, the highband encoder is configured to encode a high-frequency portion of the speech signal into at least a plurality of highband filter parameters according to at least the highband excitation signal. In this apparatus, the narrowband speech encoder is configured to output a regularization data signal describing a time-varying time warping, with respect to the speech signal, that is included in the encoded narrowband excitation signal. The apparatus includes a delay line configured to apply a plurality of different time shifts to a corresponding plurality of successive portions in time of the high-frequency portion, wherein the plurality of different time shifts are based on the regularization data signal.
[00010] In another embodiment, an apparatus includes means for encoding a low- frequency portion of a speech signal into at least an encoded lowband excitation signal and a plurality of lowband filter parameters; means for generating a highband excitation signal based on the encoded lowband excitation signal; and means for encoding a high- frequency portion of the speech signal into at least a plurality of highband filter parameters according to at least the highband excitation signal. In this apparatus, the encoded narrowband excitation signal describes a signal that is warped in time, with respect to the speech signal, according to a time-varying time warping. The apparatus includes means for applying, based on information relating to the time warping, a plurality of different time shifts to a corresponding plurality of successive portions in time of the high-frequency portion. BRIEF DESCRIPTION OF THE DRAWINGS
[00011] FIGURE Ia shows a block diagram of a wideband speech encoder AlOO according to an embodiment.
[00012] FIGURE Ib shows a block diagram of an implementation A102 of wideband speech encoder AlOO.
[00013] FIGURE 2a shows a block diagram of a wideband speech decoder BlOO according to an embodiment.
[00014] FIGURE 2b shows a block diagram of an implementation B 102 of wideband speech encoder BlOO.
[00015] FIGURE 3a shows a block diagram of an implementation Al 12 of filter bank AIlO.
[00016] FIGURE 3b shows a block diagram of an implementation B 122 of filter bank B120.
[00017] FIGURE 4a shows bandwidth coverage of the low and high bands for one example of filter bank Al 10.
[00018] FIGURE 4b shows bandwidth coverage of the low and high bands for another example of filter bank Al 10.
[00019] FIGURE 4c shows a block diagram of an implementation Al 14 of filter bank A112.
[00020] FIGURE 4d shows a block diagram of an implementation B 124 of filter bank B 122.
[00021] FIGURE 5a shows an example of a plot of frequency vs. log amplitude for a speech signal.
[00022] FIGURE 5b shows a block diagram of a basic linear prediction coding system.
[00023] FIGURE 6 shows a block diagram of an implementation A122 of narrowband encoder A 120. [00024] FIGURE 7 shows a block diagram of an implementation Bl 12 of narrowband decoder BIlO.
[00025] FIGURE 8a shows an example of a plot of frequency vs. log amplitude for a residual signal for voiced speech.
[00026] FIGURE 8b shows an example of a plot of time vs. log amplitude for a residual signal for voiced speech.
[00027] FIGURE 9 shows a block diagram of a basic linear prediction coding system that also performs long-term prediction.
[00028] FIGURE 10 shows a block diagram of an implementation A202 of highband encoder A200.
[00029] FIGURE 11 shows a block diagram of an implementation A302 of highband excitation generator A300.
[00030] FIGURE 12 shows a block diagram of an implementation A402 of spectrum extender A400.
[00031] FIGURE 12a shows plots of signal spectra at various points in one example of a spectral extension operation.
[00032] FIGURE 12b shows plots of signal spectra at various points in another example of a spectral extension operation.
[00033] FIGURE 13 shows a block diagram of an implementation A304 of highband excitation generator A302.
[00034] FIGURE 14 shows a block diagram of an implementation A306 of highband excitation generator A302.
[00035] FIGURE 15 shows a flowchart for an envelope calculation task TlOO.
[00036] FIGURE 16 shows a block diagram of an implementation 492 of combiner 490. [00037] FIGURE 17 illustrates an approach to calculating a measure of periodicity of highband signal S30.
[00038] FIGURE 18 shows a block diagram of an implementation A312 of highband excitation generator A302.
[00039] FIGURE 19 shows a block diagram of an implementation A314 of highband excitation generator A302.
[00040] FIGURE 20 shows a block diagram of an implementation A316 of highband excitation σ gevnerator A302.
[00041] FIGURE 21 shows a flowchart for a gain calculation task T200.
[00042] FIGURE 22 shows a flowchart for an implementation T210 of gain calculation task T200.
[00043] FIGURE 23a shows a diagram of a windowing function.
[00044] FIGURE 23b shows an application of a windowing function as shown in FIGURE 23a to subframes of a speech signal.
[00045] FIGURE 24 shows a block diagram for an implementation B202 of highband decoder B200.
[00046] FIGURE 25 shows a block diagram of an implementation ADlO of wideband speech encoder AlOO.
[00047] FIGURE 26a shows a schematic diagram of an implementation D122 of delay line D120.
[00048] FIGURE 26b shows a schematic diagram of an implementation D 124 of delay line D120.
[00049] FIGURE 27 shows a schematic diagram of an implementation D 130 of delay line D 120.
[00050] FIGURE 28 shows a block diagram of an implementation AD 12 of wideband speech encoder ADlO. [00051] FIGURE 29 shows a flowchart of a method of signal processing MDlOO according to an embodiment.
[00052] FIGURE 30 shows a flowchart for a method MlOO according to an embodiment.
[00053] FIGURE 31a shows a flowchart for a method M200 according to an embodiment.
[00054] FIGURE 31b shows a flowchart for an implementation M210 of method M200.
[00055] FIGURE 32 shows a flowchart for a method M300 according to an embodiment.
[00056] In the figures and accompanying description, the same reference labels refer to the same or analogous elements or signals.
DETAILED DESCRIPTION
[00057] Embodiments as described herein include systems, methods, and apparatus that may be configured to provide an extension to a narrowband speech coder to support transmission and/or storage of wideband speech signals at a bandwidth increase of only about 800 to 1000 bps (bits per second). Potential advantages of such implementations include embedded coding to support compatibility with narrowband systems, relatively easy allocation and reallocation of bits between the narrowband and highband coding channels, avoiding a computationally intensive wideband synthesis operation, and maintaining a low sampling rate for signals to be processed by computationally intensive waveform coding routines.
[00058] Unless expressly limited by its context, the term "calculating" is used herein to indicate any of its ordinary meanings, such as computing, generating, and selecting from a list of values. Where the term "comprising" is used in the present description and claims, it does not exclude other elements or operations. The term "A is based on B" is used to indicate any of its ordinary meanings, including the cases (i) "A is equal to B" and (ii) "A is based on at least B." The term "Internet Protocol" includes version 4, as described in IETF (Internet Engineering Task Force) RFC (Request for Comments) 791, and subsequent versions such as version 6.
[00059] FIGURE Ia shows a block diagram of a wideband speech encoder AlOO according to an embodiment. Filter bank AIlO is configured to filter a wideband speech signal SlO to produce a narrowband signal S20 and a highband signal S30. Narrowband encoder A 120 is configured to encode narrowband signal S20 to produce narrowband (NB) filter parameters S40 and a narrowband residual signal S50. As described in further detail herein, narrowband encoder A 120 is typically configured to produce narrowband filter parameters S40 and encoded narrowband excitation signal S50 as codebook indices or in another quantized form. Highband encoder A200 is configured to encode highband signal S30 according to information in encoded narrowband excitation signal S50 to produce highband coding parameters S60. As described in further detail herein, highband encoder A200 is typically configured to produce highband coding parameters S60 as codebook indices or in another quantized form. One particular example of wideband speech encoder AlOO is configured to encode wideband speech signal SlO at a rate of about 8.55 kbps (kilobits per second), with about 7.55 kbps being used for narrowband filter parameters S40 and encoded narrowband excitation signal S50, and about 1 kbps being used for highband coding parameters S60.
[00060] It may be desired to combine the encoded narrowband and highband signals into a single bitstream. For example, it may be desired to multiplex the encoded signals together for transmission (e.g., over a wired, optical, or wireless transmission channel), or for storage, as an encoded wideband speech signal. FIGURE Ib shows a block diagram of an implementation A102 of wideband speech encoder AlOO that includes a multiplexer A130 configured to combine narrowband filter parameters S40, encoded narrowband excitation signal S50, and highband filter parameters S60 into a multiplexed signal S70.
[00061] An apparatus including encoder A102 may also include circuitry configured to transmit multiplexed signal S70 into a transmission channel such as a wired, optical, or wireless channel. Such an apparatus may also be configured to perform one or more channel encoding operations on the signal, such as error correction encoding (e.g., rate- compatible convolutional encoding) and/or error detection encoding (e.g., cyclic redundancy encoding), and/or one or more layers of network protocol encoding (e.g., Ethernet, TCP/IP, cdma2000).
[00062] It may be desirable for multiplexer A130 to be configured to embed the encoded narrowband signal (including narrowband filter parameters S40 and encoded narrowband excitation signal S50) as a separable substream of multiplexed signal S70, such that the encoded narrowband signal may be recovered and decoded independently of another portion of multiplexed signal S70 such as a highband and/or lowband signal. For example, multiplexed signal S70 may be arranged such that the encoded narrowband signal may be recovered by stripping away the highband filter parameters S60. One potential advantage of such a feature is to avoid the need for transcoding the encoded wideband signal before passing it to a system that supports decoding of the narrowband signal but does not support decoding of the highband portion.
[00063] FIGURE 2a is a block diagram of a wideband speech decoder BlOO according to an embodiment. Narrowband decoder BIlO is configured to decode narrowband filter parameters S40 and encoded narrowband excitation signal S50 to produce a narrowband signal S90. Highband decoder B200 is configured to decode highband coding parameters S60 according to a narrowband excitation signal S80, based on encoded narrowband excitation signal S50, to produce a highband signal SlOO. In this example, narrowband decoder BIlO is configured to provide narrowband excitation signal S 80 to highband decoder B200. Filter bank B 120 is configured to combine narrowband signal S90 and highband signal SlOO to produce a wideband speech signal SIlO.
[00064] FIGURE 2b is a block diagram of an implementation B 102 of wideband speech decoder BlOO that includes a demultiplexer B 130 configured to produce encoded signals S40, S50, and S60 from multiplexed signal S70. An apparatus including decoder B 102 may include circuitry configured to receive multiplexed signal S70 from a transmission channel such as a wired, optical, or wireless channel. Such an apparatus may also be configured to perform one or more channel decoding operations on the signal, such as error correction decoding (e.g., rate-compatible convolutional decoding) and/or error detection decoding (e.g., cyclic redundancy decoding), and/or one or more layers of network protocol decoding (e.g., Ethernet, TCP/IP, cdma2000). [00065] Filter bank AIlO is configured to filter an input signal according to a split- band scheme to produce a low-frequency subband and a high-frequency subband. Depending on the design criteria for the particular application, the output subbands may have equal or unequal bandwidths and may be overlapping or nonoverlapping. A configuration of filter bank Al 10 that produces more than two subbands is also possible. For example, such a filter bank may be configured to produce one or more lowband signals that include components in a frequency range below that of narrowband signal S20 (such as the range of 50-300 Hz). It is also possible for such a filter bank to be configured to produce one or more additional highband signals that include components in a frequency range above that of highband signal S30 (such as a range of 14-20, 16-20, or 16-32 kHz). In such case, wideband speech encoder AlOO may be implemented to encode this signal or signals separately, and multiplexer A130 may be configured to include the additional encoded signal or signals in multiplexed signal S70 (e.g., as a separable portion).
[00066] FIGURE 3a shows a block diagram of an implementation Al 12 of filter bank AIlO that is configured to produce two subband signals having reduced sampling rates. Filter bank AIlO is arranged to receive a wideband speech signal SlO having a high- frequency (or highband) portion and a low-frequency (or lowband) portion. Filter bank Al 12 includes a lowband processing path configured to receive wideband speech signal SlO and to produce narrowband speech signal S20, and a highband processing path configured to receive wideband speech signal SlO and to produce highband speech signal S30. Lowpass filter 110 filters wideband speech signal SlO to pass a selected low-frequency subband, and highpass filter 130 filters wideband speech signal SlO to pass a selected high-frequency subband. Because both subband signals have more narrow bandwidths than wideband speech signal SlO, their sampling rates can be reduced to some extent without loss of information. Downsampler 120 reduces the sampling rate of the lowpass signal according to a desired decimation factor (e.g., by removing samples of the signal and/or replacing samples with average values), and downsampler 140 likewise reduces the sampling rate of the highpass signal according to another desired decimation factor.
[00067] FIGURE 3b shows a block diagram of a corresponding implementation B 122 of filter bank B 120. Upsampler 150 increases the sampling rate of narrowband signal S90 (e.g., by zero-stuffing and/or by duplicating samples), and lowpass filter 160 filters the upsampled signal to pass only a lowband portion (e.g., to prevent aliasing). Likewise, upsampler 170 increases the sampling rate of highband signal SlOO and highpass filter 180 filters the upsampled signal to pass only a highband portion. The two passband signals are then summed to form wideband speech signal SIlO. In some implementations of decoder BlOO, filter bank B 120 is configured to produce a weighted sum of the two passband signals according to one or more weights received and/or calculated by highband decoder B200. A configuration of filter bank B 120 that combines more than two passband signals is also contemplated.
[00068] Each of the filters 110, 130, 160, 180 may be implemented as a finite-impulse- response (FIR) filter or as an infinite-impulse-response (IIR) filter. The frequency responses of encoder filters 110 and 130 may have symmetric or dissimilarly shaped transition regions between stopband and passband. Likewise, the frequency responses of decoder filters 160 and 180 may have symmetric or dissimilarly shaped transition regions between stopband and passband. It may be desirable but is not strictly necessary for lowpass filter 110 to have the same response as lowpass filter 160, and for highpass filter 130 to have the same response as highpass filter 180. In one example, the two filter pairs 110, 130 and 160, 180 are quadrature mirror filter (QMF) banks, with filter pair 110, 130 having the same coefficients as filter pair 160, 180.
[00069] In a typical example, lowpass filter 110 has a passband that includes the limited PSTN range of 300-3400 Hz (e.g., the band from 0 to 4 kHz). FIGURES 4a and 4b show relative bandwidths of wideband speech signal SlO, narrowband signal S20, and highband signal S30 in two different implementational examples. In both of these particular examples, wideband speech signal SlO has a sampling rate of 16 kHz (representing frequency components within the range of 0 to 8 kHz), and narrowband signal S20 has a sampling rate of 8 kHz (representing frequency components within the range of 0 to 4 kHz).
[00070] In the example of FIGURE 4a, there is no significant overlap between the two subbands. A highband signal S30 as shown in this example may be obtained using a highpass filter 130 with a passband of 4-8 kHz. In such a case, it may be desirable to reduce the sampling rate to 8 kHz by downsampling the filtered signal by a factor of two. Such an operation, which may be expected to significantly reduce the computational complexity of further processing operations on the signal, will move the passband energy down to the range of 0 to 4 kHz without loss of information.
[00071] In the alternative example of FIGURE 4b, the upper and lower subbands have an appreciable overlap, such that the region of 3.5 to 4 kHz is described by both subband signals. A highband signal S30 as in this example may be obtained using a highpass filter 130 with a passband of 3.5-7 kHz. In such a case, it may be desirable to reduce the sampling rate to 7 kHz by downsampling the filtered signal by a factor of 16/7. Such an operation, which may be expected to significantly reduce the computational complexity of further processing operations on the signal, will move the passband energy down to the range of 0 to 3.5 kHz without loss of information.
[00072] In a typical handset for telephonic communication, one or more of the transducers (i.e., the microphone and the earpiece or loudspeaker) lacks an appreciable response over the frequency range of 7-8 kHz. In the example of FIGURE 4b, the portion of wideband speech signal SlO between 7 and 8 kHz is not included in the encoded signal. Other particular examples of highpass filter 130 have passbands of 3.5- 7.5 kHz and 3.5-8 kHz.
[00073] In some implementations, providing an overlap between subbands as in the example of FIGURE 4b allows for the use of a lowpass and/or a highpass filter having a smooth rolloff over the overlapped region. Such filters are typically easier to design, less computationally complex, and/or introduce less delay than filters with sharper or "brick-wall" responses. Filters having sharp transition regions tend to have higher sidelobes (which may cause aliasing) than filters of similar order that have smooth rolloff s. Filters having sharp transition regions may also have long impulse responses which may cause ringing artifacts. For filter bank implementations having one or more IIR filters, allowing for a smooth rolloff over the overlapped region may enable the use of a filter or filters whose poles are farther away from the unit circle, which may be important to ensure a stable fixed-point implementation.
[00074] Overlapping of subbands allows a smooth blending of lowband and highband that may lead to fewer audible artifacts, reduced aliasing, and/or a less noticeable transition from one band to the other. Moreover, the coding efficiency of narrowband encoder A120 (for example, a waveform coder) may drop with increasing frequency. For example, coding quality of the narrowband coder may be reduced at low bit rates, especially in the presence of background noise. In such cases, providing an overlap of the subbands may increase the quality of reproduced frequency components in the overlapped region.
[00075] Moreover, overlapping of subbands allows a smooth blending of lowband and highband that may lead to fewer audible artifacts, reduced aliasing, and/or a less noticeable transition from one band to the other. Such a feature may be especially desirable for an implementation in which narrowband encoder A120 and highband encoder A200 operate according to different coding methodologies. For example, different coding techniques may produce signals that sound quite different. A coder that encodes a spectral envelope in the form of codebook indices may produce a signal having a different sound than a coder that encodes the amplitude spectrum instead. A time-domain coder (e.g., a pulse-code-modulation or PCM coder) may produce a signal having a different sound than a frequency-domain coder. A coder that encodes a signal with a representation of the spectral envelope and the corresponding residual signal may produce a signal having a different sound than a coder that encodes a signal with only a representation of the spectral envelope. A coder that encodes a signal as a representation of its waveform may produce an output having a different sound than that from a sinusoidal coder. In such cases, using filters having sharp transition regions to define nonoverlapping subbands may lead to an abrupt and perceptually noticeable transition between the subbands in the synthesized wideband signal.
[00076] Although QMF filter banks having complementary overlapping frequency responses are often used in subband techniques, such filters are unsuitable for at least some of the wideband coding implementations described herein. A QMF filter bank at the encoder is configured to create a significant degree of aliasing that is canceled in the corresponding QMF filter bank at the decoder. Such an arrangement may not be appropriate for an application in which the signal incurs a significant amount of distortion between the filter banks, as the distortion may reduce the effectiveness of the alias cancellation property. For example, applications described herein include coding implementations configured to operate at very low bit rates. As a consequence of the very low bit rate, the decoded signal is likely to appear significantly distorted as compared to the original signal, such that use of QMF filter banks may lead to uncanceled aliasing. Applications that use QMF filter banks typically have higher bit rates (e.g., over 12 kbps for AMR, and 64 kbps for G.722).
[00077] Additionally, a coder may be configured to produce a synthesized signal that is perceptually similar to the original signal but which actually differs significantly from the original signal. For example, a coder that derives the highband excitation from the narrowband residual as described herein may produce such a signal, as the actual highband residual may be completely absent from the decoded signal. Use of QMF filter banks in such applications may lead to a significant degree of distortion caused by uncanceled aliasing.
[00078] The amount of distortion caused by QMF aliasing may be reduced if the affected subband is narrow, as the effect of the aliasing is limited to a bandwidth equal to the width of the subband. For examples as described herein in which each subband includes about half of the wideband bandwidth, however, distortion caused by uncanceled aliasing could affect a significant part of the signal. The quality of the signal may also be affected by the location of the frequency band over which the uncanceled aliasing occurs. For example, distortion created near the center of a wideband speech signal (e.g., between 3 and 4 kHz) may be much more objectionable than distortion that occurs near an edge of the signal (e.g., above 6 IdEIz).
[00079] While the responses of the filters of a QMF filter bank are strictly related to one another, the lowband and highband paths of filter banks Al 10 and B 120 may be configured to have spectra that are completely unrelated apart from the overlapping of the two subbands. We define the overlap of the two subbands as the distance from the point at which the frequency response of the highband filter drops to -20 dB up to the point at which the frequency response of the lowband filter drops to -20 dB. In various examples of filter bank AIlO and/or B 120, this overlap ranges from around 200 Hz to around 1 kHz. The range of about 400 to about 600 Hz may represent a desirable tradeoff between coding efficiency and perceptual smoothness. In one particular example as mentioned above, the overlap is around 500 Hz.
[00080] It may be desirable to implement filter bank Al 12 and/or B 122 to perform operations as illustrated in FIGURES 4a and 4b in several stages. For example, FIGURE 4c shows a block diagram of an implementation Al 14 of filter bank Al 12 that performs a functional equivalent of highpass filtering and downsampling operations using a series of interpolation, resampling, decimation, and other operations. Such an implementation may be easier to design and/or may allow reuse of functional blocks of logic and/or code. For example, the same functional block may be used to perform the operations of decimation to 14 kHz and decimation to 7 kHz as shown in FIGURE 4c. The spectral reversal operation may be implemented by multiplying the signal with the function eJnπ or the sequence (—1)", whose values alternate between +1 and -1. The spectral shaping operation may be implemented as a lowpass filter configured to shape the signal to obtain a desired overall filter response.
[00081] It is noted that as a consequence of the spectral reversal operation, the spectrum of highband signal S30 is reversed. Subsequent operations in the encoder and corresponding decoder may be configured accordingly. For example, highband excitation generator A300 as described herein may be configured to produce a highband excitation signal S 120 that also has a spectrally reversed form.
[00082] FIGURE 4d shows a block diagram of an implementation B 124 of filter bank B 122 that performs a functional equivalent of upsampling and highpass filtering operations using a series of interpolation, resampling, and other operations. Filter bank B 124 includes a spectral reversal operation in the highband that reverses a similar operation as performed, for example, in a filter bank of the encoder such as filter bank Al 14. In this particular example, filter bank B124 also includes notch filters in the lowband and highband that attenuate a component of the signal at 7100 Hz, although such filters are optional and need not be included. The Patent Application "SYSTEMS, METHODS, AND APPARATUS FOR SPEECH SIGNAL FILTERING" filed herewith, Attorney Docket 050551, includes additional description and figures relating to responses of elements of particular implementations of filter banks Al 10 and B 120, and this material is hereby incorporated by reference.
[00083] Narrowband encoder A120 is implemented according to a source-filter model that encodes the input speech signal as (A) a set of parameters that describe a filter and (B) an excitation signal that drives the described filter to produce a synthesized reproduction of the input speech signal. FIGURE 5a shows an example of a spectral envelope of a speech signal. The peaks that characterize this spectral envelope represent resonances of the vocal tract and are called formants. Most speech coders encode at least this coarse spectral structure as a set of parameters such as filter coefficients.
[00084] FIGURE 5b shows an example of a basic source-filter arrangement as applied to coding of the spectral envelope of narrowband signal S20. An analysis module calculates a set of parameters that characterize a filter corresponding to the speech sound over a period of time (typically 20 msec). A whitening filter (also called an analysis or prediction error filter) configured according to those filter parameters removes the spectral envelope to spectrally flatten the signal. The resulting whitened signal (also called a residual) has less energy and thus less variance and is easier to encode than the original speech signal. Errors resulting from coding of the residual signal may also be spread more evenly over the spectrum. The filter parameters and residual are typically quantized for efficient transmission over the channel. At the decoder, a synthesis filter configured according to the filter parameters is excited by a signal based on the residual to produce a synthesized version of the original speech sound. The synthesis filter is typically configured to have a transfer function that is the inverse of the transfer function of the whitening filter.
[00085] FIGURE 6 shows a block diagram of a basic implementation A122 of narrowband encoder A120. In this example, a linear prediction coding (LPC) analysis module 210 encodes the spectral envelope of narrowband signal S20 as a set of linear prediction (LP) coefficients (e.g., coefficients of an all-pole filter 1/A(z)). The analysis module typically processes the input signal as a series of nonoverlapping frames, with a new set of coefficients being calculated for each frame. The frame period is generally a period over which the signal may be expected to be locally stationary; one common example is 20 milliseconds (equivalent to 160 samples at a sampling rate of 8 kHz). In one example, LPC analysis module 210 is configured to calculate a set of ten LP filter coefficients to characterize the formant structure of each 20-millisecond frame. It is also possible to implement the analysis module to process the input signal as a series of overlapping frames.
[00086] The analysis module may be configured to analyze the samples of each frame directly, or the samples may be weighted first according to a windowing function (for example, a Hamming window). The analysis may also be performed over a window that is larger than the frame, such as a 30-msec window. This window may be symmetric (e.g. 5-20-5, such that it includes the 5 milliseconds immediately before and after the 20-millisecond frame) or asymmetric (e.g. 10-20, such that it includes the last 10 milliseconds of the preceding frame). An LPC analysis module is typically configured to calculate the LP filter coefficients using a Levinson-Durbin recursion or the Leroux-Gueguen algorithm. In another implementation, the analysis module may be configured to calculate a set of cepstral coefficients for each frame instead of a set of LP filter coefficients.
[00087] The output rate of encoder A120 may be reduced significantly, with relatively little effect on reproduction quality, by quantizing the filter parameters. Linear prediction filter coefficients are difficult to quantize efficiently and are usually mapped into another representation, such as line spectral pairs (LSPs) or line spectral frequencies (LSFs), for quantization and/or entropy encoding. In the example of HGURE 6, LP filter coefficient-to-LSF transform 220 transforms the set of LP filter coefficients into a corresponding set of LSFs. Other one-to-one representations of LP filter coefficients include parcor coefficients; log-area-ratio values; immittance spectral pairs (ISPs); and immittance spectral frequencies (ISFs), which are used in the GSM (Global System for Mobile Communications) AMR-WB (Adaptive Multirate- Wideband) codec. Typically a transform between a set of LP filter coefficients and a corresponding set of LSFs is reversible, but embodiments also include implementations of encoder A120 in which the transform is not reversible without error.
[00088] Quantizer 230 is configured to quantize the set of narrowband LSFs (or other coefficient representation), and narrowband encoder A122 is configured to output the result of this quantization as the narrowband filter parameters S40. Such a quantizer typically includes a vector quantizer that encodes the input vector as an index to a corresponding vector entry in a table or codebook.
[00089] As seen in FIGURE 6, narrowband encoder A 122 also generates a residual signal by passing narrowband signal S20 through a whitening filter 260 (also called an analysis or prediction error filter) that is configured according to the set of filter coefficients. In this particular example, whitening filter 260 is implemented as a FIR filter, although IIR implementations may also be used. This residual signal will typically contain perceptually important information of the speech frame, such as long- term structure relating to pitch, that is not represented in narrowband filter parameters S40. Quantizer 270 is configured to calculate a quantized representation of this residual signal for output as encoded narrowband excitation signal S50. Such a quantizer typically includes a vector quantizer that encodes the input vector as an index to a corresponding vector entry in a table or codebook. Alternatively, such a quantizer may be configured to send one or more parameters from which the vector may be generated dynamically at the decoder, rather than retrieved from storage, as in a sparse codebook method. Such a method is used in coding schemes such as algebraic CELP (codebook excitation linear prediction) and codecs such as 3GPP2 (Third Generation Partnership 2) EVRC (Enhanced Variable Rate Codec).
[00090] It is desirable for narrowband encoder A120 to generate the encoded narrowband excitation signal according to the same filter parameter values that will be available to the corresponding narrowband decoder. In this manner, the resulting encoded narrowband excitation signal may already account to some extent for nonidealities in those parameter values, such as quantization error. Accordingly, it is desirable to configure the whitening filter using the same coefficient values that will be available at the decoder. In the basic example of encoder A122 as shown in FIGURE 6, inverse quantizer 240 dequantizes narrowband coding parameters S40, LSF-to-LP filter coefficient transform 250 maps the resulting values back to a corresponding set of LP filter coefficients, and this set of coefficients is used to configure whitening filter 260 to generate the residual signal that is quantized by quantizer 270.
[00091] Some implementations of narrowband encoder A120 are configured to calculate encoded narrowband excitation signal S50 by identifying one among a set of codebook vectors that best matches the residual signal. It is noted, however, that narrowband encoder A120 may also be implemented to calculate a quantized representation of the residual signal without actually generating the residual signal. For example, narrowband encoder A120 may be configured to use a number of codebook vectors to generate corresponding synthesized signals (e.g., according to a current set of filter parameters), and to select the codebook vector associated with the generated signal that best matches the original narrowband signal S20 in a perceptually weighted domain.
[00092] FIGURE 7 shows a block diagram of an implementation B 112 of narrowband decoder BIlO. Inverse quantizer 310 dequantizes narrowband filter parameters S40 (in this case, to a set of LSFs), and LSF-to-LP filter coefficient transform 320 transforms the LSFs into a set of filter coefficients (for example, as described above with reference to inverse quantizer 240 and transform 250 of narrowband encoder A122). Inverse quantizer 340 dequantizes narrowband residual signal S40 to produce a narrowband excitation signal S80. Based on the filter coefficients and narrowband excitation signal S 80, narrowband synthesis filter 330 synthesizes narrowband signal S90. In other words, narrowband synthesis filter 330 is configured to spectrally shape narrowband excitation signal S 80 according to the dequantized filter coefficients to produce narrowband signal S90. Narrowband decoder Bl 12 also provides narrowband excitation signal S 80 to highband encoder A200, which uses it to derive the highband excitation signal S 120 as described herein. In some implementations as described below, narrowband decoder BIlO may be configured to provide additional information to highband decoder B200 that relates to the narrowband signal, such as spectral tilt, pitch gain and lag, and speech mode.
[00093] The system of narrowband encoder A122 and narrowband decoder Bl 12 is a basic example of an analysis-by-synthesis speech codec. Codebook excitation linear prediction (CELP) coding is one popular family of analysis-by-synthesis coding, and implementations of such coders may perform waveform encoding of the residual, including such operations as selection of entries from fixed and adaptive codebooks, error minimization operations, and/or perceptual weighting operations. Other implementations of analysis-by-synthesis coding include mixed excitation linear prediction (MELP), algebraic CELP (ACELP), relaxation CELP (RCELP), regular pulse excitation (RPE), multi-pulse CELP (MPE), and vector-sum excited linear prediction (VSELP) coding. Related coding methods include multi-band excitation (MBE) and prototype waveform interpolation (PWI) coding. Examples of standardized analysis-by-synthesis speech codecs include the ETSI (European Telecommunications Standards Institute)-GSM full rate codec (GSM 06.10), which uses residual excited linear prediction (RELP); the GSM enhanced full rate codec (ETSI-GSM 06.60); the ITU (International Telecommunication Union) standard 11.8 kb/s G.729 Annex E coder; the IS (Interim Standard)-641 codecs for IS-136 (a time-division multiple access scheme); the GSM adaptive multirate (GSM-AMR) codecs; and the 4GV™ (Fourth- Generation Vocoder™) codec (QUALCOMM Incorporated, San Diego, CA). Narrowband encoder A 120 and corresponding decoder Bl 10 may be implemented according to any of these technologies, or any other speech coding technology (whether known or to be developed) that represents a speech signal as (A) a set of parameters that describe a filter and (B) an excitation signal used to drive the described filter to reproduce the speech signal.
[00094] Even after the whitening filter has removed the coarse spectral envelope from narrowband signal S20, a considerable amount of fine harmonic structure may remain, especially for voiced speech. FIGURE 8a shows a spectral plot of one example of a residual signal, as may be produced by a whitening filter, for a voiced signal such as a vowel. The periodic structure visible in this example is related to pitch, and different voiced sounds spoken by the same speaker may have different formant structures but similar pitch structures. FIGURE 8b shows a time-domain plot of an example of such a residual signal that shows a sequence of pitch pulses in time.
[00095] Coding efficiency and/or speech quality may be increased by using one or more parameter values to encode characteristics of the pitch structure. One important characteristic of the pitch structure is the frequency of the first harmonic (also called the fundamental frequency), which is typically in the range of 60 to 400 Hz. This characteristic is typically encoded as the inverse of the fundamental frequency, also called the pitch lag. The pitch lag indicates the number of samples in one pitch period and may be encoded as one or more codebook indices. Speech signals from male speakers tend to have larger pitch lags than speech signals from female speakers.
[00096] Another signal characteristic relating to the pitch structure is periodicity, which indicates the strength of the harmonic structure or, in other words, the degree to which the signal is harmonic or nonharmonic. Two typical indicators of periodicity are zero crossings and normalized autocorrelation functions (NACFs). Periodicity may also be indicated by the pitch gain, which is commonly encoded as a codebook gain (e.g., a quantized adaptive codebook gain).
[00097] Narrowband encoder Al 20 may include one or more modules configured to encode the long-term harmonic structure of narrowband signal S20. As shown in FIGURE 9, one typical CELP paradigm that may be used includes an open-loop LPC analysis module, which encodes the short-term characteristics or coarse spectral envelope, followed by a closed-loop long-term prediction analysis stage, which encodes the fine pitch or harmonic structure. The short-term characteristics are encoded as filter coefficients, and the long-term characteristics are encoded as values for parameters such as pitch lag and pitch gain. For example, narrowband encoder A120 may be configured to output encoded narrowband excitation signal S50 in a form that includes one or more codebook indices (e.g., a fixed codebook index and an adaptive codebook index) and corresponding gain values. Calculation of this quantized representation of the narrowband residual signal (e.g., by quantizer 270) may include selecting such indices and calculating such values. Encoding of the pitch structure may also include interpolation of a pitch prototype waveform, which operation may include calculating a difference between successive pitch pulses. Modeling of the long-term structure may be disabled for frames corresponding to unvoiced speech, which is typically noise-like and unstructured.
[00098] An implementation of narrowband decoder BIlO according to a paradigm as shown in FIGURE 9 may be configured to output narrowband excitation signal S 80 to highband decoder B200 after the long-term structure (pitch or harmonic structure) has been restored. For example, such a decoder may be configured to output narrowband excitation signal S 80 as a dequantized version of encoded narrowband excitation signal S50. Of course, it is also possible to implement narrowband decoder BIlO such that highband decoder B200 performs dequantization of encoded narrowband excitation signal S50 to obtain narrowband excitation signal S80.
[00099] In an implementation of wideband speech encoder AlOO according to a paradigm as shown in FIGURE 9, highband encoder A200 may be configured to receive the narrowband excitation signal as produced by the short-term analysis or whitening filter. In other words, narrowband encoder A120 may be configured to output the narrowband excitation signal to highband encoder A200 before encoding the long-term structure. It is desirable, however, for highband encoder A200 to receive from the narrowband channel the same coding information that will be received by highband decoder B200, such that the coding parameters produced by highband encoder A200 may already account to some extent for nonidealities in that information. Thus it may be preferable for highband encoder A200 to reconstruct narrowband excitation signal S 80 from the same parametrized and/or quantized encoded narrowband excitation signal S50 to be output by wideband speech encoder AlOO. One potential advantage of this approach is more accurate calculation of the highband gain factors S60b described below.
[00010O]In addition to parameters that characterize the short-term and/or long-term structure of narrowband signal S20, narrowband encoder A120 may produce parameter values that relate to other characteristics of narrowband signal S20. These values, which may be suitably quantized for output by wideband speech encoder AlOO, may be included among the narrowband filter parameters S40 or outputted separately. Highband encoder A200 may also be configured to calculate highband coding parameters S60 according to one or more of these additional parameters (e.g., after dequantization). At wideband speech decoder BlOO, highband decoder B200 may be configured to receive the parameter values via narrowband decoder BIlO (e.g., after dequantization). Alternatively, highband decoder B200 may be configured to receive (and possibly to dequantize) the parameter values directly.
[00010I]In one example of additional narrowband coding parameters, narrowband encoder A 120 produces values for spectral tilt and speech mode parameters for each frame. Spectral tilt relates to the shape of the spectral envelope over the passband and is typically represented by the quantized first reflection coefficient. For most voiced sounds, the spectral energy decreases with increasing frequency, such that the first reflection coefficient is negative and may approach — 1. Most unvoiced sounds have a spectrum that is either flat, such that the first reflection coefficient is close to zero, or has more energy at high frequencies, such that the first reflection coefficient is positive and may approach +1.
[000102] Speech mode (also called voicing mode) indicates whether the current frame represents voiced or unvoiced speech. This parameter may have a binary value based on one or more measures of periodicity (e.g., zero crossings, NACFs, pitch gain) and/or voice activity for the frame, such as a relation between such a measure and a threshold value. In other implementations, the speech mode parameter has one or more other states to indicate modes such as silence or background noise, or a transition between silence and voiced speech.
[000103]Highband encoder A200 is configured to encode highband signal S30 according to a source-filter model, with the excitation for this filter being based on the encoded narrowband excitation signal. FIGURE 10 shows a block diagram of an implementation A202 of highband encoder A200 that is configured to produce a stream of highband coding parameters S60 including highband filter parameters S60a and highband gain factors S60b. Highband excitation generator A300 derives a highband excitation signal S 120 from encoded narrowband excitation signal S50. Analysis module A210 produces a set of parameter values that characterize the spectral envelope of highband signal S30. In this particular example, analysis module A210 is configured to perform LPC analysis to produce a set of LP filter coefficients for each frame of highband signal S30. Linear prediction filter coefficient-to-LSF transform 410 transforms the set of LP filter coefficients into a corresponding set of LSFs. As noted above with reference to analysis module 210 and transform 220, analysis module A210 and/or transform 410 may be configured to use other coefficient sets (e.g., cepstral coefficients) and/or coefficient representations (e.g., ISPs).
[000104] Quantizer 420 is configured to quantize the set of highband LSFs (or other coefficient representation, such as ISPs), and highband encoder A202 is configured to output the result of this quantization as the highband filter parameters S60a. Such a quantizer typically includes a vector quantizer that encodes the input vector as an index to a corresponding vector entry in a table or codebook.
[000105] Highband encoder A202 also includes a synthesis filter A220 configured to produce a synthesized highband signal S 130 according to highband excitation signal S 120 and the encoded spectral envelope (e.g., the set of LP filter coefficients) produced by analysis module A210. Synthesis filter A220 is typically implemented as an BDR. filter, although FIR implementations may also be used. In a particular example, synthesis filter A220 is implemented as a sixth-order linear autoregressive filter.
[000106] Highband gain factor calculator A230 calculates one or more differences between the levels of the original highband signal S30 and synthesized highband signal S 130 to specify a gain envelope for the frame. Quantizer 430, which may be implemented as a vector quantizer that encodes the input vector as an index to a corresponding vector entry in a table or codebook, quantizes the value or values specifying the gain envelope, and highband encoder A202 is configured to output the result of this quantization as highband gain factors S60b. [000107] In an implementation as shown in FIGURE 10, synthesis filter A220 is arranged to receive the filter coefficients from analysis module A210. An alternative implementation of highband encoder A202 includes an inverse quantizer and inverse transform configured to decode the filter coefficients from highband filter parameters S60a, and in this case synthesis filter A220 is arranged to receive the decoded filter coefficients instead. Such an alternative arrangement may support more accurate calculation of the gain envelope by highband gain calculator A230.
[000108] In one particular example, analysis module A210 and highband gain calculator A230 output a set of six LSFs and a set of five gain values per frame, respectively, such that a wideband extension of the narrowband signal S20 may be achieved with only eleven additional values per frame. The ear tends to be less sensitive to frequency errors at high frequencies, such that highband coding at a low LPC order may produce a signal having a comparable perceptual quality to narrowband coding at a higher LPC order. A typical implementation of highband encoder A200 may be configured to output 8 to 12 bits per frame for high-quality reconstruction of the spectral envelope and another 8 to 12 bits per frame for high-quality reconstruction of the temporal envelope. In another particular example, analysis module A210 outputs a set of eight LSFs per frame.
[000109] Some implementations of highband encoder A200 are configured to produce highband excitation signal S 120 by generating a random noise signal having highband frequency components and amplitude-modulating the noise signal according to the time- domain envelope of narrowband signal S20, narrowband excitation signal S80, or highband signal S30. While such a noise-based method may produce adequate results for unvoiced sounds, however, it may not be desirable for voiced sounds, whose residuals are usually harmonic and consequently have some periodic structure.
[000110] Highband excitation generator A300 is configured to generate highband excitation signal S 120 by extending the spectrum of narrowband excitation signal S 80 into the highband frequency range. FIGURE 11 shows a block diagram of an implementation A302 of highband excitation generator A300. Inverse quantizer 450 is configured to dequantize encoded narrowband excitation signal S50 to produce narrowband excitation signal S80. Spectrum extender A400 is configured to produce a harmonically extended signal S 160 based on narrowband excitation signal S80. Combiner 470 is configured to combine a random noise signal generated by noise generator 480 and a time-domain envelope calculated by envelope calculator 460 to produce a modulated noise signal S 170. Combiner 490 is configured to mix harmonically extended signal S60 and modulated noise signal S 170 to produce highband excitation signal S 120.
[00011I]In one example, spectrum extender A400 is configured to perform a spectral folding operation (also called mirroring) on narrowband excitation signal S 80 to produce harmonically extended signal S 160. Spectral folding may be performed by zero-stuffing excitation signal S80 and then applying a highpass filter to retain the alias. In another example, spectrum extender A400 is configured to produce harmonically extended signal S 160 by spectrally translating narrowband excitation signal S 80 into the highband (e.g., via upsampling followed by multiplication with a constant-frequency cosine signal).
[000112] Spectral folding and translation methods may produce spectrally extended signals whose harmonic structure is discontinuous with the original harmonic structure of narrowband excitation signal S 80 in phase and/or frequency. For example, such methods may produce signals having peaks that are not generally located at multiples of the fundamental frequency, which may cause tinny-sounding artifacts in the reconstructed speech signal. These methods also tend to produce high-frequency harmonics that have unnaturally strong tonal characteristics. Moreover, because a PSTN signal may be sampled at 8 kHz but bandlimited to no more than 3400 Hz, the upper spectrum of narrowband excitation signal S 80 may contain little or no energy, such that an extended signal generated according to a spectral folding or spectral translation operation may have a spectral hole above 3400 Hz.
[000113] Other methods of generating harmonically extended signal S 160 include identifying one or more fundamental frequencies of narrowband excitation signal S 80 and generating harmonic tones according to that information. For example, the harmonic structure of an excitation signal may be characterized by the fundamental frequency together with amplitude and phase information. Another implementation of highband excitation generator A300 generates a harmonically extended signal S 160 based on the fundamental frequency and amplitude (as indicated, for example, by the pitch lag and pitch gain). Unless the harmonically extended signal is phase-coherent with narrowband excitation signal S 80, however, the quality of the resulting decoded speech may not be acceptable.
[000114] A nonlinear function may be used to create a highband excitation signal that is phase-coherent with the narrowband excitation and preserves the harmonic structure without phase discontinuity. A nonlinear function may also provide an increased noise level between high-frequency harmonics, which tends to sound more natural than the tonal high-frequency harmonics produced by methods such as spectral folding and spectral translation. Typical memoryless nonlinear functions that may be applied by various implementations of spectrum extender A400 include the absolute value function (also called fullwave rectification), halfwave rectification, squaring, cubing, and clipping. Other implementations of spectrum extender A400 may be configured to apply a nonlinear function having memory.
[000115]FIGURE 12 is a block diagram of an implementation A402 of spectrum extender A400 that is configured to apply a nonlinear function to extend the spectrum of narrowband excitation signal S 80. Upsampler 510 is configured to upsample narrowband excitation signal S80. It may be desirable to upsample the signal sufficiently to minimize aliasing upon application of the nonlinear function. In one particular example, upsampler 510 upsamples the signal by a factor of eight. Upsampler 510 may be configured to perform the upsampling operation by zero-stuffing the input signal and lowpass filtering the result. Nonlinear function calculator 520 is configured to apply a nonlinear function to the upsampled signal. One potential advantage of the absolute value function over other nonlinear functions for spectral extension, such as squaring, is that energy normalization is not needed. In some implementations, the absolute value function may be applied efficiently by stripping or clearing the sign bit of each sample. Nonlinear function calculator 520 may also be configured to perform an amplitude warping of the upsampled or spectrally extended signal.
[000116] Downsampler 530 is configured to downsample the spectrally extended result of applying the nonlinear function. It may be desirable for downsampler 530 to perform a bandpass filtering operation to select a desired frequency band of the spectrally extended signal before reducing the sampling rate (for example, to reduce or avoid aliasing or corruption by an unwanted image). It may also be desirable for downsampler 530 to reduce the sampling rate in more than one stage. [000117] FIGURE 12a is a diagram that shows the signal spectra at various points in one example of a spectral extension operation, where the frequency scale is the same across the various plots. Plot (a) shows the spectrum of one example of narrowband excitation signal S80. Plot (b) shows the spectrum after signal S80 has been upsampled by a factor of eight. Plot (c) shows an example of the extended spectrum after application of a nonlinear function. Plot (d) shows the spectrum after lowpass filtering. In this example, the passband extends to the upper frequency limit of highband signal S30 (e.g., 7 kHz or 8 kHz).
[000118] Plot (e) shows the spectrum after a first stage of downsampling, in which the sampling rate is reduced by a factor of four to obtain a wideband signal. Plot (f) shows the spectrum after a highpass filtering operation to select the highband portion of the extended signal, and plot (g) shows the spectrum after a second stage of downsampling, in which the sampling rate is reduced by a factor of two. In one particular example, downsampler 530 performs the highpass filtering and second stage of downsampling by passing the wideband signal through highpass filter 130 and downsampler 140 of filter bank Al 12 (or other structures or routines having the same response) to produce a spectrally extended signal having the frequency range and sampling rate of highband signal S30.
[000119] As may be seen in plot (g), downsampling of the highpass signal shown in plot (f) causes a reversal of its spectrum. In this example, downsampler 530 is also configured to perform a spectral flipping operation on the signal. Plot (h) shows a result of applying the spectral flipping operation, which may be performed by multiplying the signal with the function eJnπ or the sequence (-l)n, whose values alternate between +1 and -1. Such an operation is equivalent to shifting the digital spectrum of the signal in the frequency domain by a distance of π, It is noted that the same result may also be obtained by applying the downsampling and spectral flipping operations in a different order. The operations of upsampling and/or downsampling may also be configured to include resampling to obtain a spectrally extended signal having the sampling rate of highband signal S30 (e.g., 7 kHz).
[000120] As noted above, filter banks Al 10 and B 120 may be implemented such that one or both of the narrowband and highband signals S20, S30 has a spectrally reversed form at the output of filter bank Al 10, is encoded and decoded in the spectrally reversed form, and is spectrally reversed again at filter bank B 120 before being output in wideband speech signal SIlO. In such case, of course, a spectral flipping operation as shown in FIGURE 12a would not be necessary, as it would be desirable for highband excitation signal S 120 to have a spectrally reversed form as well.
[000121] The various tasks of upsampling and downsampling of a spectral extension operation as performed by spectrum extender A402 may be configured and arranged in many different ways. For example, FIGURE 12b is a diagram that shows the signal spectra at various points in another example of a spectral extension operation, where the frequency scale is the same across the various plots. Plot (a) shows the spectrum of one example of narrowband excitation signal S80. Plot (b) shows the spectrum after signal S 80 has been upsampled by a factor of two. Plot (c) shows an example of the extended spectrum after application of a nonlinear function. In this case, aliasing that may occur in the higher frequencies is accepted.
[000122] Plot (d) shows the spectrum after a spectral reversal operation. Plot (e) shows the spectrum after a single stage of downsampling, in which the sampling rate is reduced by a factor of two to obtain the desired spectrally extended signal. In this example, the signal is in spectrally reversed form and may be used in an implementation of highband encoder A200 which processed highband signal S30 in such a form.
[000123] The spectrally extended signal produced by nonlinear function calculator 520 is likely to have a pronounced dropoff in amplitude as frequency increases. Spectral extender A402 includes a spectral flattener 540 configured to perform a whitening operation on the downsampled signal. Spectral flattener 540 may be configured to perform a fixed whitening operation or to perform an adaptive whitening operation. In a particular example of adaptive whitening, spectral flattener 540 includes an LPC analysis module configured to calculate a set of four filter coefficients from the downsampled signal and a fourth-order analysis filter configured to whiten the signal according to those coefficients. Other implementations of spectrum extender A400 include configurations in which spectral flattener 540 operates on the spectrally extended signal before downsampler 530.
[000124] Highband excitation generator A300 may be implemented to output harmonically extended signal S 160 as highband excitation signal S 120. In some cases, however, using only a harmonically extended signal as the highband excitation may result in audible artifacts. The harmonic structure of speech is generally less pronounced in the highband than in the low band, and using too much harmonic structure in the highband excitation signal can result in a buzzy sound. This artifact may be especially noticeable in speech signals from female speakers.
[000125] Embodiments include implementations of highband excitation generator A300 that are configured to mix harmonically extended signal S 160 with a noise signal. As shown in FIGURE 11, highband excitation generator A302 includes a noise generator 480 that is configured to produce a random noise signal. In one example, noise generator 480 is configured to produce a unit-variance white pseudorandom noise signal, although in other implementations the noise signal need not be white and may have a power density that varies with frequency. It may be desirable for noise generator 480 to be configured to output the noise signal as a deterministic function such that its state may be duplicated at the decoder. For example, noise generator 480 may be configured to output the noise signal as a deterministic function of information coded earlier within the same frame, such as the narrowband filter parameters S40 and/or encoded narrowband excitation signal S50.
[000126] Before being mixed with harmonically extended signal S 160, the random noise signal produced by noise generator 480 may be amplitude-modulated to have a time- domain envelope that approximates the energy distribution over time of narrowband signal S20, highband signal S30, narrowband excitation signal S 80, or harmonically extended signal S160. As shown in FIGURE 11, highband excitation generator A302 includes a combiner 470 configured to amplitude-modulate the noise signal produced by noise generator 480 according to a time-domain envelope calculated by envelope calculator 460. For example, combiner 470 may be implemented as a multiplier arranged to scale the output of noise generator 480 according to the time-domain envelope calculated by envelope calculator 460 to produce modulated noise signal S 170.
[000127] In an implementation A304 of highband excitation generator A302, as shown in the block diagram of FIGURE 13, envelope calculator 460 is arranged to calculate the envelope of harmonically extended signal S 160. In an implementation A306 of highband excitation generator A302, as shown in the block diagram of FIGURE 14, envelope calculator 460 is arranged to calculate the envelope of narrowband excitation signal S80. Further implementations of highband excitation generator A302 may be otherwise configured to add noise to harmonically extended signal S 160 according to locations of the narrowband pitch pulses in time.
[000128] Envelope calculator 460 may be configured to perform an envelope calculation as a task that includes a series of subtasks. FIGURE 15 shows a flowchart of an example TlOO of such a task. Subtask Tl 10 calculates the square of each sample of the frame of the signal whose envelope is to be modeled (for example, narrowband excitation signal S 80 or harmonically extended signal S 160) to produce a sequence of squared values. Subtask T120 performs a smoothing operation on the sequence of squared values. In one example, subtask T 120 applies a first-order IIR lowpass filter to the sequence according to the expression
y(n) = ax(n) + (ϊ- a)y(n -ϊ) , (1)
where x is the filter input, y is the filter output, n is a time-domain index, and a is a smoothing coefficient having a value between 0.5 and 1. The value of the smoothing coefficient a may be fixed or, in an alternative implementation, may be adaptive according to an indication of noise in the input signal, such that a is closer to 1 in the absence of noise and closer to 0.5 in the presence of noise. Subtask T130 applies a square root function to each sample of the smoothed sequence to produce the time- domain envelope.
[000129] Such an implementation of envelope calculator 460 may be configured to perform the various subtasks of task TlOO in serial and/or parallel fashion. In further implementations of task TlOO, subtask TIlO may be preceded by a bandpass operation configured to select a desired frequency portion of the signal whose envelope is to be modeled, such as the range of 3-4 kHz.
[000130] Combiner 490 is configured to mix harmonically extended signal S 160 and modulated noise signal S 170 to produce highband excitation signal S 120. Implementations of combiner 490 may be configured, for example, to calculate highband excitation signal S 120 as a sum of harmonically extended signal S 160 and modulated noise signal S 170. Such an implementation of combiner 490 may be configured to calculate highband excitation signal S 120 as a weighted sum by applying a weighting factor to harmonically extended signal S 160 and/or to modulated noise signal S 170 before the summation. Each such weighting factor may be calculated according to one or more criteria and may be a fixed value or, alternatively, an adaptive value that is calculated on a frame-by-frame or subframe-by-subframe basis.
[000131] FIGURE 16 shows a block diagram of an implementation 492 of combiner 490 that is configured to calculate highband excitation signal S 120 as a weighted sum of harmonically extended signal S 160 and modulated noise signal S 170. Combiner 492 is configured to weight harmonically extended signal S 160 according to harmonic weighting factor S 180, to weight modulated noise signal S 170 according to noise weighting factor S 190, and to output highband excitation signal S 120 as a sum of the weighted signals. In this example, combiner 492 includes a weighting factor calculator 550 that is configured to calculate harmonic weighting factor S180 and noise weighting factor S 190.
[000132] Weighting factor calculator 550 may be configured to calculate weighting factors S 180 and S 190 according to a desired ratio of harmonic content to noise content in highband excitation signal S 120. For example, it may be desirable for combiner 492 to produce highband excitation signal S 120 to have a ratio of harmonic energy to noise energy similar to that of highband signal S30. In some implementations of weighting factor calculator 550, weighting factors S 180, S 190 are calculated according to one or more parameters relating to a periodicity of narrowband signal S20 or of the narrowband residual signal, such as pitch gain and/or speech mode. Such an implementation of weighting factor calculator 550 may be configured to assign a value to harmonic weighting factor S 180 that is proportional to the pitch gain, for example, and/or to assign a higher value to noise weighting factor S 190 for unvoiced speech signals than for voiced speech signals.
[000133] In other implementations, weighting factor calculator 550 is configured to calculate values for harmonic weighting factor S 180 and/or noise weighting factor S 190 according to a measure of periodicity of highband signal S30. In one such example, weighting factor calculator 550 calculates harmonic weighting factor S 180 as the maximum value of the autocorrelation coefficient of highband signal S30 for the current frame or subframe, where the autocorrelation is performed over a search range that includes a delay of one pitch lag and does not include a delay of zero samples. FIGURE 17 shows an example of such a search range of length n samples that is centered about a delay of one pitch lag and has a width not greater than one pitch lag.
[000134] FIGURE 17 also shows an example of another approach in which weighting factor calculator 550 calculates a measure of periodicity of highband signal S30 in several stages. In a first stage, the current frame is divided into a number of subframes, and the delay for which the autocorrelation coefficient is maximum is identified separately for each subframe. As mentioned above, the autocorrelation is performed over a search range that includes a delay of one pitch lag and does not include a delay of zero samples.
[000135] In a second stage, a delayed frame is constructed by applying the corresponding identified delay to each subframe, concatenating the resulting subframes to construct an optimally delayed frame, and calculating harmonic weighting factor S 180 as the correlation coefficient between the original frame and the optimally delayed frame. In a further alternative, weighting factor calculator 550 calculates harmonic weighting factor S 180 as an average of the maximum autocorrelation coefficients obtained in the first stage for each subframe. Implementations of weighting factor calculator 550 may also be configured to scale the correlation coefficient, and/or to combine it with another value, to calculate the value for harmonic weighting factor S 180.
[000136] It may be desirable for weighting factor calculator 550 to calculate a measure of periodicity of highband signal S30 only in cases where a presence of periodicity in the frame is otherwise indicated. For example, weighting factor calculator 550 may be configured to calculate a measure of periodicity of highband signal S30 according to a relation between another indicator of periodicity of the current frame, such as pitch gain, and a threshold value. In one example, weighting factor calculator 550 is configured to perform an autocorrelation operation on highband signal S30 only if the frame's pitch gain (e.g., the adaptive codebook gain of the narrowband residual) has a value of more than 0.5 (alternatively, at least 0.5). In another example, weighting factor calculator 550 is configured to perform an autocorrelation operation on highband signal S30 only for frames having particular states of speech mode (e.g., only for voiced signals). In such cases, weighting factor calculator 550 may be configured to assign a default weighting factor for frames having other states of speech mode and/or lesser values of pitch gain. [000137] Embodiments include further implementations of weighting factor calculator 550 that are configured to calculate weighting factors according to characteristics other than or in addition to periodicity. For example, such an implementation may be configured to assign a higher value to noise gain factor S 190 for speech signals having a large pitch lag than for speech signals having a small pitch lag. Another such implementation of weighting factor calculator 550 is configured to determine a measure of harmonicity of wideband speech signal SlO, or of highband signal S30, according to a measure of the energy of the signal at multiples of the fundamental frequency relative to the energy of the signal at other frequency components.
[000138] Some implementations of wideband speech encoder AlOO are configured to output an indication of periodicity or harmonicity (e.g. a one-bit flag indicating whether the frame is harmonic or nonharmonic) based on the pitch gain and/or another measure of periodicity or harmonicity as described herein. In one example, a corresponding wideband speech decoder BlOO uses this indication to configure an operation such as weighting factor calculation. In another example, such an indication is used at the encoder and/or decoder in calculating a value for a speech mode parameter.
[000139] It may be desirable for highband excitation generator A302 to generate highband excitation signal S 120 such that the energy of the excitation signal is substantially unaffected by the particular values of weighting factors S 180 and S 190. In such case, weighting factor calculator 550 may be configured to calculate a value for harmonic weighting factor S 180 or for noise weighting factor S 190 (or to receive such a value from storage or another element of highband encoder A200) and to derive a value for the other weighting factor according to an expression such as
(Wharmonicf +(WnoiJ =l, (2)
where Whamonic denotes harmonic weighting factor S 180 and Wnoise denotes noise weighting factor S190. Alternatively, weighting factor calculator 550 may be configured to select, according to a value of a periodicity measure for the current frame or subframe, a corresponding one among a plurality of pairs of weighting factors S 180, S 190, where the pairs are precalculated to satisfy a constant-energy ratio such as expression (2). For an implementation of weighting factor calculator 550 in which expression (2) is observed, typical values for harmonic weighting factor S 180 range from about 0.7 to about 1.0, and typical values for noise weighting factor S 190 range from about 0.1 to about 0.7. Other implementations of weighting factor calculator 550 may be configured to operate according to a version of expression (2) that is modified according to a desired baseline weighting between harmonically extended signal S 160 and modulated noise signal S 170.
[000140] Artifacts may occur in a synthesized speech signal when a sparse codebook (one whose entries are mostly zero values) has been used to calculate the quantized representation of the residual. Codebook sparseness occurs especially when the narrowband signal is encoded at a low bit rate. Artifacts caused by codebook sparseness are typically quasi-periodic in time and occur mostly above 3 kHz. Because the human ear has better time resolution at higher frequencies, these artifacts may be more noticeable in the highband.
[000141] Embodiments include implementations of highband excitation generator A300 that are configured to perform anti-sparseness filtering. FIGURE 18 shows a block diagram of an implementation A312 of highband excitation generator A302 that includes an anti-sparseness filter 600 arranged to filter the dequantized narrowband excitation signal produced by inverse quantizer 450. FIGURE 19 shows a block diagram of an implementation A314 of highband excitation generator A302 that includes an anti-sparseness filter 600 arranged to filter the spectrally extended signal produced by spectrum extender A400. FIGURE 20 shows a block diagram of an implementation A316 of highband excitation generator A302 that includes an anti- sparseness filter 600 arranged to filter the output of combiner 490 to produce highband excitation signal S 120. Of course, implementations of highband excitation generator A300 that combine the features of any of implementations A304 and A306 with the features of any of implementations A312, A314, and A316 are contemplated and hereby expressly disclosed. Anti-sparseness filter 600 may also be arranged within spectrum extender A400: for example, after any of the elements 510, 520, 530, and 540 in spectrum extender A402. It is expressly noted that anti-sparseness filter 600 may also be used with implementations of spectrum extender A400 that perform spectral folding, spectral translation, or harmonic extension.
[000142] Anti-sparseness filter 600 may be configured to alter the phase of its input signal. For example, it may be desirable for anti-sparseness filter 600 to be configured and arranged such that the phase of highband excitation signal S 120 is randomized, or otherwise more evenly distributed, over time. It may also be desirable for the response of anti-sparseness filter 600 to be spectrally flat, such that the magnitude spectrum of the filtered signal is not appreciably changed. In one example, anti-sparseness filter 600 is implemented as an all-pass filter having a transfer function according to the following expression:
H{z) = -°-1 + Z\ 0-6 + Z~6 6 . (3). 1 -0.7Z"4 l + 0.6z-6
One effect of such a filter may be to spread out the energy of the input signal so that it is no longer concentrated in only a few samples.
[000143] Artifacts caused by codebook sparseness are usually more noticeable for noise- like signals, where the residual includes less pitch information, and also for speech in background noise. Sparseness typically causes fewer artifacts in cases where the excitation has long-term structure, and indeed phase modification may cause noisiness in voiced signals. Thus it may be desirable to configure anti-sparseness filter 600 to filter unvoiced signals and to pass at least some voiced signals without alteration. Unvoiced signals are characterized by a low pitch gain (e.g. quantized narrowband adaptive codebook gain) and a spectral tilt (e.g. quantized first reflection coefficient) that is close to zero or positive, indicating a spectral envelope that is flat or tilted upward with increasing frequency. Typical implementations of anti-sparseness filter 600 are configured to filter unvoiced sounds (e.g., as indicated by the value of the spectral tilt), to filter voiced sounds when the pitch gain is below a threshold value (alternatively, not greater than the threshold value), and otherwise to pass the signal without alteration.
[000144] Further implementations of anti-sparseness filter 600 include two or more filters that are configured to have different maximum phase modification angles (e.g., up to 180 degrees). In such case, anti-sparseness filter 600 may be configured to select among these component filters according to a value of the pitch gain (e.g., the quantized adaptive codebook or LTP gain), such that a greater maximum phase modification angle is used for frames having lower pitch gain values. An implementation of anti- sparseness filter 600 may also include different component filters that are configured to modify the phase over more or less of the frequency spectrum, such that a filter configured to modify the phase over a wider frequency range of the input signal is used for frames having lower pitch gain values.
[000145] For accurate reproduction of the encoded speech signal, it may be desirable for the ratio between the levels of the highband and narrowband portions of the synthesized wideband speech signal SlOO to be similar to that in the original wideband speech signal SlO. In addition to a spectral envelope as represented by highband coding parameters S60a, highband encoder A200 may be configured to characterize highband signal S30 by specifying a temporal or gain envelope. As shown in FIGURE 10, highband encoder A202 includes a highband gain factor calculator A230 that is configured and arranged to calculate one or more gain factors according to a relation between highband signal S30 and synthesized highband signal S 130, such as a difference or ratio between the energies of the two signals over a frame or some portion thereof. In other implementations of highband encoder A202, highband gain calculator A230 may be likewise configured but arranged instead to calculate the gain envelope according to such a time-varying relation between highband signal S30 and narrowband excitation signal S80 or highband excitation signal S120.
[000146] The temporal envelopes of narrowband excitation signal S80 and highband signal S30 are likely to be similar. Therefore, encoding a gain envelope that is based on a relation between highband signal S30 and narrowband excitation signal S80 (or a signal derived therefrom, such as highband excitation signal S 120 or synthesized highband signal S 130) will generally be more efficient than encoding a gain envelope based only on highband signal S30. In a typical implementation, highband encoder A202 is configured to output a quantized index of eight to twelve bits that specifies five gain factors for each frame.
[000147] Highband gain factor calculator A230 may be configured to perform gain factor calculation as a task that includes one or more series of subtasks. FIGURE 21 shows a flowchart of an example T200 of such a task that calculates a gain value for a corresponding subframe according to the relative energies of highband signal S30 and synthesized highband signal S 130. Tasks 220a and 220b calculate the energies of the corresponding subframes of the respective signals. For example, tasks 220a and 220b may be configured to calculate the energy as a sum of the squares of the samples of the respective subframe. Task T230 calculates a gain factor for the subframe as the square root of the ratio of those energies. In this example, task T230 calculates the gain factor as the square root of the ratio of the energy of highband signal S30 to the energy of synthesized highband signal S 130 over the subframe.
[000148] It may be desirable for highband gain factor calculator A230 to be configured to calculate the subframe energies according to a windowing function. FIGtORE 22 shows a flowchart of such an implementation T210 of gain factor calculation task T200. Task T215a applies a windowing function to highband signal S30, and task T215b applies the same windowing function to synthesized highband signal S 130. Implementations 222a and 222b of tasks 220a and 220b calculate the energies of the respective windows, and task T230 calculates a gain factor for the subframe as the square root of the ratio of the energies.
[000149] It may be desirable to apply a windowing function that overlaps adjacent subframes. For example, a windowing function that produces gain factors which may be applied in an overlap-add fashion may help to reduce or avoid discontinuity between subframes. In one example, highband gain factor calculator A230 is configured to apply a trapezoidal windowing function as shown in FIGURE 23 a, in which the window overlaps each of the two adjacent subframes by one millisecond. FIGURE 23b shows an application of this windowing function to each of the five subframes of a 20- millisecond frame. Other implementations of highband gain factor calculator A230 may be configured to apply windowing functions having different overlap periods and/or different window shapes (e.g., rectangular, Hamming) that may be symmetrical or asymmetrical. It is also possible for an implementation of highband gain factor calculator A230 to be configured to apply different windowing functions to different subframes within a frame and/or for a frame to include subframes of different lengths.
[000150] Without limitation, the following values are presented as examples for particular implementations. A 20-msec frame is assumed for these cases, although any other duration may be used. For a highband signal sampled at 7 kHz, each frame has 140 samples. If such a frame is divided into five subframes of equal length, each subframe will have 28 samples, and the window as shown in FIGURE 23a will be 42 samples wide. For a highband signal sampled at 8 kHz, each frame has 160 samples. If such frame is divided into five subframes of equal length, each subframe will have 32 samples, and the window as shown in FIGURE 23a will be 48 samples wide. In other implementations, subframes of any width may be used, and it is even possible for an implementation of highband gain calculator A230 to be configured to produce a different gain factor for each sample of a frame.
[00015I]FIGURE 24 shows a block diagram of an implementation B202 of highband decoder B200. Highband decoder B202 includes a highband excitation generator B300 that is configured to produce highband excitation signal S 120 based on narrowband excitation signal S80. Depending on the particular system design choices, highband excitation generator B300 may be implemented according to any of the implementations of highband excitation generator A300 as described herein. Typically it is desirable to implement highband excitation generator B300 to have the same response as the highband excitation generator of the highband encoder of the particular coding system. Because narrowband decoder BIlO will typically perform dequantization of encoded narrowband excitation signal S50, however, in most cases highband excitation generator B300 may be implemented to receive narrowband excitation signal S80 from narrowband decoder BIlO and need not include an inverse quantizer configured to dequantize encoded narrowband excitation signal S50. It is also possible for narrowband decoder BIlO to be implemented to include an instance of anti-sparseness filter 600 arranged to filter the dequantized narrowband excitation signal before it is input to a narrowband synthesis filter such as filter 330.
[000152] Inverse quantizer 560 is configured to dequantize highband filter parameters S60a (in this example, to a set of LSFs), and LSF-to-LP filter coefficient transform 570 is configured to transform the LSFs into a set of filter coefficients (for example, as described above with reference to inverse quantizer 240 and transform 250 of narrowband encoder A122). In other implementations, as mentioned above, different coefficient sets (e.g., cepstral coefficients) and/or coefficient representations (e.g., ISPs) may be used. Highband synthesis filter B200 is configured to produce a synthesized highband signal according to highband excitation signal S 120 and the set of filter coefficients. For a system in which the highband encoder includes a synthesis filter (e.g., as in the example of encoder A202 described above), it may be desirable to implement highband synthesis filter B200 to have the same response (e.g., the same transfer function) as that synthesis filter. [000153] Hi ghband decoder B202 also includes an inverse quantizer 580 configured to dequantize highband gain factors S60b, and a gain control element 590 (e.g., a multiplier or amplifier) configured and arranged to apply the dequantized gain factors to the synthesized highband signal to produce highband signal SlOO. For a case in which the gain envelope of a frame is specified by more than one gain factor, gain control element 590 may include logic configured to apply the gain factors to the respective subframes, possibly according to a windowing function that may be the same or a different windowing function as applied by a gain calculator (e.g., highband gain calculator A230) of the corresponding highband encoder. In other implementations of highband decoder B202, gain control element 590 is similarly configured but is arranged instead to apply the dequantized gain factors to narrowband excitation signal S80 or to highband excitation signal S 120.
[000154] As mentioned above, it may be desirable to obtain the same state in the highband encoder and highband decoder (e.g., by using dequantized values during encoding). Thus it may be desirable in a coding system according to such an implementation to ensure the same state for corresponding noise generators in highband excitation generators A300 and B300. For example, highband excitation generators A300 and B300 of such an implementation may be configured such that the state of the noise generator is a deterministic function of information already coded within the same frame (e.g., narrowband filter parameters S40 or a portion thereof and/or encoded narrowband excitation signal S50 or a portion thereof).
[000155] One or more of the quantizers of the elements described herein (e.g., quantizer 230, 420, or 430) may be configured to perform classified vector quantization. For example, such a quantizer may be configured to select one of a set of codebooks based on information that has already been coded within the same frame in the narrowband channel and/or in the highband channel. Such a technique typically provides increased coding efficiency at the expense of additional codebook storage.
[000156] As discussed above with reference to, e.g., FIGURES 8 and 9, a considerable amount of periodic structure may remain in the residual signal after removal of the coarse spectral envelope from narrowband speech signal S20. For example, the residual signal may contain a sequence of roughly periodic pulses or spikes over time. Such structure, which is typically related to pitch, is especially likely to occur in voiced speech signals. Calculation of a quantized representation of the narrowband residual signal may include encoding of this pitch structure according to a model of long-term periodicity as represented by, for example, one or more codebooks.
[000157] The pitch structure of an actual residual signal may not match the periodicity model exactly. For example, the residual signal may include small jitters in the regularity of the locations of the pitch pulses, such that the distances between successive pitch pulses in a frame are not exactly equal and the structure is not quite regular. These irregularities tend to reduce coding efficiency.
[000158] Some implementations of narrowband encoder A120 are configured to perform a regularization of the pitch structure by applying an adaptive time warping to the residual before or during quantization, or by otherwise including an adaptive time warping in the encoded excitation signal. For example, such an encoder may be configured to select or otherwise calculate a degree of warping in time (e.g., according to one or more perceptual weighting and/or error minimization criteria) such that the resulting excitation signal optimally fits the model of long-term periodicity. Regularization of pitch structure is performed by a subset of CELP encoders called Relaxation Code Excited Linear Prediction (RCELP) encoders.
[000159] An RCELP encoder is typically configured to perform the time warping as an adaptive time shift. This time shift may be a delay ranging from a few milliseconds negative to a few milliseconds positive, and it is usually varied smoothly to avoid audible discontinuities. In some implementations, such an encoder is configured to apply the regularization in a piecewise fashion, wherein each frame or subframe is warped by a corresponding fixed time shift. In other implementations, the encoder is configured to apply the regularization as a continuous warping function, such that a frame or subframe is warped according to a pitch contour (also called a pitch trajectory). In some cases (e.g., as described in U.S. Pat. Appl. Publ. 2004/0098255), the encoder is configured to include a time warping in the encoded excitation signal by applying the shift to a perceptually weighted input signal that is used to calculate the encoded excitation signal.
[000160] The encoder calculates an encoded excitation signal that is regularized and quantized, and the decoder dequantizes the encoded excitation signal to obtain an excitation signal that is used to synthesize the decoded speech signal. The decoded output signal thus exhibits the same varying delay that was included in the encoded excitation signal by the regularization. Typically, no information specifying the regularization amounts is transmitted to the decoder.
[000161] Regularization tends to make the residual signal easier to encode, which improves the coding gain from the long-term predictor and thus boosts overall coding efficiency, generally without generating artifacts. It may be desirable to perform regularization only on frames that are voiced. For example, narrowband encoder A124 may be configured to shift only those frames or subframes having a long-term structure, such as voiced signals. It may even be desirable to perform regularization only on subframes that include pitch pulse energy. Various implementations of RCELP coding are described in U.S. Pats. Nos. 5,704,003 (Kleijn et al.) and 6,879,955 (Rao) and in U.S. Pat. Appl. Publ. 2004/0098255 (Kovesi et al.). Existing implementations of RCELP coders include the Enhanced Variable Rate Codec (EVRC), as described in Telecommunications Industry Association (TIA) IS-127, and the Third Generation Partnership Project 2 (3GPP2) Selectable Mode Vocoder (SMV).
[000162] Unfortunately, regularization may cause problems for a wideband speech coder in which the highband excitation is derived from the encoded narrowband excitation signal (such as a system including wideband speech encoder AlOO and wideband speech decoder BlOO). Due to its derivation from a time-warped signal, the highband excitation signal will generally have a time profile that is different from that of the original highband speech signal. In other words, the highband excitation signal will no longer be synchronous with the original highband speech signal.
[000163] A misalignment in time between the warped highband excitation signal and the original highband speech signal may cause several problems. For example, the warped highband excitation signal may no longer provide a suitable source excitation for a synthesis filter that is configured according to the filter parameters extracted from the original highband speech signal. As a result, the synthesized highband signal may contain audible artifacts that reduce the perceived quality of the decoded wideband speech signal. [000164] The misalignment in time may also cause inefficiencies in gain envelope encoding. As mentioned above, a correlation is likely to exist between the temporal envelopes of narrowband excitation signal S80 and highband signal S30. By encoding the gain envelope of the highband signal according to a relation between these two temporal envelopes, an increase in coding efficiency may be realized as compared to encoding the gain envelope directly. When the encoded narrowband excitation signal is regularized, however, this correlation may be weakened. The misalignment in time between narrowband excitation signal S80 and highband signal S30 may cause fluctuations to appear in highband gain factors S60b, and coding efficiency may drop.
[000165] Embodiments include methods of wideband speech encoding that perform time warping of a highband speech signal according to a time warping included in a corresponding encoded narrowband excitation signal. Potential advantages of such methods include improving the quality of a decoded wideband speech signal and/or improving the efficiency of coding a highband gain envelope.
[000166] FIGURE 25 shows a block diagram of an implementation ADlO of wideband speech encoder AlOO. Encoder ADlO includes an implementation A124 of narrowband encoder A120 that is configured to perform regularization during calculation of the encoded narrowband excitation signal S50. For example, narrowband encoder A124 may be configured according to one or more of the RCELP implementations discussed above.
[000167] Narrowband encoder Al 24 is also configured to output a regularization data signal SDlO that specifies the degree of time warping applied. For various cases in which narrowband encoder A124 is configured to apply a fixed time shift to each frame or subframe, regularization data signal SDlO may include a series of values indicating each time shift amount as an integer or non-integer value in terms of samples, milliseconds, or some other time increment. For a case in which narrowband encoder A124 is configured to otherwise modify the time scale of a frame or other sequence of samples (e.g., by compressing one portion and expanding another portion), regularization information signal SDlO may include a corresponding description of the modification, such as a set of function parameters. In one particular example, narrowband encoder A124 is configured to divide a frame into three subframes and to calculate a fixed time shift for each subframe, such that regularization data signal SDlO indicates three time shift amounts for each regularized frame of the encoded narrowband signal.
[000168] Wideband speech encoder ADlO includes a delay line D 120 configured to advance or retard portions of highband speech signal S30, according to delay amounts indicated by an input signal, to produce time-warped highband speech signal S30a. In the example shown in FIGURE 25, delay line D 120 is configured to time warp highband speech signal S30 according to the warping indicated by regularization data signal SDlO. In such manner, the same amount of time warping that was included in encoded narrowband excitation signal S50 is also applied to the corresponding portion of highband speech signal S30 before analysis. Although this example shows delay line D 120 as a separate element from highband encoder A200, in other implementations delay line D 120 is arranged as part of the highband encoder.
[000169] Further implementations of highband encoder A200 may be configured to perform spectral analysis (e.g., LPC analysis) of the unwarped highband speech signal S30 and to perform time warping of highband speech signal S30 before calculation of highband gain parameters S60b. Such an encoder may include, for example, an implementation of delay line D 120 arranged to perform the time warping. In such cases, however, highband filter parameters S60a based on the analysis of unwarped signal S30 may describe a spectral envelope that is misaligned in time with highband excitation signal S 120.
[000170] Delay line D 120 may be configured according to any combination of logic elements and storage elements suitable for applying the desired time warping operations to highband speech signal S30. For example, delay line D 120 may be configured to read highband speech signal S30 from a buffer according to the desired time shifts. FIGURE 26a shows a schematic diagram of such an implementation D 122 of delay line D120 that includes a shift register SRl. Shift register SRl is a buffer of some length m that is configured to receive and store the m most recent samples of highband speech signal S30. The value m is equal to at least the sum of the maximum positive (or "advance") and negative (or "retard") time shifts to be supported. It may be convenient for the value m to be equal to the length of a frame or subframe of highband signal S30. [000171] Delay line D 122 is configured to output the time-warped highband signal S30a from an offset location OL of shift register SRl. The position of offset location OL varies about a reference position (zero time shift) according to the current time shift as indicated by, for example, regularization data signal SDlO. Delay line D 122 may be configured to support equal advance and retard limits or, alternatively, one limit larger than the other such that a greater shift may be performed in one direction than in the other. FIGURE 26a shows a particular example that supports a larger positive than negative time shift. Delay line D 122 may be configured to output one or more samples at a time (depending on an output bus width, for example).
[000172] A regularization time shift having a magnitude of more than a few milliseconds may cause audible artifacts in the decoded signal. Typically the magnitude of a regularization time shift as performed by a narrowband encoder A124 will not exceed a few milliseconds, such that the time shifts indicated by regularization data signal SDlO will be limited. However, it may be desired in such cases for delay line D 122 to be configured to impose a maximum limit on time shifts in the positive and/or negative direction (for example, to observe a tighter limit than that imposed by the narrowband encoder).
[000173] FIGURE 26b shows a schematic diagram of an implementation D 124 of delay line D122 that includes a shift window SW. In this example, the position of offset location OL is limited by the shift window SW. Although FIGURE 26b shows a case in which the buffer length m is greater than the width of shift window SW, delay line D 124 may also be implemented such that the width of shift window SW is equal to m.
[000174] In other implementations, delay line D 120 is configured to write highband speech signal S30 to a buffer according to the desired time shifts. FIGURE 27 shows a schematic diagram of such an implementation D 130 of delay line D 120 that includes two shift registers SR2 and SR3 configured to receive and store highband speech signal S30. Delay line D 130 is configured to write a frame or subframe from shift register SR2 to shift register SR3 according to a time shift as indicated by, for example, regularization data signal SDlO. Shift register SR3 is configured as a FIFO buffer arranged to output time-warped highband signal S30. [000175] In the particular example shown in FIGURE 27, shift register SR2 includes a frame buffer portion FBI and a delay buffer portion DB, and shift register SR3 includes a frame buffer portion FB2, an advance buffer portion AB, and a retard buffer portion RB. The lengths of advance buffer AB and retard buffer RB may be equal, or one may be larger than the other, such that a greater shift in one direction is supported than in the other. Delay buffer DB and retard buffer portion RB may be configured to have the same length. Alternatively, delay buffer DB may be shorter than retard buffer RB to account for a time interval required to transfer samples from frame buffer FB 1 to shift register SR3, which may include other processing operations such as warping of the samples before storage to shift register SR3.
[000176] In the example of FIGURE 27, frame buffer FBI is configured to have a length equal to that of one frame of highband signal S30. In another example, frame buffer FBI is configured to have a length equal to that of one subframe of highband signal S30. In such case, delay line D 130 may be configured to include logic to apply the same (e.g., an average) delay to all subframes of a frame to be shifted. Delay line D130 may also include logic to average values from frame buffer FBI with values to be overwritten in retard buffer RB or advance buffer AB. In a further example, shift register SR3 may be configured to receive values of highband signal S30 only via frame buffer FBI, and in such case delay line D 130 may include logic to interpolate across gaps between successive frames or subframes written to shift register SR3. In other implementations, delay line D130 may be configured to perform a warping operation on samples from frame buffer FBI before writing them to shift register SR3 (e.g., according to a function described by regularization data signal SDlO).
[000177] It may be desirable for delay line D 120 to apply a time warping that is based on, but is not identical to, the warping specified by regularization data signal SDlO. FIGURE 28 shows a block diagram of an implementation AD12 of wideband speech encoder ADlO that includes a delay value mapper Dl 10. Delay value mapper Dl 10 is configured to map the warping indicated by regularization data signal SDlO into mapped delay values SDlOa. Delay line D 120 is arranged to produce time-warped highband speech signal S30a according to the warping indicated by mapped delay values SDlOa. [000178] The time shift applied by the narrowband encoder may be expected to evolve smoothly over time. Therefore, it is typically sufficient to compute the average narrowband time shift applied to the subframes during a frame of speech, and to shift a corresponding frame of highband speech signal S30 according to this average. In one such example, delay value mapper Dl 10 is configured to calculate an average of the subframe delay values for each frame, and delay line D120 is configured to apply the calculated average to a corresponding frame of highband signal S30. In other examples, an average over a shorter period (such as two subframes, or half of a frame) or a longer period (such as two frames) may be calculated and applied. In a case where the average is a non-integer value of samples, delay value mapper DIlO may be configured to round the value to an integer number of samples before outputting it to delay line D 120.
[000179] Narrowband encoder A124 may be configured to include a regularization time shift of a non-integer number of samples in the encoded narrowband excitation signal. In such a case, it may be desirable for delay value mapper DIlO to be configured to round the narrowband time shift to an integer number of samples and for delay line D120 to apply the rounded time shift to highband speech signal S30.
[00018O]In some implementations of wideband speech encoder ADlO, the sampling rates of narrowband speech signal S20 and highband speech signal S30 may differ. In such cases, delay value mapper DIlO may be configured to adjust time shift amounts indicated in regularization data signal SDlO to account for a difference between the sampling rates of narrowband speech signal S20 (or narrowband excitation signal S80) and highband speech signal S30. For example, delay value mapper Dl 10 may be configured to scale the time shift amounts according to a ratio of the sampling rates. In one particular example as mentioned above, narrowband speech signal S20 is sampled at 8 kHz, and highband speech signal S30 is sampled at 7 kHz. In this case, delay value mapper Dl 10 is configured to multiply each shift amount by 7/8. Implementations of delay value mapper DIlO may also be configured to perform such a scaling operation together with an integer-rounding and/or a time shift averaging operation as described herein.
[00018I]In further implementations, delay line D 120 is configured to otherwise modify the time scale of a frame or other sequence of samples (e.g., by compressing one portion and expanding another portion). For example, narrowband encoder A124 may be configured to perform the regularization according to a function such as a pitch contour or trajectory. In such case, regularization data signal SDlO may include a corresponding description of the function, such as a set of parameters, and delay line D 120 may include logic configured to warp frames or subframes of highband speech signal S30 according to the function. In other implementations, delay value mapper Dl 10 is configured to average, scale, and/or round the function before it is applied to highband speech signal S30 by delay line D 120. For example, delay value mapper DIlO may be configured to calculate one or more delay values according to the function, each delay value indicating a number of samples, which are then applied by delay line D 120 to time warp one or more corresponding frames or subframes of highband speech signal S30.
[000182]FIGURE 29 shows a flowchart for a method MDlOO of time warping a highband speech signal according to a time warping included in a corresponding encoded narrowband excitation signal. Task TDlOO processes a wideband speech signal to obtain a narrowband speech signal and a highband speech signal. For example, task TDlOO may be configured to filter the wideband speech signal using a filter bank having lowpass and highpass filters, such as an implementation of filter bank AIlO. Task TD200 encodes the narrowband speech signal into at least a encoded narrowband excitation signal and a plurality of narrowband filter parameters. The encoded narrowband excitation signal and/or filter parameters may be quantized, and the encoded narrowband speech signal may also include other parameters such as a speech mode parameter. Task TD200 also includes a time warping in the encoded narrowband excitation signal.
[000183] Task TD300 generates a highband excitation signal based on a narrowband excitation signal. In this case, the narrowband excitation signal is based on the encoded narrowband excitation signal. According to at least the highband excitation signal, task TD400 encodes the highband speech signal into at least a plurality of highband filter parameters. For example, task TD400 may be configured to encode the highband speech signal into a plurality of quantized LSFs. Task TD500 applies a time shift to the highband speech signal that is based on information relating to a time warping included in the encoded narrowband excitation signal. [000184] Task TD400 may be configured to perform a spectral analysis (such as an LPC analysis) on the highband speech signal, and/or to calculate a gain envelope of the highband speech signal. In such cases, task TD500 may be configured to apply the time shift to the highband speech signal prior to the analysis and/or the gain envelope calculation.
[000185] Other implementations of wideband speech encoder AlOO are configured to reverse a time warping of highband excitation signal S 120 caused by a time warping included in the encoded narrowband excitation signal. For example, highband excitation generator A300 may be implemented to include an implementation of delay line D120 that is configured to receive regularization data signal SDlO or mapped delay values SDlOa, and to apply a corresponding reverse time shift to narrowband excitation signal S 80, and/or to a subsequent signal based on it such as harmonically extended signal S 160 or highband excitation signal S 120.
[000186] Further wideband speech encoder implementations may be configured to encode narrowband speech signal S20 and highband speech signal S30 independently from one another, such that highband speech signal S30 is encoded as a representation of a highband spectral envelope and a highband excitation signal. Such an implementation may be configured to perform time warping of the highband residual signal, or to otherwise include a time warping in an encoded highband excitation signal, according to information relating to a time warping included in the encoded narrowband excitation signal. For example, the highband encoder may include an implementation of delay line D120 and/or delay value mapper DIlO as described herein that are configured to apply a time warping to the highband residual signal. Potential advantages of such an operation include more efficient encoding of the highband residual signal and a better match between the synthesized narrowband and highband speech signals.
[000187] As mentioned above, embodiments as described herein include implementations that may be used to perform embedded coding, supporting compatibility with narrowband systems and avoiding a need for transcoding. Support for highband coding may also serve to differentiate on a cost basis between chips, chipsets, devices, and/or networks having wideband support with backward compatibility, and those having narrowband support only. Support for highband coding as described herein may also be used in conjunction with a technique for supporting lowband coding, and a system, method, or apparatus according to such an embodiment may support coding of frequency components from, for example, about 50 or 100 Hz up to about 7 or 8 kHz.
[000188] As mentioned above, adding highband support to a speech coder may improve intelligibility, especially regarding differentiation of fricatives. Although such differentiation may usually be derived by a human listener from the particular context, highband support may serve as an enabling feature in speech recognition and other machine interpretation applications, such as systems for automated voice menu navigation and/or automatic call processing.
[000189] An apparatus according to an embodiment may be embedded into a portable device for wireless communications such as a cellular telephone or personal digital assistant (PDA). Alternatively, such an apparatus may be included in another communications device such as a VoIP handset, a personal computer configured to support VoIP communications, or a network device configured to route telephonic or VoIP communications. For example, an apparatus according to an embodiment may be implemented in a chip or chipset for a communications device. Depending upon the particular application, such a device may also include such features as analog-to-digital and/or digital-to-analog conversion of a speech signal, circuitry for performing amplification and/or other signal processing operations on a speech signal, and/or radio- frequency circuitry for transmission and/or reception of the coded speech signal.
[00019O]It is explicitly contemplated and disclosed that embodiments may include and/or be used with any one or more of the other features disclosed in the U.S. Provisional Pat. Appls. Nos. 60/667,901 and 60/673,965 of which this application claims benefit. Such features include removal of high-energy bursts of short duration that occur in the highband and are substantially absent from the narrowband. Such features include fixed or adaptive smoothing of coefficient representations such as highband LSFs. Such features include fixed or adaptive shaping of noise associated with quantization of coefficient representations such as LSFs. Such features also include fixed or adaptive smoothing of a gain envelope, and adaptive attenuation of a gain envelope. [000191] The foregoing presentation of the described embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments are possible, and the generic principles presented herein may be applied to other embodiments as well. For example, an embodiment may be implemented in part or in whole as a hard-wired circuit, as a circuit configuration fabricated into an application-specific integrated circuit, or as a firmware program loaded into non- volatile storage or a software program loaded from or into a data storage medium as machine-readable code, such code being instructions executable by an array of logic elements such as a microprocessor or other digital signal processing unit. The data storage medium may be an array of storage elements such as semiconductor memory (which may include without limitation dynamic or static RAM (random-access memory), ROM (read-only memory), and/or flash RAM), or ferroelectric, magnetoresistive, ovonic, polymeric, or phase-change memory; or a disk medium such as a magnetic or optical disk. The term "software" should be understood to include source code, assembly language code, machine code, binary code, firmware, macrocode, microcode, any one or more sets or sequences of instructions executable by an array of logic elements, and any combination of such examples.
[000192] The various elements of implementations of highband excitation generators A300 and B300, highband encoder AlOO, highband decoder B200, wideband speech encoder AlOO, and wideband speech decoder BlOO may be implemented as electronic and/or optical devices residing, for example, on the same chip or among two or more chips in a chipset, although other arrangements without such limitation are also contemplated. One or more elements of such an apparatus may be implemented in whole or in part as one or more sets of instructions arranged to execute on one or more fixed or programmable arrays of logic elements (e.g., transistors, gates) such as microprocessors, embedded processors, IP cores, digital signal processors, FPGAs (field-programmable gate arrays), ASSPs (application-specific standard products), and ASICs (application-specific integrated circuits). It is also possible for one or more such elements to have structure in common (e.g., a processor used to execute portions of code corresponding to different elements at different times, a set of instructions executed to perform tasks corresponding to different elements at different times, or an arrangement of electronic and/or optical devices performing operations for different elements at different times). Moreover, it is possible for one or more such elements to be used to perform tasks or execute other sets of instructions that are not directly related to an operation of the apparatus, such as a task relating to another operation of a device or system in which the apparatus is embedded.
[000193] FIGURE 30 shows a flowchart of a method MlOO, according to an embodiment, of encoding a highband portion of a speech signal having a narrowband portion and the highband portion. Task XlOO calculates a set of filter parameters that characterize a spectral envelope of the highband portion. Task X200 calculates a spectrally extended signal by applying a nonlinear function to a signal derived from the narrowband portion. Task X300 generates a synthesized highband signal according to (A) the set of filter parameters and (B) a highband excitation signal based on the spectrally extended signal. Task X400 calculates a gain envelope based on a relation between (C) energy of the highband portion and (D) energy of a signal derived from the narrowband portion.
[000194] FIGURE 31a shows a flowchart of a method M200 of generating a highband excitation signal according to an embodiment. Task YlOO calculates a harmonically extended signal by applying a nonlinear function to a narrowband excitation signal , derived from a narrowband portion of a speech signal. Task Y200 mixes the harmonically extended signal with a modulated noise signal to generate a highband excitation signal. FIGURE 31b shows a flowchart of a method M210 of generating a highband excitation signal according to another embodiment including tasks Y300 and Y400. Task Y300 calculates a time-domain envelope according to energy over time of one among the narrowband excitation signal and the harmonically extended signal. Task Y400 modulates a noise signal according to the time-domain envelope to produce the modulated noise signal.
[000195] FIGURE 32 shows a flowchart of a method M300 according to an embodiment, of decoding a highband portion of a speech signal having a narrowband portion and the highband portion. Task ZlOO receives a set of filter parameters that characterize a spectral envelope of the highband portion and a set of gain factors that characterize a temporal envelope of the highband portion. Task Z200 calculates a spectrally extended signal by applying a nonlinear function to a signal derived from the narrowband portion. Task Z300 generates a synthesized highband signal according to (A) the set of filter parameters and (B) a highband excitation signal based on the spectrally extended signal. Task Z400 modulates a gain envelope of the synthesized highband signal based on the set of gain factors. For example, task Z400 may be configured to modulate the gain envelope of the synthesized highband signal by applying the set of gain factors to an excitation signal derived from the narrowband portion, to the spectrally extended signal, to the highband excitation signal, or to the synthesized highband signal.
[000196] Embodiments also include additional methods of speech coding, encoding, and decoding as are expressly disclosed herein, e.g., by descriptions of structural embodiments configured to perform such methods. Each of these methods may also be tangibly embodied (for example, in one or more data storage media as listed above) as one or more sets of instructions readable and/or executable by a machine including an array of logic elements (e.g., a processor, microprocessor, microcontroller, or other finite state machine). Thus, the present invention is not intended to be limited to the embodiments shown above but rather is to be accorded the widest scope consistent with the principles and novel features disclosed in any fashion herein, including in the attached claims as filed, which form a part of the original disclosure.

Claims

WHAT IS CLAIMED IS:
1. A method of signal processing, said method comprising:
encoding a low-frequency portion of a speech signal into at least an encoded lowband excitation signal and a plurality of lowband filter parameters;
generating a highband excitation signal based on the encoded lowband excitation signal; and
according to at least the highband excitation signal, encoding a high-frequency portion of the speech signal into at least a plurality of highband filter parameters,
wherein the encoded lowband excitation signal describes a signal that is warped in time, with respect to the speech signal, according to a time-varying time warping, and
wherein said method comprises, based on information relating to the time warping, applying a plurality of different time shifts to a corresponding plurality of successive portions in time of the high-frequency portion.
2. The method of signal processing according to claim 1, wherein the encoded excitation signal describes a signal that is warped in time according to a model of a pitch structure of the low-frequency portion.
3. The method of signal processing according to claim 2, wherein said encoding a low-frequency portion includes applying a time shift to a narrowband residual according to a model of the pitch structure of the narrowband residual,
wherein the encoded narrowband excitation signal is based on the time-shifted narrowband residual.
4. The method of signal processing according to claim 3, wherein said applying a time shift to a narrowband residual includes applying different respective time shifts to each of at least two consecutive subframes of the narrowband residual, and
wherein said applying a time shift to the high-frequency portion includes applying, to a frame of the high-frequency portion, a time shift based on an average of the respective time shifts.
5. The method of signal processing according to claim 3, wherein said applying a plurality of different time shifts comprises receiving a value indicating a time shift applied to the narrowband residual, and rounding the received value to an integer value.
6. The method of signal processing according to claim 1, wherein said applying a plurality of different time shifts is performed prior to said encoding the high- frequency portion.
7. The method of signal processing according to claim 1, wherein said encoding the high-frequency portion into at least a plurality of highband filter parameters includes encoding the high-frequency portion into at least a plurality of linear prediction filter coefficients.
8. The method of signal processing according to claim 1, wherein said encoding the high-frequency portion into at least a plurality of highband filter parameters includes encoding a gain envelope of the high-frequency portion, and
wherein said applying a plurality of different time shifts is performed prior to said encoding a gain envelope.
9. The method of signal processing according to claim 1, wherein said applying a plurality of different time shifts comprises calculating at least one of the plurality of different time shifts according to a ratio between sampling rates of the low- frequency portion and the high-frequency portion.
10. A data storage medium having machine-executable instructions describing the method of signal processing according to claim 1.
11. An apparatus comprising:
a lowband speech encoder configured to encode a low-frequency portion of a speech signal into at least an encoded lowband excitation signal and a plurality of lowband filter parameters; and
a highband speech encoder configured to generate a highband excitation signal based on the encoded lowband excitation signal;
wherein the highband encoder is configured to encode a high-frequency portion of the speech signal into at least a plurality of highband filter parameters according to at least the highband excitation signal, and
wherein said narrowband speech encoder is configured to output a regularization data signal describing a time- varying time warping, with respect to the speech signal, that is included in the encoded narrowband excitation signal, and
wherein said apparatus comprises a delay line configured to apply a plurality of different time shifts to a corresponding plurality of successive portions in time of the high-frequency portion, wherein the plurality of different time shifts are based on the regularization data signal.
12. The apparatus according to claim 11 , wherein the encoded excitation signal describes a signal that is warped in time according to a model of a pitch structure of the low-frequency portion.
13. The apparatus according to claim 11, wherein said narrowband speech encoder is configured to apply a time shift to a narrowband residual according to a model of the pitch structure of the narrowband residual and to produce the encoded narrowband excitation signal based on the time-shifted narrowband residual.
14. The apparatus according to claim 12, wherein said narrowband speech encoder is configured to apply a different respective time shift to each of at least two consecutive subframes of the narrowband residual, and
wherein said delay line is configured to apply, to a frame of the high-frequency portion, a time shift based on an average of the respective time shifts.
15. The apparatus according to claim 12, said apparatus comprising a delay value mapper configured to receive a value of a time shift of the narrowband residual and to round the received value to an integer value.
16. The apparatus according to claim 11, wherein said highband speech encoder is arranged to encode the high-frequency portion as produced by said delay line.
17. The apparatus according to claim 11, wherein said highband speech encoder is configured to encode the high-frequency portion into at least a plurality of linear prediction filter coefficients.
18. The apparatus according to claim 11, wherein said highband speech encoder is arranged to encode a gain envelope of the high-frequency portion as produced by said delay line.
19. The apparatus according to claim 11, said apparatus comprising a delay value mapper configured to calculate at least one of the plurality of different time shifts according to a ratio between sampling rates of the low-frequency portion and the high- frequency portion.
20. The apparatus according to claim 11, said apparatus comprising a cellular telephone.
21. An apparatus comprising:
means for encoding a low-frequency portion of a speech signal into at least an encoded lowband excitation signal and a plurality of lowband filter parameters;
means for generating a highband excitation signal based on the encoded lowband excitation signal; and
means for encoding a high-frequency portion of the speech signal into at least a plurality of highband filter parameters according to at least the highband excitation signal,
wherein the encoded narrowband excitation signal describes a signal that is warped in time, with respect to the speech signal, according to a time-varying time warping, and
wherein said apparatus comprises means for applying, based on information relating to the time warping, a plurality of different time shifts to a corresponding plurality of successive portions in time of the high-frequency portion.
22. The apparatus according to claim 21, said apparatus comprising a cellular telephone.
PCT/US2006/012232 2005-04-01 2006-04-03 Systems, methods, and apparatus for highband time warping WO2006107838A1 (en)

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BRPI0607691A BRPI0607691B1 (en) 2005-04-01 2006-04-03 method and equipment for broadband speech coding
JP2008504479A JP5203930B2 (en) 2005-04-01 2006-04-03 System, method and apparatus for performing high-bandwidth time axis expansion and contraction
MX2007012187A MX2007012187A (en) 2005-04-01 2006-04-03 Systems, methods, and apparatus for highband time warping.
CA2603231A CA2603231C (en) 2005-04-01 2006-04-03 Systems, methods, and apparatus for highband time warping
AU2006232362A AU2006232362B2 (en) 2005-04-01 2006-04-03 Systems, methods, and apparatus for highband time warping
EP06740356A EP1864283B1 (en) 2005-04-01 2006-04-03 Systems, methods, and apparatus for highband time warping
CN200680018212.6A CN101185126B (en) 2005-04-01 2006-04-03 Systems, methods, and apparatus for highband time warping
IL186405A IL186405A (en) 2005-04-01 2007-10-07 Systems, methods and apparatus for highband time warping
NO20075512A NO20075512L (en) 2005-04-01 2007-10-31 Hoyband's time wasting

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Cited By (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2011511311A (en) * 2008-01-31 2011-04-07 フラウンホーファー−ゲゼルシャフト・ツール・フェルデルング・デル・アンゲヴァンテン・フォルシュング・アインゲトラーゲネル・フェライン Apparatus and method for bandwidth extension of audio signal
JP2012527637A (en) * 2009-05-19 2012-11-08 エレクトロニクス アンド テレコミュニケーションズ リサーチ インスチチュート Audio signal encoding and decoding method and apparatus using hierarchical sinusoidal pulse coding
CN107527629A (en) * 2013-07-12 2017-12-29 皇家飞利浦有限公司 For carrying out the optimization zoom factor of bandspreading in audio signal decoder
US10096322B2 (en) 2013-06-21 2018-10-09 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E.V. Audio decoder having a bandwidth extension module with an energy adjusting module
US10354663B2 (en) 2014-07-28 2019-07-16 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Apparatus and method for generating an enhanced signal using independent noise-filling

Families Citing this family (320)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7987095B2 (en) * 2002-09-27 2011-07-26 Broadcom Corporation Method and system for dual mode subband acoustic echo canceller with integrated noise suppression
US7619995B1 (en) * 2003-07-18 2009-11-17 Nortel Networks Limited Transcoders and mixers for voice-over-IP conferencing
JP4679049B2 (en) 2003-09-30 2011-04-27 パナソニック株式会社 Scalable decoding device
US7668712B2 (en) 2004-03-31 2010-02-23 Microsoft Corporation Audio encoding and decoding with intra frames and adaptive forward error correction
WO2005111568A1 (en) * 2004-05-14 2005-11-24 Matsushita Electric Industrial Co., Ltd. Encoding device, decoding device, and method thereof
US8725501B2 (en) * 2004-07-20 2014-05-13 Panasonic Corporation Audio decoding device and compensation frame generation method
ATE488838T1 (en) * 2004-08-30 2010-12-15 Qualcomm Inc METHOD AND APPARATUS FOR AN ADAPTIVE DEJITTER BUFFER
US8085678B2 (en) * 2004-10-13 2011-12-27 Qualcomm Incorporated Media (voice) playback (de-jitter) buffer adjustments based on air interface
US8155965B2 (en) * 2005-03-11 2012-04-10 Qualcomm Incorporated Time warping frames inside the vocoder by modifying the residual
US8355907B2 (en) * 2005-03-11 2013-01-15 Qualcomm Incorporated Method and apparatus for phase matching frames in vocoders
EP1872364B1 (en) * 2005-03-30 2010-11-24 Nokia Corporation Source coding and/or decoding
BRPI0607646B1 (en) * 2005-04-01 2021-05-25 Qualcomm Incorporated METHOD AND EQUIPMENT FOR SPEECH BAND DIVISION ENCODING
PL1875463T3 (en) * 2005-04-22 2019-03-29 Qualcomm Incorporated Systems, methods, and apparatus for gain factor smoothing
WO2006114368A1 (en) * 2005-04-28 2006-11-02 Siemens Aktiengesellschaft Noise suppression process and device
US7177804B2 (en) * 2005-05-31 2007-02-13 Microsoft Corporation Sub-band voice codec with multi-stage codebooks and redundant coding
US7707034B2 (en) * 2005-05-31 2010-04-27 Microsoft Corporation Audio codec post-filter
US7831421B2 (en) * 2005-05-31 2010-11-09 Microsoft Corporation Robust decoder
DE102005032724B4 (en) * 2005-07-13 2009-10-08 Siemens Ag Method and device for artificially expanding the bandwidth of speech signals
ATE443318T1 (en) * 2005-07-14 2009-10-15 Koninkl Philips Electronics Nv AUDIO SIGNAL SYNTHESIS
WO2007013973A2 (en) * 2005-07-20 2007-02-01 Shattil, Steve Systems and method for high data rate ultra wideband communication
KR101171098B1 (en) * 2005-07-22 2012-08-20 삼성전자주식회사 Scalable speech coding/decoding methods and apparatus using mixed structure
US7734462B2 (en) * 2005-09-02 2010-06-08 Nortel Networks Limited Method and apparatus for extending the bandwidth of a speech signal
US8326614B2 (en) * 2005-09-02 2012-12-04 Qnx Software Systems Limited Speech enhancement system
CN101273404B (en) * 2005-09-30 2012-07-04 松下电器产业株式会社 Audio encoding device and audio encoding method
WO2007043643A1 (en) * 2005-10-14 2007-04-19 Matsushita Electric Industrial Co., Ltd. Audio encoding device, audio decoding device, audio encoding method, and audio decoding method
KR20080047443A (en) * 2005-10-14 2008-05-28 마츠시타 덴끼 산교 가부시키가이샤 Transform coder and transform coding method
JP4876574B2 (en) * 2005-12-26 2012-02-15 ソニー株式会社 Signal encoding apparatus and method, signal decoding apparatus and method, program, and recording medium
EP1852848A1 (en) * 2006-05-05 2007-11-07 Deutsche Thomson-Brandt GmbH Method and apparatus for lossless encoding of a source signal using a lossy encoded data stream and a lossless extension data stream
US8949120B1 (en) 2006-05-25 2015-02-03 Audience, Inc. Adaptive noise cancelation
US8725499B2 (en) * 2006-07-31 2014-05-13 Qualcomm Incorporated Systems, methods, and apparatus for signal change detection
US8260609B2 (en) 2006-07-31 2012-09-04 Qualcomm Incorporated Systems, methods, and apparatus for wideband encoding and decoding of inactive frames
US7987089B2 (en) * 2006-07-31 2011-07-26 Qualcomm Incorporated Systems and methods for modifying a zero pad region of a windowed frame of an audio signal
US8532984B2 (en) 2006-07-31 2013-09-10 Qualcomm Incorporated Systems, methods, and apparatus for wideband encoding and decoding of active frames
US8135047B2 (en) 2006-07-31 2012-03-13 Qualcomm Incorporated Systems and methods for including an identifier with a packet associated with a speech signal
DE602007012116D1 (en) 2006-08-15 2011-03-03 Dolby Lab Licensing Corp ARBITRARY FORMATION OF A TEMPORARY NOISE CURVE WITHOUT SIDE INFORMATION
KR101040160B1 (en) * 2006-08-15 2011-06-09 브로드콤 코포레이션 Constrained and controlled decoding after packet loss
US8239190B2 (en) * 2006-08-22 2012-08-07 Qualcomm Incorporated Time-warping frames of wideband vocoder
US8046218B2 (en) * 2006-09-19 2011-10-25 The Board Of Trustees Of The University Of Illinois Speech and method for identifying perceptual features
JP4972742B2 (en) * 2006-10-17 2012-07-11 国立大学法人九州工業大学 High-frequency signal interpolation method and high-frequency signal interpolation device
USRE50132E1 (en) 2006-10-25 2024-09-17 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E.V. Apparatus and method for generating audio subband values and apparatus and method for generating time-domain audio samples
USRE50158E1 (en) 2006-10-25 2024-10-01 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E.V. Apparatus and method for generating audio subband values and apparatus and method for generating time-domain audio samples
KR101375582B1 (en) * 2006-11-17 2014-03-20 삼성전자주식회사 Method and apparatus for bandwidth extension encoding and decoding
US8639500B2 (en) * 2006-11-17 2014-01-28 Samsung Electronics Co., Ltd. Method, medium, and apparatus with bandwidth extension encoding and/or decoding
KR101565919B1 (en) * 2006-11-17 2015-11-05 삼성전자주식회사 Method and apparatus for encoding and decoding high frequency signal
US8005671B2 (en) * 2006-12-04 2011-08-23 Qualcomm Incorporated Systems and methods for dynamic normalization to reduce loss in precision for low-level signals
GB2444757B (en) * 2006-12-13 2009-04-22 Motorola Inc Code excited linear prediction speech coding
US20080147389A1 (en) * 2006-12-15 2008-06-19 Motorola, Inc. Method and Apparatus for Robust Speech Activity Detection
FR2911031B1 (en) * 2006-12-28 2009-04-10 Actimagine Soc Par Actions Sim AUDIO CODING METHOD AND DEVICE
FR2911020B1 (en) * 2006-12-28 2009-05-01 Actimagine Soc Par Actions Sim AUDIO CODING METHOD AND DEVICE
KR101379263B1 (en) * 2007-01-12 2014-03-28 삼성전자주식회사 Method and apparatus for decoding bandwidth extension
US7873064B1 (en) 2007-02-12 2011-01-18 Marvell International Ltd. Adaptive jitter buffer-packet loss concealment
US8032359B2 (en) * 2007-02-14 2011-10-04 Mindspeed Technologies, Inc. Embedded silence and background noise compression
GB0704622D0 (en) * 2007-03-09 2007-04-18 Skype Ltd Speech coding system and method
KR101411900B1 (en) * 2007-05-08 2014-06-26 삼성전자주식회사 Method and apparatus for encoding and decoding audio signal
US9653088B2 (en) * 2007-06-13 2017-05-16 Qualcomm Incorporated Systems, methods, and apparatus for signal encoding using pitch-regularizing and non-pitch-regularizing coding
PT2186089T (en) * 2007-08-27 2019-01-10 Ericsson Telefon Ab L M Method and device for perceptual spectral decoding of an audio signal including filling of spectral holes
FR2920545B1 (en) * 2007-09-03 2011-06-10 Univ Sud Toulon Var METHOD FOR THE MULTIPLE CHARACTEROGRAPHY OF CETACEANS BY PASSIVE ACOUSTICS
EP2207166B1 (en) * 2007-11-02 2013-06-19 Huawei Technologies Co., Ltd. An audio decoding method and device
US20100250260A1 (en) * 2007-11-06 2010-09-30 Lasse Laaksonen Encoder
WO2009059633A1 (en) * 2007-11-06 2009-05-14 Nokia Corporation An encoder
CN101896968A (en) * 2007-11-06 2010-11-24 诺基亚公司 Audio coding apparatus and method thereof
KR101444099B1 (en) * 2007-11-13 2014-09-26 삼성전자주식회사 Method and apparatus for detecting voice activity
WO2009066959A1 (en) * 2007-11-21 2009-05-28 Lg Electronics Inc. A method and an apparatus for processing a signal
US8050934B2 (en) * 2007-11-29 2011-11-01 Texas Instruments Incorporated Local pitch control based on seamless time scale modification and synchronized sampling rate conversion
US8688441B2 (en) * 2007-11-29 2014-04-01 Motorola Mobility Llc Method and apparatus to facilitate provision and use of an energy value to determine a spectral envelope shape for out-of-signal bandwidth content
TWI356399B (en) * 2007-12-14 2012-01-11 Ind Tech Res Inst Speech recognition system and method with cepstral
KR101439205B1 (en) * 2007-12-21 2014-09-11 삼성전자주식회사 Method and apparatus for audio matrix encoding/decoding
WO2009084221A1 (en) * 2007-12-27 2009-07-09 Panasonic Corporation Encoding device, decoding device, and method thereof
KR101413967B1 (en) * 2008-01-29 2014-07-01 삼성전자주식회사 Encoding method and decoding method of audio signal, and recording medium thereof, encoding apparatus and decoding apparatus of audio signal
KR101413968B1 (en) * 2008-01-29 2014-07-01 삼성전자주식회사 Method and apparatus for encoding audio signal, and method and apparatus for decoding audio signal
US8433582B2 (en) * 2008-02-01 2013-04-30 Motorola Mobility Llc Method and apparatus for estimating high-band energy in a bandwidth extension system
US20090201983A1 (en) * 2008-02-07 2009-08-13 Motorola, Inc. Method and apparatus for estimating high-band energy in a bandwidth extension system
US8326641B2 (en) * 2008-03-20 2012-12-04 Samsung Electronics Co., Ltd. Apparatus and method for encoding and decoding using bandwidth extension in portable terminal
US8983832B2 (en) * 2008-07-03 2015-03-17 The Board Of Trustees Of The University Of Illinois Systems and methods for identifying speech sound features
CA2729665C (en) * 2008-07-10 2016-11-22 Voiceage Corporation Variable bit rate lpc filter quantizing and inverse quantizing device and method
MY154452A (en) * 2008-07-11 2015-06-15 Fraunhofer Ges Forschung An apparatus and a method for decoding an encoded audio signal
EP2176862B1 (en) 2008-07-11 2011-08-31 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Apparatus and method for calculating bandwidth extension data using a spectral tilt controlling framing
CN103000178B (en) * 2008-07-11 2015-04-08 弗劳恩霍夫应用研究促进协会 Time warp activation signal provider and audio signal encoder employing the time warp activation signal
KR101614160B1 (en) 2008-07-16 2016-04-20 한국전자통신연구원 Apparatus for encoding and decoding multi-object audio supporting post downmix signal
US20110178799A1 (en) * 2008-07-25 2011-07-21 The Board Of Trustees Of The University Of Illinois Methods and systems for identifying speech sounds using multi-dimensional analysis
US8463412B2 (en) * 2008-08-21 2013-06-11 Motorola Mobility Llc Method and apparatus to facilitate determining signal bounding frequencies
US8532998B2 (en) 2008-09-06 2013-09-10 Huawei Technologies Co., Ltd. Selective bandwidth extension for encoding/decoding audio/speech signal
US8352279B2 (en) * 2008-09-06 2013-01-08 Huawei Technologies Co., Ltd. Efficient temporal envelope coding approach by prediction between low band signal and high band signal
WO2010028292A1 (en) * 2008-09-06 2010-03-11 Huawei Technologies Co., Ltd. Adaptive frequency prediction
US8515747B2 (en) * 2008-09-06 2013-08-20 Huawei Technologies Co., Ltd. Spectrum harmonic/noise sharpness control
US8407046B2 (en) * 2008-09-06 2013-03-26 Huawei Technologies Co., Ltd. Noise-feedback for spectral envelope quantization
KR101178801B1 (en) * 2008-12-09 2012-08-31 한국전자통신연구원 Apparatus and method for speech recognition by using source separation and source identification
US20100070550A1 (en) * 2008-09-12 2010-03-18 Cardinal Health 209 Inc. Method and apparatus of a sensor amplifier configured for use in medical applications
WO2010031003A1 (en) * 2008-09-15 2010-03-18 Huawei Technologies Co., Ltd. Adding second enhancement layer to celp based core layer
US8577673B2 (en) * 2008-09-15 2013-11-05 Huawei Technologies Co., Ltd. CELP post-processing for music signals
WO2010036061A2 (en) * 2008-09-25 2010-04-01 Lg Electronics Inc. An apparatus for processing an audio signal and method thereof
US8364471B2 (en) * 2008-11-04 2013-01-29 Lg Electronics Inc. Apparatus and method for processing a time domain audio signal with a noise filling flag
DE102008058496B4 (en) * 2008-11-21 2010-09-09 Siemens Medical Instruments Pte. Ltd. Filter bank system with specific stop attenuation components for a hearing device
GB2466201B (en) * 2008-12-10 2012-07-11 Skype Ltd Regeneration of wideband speech
GB0822537D0 (en) 2008-12-10 2009-01-14 Skype Ltd Regeneration of wideband speech
US9947340B2 (en) 2008-12-10 2018-04-17 Skype Regeneration of wideband speech
EP2360687A4 (en) * 2008-12-19 2012-07-11 Fujitsu Ltd Voice band extension device and voice band extension method
GB2466671B (en) 2009-01-06 2013-03-27 Skype Speech encoding
GB2466673B (en) 2009-01-06 2012-11-07 Skype Quantization
GB2466672B (en) * 2009-01-06 2013-03-13 Skype Speech coding
GB2466674B (en) * 2009-01-06 2013-11-13 Skype Speech coding
GB2466675B (en) 2009-01-06 2013-03-06 Skype Speech coding
GB2466669B (en) * 2009-01-06 2013-03-06 Skype Speech coding
GB2466670B (en) * 2009-01-06 2012-11-14 Skype Speech encoding
EP2380172B1 (en) 2009-01-16 2013-07-24 Dolby International AB Cross product enhanced harmonic transposition
US8463599B2 (en) * 2009-02-04 2013-06-11 Motorola Mobility Llc Bandwidth extension method and apparatus for a modified discrete cosine transform audio coder
JP5459688B2 (en) * 2009-03-31 2014-04-02 ▲ホア▼▲ウェイ▼技術有限公司 Method, apparatus, and speech decoding system for adjusting spectrum of decoded signal
JP4921611B2 (en) * 2009-04-03 2012-04-25 株式会社エヌ・ティ・ティ・ドコモ Speech decoding apparatus, speech decoding method, and speech decoding program
JP4932917B2 (en) 2009-04-03 2012-05-16 株式会社エヌ・ティ・ティ・ドコモ Speech decoding apparatus, speech decoding method, and speech decoding program
CN101609680B (en) * 2009-06-01 2012-01-04 华为技术有限公司 Compression coding and decoding method, coder, decoder and coding device
US8000485B2 (en) * 2009-06-01 2011-08-16 Dts, Inc. Virtual audio processing for loudspeaker or headphone playback
KR20110001130A (en) * 2009-06-29 2011-01-06 삼성전자주식회사 Apparatus and method for encoding and decoding audio signals using weighted linear prediction transform
WO2011029484A1 (en) * 2009-09-14 2011-03-17 Nokia Corporation Signal enhancement processing
WO2011037587A1 (en) * 2009-09-28 2011-03-31 Nuance Communications, Inc. Downsampling schemes in a hierarchical neural network structure for phoneme recognition
US8452606B2 (en) * 2009-09-29 2013-05-28 Skype Speech encoding using multiple bit rates
JP5754899B2 (en) * 2009-10-07 2015-07-29 ソニー株式会社 Decoding apparatus and method, and program
WO2011048099A1 (en) 2009-10-20 2011-04-28 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Audio encoder, audio decoder, method for encoding an audio information, method for decoding an audio information and computer program using a region-dependent arithmetic coding mapping rule
WO2011048792A1 (en) * 2009-10-21 2011-04-28 パナソニック株式会社 Sound signal processing apparatus, sound encoding apparatus and sound decoding apparatus
ES2936307T3 (en) * 2009-10-21 2023-03-16 Dolby Int Ab Upsampling in a combined re-emitter filter bank
US8484020B2 (en) 2009-10-23 2013-07-09 Qualcomm Incorporated Determining an upperband signal from a narrowband signal
EP2502231B1 (en) * 2009-11-19 2014-06-04 Telefonaktiebolaget L M Ericsson (PUBL) Bandwidth extension of a low band audio signal
CN102714041B (en) * 2009-11-19 2014-04-16 瑞典爱立信有限公司 Improved excitation signal bandwidth extension
US8489393B2 (en) * 2009-11-23 2013-07-16 Cambridge Silicon Radio Limited Speech intelligibility
US9838784B2 (en) 2009-12-02 2017-12-05 Knowles Electronics, Llc Directional audio capture
RU2464651C2 (en) * 2009-12-22 2012-10-20 Общество с ограниченной ответственностью "Спирит Корп" Method and apparatus for multilevel scalable information loss tolerant speech encoding for packet switched networks
US20110167445A1 (en) * 2010-01-06 2011-07-07 Reams Robert W Audiovisual content channelization system
US8326607B2 (en) * 2010-01-11 2012-12-04 Sony Ericsson Mobile Communications Ab Method and arrangement for enhancing speech quality
BR122021008583B1 (en) * 2010-01-12 2022-03-22 Fraunhofer-Gesellschaft Zur Forderung Der Angewandten Forschung E.V. Audio encoder, audio decoder, method of encoding and audio information, and method of decoding audio information using a hash table that describes both significant state values and range boundaries
US8699727B2 (en) 2010-01-15 2014-04-15 Apple Inc. Visually-assisted mixing of audio using a spectral analyzer
US9525569B2 (en) * 2010-03-03 2016-12-20 Skype Enhanced circuit-switched calls
KR101445296B1 (en) * 2010-03-10 2014-09-29 프라운호퍼 게젤샤프트 쭈르 푀르데룽 데어 안겐반텐 포르슝 에. 베. Audio signal decoder, audio signal encoder, methods and computer program using a sampling rate dependent time-warp contour encoding
US8700391B1 (en) * 2010-04-01 2014-04-15 Audience, Inc. Low complexity bandwidth expansion of speech
CN102870156B (en) * 2010-04-12 2015-07-22 飞思卡尔半导体公司 Audio communication device, method for outputting an audio signal, and communication system
JP5652658B2 (en) 2010-04-13 2015-01-14 ソニー株式会社 Signal processing apparatus and method, encoding apparatus and method, decoding apparatus and method, and program
CN102971788B (en) * 2010-04-13 2017-05-31 弗劳恩霍夫应用研究促进协会 The method and encoder and decoder of the sample Precise Representation of audio signal
JP5609737B2 (en) 2010-04-13 2014-10-22 ソニー株式会社 Signal processing apparatus and method, encoding apparatus and method, decoding apparatus and method, and program
JP5850216B2 (en) 2010-04-13 2016-02-03 ソニー株式会社 Signal processing apparatus and method, encoding apparatus and method, decoding apparatus and method, and program
US9443534B2 (en) * 2010-04-14 2016-09-13 Huawei Technologies Co., Ltd. Bandwidth extension system and approach
BR112012025347B1 (en) * 2010-04-14 2020-06-09 Voiceage Corp combined innovation codebook coding device, celp coder, combined innovation codebook, celp decoder, combined innovation codebook coding method and combined innovation codebook coding method
EP2559032B1 (en) * 2010-04-16 2019-01-30 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Apparatus, method and computer program for generating a wideband signal using guided bandwidth extension and blind bandwidth extension
US8538035B2 (en) 2010-04-29 2013-09-17 Audience, Inc. Multi-microphone robust noise suppression
US8473287B2 (en) 2010-04-19 2013-06-25 Audience, Inc. Method for jointly optimizing noise reduction and voice quality in a mono or multi-microphone system
US8798290B1 (en) 2010-04-21 2014-08-05 Audience, Inc. Systems and methods for adaptive signal equalization
US8781137B1 (en) 2010-04-27 2014-07-15 Audience, Inc. Wind noise detection and suppression
US9378754B1 (en) 2010-04-28 2016-06-28 Knowles Electronics, Llc Adaptive spatial classifier for multi-microphone systems
US9558755B1 (en) 2010-05-20 2017-01-31 Knowles Electronics, Llc Noise suppression assisted automatic speech recognition
KR101660843B1 (en) 2010-05-27 2016-09-29 삼성전자주식회사 Apparatus and method for determining weighting function for lpc coefficients quantization
US8600737B2 (en) * 2010-06-01 2013-12-03 Qualcomm Incorporated Systems, methods, apparatus, and computer program products for wideband speech coding
ES2372202B2 (en) * 2010-06-29 2012-08-08 Universidad De Málaga LOW CONSUMPTION SOUND RECOGNITION SYSTEM.
CA3160488C (en) 2010-07-02 2023-09-05 Dolby International Ab Audio decoding with selective post filtering
US8447596B2 (en) 2010-07-12 2013-05-21 Audience, Inc. Monaural noise suppression based on computational auditory scene analysis
JP5589631B2 (en) * 2010-07-15 2014-09-17 富士通株式会社 Voice processing apparatus, voice processing method, and telephone apparatus
EP2593937B1 (en) * 2010-07-16 2015-11-11 Telefonaktiebolaget LM Ericsson (publ) Audio encoder and decoder and methods for encoding and decoding an audio signal
JP5777041B2 (en) * 2010-07-23 2015-09-09 沖電気工業株式会社 Band expansion device and program, and voice communication device
JP6075743B2 (en) 2010-08-03 2017-02-08 ソニー株式会社 Signal processing apparatus and method, and program
WO2012031125A2 (en) 2010-09-01 2012-03-08 The General Hospital Corporation Reversal of general anesthesia by administration of methylphenidate, amphetamine, modafinil, amantadine, and/or caffeine
KR102564590B1 (en) * 2010-09-16 2023-08-09 돌비 인터네셔널 에이비 Cross product enhanced subband block based harmonic transposition
US8924200B2 (en) 2010-10-15 2014-12-30 Motorola Mobility Llc Audio signal bandwidth extension in CELP-based speech coder
JP5707842B2 (en) 2010-10-15 2015-04-30 ソニー株式会社 Encoding apparatus and method, decoding apparatus and method, and program
WO2012053149A1 (en) * 2010-10-22 2012-04-26 パナソニック株式会社 Speech analyzing device, quantization device, inverse quantization device, and method for same
JP5743137B2 (en) * 2011-01-14 2015-07-01 ソニー株式会社 Signal processing apparatus and method, and program
US9767823B2 (en) 2011-02-07 2017-09-19 Qualcomm Incorporated Devices for encoding and detecting a watermarked signal
US9767822B2 (en) 2011-02-07 2017-09-19 Qualcomm Incorporated Devices for encoding and decoding a watermarked signal
AU2012217216B2 (en) 2011-02-14 2015-09-17 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E.V. Apparatus and method for coding a portion of an audio signal using a transient detection and a quality result
PL3471092T3 (en) 2011-02-14 2020-12-28 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Decoding of pulse positions of tracks of an audio signal
MY159444A (en) 2011-02-14 2017-01-13 Fraunhofer-Gesellschaft Zur Forderung Der Angewandten Forschung E V Encoding and decoding of pulse positions of tracks of an audio signal
CN103534754B (en) 2011-02-14 2015-09-30 弗兰霍菲尔运输应用研究公司 The audio codec utilizing noise to synthesize during the inertia stage
AU2012217153B2 (en) 2011-02-14 2015-07-16 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E.V. Apparatus and method for encoding and decoding an audio signal using an aligned look-ahead portion
CA2827000C (en) 2011-02-14 2016-04-05 Jeremie Lecomte Apparatus and method for error concealment in low-delay unified speech and audio coding (usac)
AR085895A1 (en) * 2011-02-14 2013-11-06 Fraunhofer Ges Forschung NOISE GENERATION IN AUDIO CODECS
CN102959620B (en) 2011-02-14 2015-05-13 弗兰霍菲尔运输应用研究公司 Information signal representation using lapped transform
SG192746A1 (en) 2011-02-14 2013-09-30 Fraunhofer Ges Forschung Apparatus and method for processing a decoded audio signal in a spectral domain
CA2823262C (en) 2011-02-16 2018-03-06 Dolby Laboratories Licensing Corporation Methods and systems for generating filter coefficients and configuring filters
DK3998607T3 (en) * 2011-02-18 2024-04-15 Ntt Docomo Inc VOICE CODES
WO2012122397A1 (en) 2011-03-09 2012-09-13 Srs Labs, Inc. System for dynamically creating and rendering audio objects
US10642934B2 (en) 2011-03-31 2020-05-05 Microsoft Technology Licensing, Llc Augmented conversational understanding architecture
US9244984B2 (en) 2011-03-31 2016-01-26 Microsoft Technology Licensing, Llc Location based conversational understanding
US9760566B2 (en) 2011-03-31 2017-09-12 Microsoft Technology Licensing, Llc Augmented conversational understanding agent to identify conversation context between two humans and taking an agent action thereof
JP5704397B2 (en) * 2011-03-31 2015-04-22 ソニー株式会社 Encoding apparatus and method, and program
US9842168B2 (en) 2011-03-31 2017-12-12 Microsoft Technology Licensing, Llc Task driven user intents
US9298287B2 (en) 2011-03-31 2016-03-29 Microsoft Technology Licensing, Llc Combined activation for natural user interface systems
US9064006B2 (en) 2012-08-23 2015-06-23 Microsoft Technology Licensing, Llc Translating natural language utterances to keyword search queries
CN102811034A (en) 2011-05-31 2012-12-05 财团法人工业技术研究院 Signal processing device and signal processing method
JP5986565B2 (en) * 2011-06-09 2016-09-06 パナソニック インテレクチュアル プロパティ コーポレーション オブ アメリカＰａｎａｓｏｎｉｃ Ｉｎｔｅｌｌｅｃｔｕａｌ Ｐｒｏｐｅｒｔｙ Ｃｏｒｐｏｒａｔｉｏｎ ｏｆ Ａｍｅｒｉｃａ Speech coding apparatus, speech decoding apparatus, speech coding method, and speech decoding method
US9070361B2 (en) * 2011-06-10 2015-06-30 Google Technology Holdings LLC Method and apparatus for encoding a wideband speech signal utilizing downmixing of a highband component
EP2728577A4 (en) * 2011-06-30 2016-07-27 Samsung Electronics Co Ltd Apparatus and method for generating bandwidth extension signal
US9059786B2 (en) * 2011-07-07 2015-06-16 Vecima Networks Inc. Ingress suppression for communication systems
JP5942358B2 (en) * 2011-08-24 2016-06-29 ソニー株式会社 Encoding apparatus and method, decoding apparatus and method, and program
RU2486636C1 (en) * 2011-11-14 2013-06-27 Федеральное государственное военное образовательное учреждение высшего профессионального образования "Военный авиационный инженерный университет" (г. Воронеж) Министерства обороны Российской Федерации Method of generating high-frequency signals and apparatus for realising said method
RU2486638C1 (en) * 2011-11-15 2013-06-27 Федеральное государственное военное образовательное учреждение высшего профессионального образования "Военный авиационный инженерный университет" (г. Воронеж) Министерства обороны Российской Федерации Method of generating high-frequency signals and apparatus for realising said method
RU2486637C1 (en) * 2011-11-15 2013-06-27 Федеральное государственное военное образовательное учреждение высшего профессионального образования "Военный авиационный инженерный университет" (г. Воронеж) Министерства обороны Российской Федерации Method for generation and frequency-modulation of high-frequency signals and apparatus for realising said method
RU2496222C2 (en) * 2011-11-17 2013-10-20 Федеральное государственное образовательное учреждение высшего профессионального образования "Военный авиационный инженерный университет" (г. Воронеж) Министерства обороны Российской Федерации Method for generation and frequency-modulation of high-frequency signals and apparatus for realising said method
RU2486639C1 (en) * 2011-11-21 2013-06-27 Федеральное государственное военное образовательное учреждение высшего профессионального образования "Военный авиационный инженерный университет" (г. Воронеж) Министерства обороны Российской Федерации Method for generation and frequency-modulation of high-frequency signals and apparatus for realising said method
RU2496192C2 (en) * 2011-11-21 2013-10-20 Федеральное государственное военное образовательное учреждение высшего профессионального образования "Военный авиационный инженерный университет" (г. Воронеж) Министерства обороны Российской Федерации Method for generation and frequency-modulation of high-frequency signals and apparatus for realising said method
RU2490727C2 (en) * 2011-11-28 2013-08-20 Федеральное государственное бюджетное образовательное учреждение высшего профессионального образования "Уральский государственный университет путей сообщения" (УрГУПС) Method of transmitting speech signals (versions)
RU2487443C1 (en) * 2011-11-29 2013-07-10 Федеральное государственное военное образовательное учреждение высшего профессионального образования "Военный авиационный инженерный университет" (г. Воронеж) Министерства обороны Российской Федерации Method of matching complex impedances and apparatus for realising said method
JP5817499B2 (en) * 2011-12-15 2015-11-18 富士通株式会社 Decoding device, encoding device, encoding / decoding system, decoding method, encoding method, decoding program, and encoding program
US9972325B2 (en) 2012-02-17 2018-05-15 Huawei Technologies Co., Ltd. System and method for mixed codebook excitation for speech coding
US9082398B2 (en) * 2012-02-28 2015-07-14 Huawei Technologies Co., Ltd. System and method for post excitation enhancement for low bit rate speech coding
US9437213B2 (en) * 2012-03-05 2016-09-06 Malaspina Labs (Barbados) Inc. Voice signal enhancement
CN108831501B (en) 2012-03-21 2023-01-10 三星电子株式会社 High frequency encoding/decoding method and apparatus for bandwidth extension
TR201911121T4 (en) * 2012-03-29 2019-08-21 Ericsson Telefon Ab L M Vector quantizer.
US10448161B2 (en) 2012-04-02 2019-10-15 Qualcomm Incorporated Systems, methods, apparatus, and computer-readable media for gestural manipulation of a sound field
JP5998603B2 (en) * 2012-04-18 2016-09-28 ソニー株式会社 Sound detection device, sound detection method, sound feature amount detection device, sound feature amount detection method, sound interval detection device, sound interval detection method, and program
KR101343768B1 (en) * 2012-04-19 2014-01-16 충북대학교 산학협력단 Method for speech and audio signal classification using Spectral flux pattern
RU2504894C1 (en) * 2012-05-17 2014-01-20 Федеральное государственное военное образовательное учреждение высшего профессионального образования "Военный авиационный инженерный университет" (г. Воронеж) Министерства обороны Российской Федерации Method of demodulating phase-modulated and frequency-modulated signals and apparatus for realising said method
RU2504898C1 (en) * 2012-05-17 2014-01-20 Федеральное государственное военное образовательное учреждение высшего профессионального образования "Военный авиационный инженерный университет" (г. Воронеж) Министерства обороны Российской Федерации Method of demodulating phase-modulated and frequency-modulated signals and apparatus for realising said method
US20140006017A1 (en) * 2012-06-29 2014-01-02 Qualcomm Incorporated Systems, methods, apparatus, and computer-readable media for generating obfuscated speech signal
EP3113184B1 (en) 2012-08-31 2017-12-06 Telefonaktiebolaget LM Ericsson (publ) Method and device for voice activity detection
US9460729B2 (en) 2012-09-21 2016-10-04 Dolby Laboratories Licensing Corporation Layered approach to spatial audio coding
WO2014062859A1 (en) * 2012-10-16 2014-04-24 Audiologicall, Ltd. Audio signal manipulation for speech enhancement before sound reproduction
KR101413969B1 (en) 2012-12-20 2014-07-08 삼성전자주식회사 Method and apparatus for decoding audio signal
CN103928031B (en) * 2013-01-15 2016-03-30 华为技术有限公司 Coding method, coding/decoding method, encoding apparatus and decoding apparatus
US9728200B2 (en) 2013-01-29 2017-08-08 Qualcomm Incorporated Systems, methods, apparatus, and computer-readable media for adaptive formant sharpening in linear prediction coding
CN106847297B (en) 2013-01-29 2020-07-07 华为技术有限公司 Prediction method of high-frequency band signal, encoding/decoding device
ES2768179T3 (en) * 2013-01-29 2020-06-22 Fraunhofer Ges Forschung Audio encoder, audio decoder, method of providing encoded audio information, method of providing decoded audio information, software and encoded representation using signal adapted bandwidth extension
MX347316B (en) 2013-01-29 2017-04-21 Fraunhofer Ges Forschung Apparatus and method for synthesizing an audio signal, decoder, encoder, system and computer program.
US20140213909A1 (en) * 2013-01-31 2014-07-31 Xerox Corporation Control-based inversion for estimating a biological parameter vector for a biophysics model from diffused reflectance data
US9711156B2 (en) 2013-02-08 2017-07-18 Qualcomm Incorporated Systems and methods of performing filtering for gain determination
US9741350B2 (en) 2013-02-08 2017-08-22 Qualcomm Incorporated Systems and methods of performing gain control
US9601125B2 (en) 2013-02-08 2017-03-21 Qualcomm Incorporated Systems and methods of performing noise modulation and gain adjustment
US9336789B2 (en) * 2013-02-21 2016-05-10 Qualcomm Incorporated Systems and methods for determining an interpolation factor set for synthesizing a speech signal
WO2014136629A1 (en) * 2013-03-05 2014-09-12 日本電気株式会社 Signal processing device, signal processing method, and signal processing program
EP2784775B1 (en) * 2013-03-27 2016-09-14 Binauric SE Speech signal encoding/decoding method and apparatus
CN117253498A (en) 2013-04-05 2023-12-19 杜比国际公司 Audio signal decoding method, audio signal decoder, audio signal medium, and audio signal encoding method
MX343673B (en) * 2013-04-05 2016-11-16 Dolby Int Ab Audio encoder and decoder.
EP2981955B1 (en) * 2013-04-05 2023-06-07 Dts Llc Layered audio coding and transmission
MX371425B (en) * 2013-06-21 2020-01-29 Fraunhofer Ges Forschung Apparatus and method for improved concealment of the adaptive codebook in acelp-like concealment employing improved pitch lag estimation.
FR3007563A1 (en) * 2013-06-25 2014-12-26 France Telecom ENHANCED FREQUENCY BAND EXTENSION IN AUDIO FREQUENCY SIGNAL DECODER
JP6660878B2 (en) 2013-06-27 2020-03-11 ザ ジェネラル ホスピタル コーポレイション System for tracking dynamic structures in physiological data and method of operating the system
US10383574B2 (en) 2013-06-28 2019-08-20 The General Hospital Corporation Systems and methods to infer brain state during burst suppression
CN104282308B (en) 2013-07-04 2017-07-14 华为技术有限公司 The vector quantization method and device of spectral envelope
EP2830061A1 (en) 2013-07-22 2015-01-28 Fraunhofer Gesellschaft zur Förderung der angewandten Forschung e.V. Apparatus and method for encoding and decoding an encoded audio signal using temporal noise/patch shaping
EP3503095A1 (en) 2013-08-28 2019-06-26 Dolby Laboratories Licensing Corp. Hybrid waveform-coded and parametric-coded speech enhancement
TWI557726B (en) * 2013-08-29 2016-11-11 杜比國際公司 System and method for determining a master scale factor band table for a highband signal of an audio signal
US10602978B2 (en) 2013-09-13 2020-03-31 The General Hospital Corporation Systems and methods for improved brain monitoring during general anesthesia and sedation
US9875746B2 (en) 2013-09-19 2018-01-23 Sony Corporation Encoding device and method, decoding device and method, and program
CN104517610B (en) * 2013-09-26 2018-03-06 华为技术有限公司 The method and device of bandspreading
CN105761723B (en) * 2013-09-26 2019-01-15 华为技术有限公司 A kind of high-frequency excitation signal prediction technique and device
US9224402B2 (en) 2013-09-30 2015-12-29 International Business Machines Corporation Wideband speech parameterization for high quality synthesis, transformation and quantization
US9620134B2 (en) * 2013-10-10 2017-04-11 Qualcomm Incorporated Gain shape estimation for improved tracking of high-band temporal characteristics
US10083708B2 (en) * 2013-10-11 2018-09-25 Qualcomm Incorporated Estimation of mixing factors to generate high-band excitation signal
US9384746B2 (en) * 2013-10-14 2016-07-05 Qualcomm Incorporated Systems and methods of energy-scaled signal processing
KR102271852B1 (en) * 2013-11-02 2021-07-01 삼성전자주식회사 Method and apparatus for generating wideband signal and device employing the same
EP2871641A1 (en) * 2013-11-12 2015-05-13 Dialog Semiconductor B.V. Enhancement of narrowband audio signals using a single sideband AM modulation
CN105765655A (en) 2013-11-22 2016-07-13 高通股份有限公司 Selective phase compensation in high band coding
US10163447B2 (en) * 2013-12-16 2018-12-25 Qualcomm Incorporated High-band signal modeling
CN103714822B (en) * 2013-12-27 2017-01-11 广州华多网络科技有限公司 Sub-band coding and decoding method and device based on SILK coder decoder
AU2014371411A1 (en) 2013-12-27 2016-06-23 Sony Corporation Decoding device, method, and program
FR3017484A1 (en) * 2014-02-07 2015-08-14 Orange ENHANCED FREQUENCY BAND EXTENSION IN AUDIO FREQUENCY SIGNAL DECODER
US9564141B2 (en) 2014-02-13 2017-02-07 Qualcomm Incorporated Harmonic bandwidth extension of audio signals
JP6281336B2 (en) * 2014-03-12 2018-02-21 沖電気工業株式会社 Speech decoding apparatus and program
JP6035270B2 (en) * 2014-03-24 2016-11-30 株式会社Ｎｔｔドコモ Speech decoding apparatus, speech encoding apparatus, speech decoding method, speech encoding method, speech decoding program, and speech encoding program
CN111710342B (en) * 2014-03-31 2024-04-16 弗朗霍弗应用研究促进协会 Encoding device, decoding device, encoding method, decoding method, and program
US9542955B2 (en) * 2014-03-31 2017-01-10 Qualcomm Incorporated High-band signal coding using multiple sub-bands
US9697843B2 (en) 2014-04-30 2017-07-04 Qualcomm Incorporated High band excitation signal generation
CN105336336B (en) * 2014-06-12 2016-12-28 华为技术有限公司 The temporal envelope processing method and processing device of a kind of audio signal, encoder
CN105336338B (en) * 2014-06-24 2017-04-12 华为技术有限公司 Audio coding method and apparatus
US9626983B2 (en) * 2014-06-26 2017-04-18 Qualcomm Incorporated Temporal gain adjustment based on high-band signal characteristic
US9984699B2 (en) * 2014-06-26 2018-05-29 Qualcomm Incorporated High-band signal coding using mismatched frequency ranges
CN106486129B (en) * 2014-06-27 2019-10-25 华为技术有限公司 A kind of audio coding method and device
US9721584B2 (en) * 2014-07-14 2017-08-01 Intel IP Corporation Wind noise reduction for audio reception
EP2980795A1 (en) 2014-07-28 2016-02-03 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Audio encoding and decoding using a frequency domain processor, a time domain processor and a cross processor for initialization of the time domain processor
EP2980798A1 (en) * 2014-07-28 2016-02-03 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Harmonicity-dependent controlling of a harmonic filter tool
EP2980794A1 (en) 2014-07-28 2016-02-03 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Audio encoder and decoder using a frequency domain processor and a time domain processor
EP3182412B1 (en) 2014-08-15 2023-06-07 Samsung Electronics Co., Ltd. Sound quality improving method and device, sound decoding method and device, and multimedia device employing same
CN104217730B (en) * 2014-08-18 2017-07-21 大连理工大学 A kind of artificial speech bandwidth expanding method and device based on K SVD
US9978388B2 (en) 2014-09-12 2018-05-22 Knowles Electronics, Llc Systems and methods for restoration of speech components
TWI550945B (en) * 2014-12-22 2016-09-21 國立彰化師範大學 Method of designing composite filters with sharp transition bands and cascaded composite filters
US9595269B2 (en) * 2015-01-19 2017-03-14 Qualcomm Incorporated Scaling for gain shape circuitry
WO2016123560A1 (en) 2015-01-30 2016-08-04 Knowles Electronics, Llc Contextual switching of microphones
CA2976864C (en) 2015-02-26 2020-07-14 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E.V. Apparatus and method for processing an audio signal to obtain a processed audio signal using a target time-domain envelope
WO2016142002A1 (en) 2015-03-09 2016-09-15 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E.V. Audio encoder, audio decoder, method for encoding an audio signal and method for decoding an encoded audio signal
US10847170B2 (en) 2015-06-18 2020-11-24 Qualcomm Incorporated Device and method for generating a high-band signal from non-linearly processed sub-ranges
US9837089B2 (en) * 2015-06-18 2017-12-05 Qualcomm Incorporated High-band signal generation
US9407989B1 (en) 2015-06-30 2016-08-02 Arthur Woodrow Closed audio circuit
US9830921B2 (en) * 2015-08-17 2017-11-28 Qualcomm Incorporated High-band target signal control
CN107924683B (en) * 2015-10-15 2021-03-30 华为技术有限公司 Sinusoidal coding and decoding method and device
NO339664B1 (en) 2015-10-15 2017-01-23 St Tech As A system for isolating an object
BR112017024480A2 (en) 2016-02-17 2018-07-24 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E. V. postprocessor, preprocessor, audio encoder, audio decoder, and related methods for enhancing transient processing
FR3049084B1 (en) 2016-03-15 2022-11-11 Fraunhofer Ges Forschung CODING DEVICE FOR PROCESSING AN INPUT SIGNAL AND DECODING DEVICE FOR PROCESSING A CODED SIGNAL
MX2018012490A (en) * 2016-04-12 2019-02-21 Fraunhofer Ges Forschung Audio encoder for encoding an audio signal, method for encoding an audio signal and computer program under consideration of a detected peak spectral region in an upper frequency band.
US10770088B2 (en) * 2016-05-10 2020-09-08 Immersion Networks, Inc. Adaptive audio decoder system, method and article
US10756755B2 (en) * 2016-05-10 2020-08-25 Immersion Networks, Inc. Adaptive audio codec system, method and article
CN109416913B (en) * 2016-05-10 2024-03-15 易默森服务有限责任公司 Adaptive audio coding and decoding system, method, device and medium
US10699725B2 (en) * 2016-05-10 2020-06-30 Immersion Networks, Inc. Adaptive audio encoder system, method and article
US20170330575A1 (en) * 2016-05-10 2017-11-16 Immersion Services LLC Adaptive audio codec system, method and article
US10264116B2 (en) * 2016-11-02 2019-04-16 Nokia Technologies Oy Virtual duplex operation
KR102507383B1 (en) * 2016-11-08 2023-03-08 한국전자통신연구원 Method and system for stereo matching by using rectangular window
US10786168B2 (en) 2016-11-29 2020-09-29 The General Hospital Corporation Systems and methods for analyzing electrophysiological data from patients undergoing medical treatments
WO2018109143A1 (en) 2016-12-16 2018-06-21 Telefonaktiebolaget Lm Ericsson (Publ) Methods, encoder and decoder for handling envelope representation coefficients
PL3684001T3 (en) * 2017-01-06 2022-02-07 Telefonaktiebolaget Lm Ericsson (Publ) Methods and apparatuses for signaling and determining reference signal offsets
KR102687184B1 (en) * 2017-02-10 2024-07-19 삼성전자주식회사 WFST decoding system, speech recognition system including the same and Method for stroing WFST data
US10553222B2 (en) 2017-03-09 2020-02-04 Qualcomm Incorporated Inter-channel bandwidth extension spectral mapping and adjustment
US10304468B2 (en) * 2017-03-20 2019-05-28 Qualcomm Incorporated Target sample generation
TWI807562B (en) * 2017-03-23 2023-07-01 瑞典商都比國際公司 Backward-compatible integration of harmonic transposer for high frequency reconstruction of audio signals
US10825467B2 (en) * 2017-04-21 2020-11-03 Qualcomm Incorporated Non-harmonic speech detection and bandwidth extension in a multi-source environment
US20190051286A1 (en) * 2017-08-14 2019-02-14 Microsoft Technology Licensing, Llc Normalization of high band signals in network telephony communications
US11876659B2 (en) 2017-10-27 2024-01-16 Terawave, Llc Communication system using shape-shifted sinusoidal waveforms
US10666481B2 (en) * 2017-10-27 2020-05-26 Terawave, Llc High spectral efficiency data communications system using energy-balanced modulation
CN109729553B (en) * 2017-10-30 2021-12-28 成都鼎桥通信技术有限公司 Voice service processing method and device of LTE (Long term evolution) trunking communication system
EP3483884A1 (en) 2017-11-10 2019-05-15 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Signal filtering
EP3483886A1 (en) 2017-11-10 2019-05-15 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Selecting pitch lag
WO2019091576A1 (en) 2017-11-10 2019-05-16 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Audio encoders, audio decoders, methods and computer programs adapting an encoding and decoding of least significant bits
WO2019091573A1 (en) 2017-11-10 2019-05-16 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Apparatus and method for encoding and decoding an audio signal using downsampling or interpolation of scale parameters
EP3483882A1 (en) 2017-11-10 2019-05-15 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Controlling bandwidth in encoders and/or decoders
EP3483880A1 (en) 2017-11-10 2019-05-15 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Temporal noise shaping
EP3483879A1 (en) 2017-11-10 2019-05-15 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Analysis/synthesis windowing function for modulated lapped transformation
EP3483883A1 (en) 2017-11-10 2019-05-15 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Audio coding and decoding with selective postfiltering
EP3483878A1 (en) 2017-11-10 2019-05-15 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Audio decoder supporting a set of different loss concealment tools
US10460749B1 (en) * 2018-06-28 2019-10-29 Nuvoton Technology Corporation Voice activity detection using vocal tract area information
US10957331B2 (en) 2018-12-17 2021-03-23 Microsoft Technology Licensing, Llc Phase reconstruction in a speech decoder
US10847172B2 (en) * 2018-12-17 2020-11-24 Microsoft Technology Licensing, Llc Phase quantization in a speech encoder
JP7088403B2 (en) * 2019-02-20 2022-06-21 ヤマハ株式会社 Sound signal generation method, generative model training method, sound signal generation system and program
CN110610713B (en) * 2019-08-28 2021-11-16 南京梧桐微电子科技有限公司 Vocoder residue spectrum amplitude parameter reconstruction method and system
US11380343B2 (en) 2019-09-12 2022-07-05 Immersion Networks, Inc. Systems and methods for processing high frequency audio signal
TWI723545B (en) * 2019-09-17 2021-04-01 宏碁股份有限公司 Speech processing method and device thereof
US11295751B2 (en) 2019-09-20 2022-04-05 Tencent America LLC Multi-band synchronized neural vocoder
KR102201169B1 (en) * 2019-10-23 2021-01-11 성균관대학교 산학협력단 Method for generating time code and space-time code for controlling reflection coefficient of meta surface, recording medium storing program for executing the same, and method for signal modulation using meta surface
CN114548442B (en) * 2022-02-25 2022-10-21 万表名匠(广州)科技有限公司 Wristwatch maintenance management system based on internet technology

Citations (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US6732070B1 (en) * 2000-02-16 2004-05-04 Nokia Mobile Phones, Ltd. Wideband speech codec using a higher sampling rate in analysis and synthesis filtering than in excitation searching

Family Cites Families (147)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US321993A (en) * 1885-07-14 Lantern
US525147A (en) * 1894-08-28 Steam-cooker
US526468A (en) * 1894-09-25 Charles d
US596689A (en) * 1898-01-04 Hose holder or support
US1126620A (en) * 1911-01-30 1915-01-26 Safety Car Heating & Lighting Electric regulation.
US1089258A (en) * 1914-01-13 1914-03-03 James Arnot Paterson Facing or milling machine.
US1300833A (en) * 1918-12-12 1919-04-15 Moline Mill Mfg Company Idler-pulley structure.
US1498873A (en) * 1924-04-19 1924-06-24 Bethlehem Steel Corp Switch stand
US2073913A (en) * 1934-06-26 1937-03-16 Wigan Edmund Ramsay Means for gauging minute displacements
US2086867A (en) * 1936-06-19 1937-07-13 Hall Lab Inc Laundering composition and process
US3044777A (en) * 1959-10-19 1962-07-17 Fibermold Corp Bowling pin
US3158693A (en) * 1962-08-07 1964-11-24 Bell Telephone Labor Inc Speech interpolation communication system
US3855416A (en) 1972-12-01 1974-12-17 F Fuller Method and apparatus for phonation analysis leading to valid truth/lie decisions by fundamental speech-energy weighted vibratto component assessment
US3855414A (en) * 1973-04-24 1974-12-17 Anaconda Co Cable armor clamp
JPS59139099A (en) 1983-01-31 1984-08-09 株式会社東芝 Voice section detector
US4616659A (en) * 1985-05-06 1986-10-14 At&T Bell Laboratories Heart rate detection utilizing autoregressive analysis
US4630305A (en) * 1985-07-01 1986-12-16 Motorola, Inc. Automatic gain selector for a noise suppression system
US4747143A (en) 1985-07-12 1988-05-24 Westinghouse Electric Corp. Speech enhancement system having dynamic gain control
NL8503152A (en) * 1985-11-15 1987-06-01 Optische Ind De Oude Delft Nv DOSEMETER FOR IONIZING RADIATION.
US4862168A (en) 1987-03-19 1989-08-29 Beard Terry D Audio digital/analog encoding and decoding
US4805193A (en) * 1987-06-04 1989-02-14 Motorola, Inc. Protection of energy information in sub-band coding
US4852179A (en) * 1987-10-05 1989-07-25 Motorola, Inc. Variable frame rate, fixed bit rate vocoding method
JP2707564B2 (en) 1987-12-14 1998-01-28 株式会社日立製作所 Audio coding method
US5285520A (en) 1988-03-02 1994-02-08 Kokusai Denshin Denwa Kabushiki Kaisha Predictive coding apparatus
US5077798A (en) 1988-09-28 1991-12-31 Hitachi, Ltd. Method and system for voice coding based on vector quantization
US5086475A (en) 1988-11-19 1992-02-04 Sony Corporation Apparatus for generating, recording or reproducing sound source data
JPH02244100A (en) 1989-03-16 1990-09-28 Ricoh Co Ltd Noise sound source signal forming device
DE69128772T2 (en) 1990-09-19 1998-08-06 Philips Electronics N.V., Eindhoven SYSTEM WITH A RECORDING CARRIER AND A PLAYER
JP2779886B2 (en) 1992-10-05 1998-07-23 日本電信電話株式会社 Wideband audio signal restoration method
JP3191457B2 (en) * 1992-10-31 2001-07-23 ソニー株式会社 High efficiency coding apparatus, noise spectrum changing apparatus and method
US5455888A (en) * 1992-12-04 1995-10-03 Northern Telecom Limited Speech bandwidth extension method and apparatus
PL173718B1 (en) 1993-06-30 1998-04-30 Sony Corp Apparatus for encoding digital signals, apparatus for decoding digital signals and recording medium adapted for use in conjunction with them
WO1995010760A2 (en) 1993-10-08 1995-04-20 Comsat Corporation Improved low bit rate vocoders and methods of operation therefor
US5684920A (en) 1994-03-17 1997-11-04 Nippon Telegraph And Telephone Acoustic signal transform coding method and decoding method having a high efficiency envelope flattening method therein
US5487087A (en) * 1994-05-17 1996-01-23 Texas Instruments Incorporated Signal quantizer with reduced output fluctuation
US5797118A (en) 1994-08-09 1998-08-18 Yamaha Corporation Learning vector quantization and a temporary memory such that the codebook contents are renewed when a first speaker returns
JP2770137B2 (en) 1994-09-22 1998-06-25 日本プレシジョン・サーキッツ株式会社 Waveform data compression device
US5699477A (en) 1994-11-09 1997-12-16 Texas Instruments Incorporated Mixed excitation linear prediction with fractional pitch
FI97182C (en) 1994-12-05 1996-10-25 Nokia Telecommunications Oy Procedure for replacing received bad speech frames in a digital receiver and receiver for a digital telecommunication system
JP3365113B2 (en) * 1994-12-22 2003-01-08 ソニー株式会社 Audio level control device
JP2798003B2 (en) 1995-05-09 1998-09-17 松下電器産業株式会社 Voice band expansion device and voice band expansion method
JP2956548B2 (en) 1995-10-05 1999-10-04 松下電器産業株式会社 Voice band expansion device
EP0732687B2 (en) * 1995-03-13 2005-10-12 Matsushita Electric Industrial Co., Ltd. Apparatus for expanding speech bandwidth
JP3189614B2 (en) 1995-03-13 2001-07-16 松下電器産業株式会社 Voice band expansion device
US6263307B1 (en) 1995-04-19 2001-07-17 Texas Instruments Incorporated Adaptive weiner filtering using line spectral frequencies
US5706395A (en) * 1995-04-19 1998-01-06 Texas Instruments Incorporated Adaptive weiner filtering using a dynamic suppression factor
JP3334419B2 (en) * 1995-04-20 2002-10-15 ソニー株式会社 Noise reduction method and noise reduction device
US5699485A (en) 1995-06-07 1997-12-16 Lucent Technologies Inc. Pitch delay modification during frame erasures
US5704003A (en) * 1995-09-19 1997-12-30 Lucent Technologies Inc. RCELP coder
US6097824A (en) 1997-06-06 2000-08-01 Audiologic, Incorporated Continuous frequency dynamic range audio compressor
DE69530204T2 (en) * 1995-10-16 2004-03-18 Agfa-Gevaert New class of yellow dyes for photographic materials
JP3707116B2 (en) 1995-10-26 2005-10-19 ソニー株式会社 Speech decoding method and apparatus
US5737716A (en) * 1995-12-26 1998-04-07 Motorola Method and apparatus for encoding speech using neural network technology for speech classification
JP3073919B2 (en) * 1995-12-30 2000-08-07 松下電器産業株式会社 Synchronizer
US5689615A (en) * 1996-01-22 1997-11-18 Rockwell International Corporation Usage of voice activity detection for efficient coding of speech
TW307960B (en) * 1996-02-15 1997-06-11 Philips Electronics Nv Reduced complexity signal transmission system
DE69730779T2 (en) 1996-06-19 2005-02-10 Texas Instruments Inc., Dallas Improvements in or relating to speech coding
JP3246715B2 (en) 1996-07-01 2002-01-15 松下電器産業株式会社 Audio signal compression method and audio signal compression device
DE69712537T2 (en) * 1996-11-07 2002-08-29 Matsushita Electric Industrial Co., Ltd. Method for generating a vector quantization code book
US6009395A (en) 1997-01-02 1999-12-28 Texas Instruments Incorporated Synthesizer and method using scaled excitation signal
US6202046B1 (en) * 1997-01-23 2001-03-13 Kabushiki Kaisha Toshiba Background noise/speech classification method
US6041297A (en) * 1997-03-10 2000-03-21 At&T Corp Vocoder for coding speech by using a correlation between spectral magnitudes and candidate excitations
US5890126A (en) 1997-03-10 1999-03-30 Euphonics, Incorporated Audio data decompression and interpolation apparatus and method
EP0878790A1 (en) * 1997-05-15 1998-11-18 Hewlett-Packard Company Voice coding system and method
SE512719C2 (en) * 1997-06-10 2000-05-02 Lars Gustaf Liljeryd A method and apparatus for reducing data flow based on harmonic bandwidth expansion
US6889185B1 (en) * 1997-08-28 2005-05-03 Texas Instruments Incorporated Quantization of linear prediction coefficients using perceptual weighting
US6029125A (en) * 1997-09-02 2000-02-22 Telefonaktiebolaget L M Ericsson, (Publ) Reducing sparseness in coded speech signals
US6122384A (en) * 1997-09-02 2000-09-19 Qualcomm Inc. Noise suppression system and method
US6231516B1 (en) * 1997-10-14 2001-05-15 Vacusense, Inc. Endoluminal implant with therapeutic and diagnostic capability
JPH11205166A (en) 1998-01-19 1999-07-30 Mitsubishi Electric Corp Noise detector
US6301556B1 (en) * 1998-03-04 2001-10-09 Telefonaktiebolaget L M. Ericsson (Publ) Reducing sparseness in coded speech signals
US6449590B1 (en) 1998-08-24 2002-09-10 Conexant Systems, Inc. Speech encoder using warping in long term preprocessing
US6385573B1 (en) * 1998-08-24 2002-05-07 Conexant Systems, Inc. Adaptive tilt compensation for synthesized speech residual
JP4170458B2 (en) * 1998-08-27 2008-10-22 ローランド株式会社 Time-axis compression / expansion device for waveform signals
US6353808B1 (en) * 1998-10-22 2002-03-05 Sony Corporation Apparatus and method for encoding a signal as well as apparatus and method for decoding a signal
KR20000047944A (en) 1998-12-11 2000-07-25 이데이 노부유끼 Receiving apparatus and method, and communicating apparatus and method
JP4354561B2 (en) * 1999-01-08 2009-10-28 パナソニック株式会社 Audio signal encoding apparatus and decoding apparatus
US6223151B1 (en) 1999-02-10 2001-04-24 Telefon Aktie Bolaget Lm Ericsson Method and apparatus for pre-processing speech signals prior to coding by transform-based speech coders
JP3696091B2 (en) 1999-05-14 2005-09-14 松下電器産業株式会社 Method and apparatus for extending the bandwidth of an audio signal
US6604070B1 (en) * 1999-09-22 2003-08-05 Conexant Systems, Inc. System of encoding and decoding speech signals
JP4792613B2 (en) * 1999-09-29 2011-10-12 ソニー株式会社 Information processing apparatus and method, and recording medium
US6556950B1 (en) 1999-09-30 2003-04-29 Rockwell Automation Technologies, Inc. Diagnostic method and apparatus for use with enterprise control
US6715125B1 (en) * 1999-10-18 2004-03-30 Agere Systems Inc. Source coding and transmission with time diversity
JP5220254B2 (en) * 1999-11-16 2013-06-26 コーニンクレッカ フィリップス エレクトロニクス エヌ ヴィ Wideband audio transmission system
CA2290037A1 (en) * 1999-11-18 2001-05-18 Voiceage Corporation Gain-smoothing amplifier device and method in codecs for wideband speech and audio signals
US7260523B2 (en) * 1999-12-21 2007-08-21 Texas Instruments Incorporated Sub-band speech coding system
US7167828B2 (en) 2000-01-11 2007-01-23 Matsushita Electric Industrial Co., Ltd. Multimode speech coding apparatus and decoding apparatus
US6757395B1 (en) * 2000-01-12 2004-06-29 Sonic Innovations, Inc. Noise reduction apparatus and method
US6704711B2 (en) 2000-01-28 2004-03-09 Telefonaktiebolaget Lm Ericsson (Publ) System and method for modifying speech signals
JP3681105B2 (en) 2000-02-24 2005-08-10 アルパイン株式会社 Data processing method
FI119576B (en) * 2000-03-07 2008-12-31 Nokia Corp Speech processing device and procedure for speech processing, as well as a digital radio telephone
US6523003B1 (en) * 2000-03-28 2003-02-18 Tellabs Operations, Inc. Spectrally interdependent gain adjustment techniques
US6757654B1 (en) * 2000-05-11 2004-06-29 Telefonaktiebolaget Lm Ericsson Forward error correction in speech coding
US7330814B2 (en) * 2000-05-22 2008-02-12 Texas Instruments Incorporated Wideband speech coding with modulated noise highband excitation system and method
JP2001337700A (en) 2000-05-22 2001-12-07 Texas Instr Inc <Ti> System for coding wideband speech and its method
US7136810B2 (en) * 2000-05-22 2006-11-14 Texas Instruments Incorporated Wideband speech coding system and method
JP2002055699A (en) 2000-08-10 2002-02-20 Mitsubishi Electric Corp Device and method for encoding voice
CN1279531C (en) * 2000-08-25 2006-10-11 皇家菲利浦电子有限公司 Method and apparatus for reducing the word length of a digital input signal and method and apparatus for recovering the digital input signal
US6515889B1 (en) * 2000-08-31 2003-02-04 Micron Technology, Inc. Junction-isolated depletion mode ferroelectric memory
US7386444B2 (en) 2000-09-22 2008-06-10 Texas Instruments Incorporated Hybrid speech coding and system
US6947888B1 (en) * 2000-10-17 2005-09-20 Qualcomm Incorporated Method and apparatus for high performance low bit-rate coding of unvoiced speech
JP2002202799A (en) * 2000-10-30 2002-07-19 Fujitsu Ltd Voice code conversion apparatus
JP3558031B2 (en) * 2000-11-06 2004-08-25 日本電気株式会社 Speech decoding device
EP1336175A1 (en) * 2000-11-09 2003-08-20 Koninklijke Philips Electronics N.V. Wideband extension of telephone speech for higher perceptual quality
SE0004163D0 (en) * 2000-11-14 2000-11-14 Coding Technologies Sweden Ab Enhancing perceptual performance or high frequency reconstruction coding methods by adaptive filtering
SE0004187D0 (en) * 2000-11-15 2000-11-15 Coding Technologies Sweden Ab Enhancing the performance of coding systems that use high frequency reconstruction methods
KR100910282B1 (en) 2000-11-30 2009-08-03 파나소닉 주식회사 Vector quantizing device for lpc parameters, decoding device for lpc parameters, recording medium, voice encoding device, voice decoding device, voice signal transmitting device, and voice signal receiving device
GB0031461D0 (en) 2000-12-22 2001-02-07 Thales Defence Ltd Communication sets
US20040204935A1 (en) 2001-02-21 2004-10-14 Krishnasamy Anandakumar Adaptive voice playout in VOP
JP2002268698A (en) 2001-03-08 2002-09-20 Nec Corp Voice recognition device, device and method for standard pattern generation, and program
US20030028386A1 (en) * 2001-04-02 2003-02-06 Zinser Richard L. Compressed domain universal transcoder
SE522553C2 (en) * 2001-04-23 2004-02-17 Ericsson Telefon Ab L M Bandwidth extension of acoustic signals
EP1388147B1 (en) * 2001-05-11 2004-12-29 Siemens Aktiengesellschaft Method for enlarging the band width of a narrow-band filtered voice signal, especially a voice signal emitted by a telecommunication appliance
JP2004521394A (en) * 2001-06-28 2004-07-15 コーニンクレッカ　フィリップス　エレクトロニクス　エヌ　ヴィ Broadband signal transmission system
US6879955B2 (en) * 2001-06-29 2005-04-12 Microsoft Corporation Signal modification based on continuous time warping for low bit rate CELP coding
JP2003036097A (en) * 2001-07-25 2003-02-07 Sony Corp Device and method for detecting and retrieving information
TW525147B (en) 2001-09-28 2003-03-21 Inventec Besta Co Ltd Method of obtaining and decoding basic cycle of voice
US6988066B2 (en) * 2001-10-04 2006-01-17 At&T Corp. Method of bandwidth extension for narrow-band speech
US6895375B2 (en) 2001-10-04 2005-05-17 At&T Corp. System for bandwidth extension of Narrow-band speech
TW526468B (en) 2001-10-19 2003-04-01 Chunghwa Telecom Co Ltd System and method for eliminating background noise of voice signal
JP4245288B2 (en) 2001-11-13 2009-03-25 パナソニック株式会社 Speech coding apparatus and speech decoding apparatus
KR20040066835A (en) * 2001-11-23 2004-07-27 코닌클리즈케 필립스 일렉트로닉스 엔.브이. Audio signal bandwidth extension
CA2365203A1 (en) 2001-12-14 2003-06-14 Voiceage Corporation A signal modification method for efficient coding of speech signals
US6751587B2 (en) * 2002-01-04 2004-06-15 Broadcom Corporation Efficient excitation quantization in noise feedback coding with general noise shaping
JP4290917B2 (en) * 2002-02-08 2009-07-08 株式会社エヌ・ティ・ティ・ドコモ Decoding device, encoding device, decoding method, and encoding method
JP3826813B2 (en) 2002-02-18 2006-09-27 ソニー株式会社 Digital signal processing apparatus and digital signal processing method
ES2259158T3 (en) 2002-09-19 2006-09-16 Matsushita Electric Industrial Co., Ltd. METHOD AND DEVICE AUDIO DECODER.
JP3756864B2 (en) 2002-09-30 2006-03-15 株式会社東芝 Speech synthesis method and apparatus and speech synthesis program
KR100841096B1 (en) * 2002-10-14 2008-06-25 리얼네트웍스아시아퍼시픽 주식회사 Preprocessing of digital audio data for mobile speech codecs
US20040098255A1 (en) 2002-11-14 2004-05-20 France Telecom Generalized analysis-by-synthesis speech coding method, and coder implementing such method
US7242763B2 (en) * 2002-11-26 2007-07-10 Lucent Technologies Inc. Systems and methods for far-end noise reduction and near-end noise compensation in a mixed time-frequency domain compander to improve signal quality in communications systems
CA2415105A1 (en) 2002-12-24 2004-06-24 Voiceage Corporation A method and device for robust predictive vector quantization of linear prediction parameters in variable bit rate speech coding
KR100480341B1 (en) 2003-03-13 2005-03-31 한국전자통신연구원 Apparatus for coding wide-band low bit rate speech signal
CN1820306B (en) 2003-05-01 2010-05-05 诺基亚有限公司 Method and device for gain quantization in variable bit rate wideband speech coding
WO2005004113A1 (en) 2003-06-30 2005-01-13 Fujitsu Limited Audio encoding device
US20050004793A1 (en) * 2003-07-03 2005-01-06 Pasi Ojala Signal adaptation for higher band coding in a codec utilizing band split coding
FI118550B (en) 2003-07-14 2007-12-14 Nokia Corp Enhanced excitation for higher frequency band coding in a codec utilizing band splitting based coding methods
US7428490B2 (en) 2003-09-30 2008-09-23 Intel Corporation Method for spectral subtraction in speech enhancement
US7698292B2 (en) * 2003-12-03 2010-04-13 Siemens Aktiengesellschaft Tag management within a decision, support, and reporting environment
KR100587953B1 (en) * 2003-12-26 2006-06-08 한국전자통신연구원 Packet loss concealment apparatus for high-band in split-band wideband speech codec, and system for decoding bit-stream using the same
CA2454296A1 (en) * 2003-12-29 2005-06-29 Nokia Corporation Method and device for speech enhancement in the presence of background noise
JP4259401B2 (en) 2004-06-02 2009-04-30 カシオ計算機株式会社 Speech processing apparatus and speech coding method
US8000967B2 (en) * 2005-03-09 2011-08-16 Telefonaktiebolaget Lm Ericsson (Publ) Low-complexity code excited linear prediction encoding
US8155965B2 (en) 2005-03-11 2012-04-10 Qualcomm Incorporated Time warping frames inside the vocoder by modifying the residual
BRPI0607646B1 (en) 2005-04-01 2021-05-25 Qualcomm Incorporated METHOD AND EQUIPMENT FOR SPEECH BAND DIVISION ENCODING
UA94041C2 (en) * 2005-04-01 2011-04-11 Квелкомм Инкорпорейтед Method and device for anti-sparseness filtering
PL1875463T3 (en) 2005-04-22 2019-03-29 Qualcomm Incorporated Systems, methods, and apparatus for gain factor smoothing

Patent Citations (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US6732070B1 (en) * 2000-02-16 2004-05-04 Nokia Mobile Phones, Ltd. Wideband speech codec using a higher sampling rate in analysis and synthesis filtering than in excitation searching

Non-Patent Citations (2)

* Cited by examiner, † Cited by third party
Title
KLEIJN W B ET AL: "THE RCELP SPEECH-CODING ALGORITHM", EUROPEAN TRANSACTIONS ON TELECOMMUNICATIONS AND RELATED TECHNOLOGIES, AEI, MILANO, IT, vol. 5, no. 5, 1 September 1994 (1994-09-01), pages 573 - 582, XP000470678, ISSN: 1120-3862 *
TAMMI M ET AL: "Coding distortion caused by a phase difference between the LP filter and its residual", SPEECH CODING PROCEEDINGS, 1999 IEEE WORKSHOP ON PORVOO, FINLAND 20-23 JUNE 1999, PISCATAWAY, NJ, USA,IEEE, US, 20 June 1999 (1999-06-20), pages 102 - 104, XP010345571, ISBN: 0-7803-5651-9 *

Cited By (11)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2011511311A (en) * 2008-01-31 2011-04-07 フラウンホーファー−ゲゼルシャフト・ツール・フェルデルング・デル・アンゲヴァンテン・フォルシュング・アインゲトラーゲネル・フェライン Apparatus and method for bandwidth extension of audio signal
US8996362B2 (en) 2008-01-31 2015-03-31 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E.V. Device and method for a bandwidth extension of an audio signal
JP2012527637A (en) * 2009-05-19 2012-11-08 エレクトロニクス アンド テレコミュニケーションズ リサーチ インスチチュート Audio signal encoding and decoding method and apparatus using hierarchical sinusoidal pulse coding
US10096322B2 (en) 2013-06-21 2018-10-09 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E.V. Audio decoder having a bandwidth extension module with an energy adjusting module
CN107527629A (en) * 2013-07-12 2017-12-29 皇家飞利浦有限公司 For carrying out the optimization zoom factor of bandspreading in audio signal decoder
US10354663B2 (en) 2014-07-28 2019-07-16 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Apparatus and method for generating an enhanced signal using independent noise-filling
US10529348B2 (en) 2014-07-28 2020-01-07 Fraunhofer-Gesellschaft Zur Foerderung Der Angewandten Forschung E.V. Apparatus and method for generating an enhanced signal using independent noise-filling identified by an identification vector
US10885924B2 (en) 2014-07-28 2021-01-05 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Apparatus and method for generating an enhanced signal using independent noise-filling
US11264042B2 (en) 2014-07-28 2022-03-01 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Apparatus and method for generating an enhanced signal using independent noise-filling information which comprises energy information and is included in an input signal
US11705145B2 (en) 2014-07-28 2023-07-18 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Apparatus and method for generating an enhanced signal using independent noise-filling
US11908484B2 (en) 2014-07-28 2024-02-20 Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V. Apparatus and method for generating an enhanced signal using independent noise-filling at random values and scaling thereupon

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