WO2004062087A1 - Modulateur miltimodes et emetteur - Google Patents

Modulateur miltimodes et emetteur Download PDF

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Publication number
WO2004062087A1
WO2004062087A1 PCT/CA2004/000014 CA2004000014W WO2004062087A1 WO 2004062087 A1 WO2004062087 A1 WO 2004062087A1 CA 2004000014 W CA2004000014 W CA 2004000014W WO 2004062087 A1 WO2004062087 A1 WO 2004062087A1
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WO
WIPO (PCT)
Prior art keywords
signal
input
mixer
circuit
mixing
Prior art date
Application number
PCT/CA2004/000014
Other languages
English (en)
Inventor
William Kung
Christopher Eugene Snyder
Original Assignee
Sirific Wireless Corporation
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from CA002415668A external-priority patent/CA2415668A1/fr
Application filed by Sirific Wireless Corporation filed Critical Sirific Wireless Corporation
Priority to CA002512107A priority Critical patent/CA2512107A1/fr
Priority to EP04700246A priority patent/EP1590885A1/fr
Priority to JP2006500427A priority patent/JP2006515498A/ja
Priority to US10/541,243 priority patent/US20060141952A1/en
Publication of WO2004062087A1 publication Critical patent/WO2004062087A1/fr

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/02—Transmitters
    • H04B1/04—Circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/02—Transmitters
    • H04B1/04—Circuits
    • H04B1/0475—Circuits with means for limiting noise, interference or distortion
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/16—Multiple-frequency-changing
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/16—Multiple-frequency-changing
    • H03D7/161—Multiple-frequency-changing all the frequency changers being connected in cascade
    • H03D7/163—Multiple-frequency-changing all the frequency changers being connected in cascade the local oscillations of at least two of the frequency changers being derived from a single oscillator
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/02—Transmitters
    • H04B1/04—Circuits
    • H04B2001/0491—Circuits with frequency synthesizers, frequency converters or modulators

Definitions

  • the present invention relates generally to communications, and more specifically to a method and apparatus of modulating baseband and RF (radio frequency) signals.
  • the preferred embodiment of the invention satisfies the need for an inexpensive, high-performance, fully-integrable, multi-standard transmitter.
  • the original (or baseband) signal may be. for example: data, voice or video.
  • These baseband signals may be produced by transducers such as microphones or video cameras, be computer generated, or transferred from an electronic storage device.
  • the high frequencies provide longer range and higher capacity channels than baseband signals, and because high frequency signals can effectively propagate through the air, they can be used for wireless transmissions as well as hard wired or wave guided channels.
  • RF signals are electromagnetic signals; that is, waveforms with electrical and magnetic properties within the electromagnetic spectrum normally associated with radio wave propagation.
  • Wired communication systems which employ such modulation and demodulation techniques include computer communication systems such as local area networks (LANs), point-to-point communications, and wide area networks
  • LANs local area networks
  • point-to-point communications point-to-point communications
  • wide area networks wide area networks
  • WANs such as the Internet.
  • These networks generally communicate data signals over electrically conductive or optical fibre channels.
  • Wireless communication systems which may employ modulation and demodulation include those for public broadcasting such as AM and FM radio, and UHF and VHF television.
  • Private communication systems may include cellular telephone networks, personal paging devices, HF radio systems used by taxi services, microwave backbone networks, interconnected appliances under the Bluetooth standard, and satellite communications.
  • Other wired and wireless systems which use RF modulation and demodulation would be known to those skilled in the art. There is currently a great desire to provide wireless devices which operate under multiple standards.
  • indirect modulation is a proven architecture for single-mode transmission and has the advantages of high overall performance in terms of noise, linearity and power/gain control.
  • this architecture is relatively costly to implement due to the need for IF and RF filters.
  • realization of a small and inexpensive multi-mode, multi-band transmitter is generally not possible using indirect modulation.
  • FIG. 1 presents a block diagram of a typical indirect modulation transmitter 10.
  • the mixers labelled 12 and 14 are used to translate the input signal Sin (generally a baseband signal, but could also be an RF signal) to a higher RF frequency (usually the carrier frequency of a signal being transmitted), which is labelled as output signal Sout.
  • Sin generally a baseband signal, but could also be an RF signal
  • output signal Sout usually the carrier frequency of a signal being transmitted
  • amplifier 22 buffers and amplifies the baseband signal, ensuring that it is at a level suitable to handle the subsequent processing.
  • the amplified signal is then filtered by a low pass or band pass filter 24 to remove undesirable signals which may interfere.
  • the filtered signal then enters mixer 12 which mixes the signal from filter 24 with a periodic signal generated by a local oscillator (LO1) 26. This translates the Sin signal to a higher frequency, known as the first intermediate frequency (IF1).
  • LO1 local oscillator
  • a mixer is a circuit or device that accepts as its input two different frequencies and presents at its output:
  • the typical embodiment of a mixer is a digital switch which may generate significantly more tones than stated above.
  • the IF1 signal is next filtered by a band pass filter 28 typically called a channel filter, which is centred around the IF1 frequency, thus filtering out the unwanted products of the first mixing processes; signals (a) and (c) above. This is necessary to prevent these signals from interfering with the desired signal when the second mixing process is performed.
  • a band pass filter 28 typically called a channel filter, which is centred around the IF1 frequency, thus filtering out the unwanted products of the first mixing processes; signals (a) and (c) above. This is necessary to prevent these signals from interfering with the desired signal when the second mixing process is performed.
  • the signal is then amplified by an intermediate frequency amplifier (IFA) 30, and is mixed with a second local oscillator signal using mixer 14 and local oscillator (LO2) 32.
  • the second local oscillator LO2 32 generates a periodic signal which is tuned to modulate the IF1 signal to the desired transmission or carrier frequency.
  • the signal coming from the output of 14 is now at desired transmission frequency.
  • Noise is now filtered from the desired signal using a high pass filter or band pass filter 38, and the signal is amplified by amplifier 40, so that it can now be transmitted. Note that the same process can be used to modulate or demodulate any electrical signal from one frequency to another.
  • the main problems with the in-direct conversion design are: it requires expensive off-chip components, particularly filters 24, 28 and 38; • the off-chip components require design trade-offs that increase power . consumption and reduce system gain; image rejection is limited by the off-chip components, not by the target integration technology; isolation from digital noise can be a problem; and it is not fully integratable.
  • the filters 24, 28 and 38 used in indirect conversion systems must be high quality devices, so electronically tunable filters cannot be used.
  • the only way to use the indirect conversion system in a multi-standard/multi-frequency application is to use a separate set of off-chip filters for each frequency band. Clearly this is not an effective approach to the provision of a multi-standard/multi- frequency transmitter.
  • One aspect of the invention is defined as a circuit for modulating an input signal x(t) to an output signal y(t), the circuit comprising: a first mixer having an input for an RF signal, an input for a first mixing signal f1 and an output for a mixed signal based on the two input signals; a second mixer having an input for an RF signal, an input for a second mixing signal f2 and an output for a mixed signal based on the two input signals, the output providing the output signal y(t), and the output of the first mixer being connected to the RF input of the second mixer; a switch having one input and two outputs, the input for receiving the input signal x(t) and the two outputs being connected to separate ones of the RF signal inputs of the first mixer and the second mixer, whereby the switch can be selectively controlled to direct the input signal x(t) to the input of either the first mixer or the second mixer; a first signal generator, for generating a multi-tonal mixing signal ⁇ 1 and providing the first mixing signal to
  • An alternative aspect of the invention is defined as a circuit for modulating an input signal x(t) to an output signal y(t), the circuit comprising: a first mixer having an input for an RF signal, an input for a first mixing signal f1 and an output for a mixed signal based on the two input signals; a second mixer having an input for an RF signal, an input for a second mixing signal f2 and an output for a mixed signal based on the two input signals, the output providing the output signal y(t), and the output of the first mixer being connected to the RF input of the second mixer; a first signal generator, for generating either a multi-tonal mixing signal ⁇ 1 or a constant value signal, and providing the first mixing signal to the first mixer; a second signal generator, for generating a mono-tonal mixing signal ⁇ 2 and providing the second mixing signal to the second mixer; and a control circuit for controlling the signals generated by the first signal generator and the second generator, the control circuit having two modes: a first mode in which the first signal generator
  • Figure 1 presents a block diagram of a super-heterodyne modulation topology as known in the art
  • Figure 2 presents a block diagram of a modulator topology in a broad embodiment of the invention
  • Figure 3 presents a timing diagram of a set of mixing signals in a broad embodiment of the invention
  • Figure 4 presents a block diagram of a differential modulator topology in an embodiment of the invention
  • Figure 5 presents a timing diagram of a set differential mixing signals plotted in amplitude against time, in an embodiment of the invention
  • Figure 6 presents a block diagram of a differential modulator topology in an alternative embodiment of the invention.
  • FIG. 2 A circuit which addresses a number of the objects outlined above is presented as a block diagram in Figure 2.
  • This figure presents a modulator topology 50 in which an input signal x(t) is up-converted to an output signal y(t), either by mixing it with two mixing signals ⁇ 1 and ⁇ 2 ("pseudo-direct conversion” mode), or by mixing it with only one mixing signal ⁇ 2 ("direct-conversion” mode).
  • Direct-conversion transceivers perform up and down conversion in a single step, using one mixer and one local oscillator.
  • this requires a local oscillator signal ⁇ 2 with a frequency equal to that of the desired carrier frequency.
  • these two mixing signals ⁇ 1 and ⁇ 2 use for pseudo- direct conversion are very different from mixing signals used in normal two-step conversion topologies (such as indirect conversion or superheterodyne topologies). The main difference is that two pseudo-direct conversion mixing signals are used to emulate a single direct-conversion mixing signal, without the usual short comings of direct-conversion.
  • Direct modulation has the advantages of simplified frequency planning, low cost implementation, and compatibility with multiple modulation formats. However, it suffers from limited power and gain control (while maintaining satisfactory performance) in a single, integrated circuit.
  • the proposed transmitter exploits the advantages of both direct modulation and pseudo-direct modulation. At high output/high gain control settings, the transmitter is configured as a direct modulator. At low output/low gain control settings, the transmitter is configured as a pseudo-direct modulator. The net result is an integrated, configurable, multi-mode transmitter. Virtues of the novel transmitter are simplified frequency planning, low cost of implementation, compatibility with multiple modulation formats, and wide output power/gain control range.
  • the incoming signal x(t) is fed to a switch 52 which is controlled by controller 54.
  • the controller 54 is used to select the transmitter mode of operation between direct modulation and pseudo-direct modulation.
  • switch 52 connects the incoming signal x(t) to the input of mixer 56.
  • switch 52 connects the incoming signal x(t) to the input of mixer 58.
  • the controller 54 also controls the operation of the two modulation signal generators ⁇ 1 60 and ⁇ 2 62.
  • the controller 54 sets the operating mode to direct modulation at higher output power/gain control settings and sets the operating mode to pseudo-direct modulation at lower output power/gain control settings.
  • direct modulation mode only the ⁇ 2 signal generator 62 is used, while in pseudo-direct modulation mode, both the ⁇ 1 and ⁇ 2 generators 60, 62 are required.
  • the mode of controller 54 is controlled by an input signal labelled "TXMODE".
  • the TXMODE signal could be generated in a number of ways, but typically will be generated by a digital signal processor (DSP) or an ASIC (application specific integrated circuit).
  • DSP digital signal processor
  • ASIC application specific integrated circuit
  • controller 54 In direct modulation mode, controller 54 will set the frequency of mixing signal ⁇ 2, generated by the ⁇ 2 signal generator 62 to be at the desired carrier frequency.
  • controller 54 will coordinate the ⁇ 1 and ⁇ 2 mixing signal generators 60, 62 to generate a pair of "virtual local oscillator” (VLO) signals ⁇ 1 and ⁇ 2.
  • VLO virtual local oscillator
  • These mixing signals ⁇ 1 and ⁇ 2 are generally referred to herein as VLO signals because they emulate a local oscillator signal; the product ⁇ 1 * ⁇ 2 has significant power at the frequency of a local oscillator signal being emulated.
  • neither ⁇ 1 nor ⁇ 2 have significant power at the frequency of the input signal x(t), the LO signal being emulated, or the output signal ⁇ 1 ⁇ 2 x(t).
  • Mixing signals with such characteristics greatly resolve the problem of self-mixing because the VLO signals simply do not have significant power at frequencies that will interfere with the output signal.
  • VLO signals are described in greater detail hereinafter, but an exemplary pair of ⁇ 1 and ⁇ 2 mixing signals is presented in Figure 3, plotted in amplitude versus time.
  • one of these mixing signals may be a "multi-tonal" signal (multi-tonal, or non-mono-tonal, refers to a signal having more than one fundamental frequency tone.
  • Mono-tonal signals have one fundamental frequency tone and may have other tones that are harmonically related to the fundamental tone), while the other mixing signal may be a mono-tonal signal. Both signals may also be multi-tonal.
  • the oscillator signal f1 used to generate ⁇ 1 in Figure 3 is operating at a frequency that is four times that of ⁇ 2.
  • ⁇ 1 can be generated from the simple logical operation of ⁇ 2 XOR f1.
  • the mixers 56, 58 receive separate ⁇ 1 and ⁇ 2 signals, and mix them with the input signal x(t) using different physical components. Hence, there is no LO signal which may leak into the circuit.
  • f1 does have power at the frequency of the LO signal being emulated, thus care must be taken to isolate it and to minimize any self-mixing that it might cause. This could be done using standard analogue design and layout techniques, as known in the art. These techniques could include, for example:
  • VLO mixing signals and methods of generating them are discussed in greater detail hereinafter, and in many of the Applicant's co-pending patent applications.
  • FIG. 2 implies that various elements are implemented in analogue form, they can also be implemented in digital form.
  • the mixing signals are typically presented herein in terms of binary 1s and 0s, however, bipolar waveforms, ⁇ 1 , may also be used.
  • Bipolar waveforms are typically used in spread spectrum applications because they use commutating mixers which periodically invert their inputs in step with a local control signal (this inverting process is distinct from mixing a signal with a local oscillator directly).
  • the topology of the invention allows an input signal x(t) to be down-converted effectively, using a completely integratable circuit. It is also particularly convenient when applied to the development of multi-standard/multi-frequency devices because no filters are required, and because the mixing signals can be generated and varied so easily. This advantage will become clearer from the description which follows.
  • the preferred embodiment of the invention is presented as a block diagram in Figure 4.
  • This topology is much the same as that of Figure 2, the primary differences being that the topology of Figure 4 handles in-phase (I) and quadrature (Q) signal components, and all of the signals are handled in differential mode.
  • the topology of Figure 4 also includes a number of variable-gain amplifiers, which provide greater flexibility and improved performance, particularly in multi-standard/ multi-frequency applications.
  • Differential signals are signals having positive and negative potentials with respect to ground, rather than a signal with only a single potential with respect to ground.
  • the use of a differential architecture results in a stronger output signal that is more immune to common mode noise than the architecture presented in Figures 2 and 3. If, for example, environmental noise imposes a noise signal onto the input x(t) of Figure 2, then this noise signal will propagate through the circuit. However, if this environment noise is imposed equally on the IP and IN signal inputs of the differential circuit, then the net effect will be zero.
  • Differential amplifiers, mixers and switches are well known in the art.
  • the topology 80 of Figure 4 is also designed to handle in-phase (I) and quadrature (Q) signal components.
  • I in-phase
  • Q quadrature
  • differential signalling is used throughout, generally represented by P and N labels.
  • the in-phase and quadrature components of the input signal are identified as I and Q respectively, and are handled and modulated in two separate signal channels and then merged into a combined signal after modulation has been completed.
  • the differential amplifiers A1 and A2 of Figure 4 buffer and amplify the incoming pairs of baseband signals, IP, IN and Qp, QN.
  • IP is the positive, in-phase component of the incoming signal and IN is the negative, in-phase component of the incoming signal.
  • Qp is the positive, quadrature-phase component of the incoming signal and QN is the negative, quadrature-phase component of the incoming signal. Note that these two amplifiers A1 and A2 are used in both direct modulation and pseudo-direct modulation modes of operation.
  • switches SW1 and SW2 are controlled via circuit block C1 , and are used to select the transmitter mode of operation between direct modulation and pseudo-direct modulation.
  • switches SW1 and SW2 connect the outputs of amplifiers A1 and A2, to the inputs of mixers M3 and M4 respectively.
  • switches SW1 and SW2 connect the outputs of amplifiers A1 and A2, to the inputs of mixers M1 and M2 respectively.
  • Circuit block C1 selects the transmitter mode of operation between direct modulation and pseudo-direct modulation via control of switches SW1 and SW2, and the modulation signal generators 82 and 84 generators in circuit block L1.
  • the circuit block C1 sets the operating mode to direct modulation at higher output power/gain control settings and sets the operating mode to pseudo-direct modulation at lower output power/gain control settings.
  • signal generator 84 In direct conversion mode, only signal generator 84 of circuit block L1 is used, while in pseudo-direct conversion mode, both the 82 and 84 generators are required. As noted above, in direct conversion mode signal generator 84 will generate a pair of I and Q signal components for a single ⁇ 2 modulating signal (at the carrier frequency). In pseudo-direct conversion mode, two mixing signals would have to be generated by signal generator 82, ⁇ 11 and ⁇ 1Q; and two mixing signals would have to be generated by signal generator 84, ⁇ 2l and ⁇ 2Q.
  • the incoming differential local oscillator signals LOP and LON are used by the circuit block L1 to generate the mixing signals. These local oscillator signals are preferably at a frequency which is a multiple or fraction of the actual mixing signals being used. This is desirable to minimize LO leakage into the signal path, which can interfere with useful data.
  • signals LOP and LON are at twice the frequency of the actual LO being used internally, and these signals are divided by 2, using circuit block /2.
  • the mode of circuit block C1 is controlled by an input signal labelled "TXMODE".
  • TXMODE input signal labelled "DSP”
  • DSP digital signal processor
  • ASIC application specific integrated circuit
  • Differential mixers M1 and M2 are used in the pseudo-direct modulation mode of operation. They simply mix the baseband input signals using the differential ⁇ 1 signals described herein above. The output of mixers M1 and M2 are therefore pseudo-IF signals.
  • Differential amplifiers A3 and A4 are then used in the pseudo-direct modulation mode of operation, to vary the signal gain and power of the pseudo-IF signals.
  • the degree of amplification is controlled via the external control signal GC1 to optimise the operation of the circuit in pseudo-direct modulation mode.
  • Differential mixers M3 and M4 are used in both direct modulation and pseudo-direct modulation modes of operation, mixing the signals that they receive, to the final RF frequency. If the circuit is in the direction modulation mode, then the circuit block C1 will cause the mixing signals ⁇ 2l and ⁇ 2Q to simply be oscillator signals at the desired carrier frequency. If the circuit is in pseudo-direct modulation mode, then the circuit block C1 will control the signal generators 82 and 84 to generate complementary pairs of VLO mixing signals ⁇ 1 l and ⁇ 1Q, and ⁇ 2l and ⁇ 2Q.
  • FIG. 5 A method of generating an exemplary pair of differential mixing signals ⁇ 1 P/ ⁇ 1 N and ⁇ 2P/ ⁇ 2N is shown in Figure 5.
  • the signals in Figure 5 are the same as those of Figure 3, except that complementary P and N components are required. That is, a differential oscillator signal f1 P/f1 N runs at four times the frequency of the differential mixing signal ⁇ 2P/ ⁇ 2N. This signal fl P/fl N can generate the differential mixing signal ⁇ 1 P/ ⁇ 1 N, simply using the logical operation of ⁇ 2 XOR f1.
  • the products of the mixing signals, ⁇ 1 P * ⁇ 2P and ⁇ 1 N * ⁇ 2N are clearly equal to the LO signal being emulated.
  • the generation of I and Q mixing signals follows in the same way.
  • the in-phase and quadrature signal paths are then merged using the differential summer ⁇ .
  • the differential, variable gain amplifier A5 is then used to vary the signal gain and power at RF via the external control signal GC2.
  • differential amplifier A6 is used to buffer and amplify the resultant modulated RF signal. Note, of course, that the summer ⁇ , variable gain amplifier A5 and amplifier
  • A6 are used in both the direct modulation and pseudo-direct modulation modes of operation.
  • This circuit is almost the same as that of Figure 4, all of the amplifiers, mixers and summers operating in the same way. As well, this circuit also operates in two modes: direction conversion and pseudo-direct conversion.
  • the most obvious difference between the two circuits is that switches SW1 and SW2 have been removed, and replaced with two switches SW3 and SW4, which were placed between differential amplifiers A3, A4 and differential mixers M3, M4. These two switches SW3, SW4 are used to place the new differential filters F1 , F2 in or out of the circuit. As will be explained, the new low pass filters F1, F2 are used while the circuit is in direction conversion mode.
  • circuit block C2 circuit block L2
  • differential modulation signal generators 92 and 94 are also quite different from the operation of the corresponding components in Figure 4.
  • the circuit of Figure 6 operates in one of two modes, as controlled by the "TX MODE" input to circuit block C2.
  • TX MODE input to circuit block C2.
  • circuit block C2 directs the modulation signal generator 92 to generate constant value signal (i.e. a DC signal).
  • constant value signal i.e. a DC signal.
  • the output of differential mixers M1, M2 is the product of its input signals so (input x constant) will result in an output at the same frequency as the input, making the two differential mixers M1 , M2 act simply as linear gain elements with no frequency translation;
  • circuit block C2 toggles the two switches SW3, SW4 to place the low pass filters F1 , F2 into the circuit. This is done (when required) to improve noise and spurious performance in direct modulation mode. Of course, the filters may not be necessary in all cases, and other manners of filters may also be substituted, depending on system requirements; and 3. circuit block C2 directs the modulation signal generator 94 to generate normal direct-conversion mixing signals, which are fed to differential mixers
  • circuit block C2 directs the modulation signal generators 92, 94 to generate
  • circuit block C2 toggles the two switches SW3, SW4 to place the low pass filters F1 , F2 out the circuit.
  • VLO signals An exemplary set of VLO signals were described hereinabove. The purpose of this section is to present VLO signals in a more general way, as any number of VLO signals could be generated with which the invention could be implemented.
  • VLO virtual local oscillator
  • ⁇ 1 and ⁇ 2 has minimal power around the frequency of the mixer pair output ⁇ 1(t) * ⁇ 2(t) * x(t), while the other has minimal power around the centre frequency, f RF , of the input signal x(t).
  • Minimum power means that the power should be low enough that it does not seriously degrade the performance of the RF chain in the context of the particular ' application.
  • the mixer pair is demodulating the input signal x(t) to baseband in a receiver, it is preferable that one of either ⁇ 1 and ⁇ 2 has minimal power around DC. As a result, the desired demodulation is affected, but there is little or no LO signal to leak into the signal path and appear at the output.
  • mixing two signals together generates an output with: (a) a signal equal in frequency to the sum of the frequencies of the input signals; (b) a signal equal in frequency to the difference between the frequencies of the input signals; and (c) the original input frequencies.
  • direct conversion receivers known in the art must mix the input signal x(t) with a LO signal at the carrier frequency of the input signal x(t). If the LO signal of a direct conversion receiver leaks into the signal path, it will also be demodulated to baseband along with the input signal x(t), causing interference.
  • the invention does not use an LO signal, so leakage does not generate a signal at the baseband output ⁇ 1(t) * ⁇ 2(t) * x(t).
  • the mixing signal ⁇ 2 has some amount of power within the bandwidth of the up-converted RF (output) signal, and it leaks into the signal path, then if will be suppressed by the ⁇ 1 mixing signal which has minimal power within the bandwidth of the up-converted RF (output) signal.
  • This complementary mixing suppresses interference from the mixing signals ⁇ 1 and ⁇ 2.
  • current receiver and transmitter technologies have several problems. Direct-conversion transceivers, for example, suffer from LO leakage and 1/f noise problems which limit their capabilities, while heterodyne transceivers require image-rejection techniques which are difficult to implement on-chip with high levels of performance.
  • VLO signals are complementary in that one of the ⁇ 1 and ⁇ 2 signals has minimal power around the frequency of the output signal y(t) (which is around DC if conversion is to baseband), and the other has minimal power around the centre frequency, f RF , of the input signal x(t).
  • These signals ⁇ 1 and ⁇ 2 can, in general, be:
  • PCT International Application Serial No. PCT/CA00/00994 Filed September 1 , 2000, titled: "Improved Method And Apparatus For Down-Conversion Of Radio Frequency (RF) Signals"; and 3.
  • PCT International Application Serial No. PCT/CA00/00996 Filed September 1 , 2000, titled: "Improved Method And Apparatus For Up-And-Down-Conversion Of Radio Frequency (RF) Signals”.
  • the invention provides many advantages over other up-convertors known in the art. To begin with, it offers:
  • a high level of integration results in decreased IC (integrated circuit) pin counts, decreased signal power loss, decreased IC power requirements, improved SNR (signal to noise ratio), improved NF (noise factor), and decreased manufacturing costs and complexity.
  • the design of the invention also makes the production of inexpensive, configurable, multi-standard/multi-frequency communications transmitters and receivers a reality.
  • multiple transmitters had to be designed to support multiple modes (standards). This resulted in high cost and large physical size.
  • the invention provides a topology that is extremely flexible and configurable.
  • the oscillator signals can easily be changed electronically, as can the degree of gain from the variable-gain amplifiers A3, A4 and A5.
  • the invention can be implement using bipolar technology, CMOS technology, BiCMOS technology, or another semiconductor technology including, but not limited to Silicon/Germanium (SiGe), Germanium (Ge), Gallium Arsenide
  • mixing signals can be generated in many ways, for example, using a voltage controlled oscillator (VCO). Having a VCO at the frequency of the incoming signal can allow self-mixing to occur because the tracks of the printed circuit board (PCB) and pins of integrated circuits act as antennas for the LO signal to radiate. Using a VCO at a different frequency than the incoming signal x(t), and placing a frequency divider or multiplier on chip, minimizes the possibility of self-mixing;
  • VCO voltage controlled oscillator
  • control circuit C1 and control signals GC1 and GC2 may be merged into a single circuit that controls output power/gain and mode of operation;
  • the invention may be applied to various communication protocols and formats including: amplitude modulation (AM), frequency modulation (FM), frequency shift keying (FSK), phase shift keying (PSK), cellular telephone systems including analogue and digital systems such as code division multiple access (CDMA), time division multiple access (TDMA) and frequency division multiple access (FDMA); and
  • AM amplitude modulation
  • FM frequency modulation
  • FSK frequency shift keying
  • PSK phase shift keying
  • CDMA code division multiple access
  • TDMA time division multiple access
  • FDMA frequency division multiple access
  • the mixers used in the topology of the invention could either be passive or active.
  • Active mixers are distinct from passive mixers in a number of ways: a. they provide conversion gain. Thus, an active mixer can replace the combination of a low noise amplifier and a passive mixer; . b. active mixers provide better isolation between the input and output ports because of the impedance of the active components; and c. active mixers allow a lower powered mixing signal to be used.
  • the electrical circuits of the invention may be described by computer software code in a simulation language, or hardware development language used to fabricate integrated circuits.
  • This computer software code may be stored in a variety of formats on various electronic memory media including computer diskettes, CD- ROM, Random Access Memory (RAM) and Read Only Memory (ROM).
  • electronic signals representing such computer software code may also be transmitted via a communication network.
  • Such computer software code may also be integrated with the code of other programs, implemented as a core or subroutine by external program calls, or by other techniques known in the art.
  • DSPs digital signal processors
  • FPGAs field programmable gate arrays
  • wired communication systems include computer communication systems such as local area networks (LANs), point to point signalling, and wide area networks (WANs) such as the Internet, using electrical or optical fibre cable systems.
  • wireless communication systems may include those for public broadcasting such as AM and FM radio, and UHF and VHF television; or those for private communication such as cellular telephones, personal paging devices, wireless local loops, monitoring of homes by utility companies, cordless telephones including the digital cordless European telecommunication (DECT) standard, mobile radio systems, GSM and AMPS cellular telephones, microwave backbone networks, interconnected appliances under the Bluetooth standard, and satellite communications.
  • DECT digital cordless European telecommunication

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Transmitters (AREA)

Abstract

La présente invention se relie généralement aux communications, et plus spécifiquement à un procédé et à un appareil de bande de base de modulation et de signaux de RF (fréquences radio). On met en oeuvre une topologie de modulateur dans laquelle un signal d'entrée x(t) est converti à la fréquence asendante en signal de sortie y(t), soit par mélange à deux signaux de mélange F1 et F2 (mode 'pseudo-conversion directe'), soit par mélange avec seulement un signal de mélange F2 (mode de 'conversion directe'). Dans le mode de modulation de pseudo-conversion directe, les signaux de mélange F1 et F2 émulent un signal local d'oscillateur; le produit F1 * F2 a la puissance à la fréquence d'un signal local d'oscillateur émulé, mais ni F1 ni F2 n'ont la puissance importante à la fréquence du signal d'entrée x(t), le signal LO, ou le signal de sortie F1 F2 x(t) étant émulés.
PCT/CA2004/000014 2003-01-06 2004-01-06 Modulateur miltimodes et emetteur WO2004062087A1 (fr)

Priority Applications (4)

Application Number Priority Date Filing Date Title
CA002512107A CA2512107A1 (fr) 2003-01-06 2004-01-06 Modulateur miltimodes et emetteur
EP04700246A EP1590885A1 (fr) 2003-01-06 2004-01-06 Modulateur miltimodes et emetteur
JP2006500427A JP2006515498A (ja) 2003-01-06 2004-01-06 マルチモードモジュレータ及びトランスミッタ
US10/541,243 US20060141952A1 (en) 2003-01-06 2004-01-06 Multi-mode modulator and transmitter

Applications Claiming Priority (4)

Application Number Priority Date Filing Date Title
US43820203P 2003-01-06 2003-01-06
CA2,415,668 2003-01-06
CA002415668A CA2415668A1 (fr) 2003-01-06 2003-01-06 Emetteur multi-mode integre configurable
US60/438,202 2003-01-06

Publications (1)

Publication Number Publication Date
WO2004062087A1 true WO2004062087A1 (fr) 2004-07-22

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PCT/CA2004/000014 WO2004062087A1 (fr) 2003-01-06 2004-01-06 Modulateur miltimodes et emetteur

Country Status (4)

Country Link
EP (1) EP1590885A1 (fr)
JP (1) JP2006515498A (fr)
KR (1) KR20050088491A (fr)
WO (1) WO2004062087A1 (fr)

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7672645B2 (en) 2006-06-15 2010-03-02 Bitwave Semiconductor, Inc. Programmable transmitter architecture for non-constant and constant envelope modulation
US8090316B2 (en) 2007-09-24 2012-01-03 Microsemi Semiconductor Corp. Digital FM radio transmitter
WO2013136291A3 (fr) * 2012-03-14 2013-11-21 Renesas Mobile Corporation Émetteur
EP3044870A4 (fr) * 2013-09-12 2016-11-09 Vayyar Imaging Ltd Appareil et procédés de génération, de réception et d'auto-étalonnage de signaux

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2001017121A1 (fr) * 1999-09-01 2001-03-08 Sirific Wireless Corporation Technique et dispositif ameliores pour transposition elevatrice de signaux de radiofrequence (rf)
WO2001058103A2 (fr) * 2000-02-02 2001-08-09 Interdigital Technology Corporation Modulation de transposition directe a fuite d'oscillateur local reduite

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2001017121A1 (fr) * 1999-09-01 2001-03-08 Sirific Wireless Corporation Technique et dispositif ameliores pour transposition elevatrice de signaux de radiofrequence (rf)
WO2001058103A2 (fr) * 2000-02-02 2001-08-09 Interdigital Technology Corporation Modulation de transposition directe a fuite d'oscillateur local reduite

Cited By (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7672645B2 (en) 2006-06-15 2010-03-02 Bitwave Semiconductor, Inc. Programmable transmitter architecture for non-constant and constant envelope modulation
US8090316B2 (en) 2007-09-24 2012-01-03 Microsemi Semiconductor Corp. Digital FM radio transmitter
WO2013136291A3 (fr) * 2012-03-14 2013-11-21 Renesas Mobile Corporation Émetteur
US8660503B2 (en) 2012-03-14 2014-02-25 Broadcom Corporation Transmitter
EP3044870A4 (fr) * 2013-09-12 2016-11-09 Vayyar Imaging Ltd Appareil et procédés de génération, de réception et d'auto-étalonnage de signaux
US9553621B2 (en) 2013-09-12 2017-01-24 Vayyar Imaging Ltd. Apparatus and methods for signal generation, reception, and self-calibration
US9813281B2 (en) 2013-09-12 2017-11-07 Vayyar Imaging Ltd Apparatus and methods for signal generation, reception, and self-calibration
US10116486B2 (en) 2013-09-12 2018-10-30 Vayyar Imaging Ltd. Apparatus and methods for signal generation, reception, and self-calibration

Also Published As

Publication number Publication date
EP1590885A1 (fr) 2005-11-02
JP2006515498A (ja) 2006-05-25
KR20050088491A (ko) 2005-09-06

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