Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/02—Amplitude-modulated carrier systems, e.g. using on-off keying; Single sideband or vestigial sideband modulation
  • H04L27/06—Demodulator circuits; Receiver circuits
  • H04L27/066—Carrier recovery circuits
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/06—Receivers
  • H04B1/10—Means associated with receiver for limiting or suppressing noise or interference
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/0014—Carrier regulation
  • H04L2027/0044—Control loops for carrier regulation
  • H04L2027/0053—Closed loops
  • H04L2027/0057—Closed loops quadrature phase
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L7/00—Arrangements for synchronising receiver with transmitter
  • H04L7/02—Speed or phase control by the received code signals, the signals containing no special synchronisation information
  • H04L7/027—Speed or phase control by the received code signals, the signals containing no special synchronisation information extracting the synchronising or clock signal from the received signal spectrum, e.g. by using a resonant or bandpass circuit
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
  • H04N21/00—Selective content distribution, e.g. interactive television or video on demand [VOD]
  • H04N21/40—Client devices specifically adapted for the reception of or interaction with content, e.g. set-top-box [STB]; Operations thereof
  • H04N21/41—Structure of client; Structure of client peripherals
  • H04N21/426—Internal components of the client ; Characteristics thereof

Definitions

• BACKGROUND Traditionally, local communication was done over wires, as this presented a cost-effective way of ensuring a reliable transfer of information. For long-distance communications, transmission of information over radio waves was needed. Although this was convenient from a hardware standpoint, radio frequency (RF) transmission brought with it problems related to corruption of the information and was often dependent on high-power transmitters to overcome weather conditions, large buildings, and interference from other sources of electromagnetic radiation.
• RF radio frequency
• digital modulation can use amplitude, frequency, or phase modulation with different advantages.
• frequency and phase modulation techniques offer more immunity to noise, they are the preferred techniques for the majority of services in use today.
• a simple variation from traditional analog frequency modulation can be implemented by applying a digital signal to the modulation input.
• the output takes the form of a sine wave at two distinct frequencies.
• To demodulate this waveform it is a simple matter of passing the signal through two filters and translating the resultant back into logic levels.
• this form of digital frequency modulation has been called frequency-shift keying.
• phase modulation is very similar to frequency modulation. It involves changing the phase of the transmitted waveform instead of the frequency, these finite phase changes representing digital data.
• a phase-modulated waveform can be generated by using the digital data to switch between two signals of equal frequency but opposing phase. If the resultant waveform is multiplied by a sine wave of equal frequency, two components are generated: one cosine waveform of double the received frequency and one frequency-independent term whose amplitude is proportional to the cosine of the phase shift. Thus, filtering out the higher-frequency term yields the original digital data.
• phase-shift keying a stage further, the number of possible phases can be expanded beyond two.
• the transmitted "carrier" can undergo changes among any number of phases, and multiplying the received signal by a sine wave of equal frequency, will demodulate the phase shifts into frequency-independent voltage levels.
• quadriphase-shift keying With quadriphase-shift keying, the carrier changes among four phases, and can thus represent any of four values per phase change. Although this may seem insignificant initially, it provides a modulation scheme that enables a carrier to transmit two bits of information per symbol instead of one, thus effectively doubling the data bandwidth of the carrier.
• a carrier to which a varying phase shift is applied can be demodulated into a varying output voltage by multiplying the carrier with a sinusoidal output from a local oscillator and filtering out the high-frequency component.
• the phase shift detection is limited to two quadrants; a phase shift of ⁇ /2 cannot be distinguished from a phase shift of - ⁇ /2. Therefore, to accurately decode phase shifts present in all four quadrants, the input signal needs to be multiplied by both sinusoidal and cosinusoidal waveforms, the high frequency filtered out, and the data reconstructed. Expanding on the equations above:
• removing the data from the carrier is not a simple process of low- pass filtering the output of the mixer and reconstructing four voltages back into logic levels.
• exactly synchronizing a local oscillator at the receiver with an incoming signal is not easy. If the local oscillator differs in phase from the incoming signal, the signals on the phasor diagram will undergo a phase rotation of a magnitude equal to the phase difference. Moreover, if the phase and frequency of the local oscillator are not fixed with respect to the incoming signal, there will be a continuing rotation on the phasor diagram. Therefore, the output of the front- end demodulator is normally fed into an analog-to-digital (A D) converter, and any rotation resulting from errors in the phase or frequency of the local oscillator is removed in digital signal processing.
• a D analog-to-digital
• ISI inter-symbol interference
• pulsed information such as an amplitude modulated digital transmission
• analog channel such as, for example, a phone line or an aerial broadcast.
• the original signal begins as a reasonable approximation of a discrete time sequence, but the received signal is a continuous time signal.
• the shape of the impulse train is smeared or spread by the transmission into a differentiable signal whose peaks relate to the amplitudes of the original pulses. This signal is read by digital hardware, which periodically samples the received signal.
• Each pulse produces a signal that typically approximates a sine wave.
• a sine wave is characterized by a series of peaks centered about a central peak, with the amplitude of the peaks monotonically decreasing as the distance from the central peak increases.
• the sine wave has a series of troughs having a monotonically decreasing amplitude with increasing distance from the central peak.
• the period of these peaks is on the order of the sampling rate of the receiving hardware. Therefore, the amplitude at one sampling point in the signal is affected not only by the amplitude of a pulse corresponding to that point in the transmitted signal, but by contributions from pulses corresponding to other bits in the transmission stream.
• the portion of a signal created to correspond to one symbol in the transmission stream tends to make unwanted contributions to the portion of the received signal corresponding to the arrival of other symbols in the transmission stream.
• This effect can partially be eliminated by proper shaping of the pulses, for example by generating pulses that have zero values at regular intervals corresponding to the sampling rate.
• this requires the receiver to sample at a correct time instant to have the maximum signal power and minimum inter- symbol interference. Since the transmitter and receiver normally have different crystal oscillators, a digital receiver should try its best to synchronize with the transmitter clock. In other words, the receiver must extract the clock information from the received signal and then adjust its A/D timing. This is known as symbol clock recovery.
• the Advanced Television Systems Committee has selected vestigial sideband (“VSB”) modulation as the transmission standard for digital television (“DTV”).
• VSB vestigial sideband
• DTV digital television
• 8 VSB is the standard for terrestrial broadcast, while 16 VSB is used for cable transmission.
• ITU International Telecommunications Union
• 8 VSB uses three supplementary signals for synchronization.
• a four-symbol data-segment sync is used once every 832 symbols — that is, once each segment — for synchronizing the data clock in both frequency and phase. (Typically, the four symbols are [1, -1, -1, 1], normalized.)
• an 832- symbol data-frame sync is used once every 313 segments for data framing and equalizer training.
• the data-frame sync also includes information identifying the signal as either 8 VSB, 16 VSB, or one of the other appropriate ITU modes.
• the pilot signal has 0.3 dB power. Although the pilot recovery is typically reliable, it can fail under certain circumstances, such as strong, close-in, slow- moving multipathing situations.
• Figure 2 is a block diagram of a circuit for carrier recovery according to the present invention.
• Figure 3 is a diagram of certain features of a VSB signal passing through certain points of the circuit shown in Figure 2.
• a carrier recovery system according to the present invention provides more robust capture because it can use both the pilot and the upper and lower band edges. It is therefore more reliable, especially in urban environments, where ghosts are common. A carrier recovery system according to the present invention can even capture when the pilot has been completely destroyed by a perfect null.
• Figure 1 shows certain features of the spectrum of a VSB signal, shown generally at 100.
• the primary portion 110 of the signal 100 is 5.38 MHz wide, including an unattenuated portion 105 within the 3 dB attenuated portion 110.
• the amplitude is not completely damped outside the main frequency domain.
• a substantial signal exists in this example for an additional 0.31 MHz above and below the primary portion 110 of the signal, this full band being indicated at 115.
• These "band edges" can be used for carrier recovery, as discussed hereinbelow.
• FIG. 2 is a block diagram of a circuit according to the present invention, shown generally at 200, with signals corresponding to certain points being shown in Figure 3.
• a signal is input to the circuit 200 at 201 from an A/D converter (not shown) preferably running at twice the symbol rate. It will be appreciated that sampling at twice the symbol rate is sufficient to satisfy the Nyquist condition.
• This upstream A/D converter can sample its input signal at greater than twice the symbol rate, but increases in the hardware frequency beyond this point result in increases in the hardware cost without a corresponding increase in performance.
• the circuit 200 comprises a digitally controlled oscillator ("DCO") 210, which produces two signals: sin( ⁇ n), and cos( ⁇ n), where "n" is the symbol count.
• DCO digitally controlled oscillator
• a first multiplier 202 multiplies the input signal by the cos( ⁇ n) signal
• a second multiplier 204 multiplies the input signal by the sin( ⁇ n) signal.
• the outputs from the first and second multipliers 202 and 204 are then passed through first and second root-raised cosine (“RRC") filters 220 and 230, respectively.
• the output from the first RRC filter 220 is multiplied by sin( ⁇ n/4) at a third multiplier 222, and by cos( ⁇ n/4) at a fourth multiplier 224.
• the output from the second RRC filter 230 is likewise multiplied by sin( ⁇ n/4) at a fifth multiplier 232, and by cos( ⁇ n/4) at a sixth multiplier 234.
• the output from the sixth multiplier 234 is subtracted from the output from the third multiplier 222 by a first accumulator 240 and added to the output from the third multiplier 222 by a third accumulator 260.
• the output from the fifth multiplier 232 is subtracted from the output from the fourth multiplier 224 by a second accumulator 250 and added to the output from the fourth multiplier 224 by a fourth accumulator 270.
• the output from the second accumulator 250 is passed through a first low-pass IJR filter 248, preferably having a -3 dB attenuation at 70 kHz to filter out high-frequency components beyond the band edge.
• the output from the IIR filter 248 passes through a first limiter 246.
• the first limiter 246 assigns a value of 1 to any positive input, and a value of -1 to any negative input. (Those skilled in the art will recognize this as a sign() function.)
• the output from the first limiter 246 is multiplied by the output from the first accumulator 240 using a seventh multiplier 280. It will be appreciated by those skilled in the art that the output from the seventh multiplier 280 has been multiplied by two RRC filters, so that the signal has been effectively multiplied by a plain raised cosine filter overall. Thus, the output from the seventh multiplier 280 represents the frequency and phase correction information obtained from the lower band edge.
• the output from the fourth accumulator 270 is passed through a second low-pass IIR filter 268, preferably having a -3dB attenuation at 70kHz to filter out high-frequency components beyond the band edge.
• the output from the filter 268 passes through a second limiter 266.
• the second limiter 266 assigns a value of 1 to any positive input, and a value of -1 to any negative input.
• the output from the second limiter is multiplied by the output from the third accumulator 260 using an eighth multiplier 290. It will be appreciated that the output from the eighth multiplier 290 represents the frequency and phase correction information obtained from the upper band edge.
• the output from the seventh multiplier 280 is then multiplied by a weight factor "k" using a ninth multiplier 285.
• the output from the eighth multiplier 290 is subtracted from the output from the ninth multiplier 285 using a fifth accumulator 295.
• the output from the fifth accumulator 295 is then passed through a third low-pass IIR filter 297 to generate the signal provide to the DCO controller 299, which completes the feedback loop that provides carrier recovery.
• the lower band edge of a VSB signal contains the pilot signal. This is the reason for the weight factor applied by the ninth multiplier 285.
• k is about 0.3 the upper and lower band edge contributions will be properly balanced. Variations in the implementation of the invention will occur to those of skill in the art. For example, some or all of the generation and calculation of signals can be performed by application-specific or general-purpose integrated circuits, or by discrete components, or in software.

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• Engineering & Computer Science (AREA)
• Computer Networks & Wireless Communication (AREA)
• Signal Processing (AREA)
• Digital Transmission Methods That Use Modulated Carrier Waves (AREA)

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Publications (1)

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ID=28791988

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Citations (2)

• 2003
  • 2003-04-04 KR KR10-2004-7014288A patent/KR20050007439A/ko not_active Application Discontinuation
  • 2003-04-04 WO PCT/US2003/010402 patent/WO2003085852A1/en not_active Application Discontinuation
  • 2003-04-04 BR BR0307918-0A patent/BR0307918A/pt not_active IP Right Cessation
  • 2003-04-04 AU AU2003221810A patent/AU2003221810A1/en not_active Abandoned
  • 2003-04-04 CN CN03807615A patent/CN100592644C/zh not_active Expired - Fee Related

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