WO2002052708A1 - Convertisseur elevateur de tension a correction du facteur de puissance et a commutateur unipolaire triphase - Google Patents

Convertisseur elevateur de tension a correction du facteur de puissance et a commutateur unipolaire triphase Download PDF

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Publication number
WO2002052708A1
WO2002052708A1 PCT/CN2001/001646 CN0101646W WO02052708A1 WO 2002052708 A1 WO2002052708 A1 WO 2002052708A1 CN 0101646 W CN0101646 W CN 0101646W WO 02052708 A1 WO02052708 A1 WO 02052708A1
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Prior art keywords
current
phase
output
circuit
loop
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PCT/CN2001/001646
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English (en)
French (fr)
Inventor
Xingzhu Zhang
Huajian Zhang
Yunhua Tan
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Emerson Network Power Co., Ltd.
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Publication of WO2002052708A1 publication Critical patent/WO2002052708A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4216—Arrangements for improving power factor of AC input operating from a three-phase input voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4225—Arrangements for improving power factor of AC input using a non-isolated boost converter
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02P—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN THE PRODUCTION OR PROCESSING OF GOODS
    • Y02P80/00—Climate change mitigation technologies for sector-wide applications
    • Y02P80/10—Efficient use of energy, e.g. using compressed air or pressurized fluid as energy carrier

Definitions

  • the invention relates to a three-phase single-switch rectifier converter with a power factor correction circuit, particularly a step-up converter using discontinuous mode (DCM). .
  • DCM discontinuous mode
  • the three-phase single-switch discontinuous mode boost converter is an approximate power factor correction circuit ( PFC ) (see ARprasad, PD Ziogas, and S. Manias, "An Active Power Factor Correction Technique for Three-Phase Diode Rectifiers", IEEE Power Electronic Specialists Conf. (PESC) Record, pp. 58-66, 1989.).
  • PFC power factor correction circuit
  • This type of power factor correction circuit using the traditional fixed frequency and fixed duty cycle meets the IEC-1000-3-2 Class A harmonic standard with a maximum input power of approximately 5kW. Therefore, for a communication power module with a larger input power, in order to maintain this.
  • the advantage of the simple structure of the PFC circuit can also reach the IEC-1000-3-2 level A harmonic standard. The best way is to modify the control strategy. Modulation duty cycle function.
  • Typical control technology has 6th harmonic.
  • the 6th and 6 ⁇ harmonic injection control is to superimpose the 6th or 6 ⁇ harmonics on the duty cycle control signal to reduce the 5th harmonic and improve the input current waveform.
  • 6th harmonic injection control can be found in Q. Huang and FCLee, "Harmonic Reduction in A Single- Switch, Three-Phase Boost Retifier with High Order Harmonic Injected PWM", IEEE Power Electronics Specialists Conf. (PESC) Record, pp. 1266 -1271,1996.
  • the second type of control method is to use the average current control technology, [see application number 99104662.5, the name is "single-switch three-phase power factor correction method and circuit"], it promotes the single-phase average current PFC control technology, the principle of which is control
  • the three-phase discontinuous conduction mode boost (DCM Boost) circuit rectifies the current on the side bus to follow the three-phase phase voltage wave head, so that the input current can achieve the purpose of correction, thereby reducing the input current harmonics.
  • the average current control technology overcomes the problem of poor stability of large signals in the first method, but the sampling circuit detection circuit of the three-phase voltage wave head is more complicated.
  • the purpose of the present invention is to combine the advantages of the second method and simultaneously solve its disadvantages, and provide a three-phase single-switch power factor correction boost converter, which can reduce the input current harmonics and improve the input current waveform. It also makes the circuit simpler and more reliable. Summary of the Invention
  • a three-phase single-switch power factor correction boost converter is provided, which is characterized by controlling the current (idc) of the DC-side bus of the converter rectifier circuit to be DC.
  • a three-phase single-switch power factor correction boost converter which includes a three-phase input terminal, an output terminal, a three-phase rectifier circuit, a pulse width modulation circuit, and a voltage loop.
  • the three-phase input terminal is respectively connected to a three-phase AC power source, and the output terminal is a direct current.
  • An electronic switch is connected across the output terminal of the rectifier circuit, and the control terminal of the electronic switch is connected to the output terminal of the pulse width modulation circuit.
  • the inverting input terminal of the pulse width modulation circuit is connected to an external input triangle wave signal; a positive terminal of the DC side of the rectifier circuit is connected in series with a freewheeling diode (D), which is characterized in that it further includes a current loop, and the current
  • D freewheeling diode
  • the non-inverting input of the loop is connected to the output of the voltage loop, the inverting input is connected to the output of the current sampling circuit, and the output of the current loop is connected to the non-inverting input of the pulse width modulation circuit.
  • the single-phase average current clamping control technology is applied to the three-phase single-switch discontinuous mode boost circuit and modified and integrated, so that the circuit's high power factor correction has low harmonic current.
  • the characteristics increase the power range.
  • the voltage loop bandwidth is much lower than the network frequency, and the general bandwidth
  • ⁇ 10Hz composed of voltage reference Verf, feedback voltage Vf, voltage error amplifier, and the output is connected to the B terminal of the multiplier.
  • the output is a current inner loop (consisting of a current loop error amplifier, R1R2C1C2, etc.) (fast loop, generally, Frequency> l / 10f S , fs is the reference of switching frequency).
  • the result of the control is to force the current on the DC side of the rectifier circuit to become DC. Since the bus circuit is equal to the current of the phase with the largest absolute value of the phase voltage at any time, the current of each phase has a time of 60 degrees in a half cycle (corresponding to the interval of the phase voltage of 60 degrees to 120 degrees).
  • phase current is approximately a trapezoidal wave, which reduces the input current harmonics, so it can increase the maximum power that meets the IEC-1000-3-2A harmonic standard. Experiments have also proven this. On the other hand, compared with other processing methods, the circuit is very simple and the performance is reliable, especially the input dynamic performance is improved.
  • Figure 1 is a schematic diagram of the principle of a three-phase single-switch discontinuous mode boost converter using constant frequency control in the prior art.
  • FIG. 2 is a schematic diagram of an improved three-phase single-switch discontinuous mode boost power factor correction circuit in the prior art.
  • FIG. 3 is a schematic diagram of another improved three-phase single-switch discontinuous mode boost power factor correction circuit in the prior art.
  • Figure 4 (a) is a schematic diagram of an improved single-phase discontinuous mode boost power factor correction circuit proposed by the present invention.
  • FIG. 4 (b) is a schematic diagram of another improved single-phase discontinuous mode boost power factor correction circuit proposed by the present invention.
  • the present invention detects a current signal at the DC side of a three-phase single-switch discontinuous mode boost converter and forms a current loop amplifier with the output of the voltage loop. The output of the current loop is then compared with the external carrier signal to generate the required signal. Switching duty cycle to achieve the required input current waveform and increase the input power range that meets IEC 1000—3—2 Class A harmonic standards.
  • the invented circuit control structure is similar to the traditional average current-type control technology, and it is a generalization of the single-phase current clamped average control method, so it is called the average current control scheme here. But this kind of control is realized for the first time in a three-phase single-switch discontinuous mode boost converter.
  • Figure 1 is a block diagram of a three-phase single-pass discontinuous mode boost converter using constant frequency control in the prior art. It is a traditional voltage-type control. For example, the bandwidth of a voltage loop is designed to be much lower than the frequency of the power grid (generally For example, if the bandwidth is ⁇ 10Hz), the switching duty cycle can be regarded as a constant during the rectified grid period (300Hz). This simplest three-phase PFC technology can meet the IEC-1000-3-2A harmonic standard. The maximum power is 5KW.
  • Figure 2 is a block diagram of an improved three-phase PFC in the prior art. It superimposes the 6th or 6nth harmonic signal of an input line voltage on the output of the voltage loop to modulate the switching duty cycle to make it in the power grid.
  • the period is not constant-the role of modulation is to appropriately increase the duty cycle near the peak phase voltage; and to reduce the duty cycle appropriately near 60 degrees and 120 degrees to make the input phase current closer to the sine, thereby increasing the satisfaction Maximum power of the IEC-1000-3-2A harmonic standard.
  • FIG. 3 is a block diagram of another improved three-phase power factor correction circuit in the prior art, which has the same principle as the average current control of a boost power factor correction circuit in a single-phase current continuous mode, using a chip such as UC3854, It is only that the detected current signal is the current on the DC side of the rectifier bridge, not the individual phase inductor currents.
  • the waveform signal is a six-pulse wave head that inputs three-phase phase voltages, and the phase voltage waveforms of non-phases. In principle, this control allows each phase of the inductor current to follow its input phase voltage for 60 degrees in a half cycle (corresponding to a phase voltage of 60 degrees to 120 degrees), so it can be improved to meet IEC-1000. Maximum power of -3-2A harmonic standard. But for the three-phase three-wire input, the six-pulse wave head detection circuit of the three-phase input phase voltage is more complicated, and the DC-side current detection circuit of the rectifier bridge is also very complicated.
  • Figures 4 (a) and 4 (b) are block diagrams of an improved three-phase PFC proposed by the present invention.
  • Figure 4 (a) is a very simple average current control technology.
  • the three-phase single-switch power factor correction boost converter shown in the figure includes three-phase input terminals Va, Vb, Vc, output terminals Vo, and three-phase rectification. Circuit, pulse width modulation circuit PWM and voltage loop 1, the three-phase input terminals Va, Vb, Vc are respectively connected to three-phase AC power, the output of Vo is DC, and an electron is connected across the output of the rectifier circuit.
  • a switch S a control end of the electronic switch S is connected to an output end of a pulse width modulation circuit (PWM), and an inverting input end of the pulse width modulation circuit PWM is connected to an external input triangle wave signal.
  • PWM pulse width modulation circuit
  • the non-inverting input of current loop 2 is connected to the output Vc of voltage loop 1, and the inverting input is connected.
  • To the output of the current sampling circuit, and the output of current loop 2 is connected to the non-inverting input of the PWM circuit.
  • the output of the voltage loop (the bandwidth is much lower than the network frequency, and generally, the bandwidth is ⁇ 10Hz) is the reference of the current inner loop (fast loop, generally> l / 10fs, where fs is the switching frequency).
  • the current (Idc) on the DC side of the rectifier bridge is forced to become DC, so that the inductor current of each phase has a time of 60 degrees in a half cycle (corresponding to a phase voltage of 60 degrees to 120 degrees).
  • the interval is DC, and the total phase current is approximately trapezoidal, so it can increase the maximum power to meet the IEC-1000-3-2A harmonic standard.
  • Figure 4 (b) uses the structure of Figure 3 to implement Figure 4 (a).
  • the difference from Figure 4 (a) is that there is a multiplication between the non-inverting input of current loop 2 and the output of voltage loop 1.
  • the waveform voltage output terminal A and the effective value input terminal C of the multiplier 3 are connected to a fixed voltage signal Vcon, and the signal input terminal B is connected to the output terminal of the voltage loop 1. .
  • the multiplier 3 is meaningless and can be completely omitted, but in practical applications, the circuit has a certain guiding significance.
  • a multiplier 3 is provided between the current loop 2 and the voltage loop 1.
  • the multiplier can be "through” using the connection shown in Figure 4 (b). At this time, it is equivalent to the 6-waveform voltage signal in the circuit of Fig. 3 without detection, but replaced by a fixed DC voltage. Makes the realization of control greatly simplified.
  • the circuit of the invention can directly detect the current on the DC side of the rectifier bridge by using a current transformer, so that the implementation is simpler.
  • the average current controlled three-phase single-switch discontinuous mode boost power factor correction circuit implemented by the principle of FIG. 4 can use discrete components or integrated circuits and any kind of current detection circuit, which all belong to the protection scope of this patent.
  • FIG. 4 Three current sensors are shown in FIG. 4. In an actual application circuit, as long as a pair of current sensors detecting id and is or a current sensor detecting idc are used, a corresponding sampling circuit is connected. The connection relationship is shown in Figure 4a.
  • the current sample circuit When idc is used, the current sample circuit is connected in series to the near-earth bus on the DC side of the rectifier circuit, and the sampled signal is the near-earth bus current on the DC side.
  • the superposition of id and is there are two current sampling circuits, and the output signals of the two are superimposed and input to the inverting input terminal of the current loop 2.
  • One of the current sampling circuits is connected in series to the switch of the rectifier circuit.
  • the sampled signal On the diode D, the sampled signal is the diode current id, and the other is connected in series to the switch S, and the sampled signal is the switching current is.
  • the present invention is the first time that the average current control of the current clamp is extended to a three-phase single-switch discontinuous mode boost converter, and a very stable three-phase PFC is realized with very simple detection and control.
  • This solution has been used in three-phase input (304V-456V AC), output voltage of 48V, output current of 100A, communication primary power supply, front stage, single tube discontinuous mode three-phase boost converter, within the rated input voltage and load range Can meet the harmonic requirements of IEC1000-3-2A standard. Compared with specific implementations, it is simpler, more reliable, and has better dynamic characteristics than existing control schemes.
  • the single-phase average current clamping control technology is applied to a three-phase single-switch discontinuous mode boost circuit, and after modification and synthesis, the circuit has a high power factor.
  • the correction has the characteristics of low harmonic current, which increases the power range.
  • the result of the control is to force the current on the DC side of the rectifier circuit to become DC. Since the bus circuit is equal to the current of the phase with the largest absolute value of the phase voltage at any time, the current of each phase has a time of 60 degrees in a half cycle (corresponding to the interval of the phase voltage of 60 degrees to 120 degrees) as a direct current.
  • the phase current is approximately a trapezoidal wave, which reduces the input current harmonics, so it can increase the maximum power that meets the IEC-1000-3-2A harmonic standard. Experiments have also proven this. On the other hand, compared with other processing methods, the circuit is very simple and the performance is reliable, especially the input dynamic performance is improved.

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Description

三相单开关功率因数校正升压变换器 技术领域
本发明涉及一种带有功率因数校正电路的三相单开关整流变换器， 特别是釆用 不连续模式 （DCM) 的升压变换器， 其通称为三相单开关不连续模式升压变换器。 背景技术
三相单开关不连续模式升压变换器是一种近似的功率因数校正电路 （PFC) (参见 A.R.Prasad, P.D. Ziogas, and S. Manias," An Active Power Factor Correction Technique for Three-Phase Diode Rectifiers", IEEE Power Electronic Specialists Conf. (PESC) Record, pp.58-66,1989.) 。 与其它三相功率因数校正电路比较， 它具有结构简 单、 控制方便、 续流二级管无反向恢复问题等优点。 采用传统定频定占空比的这种 功率因数校正电路， 满足 IEC-1000-3-2 A级谐波标准的最大输入功率大约是 5kW。 因而对于更大输入功率的通信电源模块， 为既能保持这种. PFC电路结构简单的优 点， 又能达到 IEC-1000-3-2 A级谐波标准， 最佳的途径就是修改控制策略， 调制占 空比函数。
近几年， 已有一些改进的定频控制方案相继被提出， 它们可以分为两大类： 第 一类是通过高次谐波注入来调制控制占空比： 典型的控制技术有 6次谐波注入控 制、 6η次谐波注入和峰值电流控制三种。 6次和 6η次谐波注入控制是通过在占空比 控制信号上叠加 6次或 6η次谐波来达到减小 5次等谐波的目的， 改善输入电流波 形。 6次谐波注入控制可参见 Q.Huang and F.C.Lee, "Harmonic Reduction in A Single- Switch, Three-Phase Boost Retifier with High Order Harmonic Injected PWM", IEEE Power Electronics Specialists Conf. (PESC) Record, pp.1266-1271,1996. 和 Qihong Huang， " Harmonic Reduction In A Single-Switch Three Phase Boost Rectifer With Harmonic -Injected PWM, Thesis submitted to the Faculty of the Virginia Polytechnic Institute and State University in Partial fulfillment of the requirements for the degree Master of Science in Electrical Engineering ,February 4,1997, Blacksburg，Virgina。 6n次 谐波注入控制可参见 Yangtack Jang, and Milan M. Jovanovie/'Robust, Harmonic Injection Method for Single-Switch, Three Phase, Discontinuous-Conduction-Mode Boost Rectifiers." United States Patent 5847944， Dec 18, 1998。 峰值电流注入控制， 则是利用 了峰值电流的 6n次谐波信息。 [参见专利号为 99121240.1名称为 《带有功率因数校 正电路的三相整流器》 的中国专利申请]等。 采用这些改进的控制策略后， 能大大减 小 5次、 7次等谐波含量， 从而保证了在主电路结构不变的情况下， 增加满足 IEC- 1000-3-2标准的最大输入功率。 尽管这些控制策略都能有效地实现电流谐波含量的减 小， 但各有缺点。 其中前两种的谐波注入控制技术要用复杂的检测电路来产生谐波 注入信号； 后一种峰值电流注入控制技术虽有非常简单的捡测电路， 但仍与前两种 技术一样， 存在大信号的稳定性较差的问题。
第二类控制方法是采用平均电流控制技术， [参见申请号为 99104662.5、 名称为 《单开关三相功率因数校正方法及电路》 ]， 它推广了单相平均电流 PFC控制技术， 其原理是控制三相不连续导电模式升压 （DCM Boost) 电路整流挢侧母线的电流跟随 三相相电压波头， 使输入电流达到校正的目的， 从而减小输入电流谐波。 平均电流 控制技术克服了第一类方法中大信号的稳定性较差的问题， 但三个相电压波头的采 样电路检测电路比较复杂。
本发明的目的就是为了结合第二类方法的优点， 同时解决它存在的弊端， 提供 一种三相单开关功率因数校正升压变换器， 既能减小输入电流谐波， 改善输入电流 波形， 又使电路更加简单可靠。 发明内容
根据本发明的一个方面， 提供一种三相单开关功率因数校正升压变换器， 其特 征在于控制变换器整流电路直流侧母线的电流 （idc) 为直流。
根据本发明的另一个方面， 实现上述目的的方案是： 一种三相单开关功率因数 校正升压变换器， 包括三相输入端、 输出端、 三相整流电路、 脉宽调制电路和电压 环， 所述三相输入端分别接三相交流电源， 输出端的输出为直流电， 在整流电路的 输出端两端跨接一个电子开关， 该电子开关的控制端和脉宽调制电路的输出端相 连， 该脉宽调制电路 （PWM) 的反相输入端与外部输入三角波信号相连； 在整流电 路直流侧的正端串接一个续流二极管 （D) ， 其特征在于： 还包括电流环， 所述电流 环的正相输入端接电压环的输出端， 反相输入端接电流采样电路的输出端， 电流环 的输出端则连接脉宽调制电路的正相输入端。
由于采用了以上的方案， 将单相的平均电流箝位控制技术推广应用于三相单开 关不连续模式升压电路并经过修改和综合， 使电路高功率因数校正具有低谐波电流 的特点， 提高了功率范围。 在这种方案中， 电压环 (带宽远低于网频， 一般带宽
<10Hz, 由电压基准 Verf, 反馈电压 Vf， 电压误差放大器组成， 输出接乘法器的 B 端)的输出是电流内环 （由电流环误差放大器、 R1R2C1C2等组成） （快环,一般来 说， 频率 >l/10fS,fs为开关频率） 的基准。 控制的结果是强制整流电路直流侧的电流 变为直流。 由于任何时刻， 母线电路均等于相电压绝对值最大一相的电流， 因此各 相电流在半周期内都有 60度的时间 （对应于相电压 60度一 120度的间隔） 为直流， 总的相电流近似为梯形波， 减小了输入电流谐波， 故可增加满足 IEC-1000-3-2A谐波 标准的最大功率， 实验也证明了这一点。 另一方面， 与其他处理方式相比， 电路十 分简单， 性能可靠， 尤其是提高了输入动态性能。 附图说明
下面结合附图说明本发明的三相单开关功率因数校正升压变换器的实施例， 以便 更好地理解本发明的目的， 特征和优点。
图 1是现有技术中采用定频控制的三相单开关不连续模式升压变换器的原理示 意图。
图 2是现有技术一种改进型三相单开关不连续模式升压功率因数校正电路示意 图。
图 3是现有技术另一种改进型三相单开关不连续模式升压功率因数校正电路示 意图。
图 4 (a) 是本发明提出的一种改进型单相不连续模式升压功率因数校正电路示 意图。
图 4 (b) 是本发明提出的另一种改进型单相不连续模式升压功率因数校正电路 示意图。
图 5是图 1的功率因数校正电路的输入相电流的示意图 （M=1.4) 。
图 6是图 3的功率因数校正电路的输入相电流的示意图 （M=1.4) 。
图 7是图 4的功率因数校正电路输入在 6kW输出条件下的相电流仿真波形 (M=1.48) 。
下面通过具体的实施例并结合附图对本发明作进一步详细的描述。
本发明是通过检测三相单开关不连续模式升压变换器直流侧的电流信号， 与电 压环的输出构成一个电流环放大器， 电流环的输出再与外部载波信号比较产生所需 的开关占空比， 以实现需要的输入电流波形， 提高满足 IEC 1000— 3— 2 A级谐波标 准的输入功率范围。 发明的电路控制结构与传统的平均电流型控制技术相似， 是单 相电流箝位的平均控制方法的推广， 故在这里叫做平均电流控制方案。 但这种控制 在三相单开关不连续模式升压变换器中的实现尚属首次。
图 1是现有技术中采用定频控制的三相单幵关不连续模式升压变换器框图， 它 是一个传统的电压型控制， 如设计电压环的带宽远远低于电网的频率（一般来说， 带宽 <10Hz) ， 则开关占空比在整流后的电网周期内 （300Hz)可看成常数， 这种最 简单的三相 PFC技术可满足 IEC-1000-3-2A谐波标准的最大功率是 5KW。
图 2是现有技术中的一种改进型三相 PFC框图， 它通过在电压环的输出迭加一 个输入线电压的 6次或 6η次谐波信号以调制开关占空比， 使之在电网周期内不为常 数——调制的作用是在相电压峰值附近， 适当增加占空比； 而在 60度和 120度附 近， 适当减小占空比， 使输入相电流更加接近正弦， 从而增加满足 IEC-1000-3-2A谐 波标准的最大功率。 但都具有大信号稳定性较差的缺点， 其中还要用复杂的检测电 路来产生谐波注入信号。
图 3是现有技术中的另一种改进型三相功率因数校正电路的框图， 它与单相电 流连续模式的升压功率因数校正电路的平均电流控制具有相同的原理， 采用 UC3854 等芯片， 只是检测的电流信号为整流桥直流侧的电流， 非单独的各相电感电流， 波 形信号是输入三相相电压的六脉波头， 非各相的相电压波形。 从原理上， 这种控制 可使各相电感电流在半周期内都有 60度的时间 （对应于相电压 60度一 120度的间 隔）跟随其输入相电压， 故可改进增加满足 IEC-1000-3-2A谐波标准的最大功率。 但 对于三相三线制输入， 三相输入相电压的六脉波头检测电路比较复杂， 整流桥直流 侧电流检测电路也很复杂。
图 4 (a) 、 4 (b)是本发明提出的改进型三相 PFC框图。 其中图 4(a)是非常简 单的一种平均电流控制技术， 该图所示三相单开关功率因数校正升压变换器包括三 相输入端 Va、 Vb、 Vc、 输出端 Vo、 三相整流电路、 脉宽调制电路 PWM和电压环 1， 所述三相输入端 Va、 Vb、 Vc分别接三相交流电源， 输出端 Vo的输出为直流 电， 在整流电路的输出端两端跨接一个电子开关 S， 该电子开关 S的控制端和脉宽调 制电路 (PWM)的输出端相连， 该脉宽调制电路 PWM的反相输入端与外部输入三角波 信号相连。 其中电流环 2的正相输入端连接到电压环 1的输出 Vc， 反相输入端连接 到电流采样电路的输出端， 电流环 2的输出端则连接到脉宽调制电路 PWM的正相输 入端。
在这种方案中， 电压环 (带宽远低于网频， 一般来说， 带宽 <10Hz)的输出是电流 内环 （快环， 一般 >l/10fs,fs为开关频率） 的基准。 通过平均电流箝位控制的结果是 强制整流桥直流侧的电流（Idc)变为直流， 从而可使各相电感电流在半周期内都有 60 度的时间 （对应于相电压 60度~120度的间隔） 为直流， 总的相电流近似为梯形波， 故可增加满足 IEC-1000-3-2A谐波标准的最大功率。
图 4(b)是采用图 3的结构来实现图 4(a), 与图 4(a)的区别是在电流环 2的正相输 入端和电压环 1的输出端之间还接有乘法器 3, 该乘法器 3的波形电压输出端 A和有 效值输入端 C与一个固定的电压信号 Vcon相连， 信号输入端 B与电压环 1的输出端 相连。 .
虽然从理论上讲， 该乘法器 3是没有意义的， 完全可以省略， 但在实际应用 中， 该电路具有一定的指导意义。 为了应用图 3所示的电路， 已有现成的集成电路 芯片， 在该芯片中在电流环 2和电压环 1之间有一个乘法器 3。 为了应用现成的芯 片， 可以用图 4(b)所示接法将乘法器 "贯通" 。 此时相当于图 3电路中 6波形电压信 号不用检测， 而用一个固定的直流电压替代。 使得控制实现大大简化。
本发明的电路可直接用电流互感器检测整流桥直流侧的电流， 使实现更加简 单。
用图 4原理实现的平均电流控制三相单开关不连续模式升压功率因数校正电 路， 可采用分离元件或集成电路以及任何种电流检测电路， 它们都属本专利的保护 范围。
图 4中示出三个电流传感器， 在实际的应用电路中， 只要采用检测 id、 is的一 对电流传感器或检测 idc的电流传感器即可， 并接上相应的取样电路。 其连接关系见 图 4a所示。
当采用 idc时， 所述电流釆样电路串接于整流电路直流侧近地端母线上， 其采 样的信号为直流侧近地端母线电流 idc。 当釆用 id、 is的迭加时， 所述电流采样电路 有两个， 二者的输出信号迭加后输入到电流环 2的反相输入端， 其中一个电流采样 电路串接于整流电路开关二极管 D上， 其采样的信号为二极管电流 id， 另一个串接 于开关管 S上， 其采样的信号为开关电流 is。 下面仅以图 4 (a) 、 4 (b) 的框图为例介绍本发明的工作原理。 传统单开关三 相不连续模式升压变换器的输入电流归一化波形如图 5所示。 采用图 3的改进型三 相 PFC， 其输入电流波形如图 6所示。 可见能大大减小输入电流的各次谐波含量。
在保证图 4 (a) 、 4 (b) 的电路参数满足条件 (1): 三个电感的电流为 DCM; 条件 (2): 电压环带宽很低 (一般来说， 带宽 <10Hz)， 即在整流后的电网周期内， 电压 环输出可近似为常数； 和条件 (3): 三个电感 La = Lb = Lc;那么当图 4 (a) 、 4 (b) 中的电流环被设计成足够快时 （一个实验例子是： R1=20K, R2=100K， Cl=470pF, C2=0.01uF) ， 其输入电流波形将如图 7所示， 与图 6相比， 此输入电流的各次谐波 含量已有明显减小。 通过调节电流环的参数， 还可使它具有最小的失真。
本发明是把电流箝位的平均电流控制首次推广到了三相单开关不连续模式升压 变换器中， 用非常简单的检测和控制实现了相当稳定的三相 PFC。 本方案已用于三 相输入（304V-456V AC) ， 输出电压 48V输出电流为 100A的通信一次电源前置级 单管不连续模式三相升压变换器中， 在额定输入电压和负载范围内都能满足 IEC1000-3-2A标准谐波要求。 从具体实现比较， 它比已有的各种控制方案都要简 单， 可靠， 且动态特性更好。 工业实用性
根据本发明的三相单开关功率因数校正升压变换器， 将单相的平均电流箝位控 制技术推广应用于三相单开关不连续模式升压电路并经过修改和综合， 使电路高功 率因数校正具有低谐波电流的特点， 提高了功率范围。 控制的结果是强制整流电路 直流侧的电流变为直流。 由于任何时刻， 母线电路均等于相电压绝对值最大一相的 电流， 因此各相电流在半周期内都有 60度的时间 （对应于相电压 60度一 120度的间 隔） 为直流， 总的相电流近似为梯形波， 减小了输入电流谐波， 故可增加满足 IEC- 1000-3-2A谐波标准的最大功率， 实验也证明了这一点。 另一方面， 与其他处理方式 相比， 电路十分简单， 性能可靠， 尤其是提高了输入动态性能。

Claims

m 妥豕 l.一种三相单开关功率因数校正升压变换器， 其特征是控制变换器整流电路 直流侧母线的电流（idc) 为直流。
2.一种三相单开关功率因数校正升压变换器， 包括三相输入端 (Va、 Vb、
Vc；)、 输出端 （Vo) 、 三相整流电路、 脉宽调制电路 （PWM) 和电压环 （1 ) ， 所述 三相输入端 (Va、 Vb、 Vc)分别接三相交流电源， 输出端 （Vo) 的输出为直流电， 在 整流电路的输出端两端跨接一个电子开关 (S)， 该电子开关 (S)的控制端和脉宽调制电 路 (PWM)的输出端相连， 该脉宽调制电路 (PWM)的反相输入端与外部输入的三角波 信号相连， 在整流电路直流侧的正端串接一个续流二极管 （D) ,其特征在于： 还包 括电流环（2) ， 所述电流环 （2) 的正相输入端接电压环 （1 ) 的输出端 （Vc) ， 反 相输入端接电流采样电路的输出端， 电流环 （2 ) 的输出端则连接脉宽调制电路 (PWM) 的正相输入端。
3.如权利要求 2所述的三相单开关功率因数校正升压变换器， 其特征在于： 在电流环 （2) 的正相输入端和电压环 （1 ) 的输出端之间还接有乘法器 （3) ， 该乘 法器 （3 ) 的波形电压输入端 （A) 和有效值输入端 （C) 与一个固定的电压信号 Vcon相连， 信号输入端 （B)与电压环 （1 ) 的输出端相连。
4.如权利要求 2或 3所述的三相单开关功率因数校正升压变换器， 其特征在 于： 所述电流采样电路串接于整流电路直流侧母线上， 其采样的信号为直流侧母线 电流 ( idc) 。
5.如权利要求 2或 3所述的三相单开关功率因数校正升压变换器， 其特征在 于： 所述电流采样电路有两个， 二者的输出信号迭加后输入到电流环 （2) 的反相输 入端， 其中一个电流采样电路串接于续流二极管 （D)上， 其釆样的信号为开关二极 管的电流 （id) ， 另一个电流采样电路串接于开关管 （S) 上， 其釆样信号为开关电 流 ( is) 。
PCT/CN2001/001646 2000-12-26 2001-12-26 Convertisseur elevateur de tension a correction du facteur de puissance et a commutateur unipolaire triphase WO2002052708A1 (fr)

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