WO1997039543A9 - Systeme de calcul de temporisation de liaison pour un systeme de repeteurs radio - Google Patents

Systeme de calcul de temporisation de liaison pour un systeme de repeteurs radio

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Publication number
WO1997039543A9
WO1997039543A9 PCT/US1997/005984 US9705984W WO9739543A9 WO 1997039543 A9 WO1997039543 A9 WO 1997039543A9 US 9705984 W US9705984 W US 9705984W WO 9739543 A9 WO9739543 A9 WO 9739543A9
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WO
WIPO (PCT)
Prior art keywords
signal
delay
link
periodic component
gated
Prior art date
Application number
PCT/US1997/005984
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English (en)
Other versions
WO1997039543A3 (fr
WO1997039543A2 (fr
Filing date
Publication date
Priority claimed from US08/631,866 external-priority patent/US5896560A/en
Priority claimed from US08/628,981 external-priority patent/US5991309A/en
Application filed filed Critical
Priority to AU27262/97A priority Critical patent/AU2726297A/en
Publication of WO1997039543A2 publication Critical patent/WO1997039543A2/fr
Publication of WO1997039543A9 publication Critical patent/WO1997039543A9/fr
Publication of WO1997039543A3 publication Critical patent/WO1997039543A3/fr

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Definitions

  • the present invention relates to a system for measuring and correcting temporal latencies in broadcast systems. More particularly, the present invention relates to a system for characterizing and correcting relative link delays using a discrete-time Fourier transform.
  • Repeater based systems employ links from a central site to coordinate transmissions by the repeaters.
  • the links introduce time delays which vary from system to system and even from link to link.
  • the link delay calculation system must provide extremely accurate measurements of link delay.
  • the link delay system should also be robust so that errors in reception are averaged out to give a precise estimation of link delay.
  • the present system provides a link delay measurement system using a discrete-time Fourier transform.
  • a pulsed tone signal is transmitted in a repeater system having a plurality of remote sites.
  • the pulsed tone signal is aligned to an external timing reference.
  • the signal is received by the remote sites, which are also aligned to the timing reference.
  • the link delay is calculated using a discrete-time Fourier transform of the pulsed tone signal which is performed on samples aligned to a timing reference signal.
  • a global positioning satellite (GPS) system is used to align the pulsed tone signal which is transmitted by the central site of the repeater system.
  • the GPS signal is also used by the remote sites to receive the pulsed tones. Compensation of a number of links is performed by introducing delays according to the link delays.
  • FIG. 1 is a block diagram showing one environment in which the present invention may be practiced
  • Figure 2 is a block diagram of a receiver for measuring the relative phase difference between different transmissions
  • Figure 3 is a diagram of one environment in which the present alignment system may operate
  • Figure 4 is a diagram of one environment in which the present alignment system may operate;
  • Figure 5 is a plot of relative phase difference as a function of axis ratio according to one embodiment of the present invention
  • Figure 6 is a plot of the relative phase difference as a function of the square of the length of the minor axis according to one embodiment of the present invention
  • Figure 7 is a flow diagram showing the alignment procedure according to one embodiment of the present invention
  • Figure 8 is a flow diagram showing a coarse alignment and fine alignment according to one embodiment of the present invention
  • Figure 9 is a flow diagram showing one example of signal comparison and transmitter alignment according to one embodiment of the present invention.
  • Figure 0 is a diagram showing a directional antenna receiver system according to one embodiment of the present invention.
  • Figure 11 A is a block diagram of the pulsed tone signal as transmitted by the control site
  • FIG. 1 IB is a block diagram of the pulsed tone signal as received by the remote site
  • Figure 12 shows prediction of the zero crossing between adjacent sample pairs
  • Figure 13 shows error in predicted or estimated zero crossings with respect to the position of samples relative to "true" zero crossing;
  • Figure 14 shows error magnitude as a function of signal to noise ratio ;
  • Figure 15 shows standard deviation as a function of signal to noise ratio
  • Figure 16 shows the calibration module and the interrelationship of Figures 16A, 16B, 16C, 16D, 16E, and 16F according to one embodiment of the present invention
  • Figure 16 A shows how the calibration module stores sample pairs according to one embodiment of the present invention
  • Figure 16B shows how the calibration module validates sample pairs according to one embodiment of the present invention
  • Figure 16C shows how the calibration module calculates zero crossings for valid transition sample pairs according to one embodiment of the present invention
  • Figure 16D shows how the calibration module averages predicted zero crossings according to one embodiment of the present invention
  • Figure 16E shows how the calibration module locates positive and negative peaks according to one embodiment of the present invention
  • Figure 16F shows how the calibration module calculates total delay according to one embodiment of the present invention
  • Figure 17 is an illustration of measurement of link delay using threshold triggering and a discrete-time Fourier transform.
  • Figure 1 shows one environment in which the present invention may be practiced.
  • two remote sites 110 and 120 contribute to the total signal received by receiver 130 in the overlap region.
  • the audio signal at the central site 100 is split and passes to each remote site over a link consisting, at each end, of a multiplexer 112 and a microwave transceiver 114.
  • Other links may be used without departing from the scope of the present invention.
  • microwave links the system could employ optical fiber links.
  • the system in Figure 1 shows two channels, A and B, to demonstrate the present system. Other numbers of channels are possible without departing from the scope of the present invention.
  • delay p .a. 2 delay Mulliptar ⁇ 2 , + delayu ⁇ + length x 5.37 ⁇ sec / mile + delay MiCTOTrave 2b + delay Mu ⁇ plora2b + delay Site2 + length 2b x 5.37 ⁇ sec / mile
  • the additional delay is added to the site delay 110 or 120, however several other embodiments exist.
  • the additional delay is introduced at the central site 100.
  • Other delay sources are possible without departing from the present invention.
  • the present system is also applicable to the case where a transmitting site is located at the central site 100, except that in this case one of the signal paths does not contain any multiplexer or microwave equipment.
  • one approach is to select one repeater of a particular channel as a reference transmitter and temporally align all of the other repeaters transmitting on a different channel to the reference transmitter. After all of the other repeaters transmitting on a different channel are temporally aligned, the remaining transmitters of the particular channel are aligned. This alignment is performed such that the simulcast situation is avoided throughout the alignment. The alignment is more reliable if simulcast is avoided, since there is no distortion due to the out-of-phase components at the receiver 130.
  • Figure 7 shows the operation of this embodiment.
  • a reference transmitter is selected (710) and a second transmitter is selected which has a different channel (720). The second transmitter is aligned to the first transmitter (730).
  • the remaining transmitters are aligned so that the simulcast condition is avoided (740). Alignment is accomplished by the method shown in Figures 8 and 9. Alignment is performed using a coarse adjustment (810) and a subsequent fine adjustment (820). .Alignment is performed by demodulating a first transmitter channel (910) and a second transmitter channel (920), comparing the signals and implementing a delay in the leading transmitter to equalize the delay times (930). The delay implementation must be consistent so that the reference channel remains at a constant phase, so that all the other transmitters are aligned to it. The reference transmitter receives an initial delay which exceeds the maximum possible delay difference between any two transmissions in the system. This maximum delay ensures that all of the remaining transmitters may be aligned to the reference by adding delay.
  • one embodiment may provide a global reduction of the excess delay imposed on the transmitters after all transmitters are aligned. This would efficiently zero the smallest delay value of each transmitter to globally reduce the latency of the transmitter system.
  • Other variations are possible without departing from the scope and spirit of the present invention.
  • channel A repeater 116 of remote site 1 110 is designated as a reference transmitter
  • the channel B transmitters of the system are temporally aligned to channel A repeater 116.
  • channel B repeater 118 and channel B repeater 128 are temporally aligned to the reference transmitter, repeater 116.
  • the remaining channel A transmitters, besides the reference transmitter are aligned to at least one of the aligned channel B transmitters.
  • the channel A repeater 126 of remote site 2 120 is aligned either to (1) the channel B repeater 118 or (2) the channel B repeater 128 using the temporal alignment technique discussed herein. In either of these possible alignments, the simulcast condition is avoided, since like-channel alignments are avoided.
  • the selection of a reference transmitter serves to provide a "starting point" for the calibration of all transmitters in the system. As each transmitter is aligned, it too can serve as a reference for another transmitter of different channel. Therefore, the order of the alignment of the transmitters may be varied. For instance, in the above example, presuming that the channel A repeater 116 is used as an original reference, the following alignment permutations are possible:
  • Channel B repeater 118 aligned relative to Channel A repeater 116
  • Channel B repeater 128 aligned relative to Channel A repeater 116
  • Channel A repeater 126 aligned relative to Channel B repeater 128
  • Channel B repeater 128 aligned relative to Channel A repeater 116
  • Channel B repeater 118 aligned relative to Channel A repeater 116
  • Channel A repeater 126 aligned relative to Channel B repeater 128
  • Channel B repeater 118 aligned relative to Channel A repeater 116
  • Channel A repeater 126 aligned relative to Channel B repeater 118
  • Channel B repeater 128 aligned relative to Channel A repeater 116
  • Channel B repeater 128 aligned relative to Channel A repeater 116
  • Channel A repeater 126 aligned relative to Channel B repeater 128
  • Channel B repeater 118 aligned relative to Channel A repeater 116
  • Channel B repeater 118 aligned relative to Channel A repeater 116
  • Channel A repeater 126 aligned relative to Channel B repeater 118
  • Channel B repeater 128 aligned relative to Channel A repeater 126
  • Channel B repeater 128 aligned relative to Channel A repeater 116 Channel A repeater 126 aligned relative to Channel B repeater 128
  • the flexibility in transmitter alignment of the present invention facilitates alignment of different repeater topologies.
  • the alignment method may be used to align the various transmitters found in Figure 3.
  • Figure 3 shows four multichannel repeaters 302, 304, 306, and 308 having reception regions 312, 314, 316, and 318, respectively, and a central station 320.
  • the present alignment system provides a consistent calibration for all transmitters in the system, since a first reference transmitter is used to align the remaining transmitters.
  • the overlapping transmission areas, K, L, M, and N present a number of alignment permutations, as demonstrated in the previous example.
  • Other topologies may be aligned using the present method and the topologies discussed herein are for illustration of alternate embodiments of the present invention and are not intended in an exclusive or limiting sense.
  • each transmitting source is aligned depends on whether an aligned transmitter of the opposite channel is present within the simulcast region of interest.
  • Figure 4 shows an alternate repeater topology having overlapping reception areas Q, R, and S. If one of the transmitters is used as a reference, the remaining transmitters may be aligned to the reference transmitter. Every transmitter in Figure 4 may be aligned to the reference transmitter since there is an overlap region linking each of the repeaters.
  • This system for alignment may be extended to applications where more than two transmitting channels are present. For example, in one embodiment, a reference channel is selected and the "other-channel" transmitters are aligned. After these transmitters are aligned, the remaining transmitters are aligned, and the simulcast situation is avoided during alignment, as described above. As illustrated above, several alignment permutations are possible without departing from the scope and spirit of the present invention.
  • the temporal alignment procedure is performed by having a special receiver
  • FIG. 130 capable of receiving the different channels present in the overlap region, as shown in Figure 2.
  • Two radios a channel X radio 210 and a channel Y radio 220, receive different channels and output a signal which is fed into comparison equipment 230 to measure the phase difference between signals received from the simulcast.
  • Figure 2 shows the signals 240 and 250 as received from different remote sites, however, it is contemplated within the present invention that the signals could originate from the same remote site for temporal alignment as described above. Additionally, Figure 2 shows only two radios, however, this block diagram is illustrative and is not intended in an exclusive or limiting sense. In alternate embodiments, several channels may be received by several radios without departing from the scope of the present invention.
  • each radio 210 and 250 are compared, giving the phase and time delay between the signals.
  • the transmitters are selectively activated such that each radio receives transmissions from a single channel source per alignment. That is, there is no simulcast condition during the alignment process.
  • repeaters are disabled as needed to ensure that the simulcast of same-frequency signals is not received by the receiver 130.
  • the alignment is performed using a coarse adjustment and a fine adjustment, as shown in Figure 8.
  • a periodic signal may be used to establish a phase relationship between the different channels during temporal alignment.
  • a pulse-like signal is used as the audio signal at the central site 100.
  • a Hewlett-Packard 3314A Function Generator has a mode in which it produces a burst of approximately one and one-half cycles of a tone. These bursts must be separated in time by a period considerably longer than the total path delays. In this way each radio in the overlap region will produce a burst which can, for example, be displayed on an oscilloscope and can provide direct and quick measurement of the difference between the path delays.
  • Other methods for phase detection may be employed, and the use of an oscilloscope is not intended to be limiting or exclusive.
  • a computer having sound analysis software measures the time delay between tone bursts.
  • audio signals may be substituted, such as the use of a digital pulsetrain for performing the coarse adjustment.
  • the digital pulsetrain has a periodicity which is longer than the total path delay to facilitate identification of the path delay.
  • the latency measured between pulse rising or falling edges is recorded and used to temporally align the system. The appropriate delay is then implemented in the transmitter which is being aligned. In this way, one transmitter is brought into approximate temporal alignment with the reference transmitter.
  • a pure tone is used as the central site audio signal for fine adjustment (820).
  • the audio signals produced by the radios in the overlap will also be tones, with some small time delay between them which is determined by the accuracy of the coarse adjustment.
  • One method of comparing the tones to determine the time delay between them is to use them as the inputs of an oscilloscope which is set on X-Y display. If the two signals are in-phase (no time delay between them) then the display will show a line. In the most general case, however, the display will show an ellipse. The ratio:
  • the time delay is calculated using:
  • time delay (seconds) (phase delay (degrees) / 360) (1 / frequency of tone (Hz))
  • Figure 5 shows how phase delay and the square of [length of minor axis] are related. Again, the appropriate delay is implemented in the channel which is being aligned.
  • the procedure is not limited to the two transmitter embodiment described above.
  • one channel e.g., channel A
  • site 1 e.g., site 1
  • All other channels of all sites are timed to it.
  • channel A of all sites other than site 1 are timed to a channel of site 1 other than A.
  • all channels at all sites have been referenced, directly or indirectly, to channel A of site 1.
  • this problem is overcome by using an averaging technique. Temporal alignment is performed as stated above. Next, the radios are tuned to the opposite channel and the measurement is repeated.
  • fine tuning is accomplished by a variable frequency generator which provides a tone whose frequency is swept across the audio spectrum instead of a single frequency tone, as discussed above. For example, for the fine adjustment, the tone frequency is swept from 300Hz to 3000Hz and the time delay information is recorded to map the time delay as a function of audio frequency. This method characterizes the time delay for various paths for different frequencies in the audio spectrum.
  • any periodic signal is used to provide the fine adjustment.
  • the periodic signal may be an audio tone or a digital pulsetrain.
  • the periodic signal has a first periodic component which has a periodicity which is larger than the anticipated relative time delays for a coarse adjustment, and a second periodic component which provides relatively fine resolution for a fine adjustment. Therefore, in this embodiment, the fine tuning is achieved by measuring the relative delay of some characteristic portion of the periodic signals arriving simultaneously at the remote receiver, and calculating a time correction factor from the relative delay.
  • Channels may be different frequencies, such as different AM, FM, or Single Sideband communications frequencies.
  • channels contemplated by this application are not limited to these examples and expressly include several other communications channels.
  • channels includes spread spectrum communications, digital communications, such as packetized communications, or any other communications mode whereby independent communications may take place, such as optical communications. Therefore, other channels are possible without departing from the scope of the present invention.
  • the link equipment may exhibit different delays at different audio frequencies.
  • one equalized path delays at one particular audio frequency then the delay at other frequencies would not, in general, be precisely equalized. This would introduce distortion into the received signal.
  • One method commonly used to alleviate frequency dependent delays is to introduce a compensating filter which has an inverse frequency dependent delay characteristic.
  • These compensating filters are typically allpass filters, which can be implemented using discrete circuit elements or through Digital Signal Processing techniques.
  • One embodiment of the present invention uses a software driven analysis program for audio analysis, such as Spectra Plus version 3.0 from Pioneer Hill Software, 24460 Mason Rd, Poulsbo, WA 98370.
  • This program combines a variety of instrument functions into one package. For example, one can look at the amplitudes of signals as a function of time, as with an oscilloscope. There are also operating modes for examining signal amplitude or phase versus frequency. Other capabilities include audio signal generator and distortion analysis modes.
  • the software analysis program provides a compact and unified analysis means for monitoring the phase data. Other sound analysis methods may be used without departing from the scope and spirit of the present invention.
  • the calibration is performed once upon the setup of the system, and then subsequently as needed. In alternate embodiments, the calibration is performed regularly to ensure system performance.
  • FIG. 10 shows a system having a receiver 1010 which has dual directional antennas.
  • the first antenna has an anisotropic reception pattern which is directed toward the transmitter at site 1.
  • the second antenna is pointed to receive signals from the transmitter at site 2.
  • This system provides alignment of the two transmitters at site 1 and site 2 using the phase detection methods described herein.
  • the transmissions are on the same channel, so the receiver is located within a simulcast overlap region, however, the signals are discriminated using the directional antennas at the receiver 1010.
  • the directional nature of the antennas provides adequate signal separation so that the signals from the different transmitters may be discriminated, even when the simulcast condition is present.
  • the receiver includes two or more radios which may be tuned to same or different channels.
  • Each antenna is pointed at a particular transmitter and the alignment procedure is performed using the phase detection and delay compensation system described above. This system works with the simulcast condition, since the decibel separation between transmitters provides the adequate signal separation to perform the delay measurements without the distortion present generally to isotropic receivers.
  • Performance of this system is improved as the directionality of the channels, and the decibel separation of each transmitter is increased.
  • Alternate embodiments may incorporate different numbers of receivers and directional antennas without departing from the scope and spirit of the present invention.
  • link delays are measured using signals from the global positioning satellite (GPS) system.
  • the first step involves the control site transmitting several cycles of a tone on a 1 pulse-per-second (pps) signal generated by the local GPS unit. This process is illustrated in Figures 11 A and 1 IB.
  • samples are collected following detection of the 1 pps signal from the remote site's GPS unit. These samples are analyzed in order to estimate the times when the received signal crosses zero.
  • the simplest method for performing this operation is to determine when sign changes occur between adjacent samples. One then interpolates, using the two samples whose signs are different, in order to estimate when the actual signal crossed zero. By having the test signal contain several cycles one is able to average many zero crossings, which improves the accuracy of this technique when noise is present.
  • the theoretical limitations of this method are best understood by considering different values of the link S/N, which is the approach taken in the following discussion.
  • a calibration module processes a buffer of data to calculate the link delay between a link transmitter and a link receiver.
  • Delay information is digitally sampled in the time domain according to the One Pulse Per Second (PPS) signal of the Global Positioning System. Alternate embodiments use any other common timing reference signal.
  • PPS Pulse Per Second
  • the calibration module begins processing information when the buffer fills with data; for example, 512 samples representing a 1 KHz tone sampled at 8000 samples second.
  • the calibration module skips over a leading delay and locates a predetermined threshold value which indicates the place where the tone starts.
  • the sample number where the data exceeds the threshold value is stored is indicative of the number of clock periods of the link delay.
  • the threshold value is approximately 0.2 volts in one system, however, other buffer sizes and threshold values may be used without departing from the scope and spirit of the present invention.
  • the calibration module finds transition sample pairs which are indicative of a zero crossing event.
  • the transition sample pairs are where the tone signal transitions from a positive value to a negative value and where the tone signal transitions from a negative value to a positive value.
  • the calibration module stores the sample numbers where transitions are located and the value of each of the transition sample pairs. This process continues until a fixed number of transition sample pairs are found or all of the samples are processed. For example, in one embodiment 18 transition sample pairs are located or all the 512 samples are processed, whichever comes first.
  • the calibration module validates the transition sample pairs by counting the number of samples between each transition sample pair.
  • the pairs which are determined to be in the proper spacing are deemed “valid" transition sample pairs and the remaining transition sample pairs are labeled "invalid.”
  • the valid transition sample pairs are flagged as such and the invalid samples are flagged as invalid. For example, in one embodiment a logic one bit is associated with the valid pairs and a logic zero value is associated with the invalid pairs.
  • the calibration module calculates and stores the predicted crossing point for each valid transition sample pair.
  • a delay for each zero crossing is found by comparing the magnitude of each sample in the pair to obtain a fractional result.
  • the result is compared to a series of entries in a delay table to indicate where the zero crossing is estimated to occur and its associated delay value.
  • the delay value is added when the second sample exceeds the first sample, and it is subtracted when the first sample exceeds the second sample of the valid transition sample pairs.
  • the delay table is precalculated and stored as a lookup table to reduce the amount of time needed to calculate the latency associated with each possible first sample:second sample ratio.
  • the samples are always associated with a fractional value and the delay is added or subtracted as demonstrated above.
  • the calibration module averages the predicted zero crossing points and stores the average value. This averaging reduces the effects of random amplitude noise.
  • the calibration module locates and stores the peak positive and peak negative values and their associated sample numbers in a buffer.
  • the peak positive and peak negative values are then validated by measuring the samples between the peaks and the valid peaks are marked.
  • the first valid peak is then used to predict the first positive to negative zero crossing using the averaged zero cross value.
  • the first zero crossing is located by subtracting 1/2 the tone period from the first positive to negative zero crossing which was just previously predicted using the averaged zero cross value.
  • the first zero crossing obtained in this last step is indicative of the overall link delay, D.
  • a coarse delay is used in combination with a fine delay to produce the overall link delay, D.
  • the link transmission is delayed to compensate for the overall link delay, D.
  • the compensation is performed at the remote site. In another embodiment, the compensation is performed at the central site. Multiple remote sites may be calibrated by equalizing overall link delays in the system.
  • Link Delay Calculation using Discrete-Time Fourier Transform is measured using synchronization signals from the global positioning satellite (GPS) system.
  • the first step involves the control site transmitting several cycles of a tone (a tone burst) on a 1 pulse-per- second (PPS) mark signal generated by the local GPS unit. This process is illustrated in Figures 11 A and 1 IB.
  • samples are collected following detection of the 1 PPS signal from the remote site's GPS unit.
  • the sampling clock used for acquisition at the remote site is frequency locked to the sampling clock used for generation at the control site. This frequency locking is accomplished via the 10 MHz GPS synchronization signal.
  • the remote site samples are stored in a buffer and analyzed in order to estimate the starting time of the received tone burst.
  • One method for calculate the time delay is to use the Discrete-Time Fourier Transform (DTFT) to find the starting phase of a segment of the tone burst.
  • DTFT Discrete-Time Fourier Transform
  • a segment is simply a particular subset of the samples that make up the tone burst. Ifthe cycle in which the segment starts is known, the starting phase can be used to estimate link delay which is accurate to the degree required for the simulcast.
  • This approach is independent of the amplitude of the tone burst and is exact in the absence of noise. In addition, it can be done in a computationally efficient manner on a DSP chip.
  • Figure 17 shows a 5 cycle tone burst with a normalized frequency of 0.125. (The leading and trailing delays are not shown in this figure.)
  • the threshold shown is a negative value.
  • the first point in the data buffer to fall below this threshold is indicated by t lhmh .
  • t start 2 cycles ahead of t lhrtsh aad represents a suitable starting point for the segment. From the ⁇ point onward, the segment must include an integer number of cycles. For this reason, it is crucial to select the tone burst frequency such that its period is an integer number of samples (e.g., a 1 kHz burst with an 8 kHz sampling frequency results in a period of 8 samples.)
  • the threshold may be adjusted for noisy environments, to decrease the possibility of noise triggering the discrete-time Fourier transform operation.
  • noisy samples include, but are not limited to, statistical analysis of a number of calculated time delays.
  • an average delay and a standard deviation are calculated and samples greater than the standard deviation are discarded.
  • One embodiment will remove noisy samples by selecting a maximum change in delay, say 5 ⁇ sec, for example, and then discarding samples which deviate more than 5 ⁇ sec from an average delay.
  • Other embodiments for treating noisy samples are possible without departing from the scope and spirit of the present invention.
  • %output results clg; plot(l :delay,tint); title('Zero Crossings for Measuring Link Delay: infinite S/N'); xIabelCposition of samples relative to "true" zero crossing ⁇ microseconds)'); ylabelCerror in zero crossing estimate ⁇ seconds ⁇ ');

Abstract

L'invention concerne un système de calcul de temporisation de liaison utilisant un signal de référence commun et un signal sonore d'impulsion ou commandé par porte pour déterminer la temporisation de la liaison. Ce système utilise une transformée de Fourier à temps discret pour mesurer des informations de phase et obtenir la temporisation de liaison. Un tel système utilise un signal d'un sytème de positionnement global par satellite pour synchroniser la génération des signaux au niveau du site de commande et la démodulation au niveau d'un site éloigné.
PCT/US1997/005984 1996-04-12 1997-04-11 Systeme de calcul de temporisation de liaison pour un systeme de repeteurs radio WO1997039543A2 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
AU27262/97A AU2726297A (en) 1996-04-12 1997-04-11 Link delay calculation system for a radio repeater system

Applications Claiming Priority (12)

Application Number Priority Date Filing Date Title
US1531196P 1996-04-12 1996-04-12
US63067396A 1996-04-12 1996-04-12
US1530996P 1996-04-12 1996-04-12
US08/631,866 1996-04-12
US60/015,311 1996-04-12
US08/631,866 US5896560A (en) 1996-04-12 1996-04-12 Transmit control system using in-band tone signalling
US08/628,981 1996-04-12
US08/630,673 1996-04-12
US08/628,981 US5991309A (en) 1996-04-12 1996-04-12 Bandwidth management system for a remote repeater network
US60/015,309 1996-04-12
US2776396P 1996-10-07 1996-10-07
US60/027,763 1996-10-07

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PCT/US1997/005985 WO1997039535A2 (fr) 1996-04-12 1997-04-11 Systeme de gestion des largeurs de bande pour un reseau a repeteurs tele-alimentes
PCT/US1997/005893 WO1997039541A1 (fr) 1996-04-12 1997-04-11 Systeme de commande d'emission utilisant la signalisation multifrequence
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PCT/US1997/005985 WO1997039535A2 (fr) 1996-04-12 1997-04-11 Systeme de gestion des largeurs de bande pour un reseau a repeteurs tele-alimentes
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Families Citing this family (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US6674855B1 (en) 1999-10-06 2004-01-06 Comverse Ltd. High performance multifrequency signal detection
CN108519738A (zh) * 2018-04-13 2018-09-11 中国科学院微电子研究所 一种车辆运动状态信息延时补偿方法和装置

Family Cites Families (18)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
NL6508593A (fr) * 1965-07-02 1967-01-03
US3867700A (en) * 1972-11-15 1975-02-18 Keith H Wycoff Tone operated single side-band communication system
GB1574599A (en) * 1977-04-20 1980-09-10 Plessey Co Ltd Fault indication in repeatered telecommunications transmission-line
US4188582A (en) * 1978-04-10 1980-02-12 Motorola, Inc. Simulcast transmission system having phase-locked remote transmitters
DE2923265A1 (de) * 1979-06-08 1980-12-11 Bosch Gmbh Robert Funknetz
JPS5815341A (ja) * 1981-07-22 1983-01-28 Nec Corp 送信ダイバーシティ信号の送信方法および装置
US4406016A (en) * 1981-11-27 1983-09-20 The United States Of America As Represented By The Secretary Of The Army VHF Sensor in-band radio relay
DE3203678A1 (de) * 1982-02-04 1983-08-11 Licentia Patent-Verwaltungs-Gmbh, 6000 Frankfurt Gleichwellenfunknetz
CA1175964A (fr) * 1982-02-22 1984-10-09 John A. Barsellotti Boucle d'abonne a deux fils pour la transmission des paroles et des donnees
DE3585502D1 (de) * 1984-12-10 1992-04-09 Nippon Telegraph & Telephone Einseitenbanduebertragungssystem.
US4903292A (en) * 1988-11-01 1990-02-20 Reliance Comm/Tec Corporation System for transmitting low frequency tones through a digital loop carrier system
US4972410A (en) * 1989-07-20 1990-11-20 Electrocom Automation, Inc. Method and apparatus for controlling signal coherency in simulcast systems
JP2611695B2 (ja) * 1989-12-15 1997-05-21 日本電気株式会社 光中継器監視回路
GB9111313D0 (en) * 1991-05-24 1991-07-17 British Telecomm Radio system
US5257404A (en) * 1991-10-04 1993-10-26 Motorola, Inc. Simulcast synchronization and equalization system and method therefor
US5280629A (en) * 1991-12-06 1994-01-18 Motorola, Inc. Technique for measuring channel delay
ES2149782T3 (es) * 1992-01-10 2000-11-16 Nec Corp Sistema de buscapersonas por radio de transmision simultanea.
US5440313A (en) * 1993-05-27 1995-08-08 Stellar Gps Corporation GPS synchronized frequency/time source

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