US7298099B2 - PFC and ballast control IC - Google Patents

PFC and ballast control IC Download PDF

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US7298099B2
US7298099B2 US11/102,603 US10260305A US7298099B2 US 7298099 B2 US7298099 B2 US 7298099B2 US 10260305 A US10260305 A US 10260305A US 7298099 B2 US7298099 B2 US 7298099B2
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current
ignition
voltage
pin
preheat
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US20050225265A1 (en
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Thomas J. Ribarich
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Infineon Technologies North America Corp
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International Rectifier Corp USA
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    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/282—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/295—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices and specially adapted for lamps with preheating electrodes, e.g. for fluorescent lamps
    • H05B41/298—Arrangements for protecting lamps or circuits against abnormal operating conditions
    • H05B41/2988—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the lamp against abnormal operating conditions
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/24—Circuit arrangements in which the lamp is fed by high frequency ac, or with separate oscillator frequency
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/295—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices and specially adapted for lamps with preheating electrodes, e.g. for fluorescent lamps
    • H05B41/298—Arrangements for protecting lamps or circuits against abnormal operating conditions
    • H05B41/2981—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/295—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices and specially adapted for lamps with preheating electrodes, e.g. for fluorescent lamps
    • H05B41/298—Arrangements for protecting lamps or circuits against abnormal operating conditions
    • H05B41/2981—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions
    • H05B41/2985—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions against abnormal lamp operating conditions
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/295—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices and specially adapted for lamps with preheating electrodes, e.g. for fluorescent lamps
    • H05B41/298—Arrangements for protecting lamps or circuits against abnormal operating conditions
    • H05B41/2981—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions
    • H05B41/2986—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions against internal abnormal circuit conditions

Definitions

  • the present invention relates to a ballast control IC, particularly for driving fluorescent lamps, and more particularly with additional PFC circuitry on the IC.
  • the IRS21681D is a fully integrated, fully protected 600V ballast control IC designed to drive all types of fluorescent lamps.
  • the IRS21681D is based on the popular IR2166 control IC with additional improvements to increase ballast performance.
  • PFC circuitry operates in critical conduction mode and provides high PF, low THD and DC bus regulation.
  • the IRS21681D features include programmable preheat and run frequencies, programmable preheat time, programmable ignition ramp, programmable PFC over-current protection, and programmable end-of-life protection.
  • Comprehensive protection features such as protection from failure of a lamp to strike, filament failures, end-of-life protection, DC bus under-voltage reset as well as an automatic restart function, have been included in the design.
  • the IRS2168D has, in addition, closed-loop half-bridge ignition current regulation and a novel fault counter.
  • the IRS21681D unlike the IRS2168D, ramps up during ignition and shuts down at the first over-current fault.
  • FIG. 4 it is seen that only a single event of CS pin >1.25V is needed to go to fault mode from ignition or run mode. In the preheat mode, the CS pin over-current is disabled.
  • FIG. 8 see the zoomed images at the bottom. The middle image shows the ignition ramp and it can be seen that the current ramps up and the ballast shuts off (fault mode) as soon as CS>1.25V.
  • FIG. 5 it can be seen that the CS pin over-current is enabled in preheat mode and run mode, but that 60 cycles of consecutive faults (internal fault counter) are needed in order to go to fault mode.
  • fault mode is disabled. Instead, the ignition regulation circuit keeps the CS pin limited to 1.25V, and therefore limits the maximum ignition current and voltage of the ballast output stage. See also the timing diagram, FIG. 9 , which shows that the current is regulated for the duration of ignition.
  • the IRS21681D and IRS2168D are both available in either 16-pin PDIP or 16-pin narrow body SOIC packages.
  • the IRS2168D has, in addition:
  • FIG. 1 is a schematic diagram showing a typical application of the IC's.
  • FIGS. 2 and 3 are schematic block diagrams of the IRS21681D and IRS2168D chips, respectively.
  • FIGS. 4 and 5 are state diagrams showing operating modes of the IRS21681D and IRS2168D, respectively.
  • FIGS. 6 and 7 show lead assignments and definitions in the IRS21681D and IRS2168D, respectively.
  • FIG. 8 shows timing diagrams for the ballast section of the IRS21681D.
  • FIG. 9 shows timing diagrams for the ballast section of the IRS2168D.
  • FIG. 10 shows start-up and supply circuitry.
  • FIG. 11 is a graph showing Vcc supply voltage versus time during start-up.
  • FIG. 12 is a schematic block diagram showing preheat circuitry.
  • FIG. 13 is a timing diagram relative to the preheat and oscillator functions.
  • FIG. 14 shows ignition circuitry
  • FIG. 15 is a timing diagram relative to ignition regulation.
  • FIG. 16 is a timing diagram for the fault counter.
  • FIG. 17 is a schematic diagram of a boost converter.
  • FIG. 18 is a graph showing sinusoidal line input voltage (solid line), smoothed sinusoidal line input current (dashed line), and triangular PFC inductor current, over one-half cycle of the line input voltage.
  • FIG. 19 is a simplified schematic of a PFC control circuit.
  • FIG. 20 is a detailed block diagram of the PFC control circuit.
  • FIG. 21 is a timing diagram showing inductor current, and PFC pin, ZX pin and OC pin signals.
  • FIG. 22 is a timing diagram showing on-time modulation near the AC line zero-crossings.
  • FIG. 23 is a graph of RFMIN vs. frequency for use in selecting component values.
  • UVLO Under-Voltage Lock-Out Mode
  • the under-voltage lock-out mode is defined as the state the IC is in when VCC is below the turn-on threshold of the IC.
  • the IRS2168D undervoltage lock-out is designed to maintain an ultra low supply current of less than 400 ⁇ A, and to guarantee the IC is fully functional before the high- and low-side output drivers are activated.
  • FIG. 10 shows an efficient voltage supply using the micro-power start-up current of the IRS2168D together with a snubber charge pump from the half-bridge output (R VCC , C VCC1 , C VCC2 , C SNUB , D CP1 and D CP2 ).
  • the VCC capacitors (C VCC1 and C VCC2 ) are charged by the current through supply resistor (R VCC ) minus the start-up current drawn by the IC. This resistor is chosen to set the desired AC line input voltage turn-on threshold for the ballast.
  • the capacitors at VCC begin to discharge due to the increase in IC operating current ( FIG. 11 ).
  • the high-side supply voltage, VB-VS begins to increase as capacitor C BS is charged through the internal bootstrap MOSFET during the LO on-time of each LO switching cycle.
  • VB-VS voltage exceeds the high-side start-up threshold (UVBS+)
  • HO begins to oscillate. This may take several cycles of LO to charge VB-VS above UVBS+ due to RDSon of the internal bootstrap MOSFET.
  • Capacitor C VCC2 supplies the IC current during the VCC discharge time and should be large enough such that VCC does not decrease below UVLO ⁇ before the charge pump takes over.
  • Capacitor C VCC1 is provided for noise filtering and is placed as close as possible and directly between VCC and COM, and should not be lower than 0.1 ⁇ F.
  • Resistors R 1 and R 2 are recommended for limiting high currents that can flow to VCC from the charge pump during hard-switching of the half-bridge or during lamp ignition.
  • the internal bootstrap MOSFET and supply capacitor (C BS ) comprise the supply voltage for the high side driver circuitry. During UVLO mode, the high- and low-side driver outputs HO and LO are both low, the internal oscillator is disabled, and pin CPH is connected internally to COM for resetting the preheat time.
  • the IRS2168D enters preheat mode when VCC exceeds the UVLO positive-going threshold (UVLO+).
  • the internal MOSFET that connects pin CPH to COM is turned off and an external resistor ( FIG. 12 ) begins to charge the external preheat timing capacitor (CPH).
  • LO and HO begin to oscillate at a higher soft-start frequency and ramp down quickly to the preheat frequency.
  • the VCO pin is connected to COM through an internal MOSFET so the preheat frequency is determined by the equivalent resistance at the FMIN pin formed by the parallel combination of resistors RFMIN and RPH. The frequency remains at the preheat frequency until the voltage on pin CPH exceeds 2 ⁇ 3*VCC and the IC enters Ignition Mode.
  • the over-current protection on pin CS and the 60-cycle consecutive over-current fault counter are both enabled.
  • the PFC circuit is working in high-gain mode (see PFC section) and keeps the DC bus voltage regulated at a constant level.
  • the IRS2168D ignition mode is defined by the second time CPH charges from 1 ⁇ 3*VCC to 2 ⁇ 3*VCC.
  • pin CPH When the voltage on pin CPH exceeds 2 ⁇ 3*VCC for the first time, pin CPH is discharged quickly through an internal MOSFET down to 1 ⁇ 3*VCC (see FIGS. 13 and 14 ).
  • the internal MOSFET turns off and the voltage on pin CPH begins to increase again.
  • the internal MOSFET at pin VCO turns off and resistor RPH is disconnected from COM.
  • the equivalent resistance at the FMIN pin increases from the parallel combination (RPH//RFMIN) to RFMIN at a rate programmed by the external capacitor at pin VCO (CVCO) and resistor RPH. This causes the operating frequency to ramp down smoothly from the preheat frequency through the ignition frequency to the final run frequency. During this ignition ramp, the frequency sweeps through the resonance frequency of the lamp output stage to ignite the lamp.
  • the over-current threshold on pin CS will protect the ballast against a non-strike or open-filament lamp fault condition.
  • the voltage on pin CS is defined by the lower half-bridge MOSFET current flowing through the external current sensing resistor RCS. This resistor programs the maximum peak ignition current (and therefore peak ignition voltage) of the ballast output stage. Should this voltage exceed the internal threshold of 1.25V, the ignition regulation circuit discharges the VCO voltage slightly to increase the frequency slightly (see FIG. 15 ). This cycle-by-cycle feedback from the CS pin to the VCO pin will adjust the frequency each cycle to limit the amplitude of the current for the entire duration of ignition mode. When CPH exceeds 2 ⁇ 3*VCC for the second time, the IC enters run mode and the fault counter becomes enabled.
  • the ignition regulation remains active in run mode but the IC will enter fault mode after 60 consecutive over-current faults and gate driver outputs HO, LO and PFC will be latched low.
  • the PFC circuit is working in high-gain mode and keeps the DC bus voltage regulated at a constant level. The high-gain mode prevents the DC bus from decreasing during lamp ignition or ignition regulation.
  • the IC Once VCC has exceeded 2 ⁇ 3*VCC for the second time, the IC enters run mode. CPH continues to charge up to VCC.
  • the operating frequency is at the minimum frequency (after the ignition ramp) and is programmed by the external resistor (RFMIN) at the FMIN pin. Should hard-switching occur at the half-bridge at any time (open-filament, lamp removal, etc.), the voltage across the current sensing resistor (RCS) will exceed the internal threshold of 1.25 volts and the fault counter will begin counting (see FIG. 14 ). Should the number of consecutive over-current faults exceed 60, the IC will enter fault mode and the HO, LO and PFC gate driver outputs will be latched low. During run mode, the end-of-life (EOL) window comparator and the DC bus under-voltage reset are both enabled.
  • EOL end-of-life
  • the VBUS pin includes a 3.0V under-voltage reset threshold. When the IC is in run mode and the voltage at the VBUS pin decreases below 3.0V, VCC will be discharged through an internal MOSFET down to the UVLO ⁇ threshold and all gate driver outputs will be latched low.
  • the designer should set the over-current limit of the PFC section such that the DC bus does not drop until the AC line input voltage falls below the minimum rated input voltage of the ballast (see PFC section).
  • the DC bus voltage will start to decrease when over-current is reached during low-line conditions.
  • the voltage measured at the VBUS pin will decrease below the internal 3.0V threshold and the ballast will turn off cleanly.
  • the pull-up resistor to VCC (R VCC ) will then turn the ballast on again when the AC input line voltage increases high enough again where VCC exceeds UVLO+.
  • R VCC should be set to turn the ballast on at the minimum specified ballast input voltage and the PFC over-current should be set somewhere below this level. This hysteresis will result in clean turn-on and turn-off of the ballast.
  • the current sense function will force the IC to enter fault mode only after the voltage at the CS pin has been greater than 1.25V for 60 consecutive cycles of LO.
  • the voltage at the CS pin is AND-ed with LO (see FIG. 16 ) so it will work with pulses that occur during the LO on-time or DC. If the over-current faults are not consecutive, then the internal fault counter will count down each cycle when there is no fault. Should an over-current fault occur only for a few cycles and then not occur again, the counter will eventually reset to zero.
  • the over-current fault counter is enabled during preheat and run modes and disabled during ignition mode.
  • the run frequency is programmed with the timing resistor RFMIN at the FMIN pin.
  • the run frequency is given as:
  • f RUN 1 ( 4.8 ⁇ e - 10 ) ⁇ R FMIN ⁇ [ Hertz ] ⁇ ⁇ or ( 1 )
  • R FMIN 1 ( 4.8 ⁇ e - 10 ) ⁇ f RUN ⁇ [ Ohms ] ⁇ ⁇ or ( 2 )
  • the preheat frequency is programmed with timing resistors RFMIN and RPH.
  • the timing resistors are connected in parallel for the duration of the preheat time.
  • the preheat frequency is therefore given as:
  • the preheat time is defined by the time it takes for the external capacitor on pin CPH to charge up to 2 ⁇ 3*VCC.
  • An external resistor (RCPH) connected to VCC charges capacitor CPH. The preheat time is therefore given as:
  • the ramp time is defined by the time it takes for the external capacitor on pin VCO to charge up to 2V.
  • the external timing resistor (RPH) connected to FMIN charges capacitor CVCO.
  • the ignition ramp time is therefore given as:
  • the maximum ignition current is programmed with the external resistor RCS and an internal threshold of 1.25V. This threshold determines the over-current limit of the ballast, which will be reached when the frequency ramps down towards resonance during ignition and the lamp does not ignite.
  • the maximum ignition current is given as:
  • i PK 2 ⁇ 2 ⁇ P OUT VAC MIN ⁇ ⁇ ⁇ [ Amps ⁇ ⁇ Peak ] ( 2 )
  • the circuit acts as a pure resistive load to the AC input line voltage.
  • the degree to which the circuit matches a pure resistor is measured by the phase shift between the input voltage and input current and how well the shape of the input current waveform matches the shape of the sinusoidal input voltage.
  • the cosine of the phase angle between the input voltage and input current is defined as the power factor (PF), and how well the shape of the input current waveform matches the shape of the input voltage is determined by the total harmonic distortion (THD).
  • PF power factor
  • THD total harmonic distortion
  • a power factor of 1.0 (maximum) corresponds to zero phase shift and a THD of 0% and represents a pure sinusoidal waveform (no distortion). For this reason it is desirable to have a high PF and a low THD.
  • the IR2168D includes an active power factor correction (PFC) circuit.
  • PFC active power factor correction
  • the control method implemented in the IR2168D is for a boost-type converter ( FIG. 17 ) running in critical-conduction mode (CCM). This means that during each switching cycle of the PFC MOSFET, the circuit waits until the inductor current discharges to zero before turning the PFC MOSFET on again.
  • the PFC MOSFET is turned on and off at a much higher frequency (>10 KHz) than the line input frequency (50 to 60 Hz).
  • the inductor LPFC When the switch MPFC is turned on, the inductor LPFC is connected between the rectified line input (+) and ( ⁇ ) causing the current in LPFC to charge up linearly.
  • MPFC When MPFC is turned off, LPFC is connected between the rectified line input (+) and the DC bus capacitor CBUS (through diode DPFC) and the stored current in LPFC flows into CBUS.
  • MPFC is turned on and off at a high frequency and the voltage on CBUS charges up to a specified voltage.
  • the feedback loop of the IR2168D regulates this voltage to a fixed value by continuously monitoring the DC bus voltage and adjusting the on-time of MPFC accordingly. For an increasing DC bus the on-time is decreased, and for a decreasing DC bus the on-time is increased.
  • This negative feedback control is performed with a slow loop speed and a low loop gain such that the average inductor current smoothly follows the low-frequency line input voltage for high power factor and low THD.
  • the on-time of MPFC therefore appears to be fixed (with an additional modulation to be discussed later) over several cycles of the line voltage. With a fixed on-time, and an off-time determined by the inductor current discharging to zero, the result is a system where the switching frequency is free-running and constantly changing from a high frequency near the zero crossing of the AC input line voltage, to a lower frequency at the peaks ( FIG. 18 ).
  • the inductor current When the line input voltage is low (near the zero crossing), the inductor current will charge up to a small amount and the discharge time will be fast resulting in a high switching frequency. When the input line voltage is high (near the peak), the inductor current will charge up to a higher amount and the discharge time will be longer giving a lower switching frequency.
  • the PFC control circuit of the IR2168D ( FIG. 19 ) includes five control pins: VBUS, COMP, ZX, PFC and OC.
  • the VBUS pin measures the DC bus voltage via an external resistor voltage divider.
  • the COMP pin programs the on-time of MPFC and the speed of the feedback loop with an external capacitor.
  • the ZX pin detects when the inductor current discharges to zero each switching cycle using a secondary winding from the PFC inductor.
  • the PFC pin is the low-side gate driver output for the external MOSFET, MPFC.
  • the OC pin senses the current flowing through MPFC and performs cycle-by-cycle over-current protection.
  • the VBUS pin is regulated against a fixed internal 4V reference voltage for regulating the DC bus voltage ( FIG. 20 ).
  • the feedback loop is performed by an operational transconductance amplifier (OTA) that sinks or sources a current to the external capacitor at the COMP pin.
  • OTA operational transconductance amplifier
  • the resulting voltage on the COMP pin sets the threshold for the charging of the internal timing capacitor (C1, FIG. 20 ) and therefore programs the on-time of MPFC.
  • the gain of the OTA is set to a high level to raise the DC bus level quickly and to minimize the transient on the DC bus that can occur during ignition.
  • run mode the gain is then decreased to a lower level necessary for a slower loop speed for achieving high power factor and low THD.
  • the off-time of MPFC is determined by the time it takes the LPFC current to discharge to zero.
  • the zero current level is detected by a secondary winding on LPFC that is connected to the ZX pin through an external current limiting resistor RZX.
  • a positive-going edge exceeding the internal 2V threshold signals the beginning of the off-time.
  • a negative-going edge on the ZX pin falling below 1.7V will occur when the LPFC current discharges to zero which signals the end of the off-time and MPFC is turned on again ( FIG. 21 ).
  • the cycle repeats itself indefinitely until the PFC section is disabled due to a fault detected by the ballast section (Fault Mode), an over-voltage or under-voltage condition on the DC bus, or, the negative transition of ZX pin voltage does not occur.
  • MPFC will remain off until the watch-dog timer forces a turn-on of MPFC for an on-time duration programmed by the voltage on the COMP pin.
  • the watch-dog pulses occur every 400 ⁇ s indefinitely until a correct positive- and negative-going signal is detected on the ZX pin and normal PFC operation is resumed.
  • the PFC output will turn off. The circuit will then wait for a negative-going transition on the ZX pin or a forced turn-on from the watch-dog timer to turn the PFC output on again.
  • a fixed on-time of MPFC over an entire cycle of the line input voltage produces a peak inductor current which naturally follows the sinusoidal shape of the line input voltage.
  • the smoothed averaged line input current is in phase with the line input voltage for high power factor but the total harmonic distortion (THD), as well as the individual higher harmonics, of the current can still be too high. This is mostly due to cross-over distortion of the line current near the zero-crossings of the line input voltage.
  • TDD total harmonic distortion
  • an additional on-time modulation circuit has been added to the PFC control. This circuit dynamically increases the on-time of MPFC as the line input voltage nears the zero-crossings ( FIG. 22 ).
  • the on-time of MPFC increases to keep the DC bus constant.
  • the on-time will continue to increase as the line voltage continues to decrease until the OC pin exceeds the internal 1.2V over-current threshold.
  • the on-time can no longer increase and the PFC can no longer supply enough current to keep the DC bus fixed for the given load power. This will cause the DC bus to begin to decrease.
  • the decreasing DC bus will cause the VBUS pin to decrease below the internal 3V threshold ( FIG. 20 ).
  • VCC is discharged internally to UVLO ⁇ .
  • the IR2168D enters UVLO mode and both the PFC and ballast sections are disabled.
  • the start-up supply resistor to VCC should be set such that the ballast turns on at an AC line input voltage above the level at which the DC bus begins to drop.
  • the current-sensing resistor at the OC pin sets the maximum PFC current and therefore sets the maximum on-time of MPFC. This prevents saturation of the PFC inductor and programs the minimum low-line input voltage for the ballast.
  • the micro-power supply resistor to VCC and the current-sensing resistor at the OC pin program the on and off input line voltage thresholds for the ballast. With these thresholds correctly set, the ballast will turn off due to the 3V under-voltage threshold on the VBUS pin, and on again at a higher voltage (hysteresis) due to the supply resistor to VCC.

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US11/102,603 US7298099B2 (en) 2004-04-08 2005-04-08 PFC and ballast control IC
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US11/102,603 US7298099B2 (en) 2004-04-08 2005-04-08 PFC and ballast control IC

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US20120019146A1 (en) * 2009-01-16 2012-01-26 Osram Gesellschaft Mit Beschraenkter Haftung Detector circuit and method for controlling a fluorescent lamp
US20120086355A1 (en) * 2010-10-06 2012-04-12 Osram Ag Circuit and method for driving a lamp
US8482213B1 (en) 2009-06-29 2013-07-09 Panasonic Corporation Electronic ballast with pulse detection circuit for lamp end of life and output short protection
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TWI425877B (zh) * 2010-10-04 2014-02-01 Delta Electronics Inc 氣體放電燈管的安定控制電路與安定電路的控制方法
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US20090116161A1 (en) * 2005-12-26 2009-05-07 Autonetworks Technologies , Ltd. Power Supply Controller and Threshold Adjustment Method Thereof
US8270138B2 (en) * 2005-12-26 2012-09-18 Autonetworks Technologies, Limited Power supply controller and threshold adjustment method thereof
US20090021177A1 (en) * 2007-07-04 2009-01-22 Gye-Hyun Cho Diagnosis circuit apparatus and lamp ballast circuit using the same
US8193718B2 (en) * 2007-07-04 2012-06-05 Fairchild Korea Semiconductor, Ltd. Diagnosis circuit apparatus and lamp ballast circuit using the same
US9621061B2 (en) * 2008-12-15 2017-04-11 Canon Kabushiki Kaisha Power supply apparatus and image forming apparatus
US20130307917A1 (en) * 2008-12-15 2013-11-21 Canon Kabushiki Kaisha Power supply apparatus and image forming apparatus
US8736188B2 (en) * 2009-01-16 2014-05-27 Osram Gesellschaft Mit Beschraenkter Haftung Detector circuit and method for controlling a fluorescent lamp
US20120019146A1 (en) * 2009-01-16 2012-01-26 Osram Gesellschaft Mit Beschraenkter Haftung Detector circuit and method for controlling a fluorescent lamp
US20100270932A1 (en) * 2009-04-24 2010-10-28 Naoki Onishi Fault detection and shutdown control circuits and methods for electronic ballasts
US8482213B1 (en) 2009-06-29 2013-07-09 Panasonic Corporation Electronic ballast with pulse detection circuit for lamp end of life and output short protection
US8350488B2 (en) * 2009-06-30 2013-01-08 Microsemi Corporation Integrated backlight control system
US20100327761A1 (en) * 2009-06-30 2010-12-30 Microsemi Corporation Integrated backlight control system
US20110204815A1 (en) * 2010-02-19 2011-08-25 Gye-Hyun Cho Preheating control device, lamp driving device including the same, and preheating control method
US8957596B2 (en) * 2010-02-19 2015-02-17 Fairchild Korea Semiconductor Ltd. Preheating control device, lamp driving device including the same, and preheating control method
TWI425877B (zh) * 2010-10-04 2014-02-01 Delta Electronics Inc 氣體放電燈管的安定控制電路與安定電路的控制方法
CN102573250A (zh) * 2010-10-06 2012-07-11 欧司朗股份有限公司 用于激励灯的电路和方法
US20120086355A1 (en) * 2010-10-06 2012-04-12 Osram Ag Circuit and method for driving a lamp
US8947020B1 (en) 2011-11-17 2015-02-03 Universal Lighting Technologies, Inc. End of life control for parallel lamp ballast
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WO2005101921A2 (en) 2005-10-27
WO2005101921A8 (en) 2008-01-10
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GB2428526B (en) 2007-12-19
GB2428526A (en) 2007-01-31
EP1736037A4 (en) 2009-03-04
FI20065645L (fi) 2006-10-06
CN101208997A (zh) 2008-06-25
DE112005000783T5 (de) 2007-03-08
KR20070009679A (ko) 2007-01-18
KR100872897B1 (ko) 2008-12-10
GB0621355D0 (en) 2006-12-13
US20050225265A1 (en) 2005-10-13
WO2005101921A3 (en) 2007-11-08
EP1736037A2 (en) 2006-12-27

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