US11855564B2 - Control device and electric vehicle - Google Patents

Control device and electric vehicle Download PDF

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US11855564B2
US11855564B2 US17/611,301 US202017611301A US11855564B2 US 11855564 B2 US11855564 B2 US 11855564B2 US 202017611301 A US202017611301 A US 202017611301A US 11855564 B2 US11855564 B2 US 11855564B2
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inverter circuit
output voltage
control device
power supply
current
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US20220311370A1 (en
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Takahiro Araki
Daiki Komatsu
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Hitachi Ltd
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Hitachi Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0016—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters
    • H02M1/0022—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters the disturbance parameters being input voltage fluctuations
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/16—Estimation of constants, e.g. the rotor time constant
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop

Definitions

  • the present invention relates to a control device of an inverter circuit and an electric vehicle including the control device.
  • the control device of an inverter circuit mounted on a hybrid automobile, an electric automobile, and the like is desired to be highly responsive.
  • a technology for securing control responsiveness of the output voltage reflecting change in the internal resistance of DC power supply is proposed.
  • Patent Literatures 1 As a background art of the present technical field, Patent Literatures 1 is known.
  • an output voltage control system includes a feedback arithmetic unit executing feedback control using feedback gains from a feedback gain determination unit, the feedback control being for making deviation of the output voltage of the DC/DC converter to the target voltage to be zero, and the feedback gain determination unit determines the feedback gains so as to reflect change in internal resistance corresponding to the charging rate at a DC power supply (battery).
  • a DC power supply battery
  • the main object of the present invention is to improve the response performance of control of an inverter circuit.
  • a control device executes control of an inverter circuit, calculates input current of the inverter circuit based on an output current command value that is for controlling an output current of the inverter circuit, and calculates an output voltage compensation amount depending on a variation amount of an input voltage of the inverter circuit based on the input current calculated.
  • An electric vehicle includes the control device, an inverter circuit, and a motor, the inverter circuit being controlled by the control device and converting DC power to AC power, the motor being driven using the AC power outputted from the inverter circuit.
  • the response performance of control of the inverter circuit can be improved.
  • FIG. 1 is a drawing showing a configuration of a motor drive system including a control device related to the first embodiment of the present invention.
  • FIG. 2 is a drawing showing an example of a waveform of input/output voltage and input/output current of an inverter circuit.
  • FIG. 3 is a block diagram showing a functional configuration of the control device related to the first embodiment of the present invention.
  • FIG. 4 is a flowchart showing a process procedure of an output voltage error calculation unit.
  • FIG. 5 is a table showing the relation between output voltage vector V x and input current i dc .
  • FIG. 6 is a voltage vector diagram showing an example of output voltage error vector ⁇ V x .
  • FIG. 7 is a drawing showing a configuration of a motor drive system including a control device related to the second embodiment of the present invention.
  • FIG. 8 is a drawing showing a configuration of an electric vehicle system related to the third embodiment of the present invention.
  • FIG. 1 is a drawing showing a configuration of a motor drive system including a control device related to the first embodiment of the present invention.
  • the motor drive system shown in FIG. 1 includes an inverter circuit 100 , a control device 1 for controlling the inverter circuit 100 , a motor 200 , a position sensor 210 , a current sensor 220 , and a DC power supply 300 .
  • the motor 200 is a three-phase AC motor, and is driven using three-phase AC power outputted from the inverter circuit 100 .
  • the position sensor 210 detects the position of a rotor of the motor 200 , and outputs rotor position ⁇ detected.
  • the current sensor 220 detects current of each phase flowing through the motor 200 , and outputs three-phase current values i u , i v , i w detected.
  • the control device 1 executes PWM control that is for controlling the inverter circuit 100 based on a torque command T* outputted from the outside, the three-phase current values i u , i v , i w detected by the current sensor 220 , and the rotor position ⁇ detected by the position sensor 210 .
  • the control device 1 generates a switching signal that is for controlling respective switching elements included in the inverter circuit 100 , and outputs the switching signal to the inverter circuit 100 . Also, the detail of the PWM control executed by the control device 1 will be described below.
  • the inverter circuit 100 includes switching elements 110 a to 110 f .
  • the switching elements 110 a is arranged in the U-phase upper arm
  • the switching elements 110 b is arranged in the U-phase lower arm
  • the switching elements 110 c is arranged in the V-phase upper arm
  • the switching elements 110 d is arranged in the V-phase lower arm
  • the switching elements 110 e is arranged in the W-phase upper arm
  • the switching elements 110 f is arranged in the W-phase lower arm respectively.
  • the switching elements 110 a to 110 f are configured respectively by combining a semiconductor element and a diode, the semiconductor element being capable of on/off operation such as a metal oxide semiconductor field effect transistor (MOSFET) and an insulated gate bipolar transistor (IGBT) for example.
  • MOSFET metal oxide semiconductor field effect transistor
  • IGBT insulated gate bipolar transistor
  • the switching elements 110 a to 110 f are turned on or off individually based on a switching signal inputted from the control device 1 , and convert DC power supplied from the DC power supply 300 to three-phase AC power.
  • U-phase AC power is generated by the switching elements 110 a , 110 b
  • V-phase AC power is generated by the switching elements 110 c , 110 d
  • W-phase AC power is generated by the switching elements 110 e , 110 f respectively.
  • the three-phase AC power generated thus is outputted from the inverter circuit 100 to a stator of the motor 200 , and generates three-phase AC current in the motor 200 .
  • This three-phase AC current generates a rotating magnetic field in the motor 200 , thereby the rotor rotates and the motor 200 is driven.
  • the DC power supply 300 is connected to an input terminal of the inverter circuit 100 , and supplies DC power required for driving the motor 200 .
  • the DC power supply 300 includes an internal resistance 310 and a voltage supply 320 .
  • DC voltage applied from the DC power supply 300 to the inverter circuit 100 changes according to the state of the DC power supply 300 .
  • the DC power supply 300 is a secondary battery such as a lead-acid storage battery and a lithium-ion battery
  • the output voltage of the DC power supply 300 also changes.
  • FIG. 2 is a drawing showing an example of a waveform of input/output voltage and input/output current of the inverter circuit 100 .
  • FIG. 2 ( a ) expresses an example of DC input voltage V dc and line output voltage v uv , v vw , v wu of each phase of the inverter circuit 100
  • FIG. 2 ( b ) expresses an example of DC input current i dc and output current i u , i v , i w of each phase of the inverter circuit 100
  • the interval between each point set at an equal interval on the input voltage V dc in FIG. 2 ( a ) corresponds to the control frequency of the inverter circuit 100 by the control device 1 .
  • FIG. 3 is a block diagram showing a functional configuration of the control device 1 related to the first embodiment of the present invention.
  • the control device 1 includes respective functional blocks of a current command arithmetic unit 10 , a dq axis current control unit 20 , coordinate conversion units 30 , 31 , a PWM signal generation unit 40 , a dq conversion unit 50 , a velocity conversion unit 60 , and an output voltage error calculation unit 70 .
  • the control device 1 is configured of a microcomputer for example, and can achieve these functional blocks by executing a predetermined program in the microcomputer. Alternatively, a part or all of these functional blocks may be achieved by using a hardware circuit such as a logic IC and an FPGA.
  • the current command arithmetic unit 10 calculates a d-axis current command value i d * and a q-axis current command value i q *which are for controlling the output current of the inverter circuit 100 based on the torque command value T* inputted and an angular velocity ⁇ calculated by the velocity conversion unit 60 .
  • dq axis current control unit 20 there are inputted difference of the d-axis current command value i d * calculated by the current command arithmetic unit 10 and a d-axis current detection value i d outputted from the dq conversion unit 50 based on the three-phase current i u , i v , i w detected by the current sensor 220 and difference of the q-axis current command value i q * calculated by the current command arithmetic unit 10 and a q-axis current detection value i q outputted from the dq conversion unit 50 based on the three-phase current i u , i v , i w detected by the current sensor 220 .
  • the dq axis current control unit 20 calculates a d-axis voltage command value v d * and a q-axis voltage command value v q * by executing proportional control and integral control based on a predetermined control gain command value based on the respective differences having been inputted.
  • the coordinate conversion unit 30 To the coordinate conversion unit 30 , there are inputted the d-axis voltage command value v d * and the q-axis voltage command value v q * calculated by the dq axis current control unit 20 and the rotor position ⁇ detected by the position sensor 210 .
  • the coordinate conversion unit 30 outputs a U-phase voltage command value v u *, a V-phase voltage command value v v *, and a W-phase voltage command value v w * by subjecting the d-axis voltage command value v d * and the q-axis voltage command value v q * to rotated coordinate conversion based on the rotor position ⁇ .
  • the coordinate conversion unit 31 To the coordinate conversion unit 31 , there are inputted the d-axis current command value i d * and the q-axis current command value i q * calculated by the current command arithmetic unit 10 and the rotor position ⁇ detected by the position sensor 210 .
  • the coordinate conversion unit 31 outputs a U-phase current command value i u *, a V-phase current command value v v *, and a W-phase current command value i w * by subjecting the d-axis current command value i d * and the q-axis current command value i q * to rotated coordinate conversion based on the rotor position ⁇ .
  • the PWM signal generation unit 40 there are inputted the U-phase voltage command value v u *, the V-phase voltage command value v v *, and the W-phase voltage command value v w * outputted from the coordinate conversion unit 30 based on the d-axis voltage command value v d * and the q-axis voltage command value v q * calculated by the dq axis current control unit 20 and a U-phase voltage command compensation amount V ucomp , a V-phase voltage command compensation amount V vcomp , and a W-phase voltage command compensation amount V wcomp calculated by the output voltage error calculation unit 70 .
  • the PWM signal generation unit 40 Based on a sum of these voltage command values and the voltage command compensation amounts of each of the U, V, and W phases, the PWM signal generation unit 40 generates a switching signal (PWM signal) which is for controlling turning on/off of the switching elements 110 a to 110 f of each phase included in the inverter circuit 100 .
  • PWM signal a switching signal
  • the voltage command values v u *, v v *, v w * are subjected to compensation respectively based on the voltage command compensation amounts V ucomp , V vcomp , V wcomp and the switching signal is generated. Therefore, feedforward compensation of the output voltage command value with respect to the inverter circuit 100 can be achieved.
  • the dq conversion unit 50 To the dq conversion unit 50 , there are inputted the three-phase current i u , i v , i w detected by the current sensor 220 and the rotor position ⁇ detected by the position sensor 210 . Based on these values having been inputted, the dq conversion unit 50 outputs the d-axis current detection value i d and the q-axis current detection value i q .
  • the velocity conversion unit 60 To the velocity conversion unit 60 , the rotor position ⁇ detected by the position sensor 210 is inputted. The velocity conversion unit 60 outputs the angular velocity ⁇ based on the rotor position ⁇ , the rotor of the motor 200 rotating with the angular velocity ⁇ .
  • the output voltage error calculation unit 70 there are inputted the U-phase current command value i u *, the V-phase current command value i v *, and the W-phase current command value i w * outputted from the coordinate conversion unit 31 based on the d-axis current command value i d * and the q-axis current command value i q * calculated by the current command arithmetic unit 10 and the U-phase voltage command value v u *, the V-phase voltage command value v v *, and the W-phase voltage command value v w * outputted from the coordinate conversion unit 30 based on the d-axis voltage command value v d * and the q-axis voltage command value v q * calculated by the dq axis current control unit 20 .
  • the output voltage error calculation unit 70 calculates the U-phase voltage command compensation amount V ucomp , the V-phase voltage command compensation amount V ucomp , and the W-phase voltage command compensation amount V wcomp according to the variation amount of the input voltage V dc of the inverter circuit 100 . Also, with respect to the calculation method of the voltage command compensation amounts V ucomp , V vcomp , V wcomp of each of the U, V, and W phases by the output voltage error calculation unit 70 will be explained below referring to FIGS. 4 , 5 , and 6 .
  • FIG. 4 is a flowchart showing a process procedure of the output voltage error calculation unit 70 .
  • step S 1 the output voltage error calculation unit 70 calculates the output voltage vector V x according to the state of the switching elements 110 a to 110 f of the inverter circuit 100 from the voltage command values v u *, v v *, v w * of each of the U, V, and W phases having been inputted and a period T x of each output voltage vector V x .
  • step S 2 the output voltage error calculation unit 70 calculates the input current i dc flowing through the inverter circuit 100 from the DC power supply 300 for each output voltage vector V x calculated in step S 1 and the period T x when each output voltage vector V x is outputted from the current command values i u *, i v *, i w * of each of the U, V, and W phases having been inputted.
  • the input current i dc corresponding to each output voltage vector V x is calculated by referring a table shown in FIG. 5 described below for example.
  • step S 3 the output voltage error calculation unit 70 presumes the resistance value R ESR of the internal resistance 310 included in the DC power supply 300 .
  • the resistance value R ESR corresponding to the present state of the DC power supply 300 can be presumed.
  • the DC power supply 300 is a secondary battery such as a lead-acid storage battery and a lithium-ion battery for example
  • the output voltage of the DC power supply 300 changes according to the charging rate and the internal temperature as described above. Therefore, by setting beforehand corresponding internal resistance value as the table data for each of the charging rate and the internal temperature of the DC power supply 300 , an appropriate resistance value R ESR can be presumed based on these table data.
  • the output voltage error calculation unit 70 calculates the output voltage error vector ⁇ V x according to the error of the output voltage of the inverter circuit 100 generated due to the variation of the input voltage V dc for each period T x when each output voltage vector V x is outputted.
  • the output voltage error vector ⁇ V x is calculated by the expression (1) below for example.
  • R ESR expresses the resistance value of the internal resistance 310 presumed in step S 3
  • i dc expresses the input current calculated in step S 2 .
  • the direction of the output voltage error vector ⁇ V x obtained by the expression (1) is defined to be opposite of that of the output voltage vector V x .
  • the calculation accuracy of it can be improved by executing the calculation repeatedly.
  • step S 5 the output voltage error calculation unit 70 calculates the voltage command compensation amounts V ucomp , V vcomp , V wcomp of each of the U, V, and W phases by subjecting the output voltage error vector ⁇ V x calculated in step S 4 to coordinate conversion to a three-phase voltage value.
  • the variation amount of the input voltage V dc to the inverter circuit 100 is calculated, and the voltage command compensation amounts V ucomp , V vcomp , V wcomp for the output voltage are calculated.
  • FIG. 5 is a table showing the relation between the output voltage vector V x and the input current i dc used when the output voltage error calculation unit 70 calculates the input current i dc in step S 2 described above.
  • the on/off state of the switching element of each of the U, V, and W phases is shown, and the correspondence relation of the phase current and the input current i dc is shown.
  • the on/off state of the switching element of each of the U, V, and W phases is expressed by “0” or “1”.
  • the input current i dc of the inverter circuit 100 can be obtained by any one of the output currents i u , i v , i w of three phases.
  • the control block diagram of FIG. 1 In the period when other voltage vectors are outputted, in a similar manner, the input current i dc of the inverter circuit 100 can be obtained by any one of the output currents i u , i v , i w of three phases.
  • the output voltage error calculation unit 70 presumes the output currents i u , i v , i w in the next control period using the current command values i u *, i v *, i w * of each of the U, V, and W phases outputted from the coordinate conversion unit 31 by calculating, with the current command arithmetic unit 10 , the d-axis current command value i d * and the q-axis current command value i q *, and calculates the input current i dc in the next control period using them.
  • FIG. 6 is a voltage vector diagram showing an example of the output voltage error vector ⁇ V x calculated by the output voltage error calculation unit 70 in step S 4 described above. Also, in the example of FIG. 6 , such case is assumed that an output voltage command vector v out * which is the vector sum of the voltage command values v u *, v v *, v w * of each of the U, V, and W phases in a certain period is within a range surrounded by the V 1 vector and the V 2 vector.
  • ⁇ V 1 R ESR ⁇ i u ⁇ T 1 (2)
  • the output voltage error ⁇ v out generated during the control period in question becomes the vector sum of the output voltage error vector ⁇ V 1 and the output voltage error vector ⁇ V 2 .
  • the output voltage v out of the inverter circuit 100 becomes the vector sum of the output voltage command vector v out * and the output voltage error ⁇ v out , and, as it is, the voltage in accordance with the command is not outputted.
  • the voltage command compensation amounts V ucomp , V vcomp , V wcomp of each phase calculated by the output voltage error calculation unit 70 in step S 5 of FIG. 4 are added to the voltage command values v u *, v v *, v w * of each phase respectively in the PWM signal generation unit 40 .
  • the output voltage error ⁇ v out which is the vector sum of the output voltage error vector ⁇ V x calculated in step S 4 is added to the output voltage command vector v out * beforehand. Therefore, feedforward compensation of the output voltage command value can be achieved.
  • FIG. 7 is a drawing showing a configuration of a motor drive system including a control device related to the second embodiment of the present invention.
  • the motor drive system shown in FIG. 7 is different in terms that a power supply control device 2 controlling the DC power supply 300 is arranged.
  • the power supply control device 2 controls charging/recharging of the DC power supply 300 , detects the resistance value R ESR of the internal resistance 310 in the DC power supply 300 , and outputs an internal resistance value signal expressing the resistance value R ESR to the control device 1 .
  • the control device 1 presumes the resistance value R ESR of the internal resistance 310 included in the DC power supply 300 based on the internal resistance value signal outputted from the power supply control device 2 in step 3 of FIG. 4 when the voltage command compensation amounts V ucomp , V vcomp , V wcomp of each of the U, V, and W phases are calculated by the output voltage error calculation unit 70 . With respect to the points other than this, processes similar to those explained in the first embodiment are executed.
  • step S 3 the control device 1 presumes the internal resistance value R ESR of the DC power supply 300 based on the internal resistance value signal outputted from the power supply control device 2 controlling the DC power supply 300 .
  • the internal resistance value R ESR of the DC power supply 300 detected by the power supply control device 2 is transferred to the control device 1 as the internal resistance value signal, even when the internal resistance value R ESR may change by the operation state of the DC power supply 300 , the internal resistance value R ESR can be presumed precisely, and the variation amount of the input voltage V dc can be calculated accurately.
  • FIG. 8 is a drawing showing a configuration of an electric vehicle system related to the third embodiment of the present invention.
  • the electric vehicle system shown in FIG. 8 is mounted on a body 700 of a hybrid electric automobile, and includes the motor drive system explained in the first and second embodiments respectively.
  • the inverter circuit 100 is operated based on the switching signal outputted from the control device 1 , and executes power conversion from DC power to AC power.
  • the motor 200 is driven using AC power outputted from the inverter circuit 100 .
  • the electric vehicle system can travel using a drive force of the motor 200 .
  • the motor 200 acts not only as a motor generating a rotational drive force but also as a generator receiving a drive force and generating power. That is to say, the electric vehicle system of FIG. 8 includes a power train to which the motor 200 is applied as a motor/generator.
  • a front wheel axle 701 is pivotally supported by the front portion of the body 700 in a rotatable manner, and front wheels 702 , 703 are arranged at both ends of the front wheel axle 701 .
  • a rear wheel axle 704 is pivotally supported by the rear portion of the body 700 in a rotatable manner, and rear wheels 705 , 706 are arranged at both ends of the rear wheel axle 704 .
  • a differential gear 711 that is a power distribution mechanism is arranged, and it is configured that a rotational drive force transmitted from an engine 710 through a transmission 712 is distributed to the front wheel axle 701 of the left and right.
  • a pulley arranged on a crankshaft of the engine 710 and a pulley arranged on a rotary shaft of the motor 200 are mechanically connected to each other through a belt 730 .
  • a rotational drive force of the motor 200 is transmitted to the engine 710 and a rotational drive force of the engine 710 is transmitted to the motor 200 respectively.
  • the motor 200 With respect to the motor 200 , by supply of three-phase AC power controlled by the inverter circuit 100 to a stator coil of a stator, the rotor rotates, and a rotational drive force according to the three-phase AC power is generated. That is to say, the motor 200 is controlled by the inverter circuit 100 and operates as a motor. On the other hand, the rotor rotates receiving the rotational drive force of the engine 710 , thereby an electromotive force is induced in the stator coil of the stator, and the motor 200 operates as a generator generating three-phase AC power.
  • the inverter circuit 100 is a power conversion device converting DC power supplied from the DC power supply 300 that is a high-tension (42V or 300V for example) system power supply to three-phase AC power, and controls three-phase AC current flowing through the stator coil of the motor 200 matching the magnetic pole position of the rotor according to the operation command value.
  • the three-phase AC power generated by the motor 200 is converted to DC power by the inverter circuit 100 , and charges the DC power supply 300 .
  • the DC power supply 300 is electrically connected to a low-tension battery 723 through a DC-DC converter 724 .
  • the low-tension battery 723 configures a low-tension (12V for example) system power supply of an automobile, and is used for a power supply of a starter 725 for initial start-up (cold start-up) of the engine 710 , an auxiliary group such as a radio and a light, and so on.
  • a vehicle is stopping (idle stop mode) such as waiting at a traffic light
  • the engine 710 is to be stopped and is to be restarted (hot start-up) in restarting the vehicle
  • the motor 200 is driven by the inverter circuit 100 , and the engine 710 is restarted.
  • the amount of charge of the DC power supply 300 is insufficient or when the engine 710 has not been warmed up sufficiently and so on, even in the idle stop mode, it is preferable not to stop the engine 710 but to continue driving of the engine 710 .
  • a drive source of an auxiliary machine group whose drive source is the engine 710 such as a compressor of an air conditioner. In this case, it is also possible to drive the motor 200 instead of the engine 710 to be used as the drive source of the auxiliary machine group.
  • the motor 200 when a vehicle is in the acceleration mode or the high load operation mode, the motor 200 is driven to assist driving of the engine 710 .
  • the charging mode when charging of the DC power supply 300 is necessary, the motor 200 is made to generate power by the engine 710 and the DC power supply 300 is charged.
  • the regeneration mode is applied and it is possible that the motor 200 is made to generate power by kinetic energy of the vehicle and the DC power supply 300 is charged.
  • the electric vehicle system includes the control device 1 , the inverter circuit 100 controlled by the control device 1 and converting DC power to AC power, and the motor 200 that is driven using AC power outputted from the inverter circuit 100 .
  • the control device 1 the inverter circuit 100 controlled by the control device 1 and converting DC power to AC power
  • the motor 200 that is driven using AC power outputted from the inverter circuit 100 .

Abstract

This control device for controlling an inverter circuit calculates an input current of the inverter circuit on the basis of an output current instruction value for controlling an output current of the inverter circuit, and calculates, on the basis of the calculated input current, an output voltage compensation amount according to the fluctuation amount of an input voltage of the inverter circuit.

Description

TECHNICAL FIELD
The present invention relates to a control device of an inverter circuit and an electric vehicle including the control device.
BACKGROUND ART
The control device of an inverter circuit mounted on a hybrid automobile, an electric automobile, and the like is desired to be highly responsive. With respect to the request, there is proposed a technology for securing control responsiveness of the output voltage reflecting change in the internal resistance of DC power supply.
As a background art of the present technical field, Patent Literatures 1 is known. In Patent Literatures 1, it is described that an output voltage control system includes a feedback arithmetic unit executing feedback control using feedback gains from a feedback gain determination unit, the feedback control being for making deviation of the output voltage of the DC/DC converter to the target voltage to be zero, and the feedback gain determination unit determines the feedback gains so as to reflect change in internal resistance corresponding to the charging rate at a DC power supply (battery). Thus, control responsiveness of output voltage is secured properly reflecting change in the internal resistance of DC power supply.
CITATION LIST Patent Literature
  • Patent Literature 1: JP-A No. 2007-068290
SUMMARY OF INVENTION Technical Problem
According to the method described in Patent Literatures 1, there was a concern that, when the control deviation increased/decreased sharply, response of the feedback control amount delayed and sufficient response performance was not secured.
In view of the problem described above, the main object of the present invention is to improve the response performance of control of an inverter circuit.
Solution to Problem
A control device according to the present invention executes control of an inverter circuit, calculates input current of the inverter circuit based on an output current command value that is for controlling an output current of the inverter circuit, and calculates an output voltage compensation amount depending on a variation amount of an input voltage of the inverter circuit based on the input current calculated.
An electric vehicle according to the present invention includes the control device, an inverter circuit, and a motor, the inverter circuit being controlled by the control device and converting DC power to AC power, the motor being driven using the AC power outputted from the inverter circuit.
Advantageous Effects of Invention
According to the present invention, the response performance of control of the inverter circuit can be improved.
BRIEF DESCRIPTION OF DRAWINGS
FIG. 1 is a drawing showing a configuration of a motor drive system including a control device related to the first embodiment of the present invention.
FIG. 2 is a drawing showing an example of a waveform of input/output voltage and input/output current of an inverter circuit.
FIG. 3 is a block diagram showing a functional configuration of the control device related to the first embodiment of the present invention.
FIG. 4 is a flowchart showing a process procedure of an output voltage error calculation unit.
FIG. 5 is a table showing the relation between output voltage vector Vx and input current idc.
FIG. 6 is a voltage vector diagram showing an example of output voltage error vector ΔVx.
FIG. 7 is a drawing showing a configuration of a motor drive system including a control device related to the second embodiment of the present invention.
FIG. 8 is a drawing showing a configuration of an electric vehicle system related to the third embodiment of the present invention.
 DESCRIPTION OF EMBODIMENTS
Embodiments of the present invention will be hereinafter explained referring to the drawings. However, the present invention is not to be interpreted to be limited to the embodiments described below, and technological thoughts of the present invention may be achieved combining known other configuration elements. Also, in each drawing, identical elements will be marked with a same reference sign and duplicated explanation thereof will be omitted.
First Embodiment
FIG. 1 is a drawing showing a configuration of a motor drive system including a control device related to the first embodiment of the present invention. The motor drive system shown in FIG. 1 includes an inverter circuit 100, a control device 1 for controlling the inverter circuit 100, a motor 200, a position sensor 210, a current sensor 220, and a DC power supply 300.
The motor 200 is a three-phase AC motor, and is driven using three-phase AC power outputted from the inverter circuit 100.
The position sensor 210 detects the position of a rotor of the motor 200, and outputs rotor position θ detected.
The current sensor 220 detects current of each phase flowing through the motor 200, and outputs three-phase current values iu, iv, iw detected.
The control device 1 executes PWM control that is for controlling the inverter circuit 100 based on a torque command T* outputted from the outside, the three-phase current values iu, iv, iw detected by the current sensor 220, and the rotor position θ detected by the position sensor 210. Thus, the control device 1 generates a switching signal that is for controlling respective switching elements included in the inverter circuit 100, and outputs the switching signal to the inverter circuit 100. Also, the detail of the PWM control executed by the control device 1 will be described below.
The inverter circuit 100 includes switching elements 110 a to 110 f. The switching elements 110 a is arranged in the U-phase upper arm, the switching elements 110 b is arranged in the U-phase lower arm, the switching elements 110 c is arranged in the V-phase upper arm, the switching elements 110 d is arranged in the V-phase lower arm, the switching elements 110 e is arranged in the W-phase upper arm, and the switching elements 110 f is arranged in the W-phase lower arm respectively. The switching elements 110 a to 110 f are configured respectively by combining a semiconductor element and a diode, the semiconductor element being capable of on/off operation such as a metal oxide semiconductor field effect transistor (MOSFET) and an insulated gate bipolar transistor (IGBT) for example.
The switching elements 110 a to 110 f are turned on or off individually based on a switching signal inputted from the control device 1, and convert DC power supplied from the DC power supply 300 to three-phase AC power. U-phase AC power is generated by the switching elements 110 a, 110 b, V-phase AC power is generated by the switching elements 110 c, 110 d, and W-phase AC power is generated by the switching elements 110 e, 110 f respectively. The three-phase AC power generated thus is outputted from the inverter circuit 100 to a stator of the motor 200, and generates three-phase AC current in the motor 200. This three-phase AC current generates a rotating magnetic field in the motor 200, thereby the rotor rotates and the motor 200 is driven.
The DC power supply 300 is connected to an input terminal of the inverter circuit 100, and supplies DC power required for driving the motor 200. The DC power supply 300 includes an internal resistance 310 and a voltage supply 320. DC voltage applied from the DC power supply 300 to the inverter circuit 100 changes according to the state of the DC power supply 300. For example, when the DC power supply 300 is a secondary battery such as a lead-acid storage battery and a lithium-ion battery, since the voltage of the voltage supply 320 changes according to the charging rate of the DC power supply 300, the output voltage of the DC power supply 300 also changes. Further, when current flows through the DC power supply 300, voltage drop according to a resistance value RESR of the internal resistance 310 is generated, and the output voltage of the DC power supply 300 thereby changes. Since this resistance value RESR generally changes according to the temperature, variation occurs in the output voltage of the DC power supply 300 also by the internal temperature of the DC power supply 300.
FIG. 2 is a drawing showing an example of a waveform of input/output voltage and input/output current of the inverter circuit 100. FIG. 2 (a) expresses an example of DC input voltage Vdc and line output voltage vuv, vvw, vwu of each phase of the inverter circuit 100, and FIG. 2 (b) expresses an example of DC input current idc and output current iu, iv, iw of each phase of the inverter circuit 100. Also, the interval between each point set at an equal interval on the input voltage Vdc in FIG. 2 (a) corresponds to the control frequency of the inverter circuit 100 by the control device 1.
To the inverter circuit 100, input voltage Vdc according to the output voltage of the DC power supply 300 is applied. At this time, as shown in FIG. 2 , the input voltage Vdc of a period when input current idc does not flow through the inverter circuit 100 is same to the voltage of the voltage supply 320 of a case of opening the output terminal of the DC power supply 300. On the other hand, the input voltage Vdc of a period when the input current idc flows through the inverter circuit 100 drops compared to that in the period when the input current idc does not flow through the inverter circuit 100 due to the voltage drop according to the resistance value RESR of the internal resistance 310 described above. As a result, in the period when the input current idc flows through the inverter circuit 100, the amplitude of the line output voltage vuv, vvw, vwu of each phase also lowers. This event becomes conspicuous as the capacitance of a smoothing capacitor mounted on the inverter circuit 100 is smaller.
In a control device of an inverter circuit according to a prior art, the fact that the amplitude of the line output voltage vuv, vvw, vwu of each phase changes according to presence/absence of the input current of the inverter circuit as described above has not been considered. Therefore, the inverter circuit could not output voltage in accordance with the command value form the control device, and there was a concern of causing deterioration of response performance and instability of control. On the other hand, in the embodiment of the present invention described below, in the control device 1, the response performance is improved than before by applying the control considering the point described above to the inverter circuit 100.
FIG. 3 is a block diagram showing a functional configuration of the control device 1 related to the first embodiment of the present invention. In FIG. 3 , the control device 1 includes respective functional blocks of a current command arithmetic unit 10, a dq axis current control unit 20, coordinate conversion units 30, 31, a PWM signal generation unit 40, a dq conversion unit 50, a velocity conversion unit 60, and an output voltage error calculation unit 70. The control device 1 is configured of a microcomputer for example, and can achieve these functional blocks by executing a predetermined program in the microcomputer. Alternatively, a part or all of these functional blocks may be achieved by using a hardware circuit such as a logic IC and an FPGA.
The current command arithmetic unit 10 calculates a d-axis current command value id* and a q-axis current command value iq*which are for controlling the output current of the inverter circuit 100 based on the torque command value T* inputted and an angular velocity ω calculated by the velocity conversion unit 60.
To the dq axis current control unit 20, there are inputted difference of the d-axis current command value id* calculated by the current command arithmetic unit 10 and a d-axis current detection value id outputted from the dq conversion unit 50 based on the three-phase current iu, iv, iw detected by the current sensor 220 and difference of the q-axis current command value iq* calculated by the current command arithmetic unit 10 and a q-axis current detection value iq outputted from the dq conversion unit 50 based on the three-phase current iu, iv, iw detected by the current sensor 220. The dq axis current control unit 20 calculates a d-axis voltage command value vd* and a q-axis voltage command value vq* by executing proportional control and integral control based on a predetermined control gain command value based on the respective differences having been inputted.
To the coordinate conversion unit 30, there are inputted the d-axis voltage command value vd* and the q-axis voltage command value vq* calculated by the dq axis current control unit 20 and the rotor position θ detected by the position sensor 210. The coordinate conversion unit 30 outputs a U-phase voltage command value vu*, a V-phase voltage command value vv*, and a W-phase voltage command value vw* by subjecting the d-axis voltage command value vd* and the q-axis voltage command value vq* to rotated coordinate conversion based on the rotor position θ.
To the coordinate conversion unit 31, there are inputted the d-axis current command value id* and the q-axis current command value iq* calculated by the current command arithmetic unit 10 and the rotor position θ detected by the position sensor 210. The coordinate conversion unit 31 outputs a U-phase current command value iu*, a V-phase current command value vv*, and a W-phase current command value iw* by subjecting the d-axis current command value id* and the q-axis current command value iq* to rotated coordinate conversion based on the rotor position θ.
To the PWM signal generation unit 40, there are inputted the U-phase voltage command value vu*, the V-phase voltage command value vv*, and the W-phase voltage command value vw* outputted from the coordinate conversion unit 30 based on the d-axis voltage command value vd* and the q-axis voltage command value vq* calculated by the dq axis current control unit 20 and a U-phase voltage command compensation amount Vucomp, a V-phase voltage command compensation amount Vvcomp, and a W-phase voltage command compensation amount Vwcomp calculated by the output voltage error calculation unit 70. Based on a sum of these voltage command values and the voltage command compensation amounts of each of the U, V, and W phases, the PWM signal generation unit 40 generates a switching signal (PWM signal) which is for controlling turning on/off of the switching elements 110 a to 110 f of each phase included in the inverter circuit 100. Thus, with respect to each of the U, V, and W phases, the voltage command values vu*, vv*, vw* are subjected to compensation respectively based on the voltage command compensation amounts Vucomp, Vvcomp, Vwcomp and the switching signal is generated. Therefore, feedforward compensation of the output voltage command value with respect to the inverter circuit 100 can be achieved.
To the dq conversion unit 50, there are inputted the three-phase current iu, iv, iw detected by the current sensor 220 and the rotor position θ detected by the position sensor 210. Based on these values having been inputted, the dq conversion unit 50 outputs the d-axis current detection value id and the q-axis current detection value iq.
To the velocity conversion unit 60, the rotor position θ detected by the position sensor 210 is inputted. The velocity conversion unit 60 outputs the angular velocity ω based on the rotor position θ, the rotor of the motor 200 rotating with the angular velocity ω.
To the output voltage error calculation unit 70, there are inputted the U-phase current command value iu*, the V-phase current command value iv*, and the W-phase current command value iw* outputted from the coordinate conversion unit 31 based on the d-axis current command value id* and the q-axis current command value iq* calculated by the current command arithmetic unit 10 and the U-phase voltage command value vu*, the V-phase voltage command value vv*, and the W-phase voltage command value vw* outputted from the coordinate conversion unit 30 based on the d-axis voltage command value vd* and the q-axis voltage command value vq* calculated by the dq axis current control unit 20. Based on these values having been inputted, the output voltage error calculation unit 70 calculates the U-phase voltage command compensation amount Vucomp, the V-phase voltage command compensation amount Vucomp, and the W-phase voltage command compensation amount Vwcomp according to the variation amount of the input voltage Vdc of the inverter circuit 100. Also, with respect to the calculation method of the voltage command compensation amounts Vucomp, Vvcomp, Vwcomp of each of the U, V, and W phases by the output voltage error calculation unit 70 will be explained below referring to FIGS. 4, 5, and 6.
FIG. 4 is a flowchart showing a process procedure of the output voltage error calculation unit 70.
First, in step S1, the output voltage error calculation unit 70 calculates the output voltage vector Vx according to the state of the switching elements 110 a to 110 f of the inverter circuit 100 from the voltage command values vu*, vv*, vw* of each of the U, V, and W phases having been inputted and a period Tx of each output voltage vector Vx. The output voltage vector Vx is one obtained by expressing the combination of the switching elements 110 a to 110 f each of which being in a state of being turned on or off using 8 kinds of vector (x=0 to 7). This can be obtained by a known method such as the space vector modulation.
Next, in step S2, the output voltage error calculation unit 70 calculates the input current idc flowing through the inverter circuit 100 from the DC power supply 300 for each output voltage vector Vx calculated in step S1 and the period Tx when each output voltage vector Vx is outputted from the current command values iu*, iv*, iw* of each of the U, V, and W phases having been inputted. Here, the input current idc corresponding to each output voltage vector Vx is calculated by referring a table shown in FIG. 5 described below for example.
Next, in step S3, the output voltage error calculation unit 70 presumes the resistance value RESR of the internal resistance 310 included in the DC power supply 300. Here, for example, by setting beforehand table data determining the internal resistance value for each state of the DC power supply 300 and referring to these table data, the resistance value RESR corresponding to the present state of the DC power supply 300 can be presumed. When the DC power supply 300 is a secondary battery such as a lead-acid storage battery and a lithium-ion battery for example, the output voltage of the DC power supply 300 changes according to the charging rate and the internal temperature as described above. Therefore, by setting beforehand corresponding internal resistance value as the table data for each of the charging rate and the internal temperature of the DC power supply 300, an appropriate resistance value RESR can be presumed based on these table data.
Thereafter, in step S4, the output voltage error calculation unit 70 calculates the output voltage error vector ΔVx according to the error of the output voltage of the inverter circuit 100 generated due to the variation of the input voltage Vdc for each period Tx when each output voltage vector Vx is outputted. Here, the output voltage error vector ΔVx is calculated by the expression (1) below for example. In the expression (1), RESR expresses the resistance value of the internal resistance 310 presumed in step S3, and idc expresses the input current calculated in step S2. Also, the direction of the output voltage error vector ΔVx obtained by the expression (1) is defined to be opposite of that of the output voltage vector Vx.
ΔV x =R ESR ×i dc ×T x (x=0,1,2,3,4,5,6,7)  (1)
Further, with respect to the output voltage error vector ΔVx calculated in step S4, the calculation accuracy of it can be improved by executing the calculation repeatedly.
Lastly, in step S5, the output voltage error calculation unit 70 calculates the voltage command compensation amounts Vucomp, Vvcomp, Vwcomp of each of the U, V, and W phases by subjecting the output voltage error vector ΔVx calculated in step S4 to coordinate conversion to a three-phase voltage value. Thus, with respect to each combination of the state of turning on or off of the switching elements 110 a to 110 f of each of the U, V, and W phases included in the inverter circuit 100, the variation amount of the input voltage Vdc to the inverter circuit 100 is calculated, and the voltage command compensation amounts Vucomp, Vvcomp, Vwcomp for the output voltage are calculated.
FIG. 5 is a table showing the relation between the output voltage vector Vx and the input current idc used when the output voltage error calculation unit 70 calculates the input current idc in step S2 described above. In the table of FIG. 5 , with respect to each of 8 kinds of the output voltage vector Vx (x=0 to 7), the on/off state of the switching element of each of the U, V, and W phases is shown, and the correspondence relation of the phase current and the input current idc is shown. Also, in FIG. 5 , the on/off state of the switching element of each of the U, V, and W phases is expressed by “0” or “1”. “0” expresses the state that the upper arm switching element of a phase in question is turned off and the lower arm switching element of the phase in question is turned on, and “1” expresses the state that the upper arm switching element of a phase in question is turned on and the lower arm switching element of the phase in question is turned off.
In the V0 vector period when the lower arm switching elements of all phases are turned on, the output currents iu, iv, iw of three phases recirculate without going through the DC power supply 300. Therefore, as shown in the table of FIG. 5 , the input current idc becomes 0. In the V1 vector period when only the upper arm switching element of the U phase is turned on, all of the input current idc flows in to the U phase in the inverter circuit 100. Therefore, as shown in the table of FIG. 5 , idc=iu is effected. In the V2 vector period when the upper arm switching elements of the U phase and the V phase are turned on and the upper arm switching element of the W phase is turned off, the W phase current iw flows toward the DC power supply 300 from the inverter circuit 100. Therefore, as shown in the table of FIG. 5 , idc=−iw is effected.
In the period when other voltage vectors are outputted, in a similar manner, the input current idc of the inverter circuit 100 can be obtained by any one of the output currents iu, iv, iw of three phases. Here, in the control block diagram of FIG. 3 , it is configured that the output voltage error calculation unit 70 presumes the output currents iu, iv, iw in the next control period using the current command values iu*, iv*, iw* of each of the U, V, and W phases outputted from the coordinate conversion unit 31 by calculating, with the current command arithmetic unit 10, the d-axis current command value id* and the q-axis current command value iq*, and calculates the input current idc in the next control period using them.
FIG. 6 is a voltage vector diagram showing an example of the output voltage error vector ΔVx calculated by the output voltage error calculation unit 70 in step S4 described above. Also, in the example of FIG. 6 , such case is assumed that an output voltage command vector vout* which is the vector sum of the voltage command values vu*, vv*, vw* of each of the U, V, and W phases in a certain period is within a range surrounded by the V1 vector and the V2 vector. In this case, first, the magnitude |ΔV1| of the output voltage error vector generated in the period T1 when the inverter circuit 100 outputs the V1 vector as the output voltage vector Vx is obtained by the expression (2) below by substituting the value of the input current idc (idc=iu) obtained from the table of FIG. 5 into the expression (1) described above.
ΔV 1 =R ESR ×i u ×T 1  (2)
In a similar manner, the magnitude ≡ΔV2| of the output voltage error vector generated in the period T2 when the inverter circuit 100 outputs the V2 vector as the output voltage vector Vx is obtained by the expression (3) below.
ΔV 2 =R ESR×(−i w)×T 2  (3)
On the other hand, in the period T0 when the V0 vector is outputted and the period T7 when the V7 vector is outputted, since the current does not flow through the DC power supply 300, the input voltage Vdc of the inverter circuit 100 does not change, and the output voltage error vector is not generated. Therefore, as shown in FIG. 6 , the output voltage error Δvout generated during the control period in question (during the carrier period) becomes the vector sum of the output voltage error vector ΔV1 and the output voltage error vector ΔV2. As a result, the output voltage vout of the inverter circuit 100 becomes the vector sum of the output voltage command vector vout* and the output voltage error Δvout, and, as it is, the voltage in accordance with the command is not outputted.
In the control device 1 of the present embodiment, the voltage command compensation amounts Vucomp, Vvcomp, Vwcomp of each phase calculated by the output voltage error calculation unit 70 in step S5 of FIG. 4 are added to the voltage command values vu*, vv*, vw* of each phase respectively in the PWM signal generation unit 40. Thus, the output voltage error Δvout which is the vector sum of the output voltage error vector ΔVx calculated in step S4 is added to the output voltage command vector vout* beforehand. Therefore, feedforward compensation of the output voltage command value can be achieved.
According to the first embodiment of the present invention explained above, actions and effects described below are exhibited.
    • (1) The control device 1 controls the inverter circuit 100, calculates the input current idc of the inverter circuit 100 based on the d-axis current command value id* and the q-axis current command value iq* which are the output current command values for controlling the output current of the inverter circuit 100, and calculates the U-phase voltage command compensation amount Vucomp, the V-phase voltage command compensation amount Vvcomp, and the W-phase voltage command compensation amount Vwcomp which are the output voltage compensation amounts according to the variation amount of the input voltage Vdc of the inverter circuit 100 based on the input current idc having been calculated. By doing so, the response performance of control of the inverter circuit 100 can be improved.
    • (2) The control device 1 presumes the internal resistance value RESR of the DC power supply 300 supplying DC power to the inverter circuit 100 (Step S3), and calculates the output voltage error vector ΔVx expressing the variation amount of the input voltage Vdc based on the product of the internal resistance value RESR having been presumed and the input current ide (step S4). By doing so, even when the internal resistance value RESR changes according to the state of the DC power supply 300 and the variation amount of the input voltage Vdc changes accompanying it, the variation amount of the input voltage Vdc can be obtained precisely.
    • (3) In step S3, the control device 1 presumes the internal resistance value RESR of the DC power supply 300 based on the internal resistance value set beforehand for each state of the DC power supply 300. By doing so, the internal resistance value RESR in accordance with the state of the DC power supply 300 can be presumed precisely.
    • (4) The control device 1 calculates the output voltage error vector ΔVx expressing the variation amount of the input voltage Vdc in step S4 for each output voltage vector Vx expressing the combination of the state of turning on or off of the switching elements 110 a to 110 f of each phase included in the inverter circuit 100, and calculates the voltage command compensation amounts Vucomp, Vvcomp, Vwcomp of each phase (step S5). By doing so, the output voltage compensation amount can be calculated so that the error of the output voltage generated during each control period according to the operation state of the inverter circuit 100 can be surely compensated.
    • (5) The control device 1 includes the current command arithmetic unit 10, the dq axis current control unit 20, the coordinate conversion unit 30, the PWM signal generation unit 40 generating a switching signal that is for controlling turning on/off of the switching elements 110 a to 110 f of each phase included in the inverter circuit 100, and the output voltage error calculation unit 70. The current command arithmetic unit 10 calculates the d-axis current command value id* and the q-axis current command value iq* which are the output current command values of the inverter circuit 100. The dq axis current control unit 20 and the coordinate conversion unit 30 calculates the U-phase voltage command value vu*, the V-phase voltage command value vv*, and the W-phase voltage command value vw* which are the output voltage command values of the inverter circuit 100 based on the d-axis current command value id* and the q-axis current command value iq* calculated by the current command arithmetic unit 10. The PWM signal generation unit 40 generates a switching signal based on the voltage command values vu*, vv*, and vw* of each phase calculated by the dq axis current control unit 20 and the coordinate conversion unit 30. The output voltage error calculation unit 70 calculates the voltage command compensation amounts Vucomp, Vvcomp, and Vwcomp of each phase based on the current command values iu*, iv*, iw* of each phase and the voltage command values vu*, vv*, and vw* of each phase outputted from the coordinate conversion unit 31 based on the d-axis current command value id* and the q-axis current command value iq*. Here, the PWM signal generation unit 40 compensates the voltage command values vu*, vv*, vw* of each phase based on the voltage command compensation amounts Vucomp, Vvcomp, Vwcomp of each phase calculated by the output voltage error calculation unit 70, and generates a switching signal. By doing so, feedforward compensation of the output voltage command value with respect to the inverter circuit 100 can be achieved while properly controlling the inverter circuit 100.
Second Embodiment
FIG. 7 is a drawing showing a configuration of a motor drive system including a control device related to the second embodiment of the present invention. Compared to the motor drive system of FIG. 1 explained in the first embodiment, the motor drive system shown in FIG. 7 is different in terms that a power supply control device 2 controlling the DC power supply 300 is arranged. The power supply control device 2 controls charging/recharging of the DC power supply 300, detects the resistance value RESR of the internal resistance 310 in the DC power supply 300, and outputs an internal resistance value signal expressing the resistance value RESR to the control device 1.
In the present embodiment, the control device 1 presumes the resistance value RESR of the internal resistance 310 included in the DC power supply 300 based on the internal resistance value signal outputted from the power supply control device 2 in step 3 of FIG. 4 when the voltage command compensation amounts Vucomp, Vvcomp, Vwcomp of each of the U, V, and W phases are calculated by the output voltage error calculation unit 70. With respect to the points other than this, processes similar to those explained in the first embodiment are executed.
According to the second embodiment of the present invention explained above, in step S3, the control device 1 presumes the internal resistance value RESR of the DC power supply 300 based on the internal resistance value signal outputted from the power supply control device 2 controlling the DC power supply 300. By doing so, since the internal resistance value RESR of the DC power supply 300 detected by the power supply control device 2 is transferred to the control device 1 as the internal resistance value signal, even when the internal resistance value RESR may change by the operation state of the DC power supply 300, the internal resistance value RESR can be presumed precisely, and the variation amount of the input voltage Vdc can be calculated accurately.
Third Embodiment
Next, the third embodiment of the present invention will be explained. In the present embodiment, an application example to an electric vehicle system will be explained.
FIG. 8 is a drawing showing a configuration of an electric vehicle system related to the third embodiment of the present invention. The electric vehicle system shown in FIG. 8 is mounted on a body 700 of a hybrid electric automobile, and includes the motor drive system explained in the first and second embodiments respectively. In the electric vehicle system of the present embodiment, the inverter circuit 100 is operated based on the switching signal outputted from the control device 1, and executes power conversion from DC power to AC power. The motor 200 is driven using AC power outputted from the inverter circuit 100. Thus, the electric vehicle system can travel using a drive force of the motor 200.
Also, in the electric vehicle system of the present embodiment, the motor 200 acts not only as a motor generating a rotational drive force but also as a generator receiving a drive force and generating power. That is to say, the electric vehicle system of FIG. 8 includes a power train to which the motor 200 is applied as a motor/generator.
A front wheel axle 701 is pivotally supported by the front portion of the body 700 in a rotatable manner, and front wheels 702, 703 are arranged at both ends of the front wheel axle 701. A rear wheel axle 704 is pivotally supported by the rear portion of the body 700 in a rotatable manner, and rear wheels 705, 706 are arranged at both ends of the rear wheel axle 704. At the center portion of the front wheel axle 701, a differential gear 711 that is a power distribution mechanism is arranged, and it is configured that a rotational drive force transmitted from an engine 710 through a transmission 712 is distributed to the front wheel axle 701 of the left and right.
With respect to the engine 710 and the motor 200, a pulley arranged on a crankshaft of the engine 710 and a pulley arranged on a rotary shaft of the motor 200 are mechanically connected to each other through a belt 730. Thus, it is configured that a rotational drive force of the motor 200 is transmitted to the engine 710 and a rotational drive force of the engine 710 is transmitted to the motor 200 respectively.
With respect to the motor 200, by supply of three-phase AC power controlled by the inverter circuit 100 to a stator coil of a stator, the rotor rotates, and a rotational drive force according to the three-phase AC power is generated. That is to say, the motor 200 is controlled by the inverter circuit 100 and operates as a motor. On the other hand, the rotor rotates receiving the rotational drive force of the engine 710, thereby an electromotive force is induced in the stator coil of the stator, and the motor 200 operates as a generator generating three-phase AC power.
The inverter circuit 100 is a power conversion device converting DC power supplied from the DC power supply 300 that is a high-tension (42V or 300V for example) system power supply to three-phase AC power, and controls three-phase AC current flowing through the stator coil of the motor 200 matching the magnetic pole position of the rotor according to the operation command value.
The three-phase AC power generated by the motor 200 is converted to DC power by the inverter circuit 100, and charges the DC power supply 300. The DC power supply 300 is electrically connected to a low-tension battery 723 through a DC-DC converter 724. The low-tension battery 723 configures a low-tension (12V for example) system power supply of an automobile, and is used for a power supply of a starter 725 for initial start-up (cold start-up) of the engine 710, an auxiliary group such as a radio and a light, and so on.
In a case a vehicle is stopping (idle stop mode) such as waiting at a traffic light, when the engine 710 is to be stopped and is to be restarted (hot start-up) in restarting the vehicle, the motor 200 is driven by the inverter circuit 100, and the engine 710 is restarted. However, when the amount of charge of the DC power supply 300 is insufficient or when the engine 710 has not been warmed up sufficiently and so on, even in the idle stop mode, it is preferable not to stop the engine 710 but to continue driving of the engine 710. Also, during the idle stop mode, it is necessary to secure a drive source of an auxiliary machine group whose drive source is the engine 710 such as a compressor of an air conditioner. In this case, it is also possible to drive the motor 200 instead of the engine 710 to be used as the drive source of the auxiliary machine group.
On the other hand, when a vehicle is in the acceleration mode or the high load operation mode, the motor 200 is driven to assist driving of the engine 710. To the contrary, in the charging mode when charging of the DC power supply 300 is necessary, the motor 200 is made to generate power by the engine 710 and the DC power supply 300 is charged. Also, when the vehicle is to be braked or decelerated and so on, the regeneration mode is applied and it is possible that the motor 200 is made to generate power by kinetic energy of the vehicle and the DC power supply 300 is charged.
According to the third embodiment of the present invention explained above, the electric vehicle system includes the control device 1, the inverter circuit 100 controlled by the control device 1 and converting DC power to AC power, and the motor 200 that is driven using AC power outputted from the inverter circuit 100. By doing so, it is possible to construct an electric vehicle system having high response performance of control in being driven by the motor 200.
The embodiments and various modifications explained above are examples only, and the present invention is not to be limited to the content of them as far as the feature of the present invention is not spoiled. Further, although various embodiments and modifications were explained above, the present invention is not to be limited to the content of them. Other aspects that can be considered within the range of the technical thought of the present invention also are to be included within the range of the present invention.
The disclosure of the following priority right basic application is hereby incorporated herein by reference.
  • Japanese Patent Application No. 2019-137903 (applied on Jul. 26, 2019)
LIST OF REFERENCE SIGNS
    • 1: Control device
    • 2: Power supply control device
    • 10: Current command arithmetic unit
    • 20: dq axis current control unit
    • 30, 31: Coordinate conversion unit
    • 40: PWM signal generation unit
    • 50: dq conversion unit
    • 60: Velocity conversion unit
    • 70: Output voltage error calculation unit
    • 100: Inverter circuit
    • 110 a: U-phase upper arm switching element
    • 110 b: U-phase lower arm switching element
    • 110 c: V-phase upper arm switching element
    • 110 d: V-phase lower arm switching element
    • 110 e: W-phase upper arm switching element
    • 110 f: W-phase lower arm switching element
    • 200: Motor
    • 210: Position sensor
    • 220: Current sensor
    • 300: DC power supply
    • 310: Internal resistance
    • 320: Voltage source

Claims (7)

The invention claimed is:
1. A control device executing control of an inverter circuit, wherein
an input current of the inverter circuit is calculated based on an output current command value that is for controlling an output current of the inverter circuit, and
an output voltage compensation amount depending on a variation amount of input voltage of the inverter circuit is calculated based on the input current calculated.
2. The control device according to claim 1, wherein
an internal resistance value of a power supply supplying DC power to the inverter circuit is presumed, and
a variation amount of the input voltage is calculated based on a product of the internal resistance value presumed and the input current.
3. The control device according to claim 2, wherein
an internal resistance value of the power supply is presumed based on an internal resistance value having been set beforehand for each state of the power supply.
4. The control device according to claim 2, wherein
an internal resistance value of the power supply is presumed based on an internal resistance value signal outputted from a power supply control device controlling the power supply.
5. The control device according to claim 1, wherein
a variation amount of the input voltage is calculated and the output voltage compensation amount is calculated for each combination of a state of turning on or turning off of switching elements of respective phases included in the inverter circuit.
6. The control device according to claim 1, comprising:
a current command arithmetic unit calculating the output current command value;
a current control unit calculating an output voltage command value of the inverter circuit based on the output current command value calculated by the current command arithmetic unit;
a PWM signal generation unit generating a switching signal that is for controlling of on/off of switching elements of respective phases included in the inverter circuit based on the output voltage command value calculated by the current control unit, and
an output voltage error calculation unit calculating the output voltage compensation amount based on the output current command value and the output voltage command value, wherein
the PWM signal generation unit compensates the output voltage command value based on the output voltage compensation amount calculated by the output voltage error calculation unit, and generates the switching signal.
7. An electric vehicle, comprising:
the control device according to claim 1;
an inverter circuit that is controlled by the control device and converts DC power to AC power; and
a motor that is driven using the AC power outputted from the inverter circuit.
US17/611,301 2019-07-26 2020-07-14 Control device and electric vehicle Active 2041-02-17 US11855564B2 (en)

Applications Claiming Priority (3)

Application Number Priority Date Filing Date Title
JP2019137903A JP7148463B2 (en) 2019-07-26 2019-07-26 Control devices, electric vehicles
JP2019-137903 2019-07-26
PCT/JP2020/027404 WO2021020115A1 (en) 2019-07-26 2020-07-14 Control device and electric vehicle

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Citations (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2007068290A (en) 2005-08-30 2007-03-15 Toyota Motor Corp Voltage conversion system

Family Cites Families (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP5478190B2 (en) * 2009-10-23 2014-04-23 株式会社豊田中央研究所 DCDC converter system
JP5488097B2 (en) * 2010-03-24 2014-05-14 トヨタ自動車株式会社 Current estimation device and DCDC converter control system

Patent Citations (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2007068290A (en) 2005-08-30 2007-03-15 Toyota Motor Corp Voltage conversion system

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
International Search Report of PCT/JP2020/027404 dated Sep. 24, 2020.

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