TWI477047B - High boost power conversion device - Google Patents

Description

High step-up power conversion device

The present invention relates to a step-up type power conversion device, and more particularly to a high-boost type power conversion device that combines a charge pump and a coupled inductor to obtain a high voltage conversion ratio.

In many power generation systems, such as wind power, solar power, fuel cells, and hybrid vehicles, power converters with high voltage conversion ratios are needed to boost the output voltage of the front-end power generation to higher voltages for later stages. The load is powered. In addition to the above power supply systems, power converters with high voltage conversion ratios are also commonly used in other applications, such as uninterruptible power systems (UPS), gas discharge headlamps (HID) for vehicles, and the like.

In view of this, many scholars have proposed a number of new high-boost converters, such as using the turns ratio of the coupled inductor to increase the voltage-gain ratio; there are also coupling inductors with a voltage doubler circuit, or with switched capacitors. The voltage is superimposed to further increase the voltage conversion ratio; or, the magnetic energy is simultaneously stored by using a plurality of magnetic elements, and simultaneously discharged to the output terminal to increase the voltage gain ratio.

However, the methods for improving the voltage conversion ratio mentioned above have their own shortcomings, including: some technologies use too many circuit components to cause complicated design, resulting in increased circuit cost, and some technologies use floating-type switches and must be isolated and driven; Some technologies require additional switches to implement active clamping circuits, which makes circuit analysis difficult.

Based on the above, there is a need for a technical solution that can increase the voltage conversion ratio and can simplify circuit components.

It is an object of the present invention to provide a high step-up type power conversion device which can improve efficiency and simplify circuit components.

Therefore, the high step-up power conversion device of the present invention comprises a charge pump, a conversion circuit and an output circuit.

The charge pump is configured to receive an input voltage, including a first switch having a first end and a second end, and a second switch connected to the first end of the first switch by a first end, An anode is connected to the pump diode of the second end of the first switch, and a pump capacitor having a first end and a second end, the first end of the pump capacitor is electrically connected to the pump diode a cathode, the second end of the pump capacitor is electrically connected between the first end of the first switch and the first end of the second switch.

The conversion circuit includes a coupled inductor electrically connected to the charge pump and a third switch, the coupled inductor includes a primary winding having a first end and a second end, and a first end and a first end a secondary winding of the second end, the primary winding is coupled at a first end thereof to a cathode of the pump diode and a first end of the pump capacitor, the secondary winding having the first end and the first end The second end of the side winding is coupled to the first end and the second end, and the first end is electrically connected to the second end of the primary winding and the second winding The second end is electrically connected to the second end of the second switch.

The output circuit has an output diode and an output capacitor, the anode of the output diode is coupled to the second end of the secondary winding, and the output capacitor is connected in parallel with the cathode of the output diode, and a first switch, the second switch, and the The third switch is respectively driven by a wave width adjustment control signal to be turned on or off, and the input voltage is boosted and output by the output circuit.

Preferably, if the high-boost power conversion device operates in the continuous conduction mode, the duty cycle intervals of the bandwidth adjustment control signals are D and 1-D, respectively, wherein the interval D is the first switch and the first The third switch is turned on and the second switch is turned off, the interval 1-D is that the second switch is turned on and the first switch and the third switch are turned off; if the operation is in the discontinuous conduction mode, there is still one in addition to the above two intervals In the third interval, the current flowing through the primary winding in the third interval is zero, and the second switch is turned on, the first switch is turned off, and the third switch is turned off.

Preferably, the first switch, the second switch and the third switch are respectively composed of an N-type MOS field-effect transistor and a back-connected diode, and the cathode of the back-connected diode is coupled to the N The source of the MOS field-effect transistor, the anode of the back-connected diode is coupled to the drain of the N-type MOS field-effect transistor.

Preferably, the high-boost power conversion device further includes a clamp diode, the anode of the clamp diode is coupled to the first end of the secondary winding, and the cathode of the clamp diode is coupled. The cathode of the output diode.

Preferably, the high-boost power conversion device further includes a buffer diode and a buffer capacitor. The anode of the buffer diode is coupled to the second end of the primary winding, and the cathode of the buffer diode is coupled. The first end of the secondary winding, the snubber capacitor is connected in parallel between the cathode of the buffer diode and the first end of the secondary winding.

The effect of the high-boost power conversion device of the present invention is that a high rise can be achieved by a coupled inductor, two diodes, a capacitor and three switches. The purpose of the pressure conversion is that the component composition is simple and easy to implement, and the conversion performance of the present invention is also excellent.

The above and other technical contents, features and advantages of the present invention will be apparent from the following detailed description of FIG. Before the present invention is described in detail, it is noted that in the following description, similar elements are denoted by the same reference numerals.

Referring to FIG. 1, in a first preferred embodiment of the present invention, a high step-up power conversion device 100 includes a charge pump 11, a conversion circuit 12, and an output circuit 13.

The charge pump 11 is configured to receive an input voltage v _i , including a first switch S ₁ having a first end 111 and a second end 112 , and a first switch 121 connected to the first switch S ₁ . a second switch S ₂ at one end 111 , a pump diode D _b connected to the second end 112 of the first switch S ₁ by an anode 131 , and a gang having a first end 141 and a second end 142 Pu capacitance C _b, a first end of the pump 141 is electrically connected to the cathode of the capacitance C _b 132 pump diode D _b, the capacitance C _b of the pump 142 is electrically connected to a second terminal of the first switch S ₁ of the first Between the end 111 and the first end 121 of the second switch S ₂ .

The conversion circuit 12 includes a coupled inductor 120 electrically coupled to the charge pump 11 and a third switch S ₃ . The coupled inductor 120 includes a primary winding L _p having a first end 151 and a second end 152 , and a a secondary winding L _s having a first end 161 and a second end 162, the primary side winding L _{p having a} first end 151 and a cathode 132 of the pump diode D _b and a pump capacitor C _b one end 141 coupled to the secondary winding L _s its first end 161 and the primary winding L _p of the second end 152 is coupled to the third switch S ₃ has a first end 171 and a second end 172, And the first end 171 is electrically connected between the second end 152 of the primary side winding L _p and the first end 161 of the secondary side winding L _s , and the second end 172 is electrically connected to the second switch S The second end 122 of ₂ .

The output circuit 13 has an output diode D _o , an output capacitor C _o and an output resistor R _o . The anode 181 of the output diode D _o is coupled to the second end 162 of the secondary winding L _s , and the output capacitor C _o and the output resistor R _{o is} connected in parallel with the cathode 182 of the output diode D _o , and is turned on by the first switch S ₁ , the second switch S ₂ and the third switch S ₃ respectively by receiving a wave width adjustment control signal off and the input voltage or the output V _i boosting an output voltage V _o by the output circuit 13.

In this embodiment, if the high-boost power conversion device 100 operates in the continuous conduction mode, the duty cycle intervals of the bandwidth adjustment control signals are D and 1-D, respectively, wherein the interval D is the first switch S ₁ and The third switch S _{3 is} turned on and the second switch S _{2 is} turned off, the interval 1-D is that the second switch S _{2 is} turned on and the first switch S ₁ and the third switch S _{3 are} turned off; and, the first switch S ₁ , the second switch S ₂ and the third switch S ₃ are respectively composed of an N-type MOSFET and a back-connected diode D ₁ , D ₂ , D ₃ , and are connected back to the diodes D ₁ , D ₂ , D ₃ The cathode is coupled to the source (S) of the N-type MOS field-effect transistor, and the anode connected to the diodes D ₁ , D ₂ , and D ₃ is coupled to the drain (D) of the N-type MOS field-effect transistor.

Referring to FIGS. 2 and 3, the high boost power converter apparatus 100 is operable to inductor current i _Lp continuous conduction mode with the inductor current i _Lp discontinuous conduction mode, the waveform of FIG inductor current i _Lp continuous conduction mode in FIG. 2 and an inductor The waveform of the current i _Lp discontinuous conduction mode is shown in Figure 3.

Before performing the circuit action analysis, a brief description of its associated symbol definition and its required assumptions is made: the input voltage is a fixed value V _i , and the pump capacitor C _b is large enough to make its voltage across the V equal to V. _i . The output capacitor C _o is large enough to have its output voltage at a certain value V _o . i _i is the input current, i _b is the current flowing through the pump capacitor C _b , i _Lp is the current flowing through the primary winding L _p , i _Ls is the current flowing through the secondary winding L _s , i _Co is the current The current through the output capacitor C _o . To couple the magnetic flux of the inductor 120, the coupling coefficient k of the coupled inductor 120 is =1, that is, the leakage inductance is not considered. N _p and N _s are the number of turns of the primary side winding and the secondary side winding of the coupled inductor, respectively, and the turns ratio is n = N _s / N _p . V _gs1 first switch upper arm gate drive signals of S _1, V _gs2 of the lower arm of the second switch S ₂ gate drive signal, V _{gs3 S. 3} of the third switching gate drive signals. T _s is the switching period. The blank time of the two power switches is negligible, and the on-time of the first switch S ₁ and the third switch S ₃ is DT _s , and the on-time of the second switch S ₂ is (1- D ) T _s . When the inductor current i _Lp operated in discontinuous conduction mode, the first switch S ₁ is turned off to the moment the current flowing through the primary inductor i _Lp degaussing winding L _p of the time it takes to zero △ ₁ T _s; and inductor current The time taken for i _{Lp to} be zero to the first switch S _{1 to be} turned on is Δ ₂ T _s , where Δ ₁ T _s + Δ ₂ T _s = (1 - D ) T _s . All power switches, diodes, capacitors and inductors are considered ideal components.

Referring to Figure 4, the equivalent model of the coupled inductor 120, so that the primary winding inductance L is equal to _{L. 1} (not shown) _p of, when the third switch S ₃ is at OFF, by inductive coupling characteristics, so that the primary side The inductance value of the winding L _p and the secondary winding L _{s in} series is equivalent to L ₂ , which is shown in Equation 1.

When the third switch S ₃ is off, the equivalent inductance L _{2 of the} coupled inductor 120 is known by the continuity of the magnetic flux and the ampere theorem. The inductor current i _{Lp of the} circuit is current transient at the switching instant because of the coupled inductor. 120 caused by the difference in the number of turns corresponding to the excitation and demagnetization, as in Equation 2, where It is the reluctance of the coupled inductor core.

Referring to FIG. 5, FIG. 5(a) shows the relationship between the currents i _L1 and i _L2 and i _{Lp in} the continuous conduction mode, and FIG. 5(b) shows the currents i _L1 and i _{L2 in the} discontinuous conduction mode. Diagram of relationship with i _Lp .

The inductor current i _Lp is algebraically converted in different operating states via Equation 2, that is, as shown in Equation 3 and FIG. 5, if the number of turns of the coupled inductor during excitation and demagnetization is fixed to N _p , A continuous current i _{L1 is obtained} . Similarly, if the number of turns of the coupled inductor during excitation and demagnetization is fixed to N _p +N _s , another continuous current i _L2 can be obtained.

Among them, the relationship between i _Lp and i _L1 , i _L2 is obtained by Equation 2 and Equation 3.

In the figure, I _L1 and I _L2 are the average currents of i _L1 and i _L2 , respectively, and satisfy the following relationship:

Referring to FIG. 6, the equivalent current path of state one when the inductor current i _{Lp is} continuously turned on is described as follows: state one is a time interval ( t ₀ t t ₁ ), at this time, the first switch S ₁ and the third switch S _{3 are} turned on, the second switch S _{2 is} turned off, and the pump diode D _b and the output diode D _{o are} turned off. At this time, the inductance L _{p is} excited, and its voltage across is twice the input voltage V _i , as in Equation 6.

v _Lp =2 V _i Equation 6

And the current i _Co flowing through the output capacitor C _o is equal to the negative output current

This time interval ( t ₀ t t ₁ ) ends when the first switch S ₁ and the third switch S _{3 are} turned off and the second switch S _{2 is} turned on.

Referring to FIG. 7, the equivalent current path of the state 2 of the continuous current mode of the inductor current i _Lp is described as follows; the state 2 is the time interval ( t ₁ t t ₀ + T _s ), at this time, the first switch S ₁ and the third switch S ₃ are turned off, the second switch S ₂ is turned on, and the pump diode D _b and the output diode D _{o are} turned on. FIG 7 can be learned from the case demagnetization equivalent inductance L _2, L and cross voltage V _{L2 2} of the input voltage V _i by subtracting the output voltage V _o, as shown in Equation 8.

v _{L 2} = V _i - V _o Equation 8

The above formula can be used to find the voltage across the inductor L _p via the partial pressure theorem, as in Equation 9.

The current i _Co flowing through the output capacitor C _o is equal to the inductor current i _Lp minus the output current, as in Equation 10.

Substituting Equation 4 into Equation 10 yields Equation 11.

This interval is turned on when the first switch S ₁ and the third switch S _{3 are} turned on, and ends when the second switch S _{2 is} turned off.

Through the above analysis and by the inductance across the pressure must meet the volt-second balance, as in Equation 12.

Equation 12 is organized to obtain a voltage conversion when the circuit operates in the continuous current mode of the inductor current i _Lp , such as Equation 13.

In addition, Equation 14 can be obtained by the capacitor current having to meet the second-second balance.

After formula 14 is compiled, the relationship between I _L2 and the output current is obtained as Equation 15.

Referring to Figure 8, the waveform timing diagram of the inductor current at the boundary condition, when the circuit operates in the boundary condition, the current i _Lp and its corresponding i _L2 waveform are as shown in FIG.

When the end of a switching period T _s instant, the current through the inductor L _p i _Lp is zero, the boundary at this time through the inductor L _p of the average current I _LB by Equation 15 Equation 16 is obtained.

The chopping value of the inductor current i _Lp corresponding to i _L2 during the off period of the first switch S ₁ is Equation 17, where i _{L 2, peak} is the peak-to-peak value of i _L2 .

Substituting Formula 1 into Equation 17 is organized to obtain Equation 18.

The condition required for the circuit to operate in the discontinuous conduction mode of the inductor current i _Lp is Equation 19.

I _LB <△ i _{L 2} formula 19

Substituting Equation 13, Equation 16, and Formula 18 into the rewritable Formula 19 is Equation 20.

Among them, order

Referring to Figure 9, a demarcation graph is obtained in which n = 1 is obtained.

Substituting Equation 21 into Equation 20 yields Equation 22.

K < K _crit ( D ) Formula 22

It can be seen from the above equation that when K is less than K _crit ( D ), the circuit will operate in the discontinuous conduction mode of the inductor current i _Lp .

Referring to FIG. 3 and FIG. 5(b), when the circuit operates in the discontinuous conduction mode of the inductor current i _Lp , the circuit action can be divided into three states.

As shown in FIG. 6, the state of the continuous current conduction mode of the inductor current i _Lp is a time interval ( t ₀ t t ₁ ), the first switch S ₁ and the third switch S ₃ are turned on, the second switch S ₂ is turned off, the pump diode D _b and the output diode D _o are turned off; at this time, the voltage across the inductor L _p is Same as Equation 6, and the current i _Co flowing through the output capacitor C _{o is} the same as Equation 7. This interval is ended when the first switch S ₁ and the third switch S _{3 are} turned off, and when the second switch S _{2 is} turned on.

As shown in Figure 7, the state of the inductor current i _Lp continuous conduction mode is the time interval ( t ₁ t t ₂ ), the first switch S ₁ and the third switch S ₃ are off, the second switch S ₂ is turned on, the pump diode D _b , and the output diode D _{o are} turned on; at this time, the voltage across the inductor L _p Same as Equation 9, and the current i _Co flowing through the output capacitor C _{o is} the same as Equation 11. This interval ends when the equivalent inductance L _{2 is} demagnetized until i _{Lp is} equal to zero.

As shown in Figure 7, the state of the inductor current i _Lp continuous conduction mode is the time interval ( t ₂ t t ₀ + T _s ), the first switch S ₁ and the third switch S ₃ are turned off, the second switch S ₂ is turned on, the pump diode D _{b is} turned on, and the output diode D _{o is} turned off. At this time, the voltage across the equivalent inductance L ₂ is zero, so the voltage across the voltage V _{Lp of the} inductance L _p is also zero through the partial pressure theorem.

v _Lp =0 Equation 23

And the current i _Co flowing through the output capacitor C _o is equal to the negative output current

This interval is turned on when the first switch S ₁ and the third switch S _{3 are} turned on, and ends when the second switch S _{2 is} turned off.

Referring to Figure 10, the current path of state three of the inductor current i _Lp discontinuous conduction mode.

Equation 25 is obtained by the above analysis and by the inductance across the pressure whisker in accordance with the volt-second balance.

Formula 26 can be obtained by arranging the above formula.

In addition, Equation 27 can be obtained by the capacitance current to be in accordance with the amperometric balance and Figure 5(b).

After the above formula is sorted, the formula 28 can be obtained.

Can be obtained by the basic formula of inductance As shown in Equation 29, where i _{L 2, peak} is the _peak of current i _L2 .

Substituting Equation 29 into Equation 28 yields Equation 30.

Substituting Equation 26 into Equation 30 yields Equation 31.

Will be in formula 21 Substituting into Equation 31, Equation 32 is obtained.

The output voltage V _o can be obtained by the above equation as Equation 33.

Finally, by dividing Equation 33 by the input voltage V _i , the voltage conversion when the circuit operates in the discontinuous conduction mode of the inductor current i _Lp can be obtained, for example, Equation 34.

In actual operation, the coupling inductance 120 must have a leakage inductance L _LK , so that the third switch S ₃ generates a great voltage surge at the moment of the cutoff. Therefore, the second embodiment of the present invention further includes a passive clamp circuit or a third embodiment of the present invention, and a passive buffer circuit is added to the main circuit of the first embodiment, as shown in FIG. 11(a) as a passive type. A high-boost type power conversion device 200 of a clamp circuit; and FIG. 11(b) is a high-boost type power conversion device 300 having a passive snubber circuit.

Referring to FIG. 11(a) and FIG. 12, in the second embodiment of the present invention, the high-boost type power conversion device 200 includes a clamp diode in addition to the components of FIG. 1 and the same control technology. D _c1 , the anode 201 of the clamp diode D _c1 is coupled to the first end 161 of the secondary winding L _s , and the cathode 202 of the clamp diode D _c1 is coupled to the cathode 182 of the output diode D _o .

The advantage of this architecture is that only one additional clamp diode D _c1 is needed, and when the third switch S _{3 is} turned off, the back electromotive force generated by the leakage inductance L _LK causes the crossover voltage V _{ds3 of the} third switch S _{3 to} be greater than When the voltage V _o is output, the energy of the leakage inductance L _LK will be released to the output via the diode D _c1 .

Referring to FIG. 11(b), in the third embodiment of the present invention, the high-boost type power conversion device 300 includes a buffer diode D _sn and a unit in addition to the components of FIG. 1 and the same control technology. The snubber capacitor C _sn , the anode 301 of the buffer diode D _sn is coupled to the second end 152 of the primary side winding L _p , and the cathode 302 of the buffer diode D _sn is coupled to the first end 161 of the secondary side winding L _s , The snubber capacitor C _{sn is} connected in parallel between the cathode 302 of the buffer diode D _sn and the first end 161 of the secondary winding L _s .

This architecture only requires additional buffer diode D _sn and snubber capacitor C _sn , and its operating state can be divided into state one and state two, as described below.

Referring to Figure 13, a current path state diagram when the third switch S ₃ is turned off instantaneously, the leakage inductance L _LK counter electromotive force is generated such that the diode D _sn buffer and an output diode D _o is turned on, then the leakage inductance L _{The LK is} demagnetized and charges the snubber capacitor C _sn . This state ends when the leakage inductance current i _LK is zero.

Referring to FIG. 14, it is a current path diagram of state 2. At this time, the buffer diode D _{sn is} turned off, the output diode D _{o is} turned on, the snubber capacitor C _{sn is} discharged to the output terminal, and when the snubber capacitor C _{sn is} discharged to a voltage equal to v _{Csn , min} , and the buffer diode D _{sn is} turned on, this state ends.

The system configurations of the high-boost type power conversion devices 200 and 300 are the same, as shown in Table 1, and both operate in the continuous conduction mode of the inductor current i _Lp .

The parameters and selections of the components of the high-boost type power conversion devices 200 and 300 are shown in Table 2 and Table 3, respectively.

15 to 18 are waveform diagrams of the high-boost type power conversion device 200 according to the second embodiment of the present invention under different loads.

Referring to Figure 15, when a high boost at full power conversion device 200, the first gate switch S ₁ of the drive signal V _GS1, the second switch S ₂ of the gate driving signal V _GS2, the third switch S ₃ of the shutter The waveform of the pole drive signal and the voltage across the voltage V _Cb of the v _gs3 and the pump capacitor C _b . As can be seen from Fig. 15, the voltage across the voltage of the pump capacitor v _{Cb is} approximately equal to the input voltage of 5V.

Referring to Figure 16, a high boost power converter apparatus 200 at the time of full load, the first switch S ₁ of the gate electrode drive signals V _GS1, the second switch S ₂ of the gate electrode driving signal V _GS2, flowing through the primary side of the coupled inductor The current i _Lp and the waveform of the current i _Ls flowing through the secondary side of the coupled inductor.

Referring to Figure 17, a high cross-boost power converter apparatus 200 at the time of full load, a first gate switch S ₁ of the drive signal V _GS1, the second switch S ₂ of the gate driving signal V _GS2, the third switch _{S. 3} The waveform of the voltage v _ds3 and the voltage across the voltage of the clamp diode D _c1 v _Dc1 .

18 is a partial enlarged waveform diagram of FIG. 17. As can be seen from FIG. 17 and FIG. 18, when the third switch S ₃ crosses the surge voltage, the clamp diode D _c1 will be simultaneously turned on so that the third switch S ₃ The surge voltage can be clamped to an output voltage of 48V. The ringing phenomenon of the third switch S ₃ and the clamp diode D _c1 in FIG. 18 is caused by the back-connection capacitance of the power component resonating with the stray inductance on the line.

19 to 22 are waveform diagrams of the high-boost type power conversion device 300 according to the third embodiment of the present invention under different loads.

Referring to Figure 19, a high boost power converter 300 at the time of full load, a first gate switch S ₁ of the drive signal V _GS1, the second switch S ₂ of the gate driving signal V _GS2, the third switch S ₃ of the shutter The waveform of the pole drive signal and the voltage across the voltage V _Cb of the v _gs3 and the pump capacitor C _b . As for the high-boost type power conversion device 300, as shown in Fig. 19, the voltage across the voltage of the pump capacitor v _{Cb is} at the input voltage, that is, 5V.

Referring to Figure 20, the boost converter 300 to a high power at full load type, a first switch S ₁ of the gate electrode drive signals V _GS1, the second switch S ₂ of the gate electrode driving signal V _GS2, flowing through the primary side of the coupled inductor The current i _Lp and the waveform of the current i _Ls flowing through the secondary side of the coupled inductor. As can be seen from the comparison of FIG. 20, the primary and secondary currents i _Lp and i _Ls have a ringing phenomenon, because the back-connection capacitance of the third switch S ₃ is also involved in the leakage inductance L _LK when the diode D _{sn is} turned on twice. Resonance behavior with snubber capacitor C _sn .

Referring to Fig. 22, a partially enlarged waveform diagram of Fig. 21 is shown. Referring to Figure 21, a high boost power converter 300 at the time of full load, a first gate switch S ₁ of the drive signal V _GS1, the second switch S ₂ of the gate driving signal V _GS2, the third switch S ₃ Cross The waveform of the voltage v _ds3 and the voltage across the buffer diode D _sn across the voltage V _Dsn . Seen from FIG. 21, the third switch S ₃ is turned off to a surge voltage of about 20V at full load, and is seen from FIG. 22, the instant the first switch S ₁ is turned off, the leakage inductance of the energy by conduction D _sn Transfer to the snubber capacitor C _sn . The reason why the third switch S ₃ in FIG. 22 and the buffer diode D _sn appear to be different from the boost type power conversion device 200 is that, in addition to the back-up capacitance of the power component and the stray inductance on the line In addition, the snubber capacitor C _sn also participates in resonance.

Referring to FIG. 23, there is a graph of efficiency versus load current for both the high step-up power conversion device 200 and the first embodiment, wherein the conversion efficiency is higher than that of the first embodiment regardless of any load current. The conversion efficiency is 89.55%, and the highest conversion efficiency can reach 92.79%.

Referring to FIG. 24, there is a graph of efficiency versus load current for both the high step-up power conversion device 300 and the first embodiment, wherein the conversion efficiency is higher than the first implementation as the current load increases after the middle load. For example, the minimum efficiency is 90.38% and the highest efficiency is 92.29%.

Comparing FIG. 23 with FIG. 24, it can be found that the efficiency of the high-boost power converter 300 is lower than that of the first embodiment and the high-boost power converter 200, which is high. When the boost type power conversion device 300 demagnetizes the coupled inductor, the inductor current i _Ls flows through the buffer diode D _{sn to} increase the conduction loss. However, when the load is gradually loaded by the medium load, the efficiency of the high-boost type power conversion device 300 is higher than that of the high-boost type power conversion device 200 because the high-boost type power conversion device 300 is boosted higher. The type power conversion device 200 can more effectively reduce the withstand voltage of the third switch S ₃ , so the third switch S _{3 of} the selected switch has a lower on-resistance R _{on the} higher step-up type power conversion device 200, so that the load current is gradually reduced. When the medium load is increased to the full load, the conduction loss of the switch S ₃ of the high step-up type power conversion device 300 is much lower than that of the high step-up power conversion device 200.

Therefore, as a result of the above-described implementation, it is understood that the high-boost power converter 300 can reduce the surge voltage of the third switch S ₃ more effectively than the high-boost power converter 200, and after the load current reaches the middle load The efficiency of the step-up power conversion device 200 is also higher. In contrast, the high-boost power conversion device 200 has a higher efficiency when the load current is between light load and medium load. 300 is preferable, and in terms of circuit architecture, the number of components of the high-boost type power conversion device 200 is higher than that of the step-up power conversion device 300, so that the higher-voltage-type power conversion device 300 has a low circuit cost. And the advantage of good reliability.

In summary, the high-boost power conversion device 100 of the present invention mainly combines the charge pump 11 and the coupled inductor 120 to obtain a high voltage conversion ratio, and proposes passive clamping for the leakage inductance problem. The high-boost type power conversion device 200 and the passively buffer-damped high-boost type power conversion device 300, therefore, the voltage conversion ratio of the present invention can be greatly improved compared to the prior art, and the circuit is simple and easy to control. The characteristics of the present invention are indeed achieved.

The above is only the preferred embodiment of the present invention, and the scope of the invention is not limited thereto, that is, the simple equivalent changes and modifications made by the scope of the invention and the description of the invention are All remain within the scope of the invention patent.

100‧‧‧High-boost power conversion device

11‧‧‧Charge pump

12‧‧‧Transition circuit

120‧‧‧coupled inductor

13‧‧‧Output circuit

111‧‧‧First end of the first switch

112‧‧‧second end of the first switch

121‧‧‧ the first end of the second switch

122‧‧‧second end of the second switch

131‧‧‧The first end of the pump diode

132‧‧‧Second end of the pump diode

141‧‧‧The first end of the pump capacitor

142‧‧‧The second end of the pump capacitor

151‧‧‧First end of the primary side winding

152‧‧‧second end of the primary side winding

161‧‧‧ the first end of the secondary winding

162‧‧‧second end of the secondary winding

171‧‧‧ the first end of the third switch

172‧‧‧second end of the third switch

181‧‧‧ The first end of the output diode

182‧‧‧ second end of the output diode

C _b ‧‧‧ pump capacitor

C _o ‧‧‧output capacitor

C _sn ‧‧‧ snubber capacitor

D ₁ , D ₂ , D ₃ ‧‧‧ back contact diode

D _b ‧‧‧ pumping diode

D _o ‧‧‧ output diode

D _c1 ‧‧‧Clamping diode

D _sn ‧‧‧ Buffer diode

L _p ‧‧‧ primary winding

L _s ‧‧‧ secondary winding

R _o ‧‧‧ output resistance

S ₁ ‧‧‧first switch

S ₂ ‧‧‧second switch

S ₃ ‧‧‧third switch

v _i ‧‧‧ input voltage

v _o ‧‧‧output voltage

1 is a circuit diagram for explaining a first preferred embodiment of a high-boost power conversion device of the present invention; and FIG. 2 is a circuit diagram for explaining a high-boost power conversion device of the present invention operating in an inductor current continuous conduction mode; The circuit diagram for explaining the operation of the high-boost type power conversion device of the present invention in the discontinuous conduction mode of the inductor current; FIG. 4 is a diagram illustrating the equivalent model of the coupled inductor of the present invention; FIGS. 5(a) and 5(b) are respectively Description displays the current in the continuous conduction mode i _L1 and and display current in conduction mode discontinuous conduction i _L1 and graph i _L2 and i _Lp of i _L2 and i _Lp of the diagram; Figure 6 is a diagram illustrating inductor current continuous conduction of The equivalent current path of state one; FIG. 7 is an equivalent current path illustrating state two when the inductor current is continuously turned on; FIG. 8 is a waveform timing diagram illustrating the inductor current at the boundary condition; FIG. 9 is a diagram illustrating the correlation formula substituting n= 1 shows the boundary curve of the operation mode; FIG. 10 is a current path illustrating the state 3 of the inductor current discontinuous conduction mode; FIGS. 11(a) and 11(b) respectively illustrate a high rise of the passive clamp circuit Profiled power A switching device and a high-boost type power conversion device having a passive snubber circuit; FIG. 12 is a current path diagram illustrating FIG. 11(a); and FIG. 13 is a current path diagram illustrating a state of operation of FIG. 11(b) 14 is a current path diagram illustrating the operational state of FIG. 11(b) as state 2; FIGS. 15 to 18 are waveform diagrams of the second embodiment of the present invention under different loads; FIGS. 19 to 22 are the present invention. FIG. 23 is a graph illustrating the efficiency versus load current of the second embodiment of the present invention and the first embodiment; and FIG. 24 is a third embodiment of the present invention. A graph of the efficiency versus load current for both the embodiment and the first embodiment.

100‧‧‧High-boost power conversion device

11‧‧‧Charge pump

12‧‧‧Transition circuit

120‧‧‧coupled inductor

13‧‧‧Output circuit

111‧‧‧First end of the first switch

112‧‧‧second end of the first switch

121‧‧‧ the first end of the second switch

122‧‧‧second end of the second switch

131‧‧‧The first end of the pump diode

132‧‧‧Second end of the pump diode

141‧‧‧The first end of the pump capacitor

142‧‧‧The second end of the pump capacitor

151‧‧‧First end of the primary side winding

152‧‧‧second end of the primary side winding

161‧‧‧ the first end of the secondary winding

162‧‧‧second end of the secondary winding

171‧‧‧ the first end of the third switch

172‧‧‧second end of the third switch

181‧‧‧ The first end of the output diode

182‧‧‧ second end of the output diode

C _b ‧‧‧ pump capacitor

C _o ‧‧‧output capacitor

D ₁ , D ₂ , D ₃ ‧‧‧ back contact diode

D _b ‧‧‧ pumping diode

D _o ‧‧‧ output diode

L _p ‧‧‧ primary winding

L _s ‧‧‧ secondary winding

R _o ‧‧‧ output resistance

S ₁ ‧‧‧first switch

S ₂ ‧‧‧second switch

S ₃ ‧‧‧third switch

V _i ‧‧‧ input voltage

V _o ‧‧‧output voltage

Claims (4)

A high-boost type power conversion device includes: a charge pump for receiving an input voltage, comprising a first switch having a first end and a second end, and a first end connected in series with the first end a second switch of the first end of the switch, a pump diode connected to the second end of the first switch by an anode, and a pump capacitor having a first end and a second end, the pump capacitor One end is electrically connected to the cathode of the pump diode, and the second end of the pump capacitor is electrically connected between the first end of the first switch and the first end of the second switch; a conversion circuit includes a coupling inductor electrically connected to the charge pump and a third switch, the coupled inductor comprising a primary side winding having a first end and a second end, and a second having a first end and a second end a side winding having a first end coupled to a cathode of the pump diode and a first end of the pump capacitor, the second side winding having a first end and a first side winding The second end is coupled to the first end and the second end, and the first end is electrically Connected between the second end of the primary side winding and the first end of the secondary side winding, and the second end thereof is electrically connected to the second end of the second switch; and an output circuit having an output two An anode and an output capacitor, the anode of the output diode is coupled to the second end of the secondary winding, the output capacitor is connected in parallel with the cathode of the output diode, and the first switch, the second The switch and the third switch are respectively driven by a wave width adjustment control signal to be turned on or off, and the input voltage is boosted and output by the output circuit; wherein, if the high step-up power conversion device operates in a continuous conduction mode In the formula, the duty cycle interval of the wave width adjustment control signal is D and 1-D, respectively, wherein the interval D is that the first switch is turned on and the third switch is turned on, and the interval 1-D is the first The second switch is turned on and the first switch and the third switch are turned off; if the operation is in the discontinuous conduction mode, in addition to the two intervals, there is still a third interval in which the primary winding flows through the primary winding The current is zero, and the second switch is turned on, the first switch is turned off, and the third switch is turned off. The high-boost type power conversion device according to claim 1, wherein the first switch, the second switch, and the third switch are respectively an N-type MOSFET and a back-connection The anode body is configured, and the cathode of the back-connected diode is coupled to the source of the N-type MOS field-effect transistor, and the anode of the back-connected diode is coupled to the drain of the N-type MOS field-effect transistor. The high-boost type power conversion device of claim 1, further comprising a clamp diode, an anode of the clamp diode coupled to the first end of the secondary winding, the clamp The cathode of the diode is coupled to the cathode of the output diode. The high-boost type power conversion device of claim 1, further comprising a buffer diode and a buffer capacitor, the anode of the buffer diode being coupled to the second end of the primary winding, the buffer The cathode of the diode is coupled to the first end of the secondary winding, and the snubber capacitor is connected between the cathode of the buffer diode and the first end of the secondary winding.