I2959〇97pif.d〇c 九、發明說明： 【發明所屬之技術領域】 的片’步及—種對各種鱗的負載進行高頻加熱所用 本發明特別是涉及-種對應於負載的 種"員來進仃加熱所用的感應加熱調理器。 【先前技術】 洛心加熱㉟理$由於不需使用火即具有安全地達成溫 ^工，功I等優點’因此急速地普及成可組裝至訂製式廚 custom htehen)中的感應加熱調理器。在此種情況下， 除了鐵製鋼之_高導磁率或高電阻率_之加熱之外， 亦可對|g製銷之_低導辦或低電阻率的鋼加熱而予以 使用之廚房亦期望有良好的感應加熱調理器。依據此種期 望而製成的感應加熱調理器的―例揭示在以下的專利文獻 1中。 【專利文獻1】特開昭61-16491號公報 【發明内容】 決的問題 圖11顯示一種可對鐵製鍋和鋁製鍋此二者進行加熱 所用之先前之感應加熱調理器的電路之構成。 圖11中’全波整流電路1〇〇的交流輸入端經由雜訊濾 波器(noise filter)i〇i而連接至200伏(v)之單相交流電源 102。全波整流電路1〇〇的直流輸出端之間連接著電抗器 121和正平電谷器所形成的串列電路。整平電容号IQ] 與一種由絕緣閘極雙載子電晶體（IGBT，insulated gate I2959ft7pifd〇c biP〇lar transistor)104和1〇5串列連接而成之換流器 (inVerter)106並列連接著。即，換流器1〇6是以一種由 IGBT1()4和105的串列電路的1臂部份 路所構成。IGBT104和1〇5是一種切換(switching)元件。 感應加熱調理器的負载處的鋼1〇7加熱用的感應加熱 線圈108疋由圈數20圈(turn)之内側線圈1〇8a和圈數6〇 圈之外側線圈108b所構成。共振電容器1〇9是由容量〇 之第1電容态109a和容量C2之第2電容器10%所構成。 • 容量C2設定成較容量C1有更充份大的容量。然而，換流 态106的IGBT105並列連接著内側線圈1〇8a，切換用繼電 态開關（relay switch)110的可動接點固定接點七間的電 路，外側線圈108b，第1電容器109a和第2電容器1〇卯 所構成的串列電路。切換用繼電器開關11〇的固定接點a 連接至電容器109a和電容器i〇9b之共同接點。 控制電路111是8位元之一般微電腦所構成，輸入電 力控制部112具備一負載判定部113和頻率指令信號產生 φ 部114。輸入電壓檢出電路115檢出交流輸入電壓，已檢 出的交流輸入電壓施加至輸入電力控制部112。輸入電流 檢出電路116藉由變流器122而檢出交流輸入電流。此已 檢出的交流輸入電流施加至輸入電力控制部112和負載判 定部113。換流器電流檢出電路117藉由變流器123而檢 出该流過此換流器106中的電流。此已檢出的換流器電流 施加至負載判定部113。 輸入電力控制部112對由交流輸入電壓和交流輸入電 9 Ι2959Θ^7ρμ〇。 流而來的輸入電力進行演算，以控制換流器106的驅動頻 率使成為使用者所設定的設定輸入電力。此頻率已受控制 的驅動信號施加至頻率指令信號產生部114。'頻率指令信 號產生部114產生一種與該驅動頻率相當之頻率指令信號 且施加此信號至-種由類比IC所構成的換流器驅動脈波 產生電路(VCO; voltage c〇_lled GsdliatGr)118。換流器驅 動脈波產生電路118依據該頻率指令信號而產生一種驅 脈波。此已產生的驅動脈波經由驅動器119而施加至 IGBTHM和1〇5以作為閘極控制信號彻和vg2。負 判定部⑴衫流輸人電_量和換 判 :咖之種類。若判定此細是 邰113错由繼電器切換電路12〇使切換 開 的接點&和接點b間成為導通⑽non)。若 113藉由繼電器切換電路⑶使 切換用^電益開關i 10的接點心和接點 從 述4=的電路構成，則崎鋼的=依下 會有過大的m過，因此枝進行高火力^由於換流器 在對鋁製鍋進行加熱時 ϋ…、。於是， 熱時更多，其電感亦4感且應=，圈之圈數較鐵製銷加 動換流器。 “且⑽60仟赫程度的高頻來驅 負載判定部⑴若判定此鋼107是紹·時，則藉由 1295觸 _oc 繼電器切換電路120使切換用繼電器開關11()的接點心接 點b之間成為導通。於是，内側線圈i〇8a，外側線圈i〇8b， 第1電容器109a和第2電容器i〇9b串列地相連接而形成 共振電路。因此’感應加熱線圈1〇8之圈數成為8〇圈（内 側線圈108a的20圈+外側線圈1〇8b的6〇圈），而且，共 振電容器109之容量大約等於第1電容器1〇9a的容量ci。 感應加熱線圈108之圈數是8〇圈時，此容量ci設定成使 共振頻率成為在59仟赫附近之值。 輸入電力控制部112控制該換流器ι〇6之動作頻率而 進行一種輸入一定之控制，以得到使用者所設定的調理火 力。鍋107若是一種鋁製鍋時，則定額輸入是2仟瓦 (2kW) ’此時的換流器1〇6的切換頻率是6〇仟赫。此時， 各部的時序波形成為如圖12中所示者。圖12中，iq是換 流器電流(共振電路中流過的電流）、VQ是換流器ι〇6所輸 出的咼頻電壓（共振電路中所施加的高頻電壓）、VG1是施 加至IGBT104中的閘極信號、VG2是施加至IGBT1〇5中 的閘極信號。 負載判定部113若判定此鍋1〇7是鐵製鍋時，則藉由 繼電器切換電路120使切換用繼電器開關11〇的接點心接 點a之間成為導通。於是，内側線圈1〇8a和第2電容器 109b串列地相連接而形成共振電路。因此，感應加熱線圈 108的圈數成為20圈（内側線圈1〇8a之圈數），而且，共振 電容裔109之容量成為與第2電容器1〇9b的容量C2相 等。感應加熱線圈108之圈數是2〇圈時，此容量〇2設定I2959〇97pif.d〇c IX. Description of the invention: [The technical field of the invention] The invention of the invention relates to a type of step corresponding to a load of a variety of scales. Into the induction heating conditioner used for heating. [Prior Art] Luoxin Heating 35 is an inductive heating conditioner that can be quickly assembled into a custom-made kitchen custom htehen because it does not require the use of fire. . In this case, in addition to the heating of the high-permeability or high-resistivity of iron-steel steel, the kitchen that can be used for heating the steel with low-conductivity or low-resistivity is also expected. There is a good induction heating conditioner. An example of an induction heating conditioner manufactured in accordance with such a expectation is disclosed in the following Patent Document 1. [Patent Document 1] JP-A-61-16491 SUMMARY OF INVENTION Problem Figure 11 shows a circuit configuration of a conventional induction heating conditioner for heating both an iron pot and an aluminum pot. . In Fig. 11, the AC input terminal of the 'full-wave rectifying circuit 1' is connected to a single-phase AC power source 102 of 200 volts (v) via a noise filter i〇i. A series circuit formed by the reactor 121 and the positive-level electric grid is connected between the DC output terminals of the full-wave rectifier circuit 1〇〇. The leveling capacitor number IQ] is connected in parallel with an inverter (inVerter) 106 which is connected by a series of insulated gate I 2 959 ft 7 pifd 〇 c biP lar transistors 104 and 1 〇 5 With. Namely, the inverter 1〇6 is constituted by a one-arm partial path of the tandem circuit of the IGBTs 1 and 105. The IGBTs 104 and 1〇5 are a switching element. The induction heating coil 108 for heating the steel 1〇7 at the load of the induction heating conditioner is composed of an inner coil 1〇8a having a number of turns of 20 turns and an outer coil 108b having a number of turns of 6 turns. The resonant capacitor 1〇9 is composed of a first capacitor state 109a of a capacity 〇 and a second capacitor 10% of a capacity C2. • Capacity C2 is set to have a larger capacity than capacity C1. However, the IGBT 105 of the commutation state 106 is connected in parallel with the inner coil 1〇8a, the circuit of the movable contact of the relay switch 110 is fixed between the seven contacts, the outer coil 108b, the first capacitor 109a and the 2 A series circuit of capacitors 1〇卯. The fixed contact a of the switching relay switch 11A is connected to the common contact of the capacitor 109a and the capacitor i〇9b. The control circuit 111 is constituted by an 8-bit general microcomputer, and the input power control unit 112 includes a load determination unit 113 and a frequency command signal generation φ unit 114. The input voltage detecting circuit 115 detects the AC input voltage, and the detected AC input voltage is applied to the input power control unit 112. The input current detecting circuit 116 detects the AC input current by the converter 122. This detected AC input current is applied to the input power control unit 112 and the load determination unit 113. The inverter current detecting circuit 117 detects the current flowing through the converter 106 by the converter 123. This detected inverter current is applied to the load determining unit 113. The input power control unit 112 is connected to the AC input voltage and the AC input voltage 9 Ι 2959 Θ 7 ρ μ 〇. The incoming input power is calculated to control the drive frequency of the inverter 106 to be the set input power set by the user. The drive signal whose frequency has been controlled is applied to the frequency command signal generating portion 114. The frequency command signal generating unit 114 generates a frequency command signal corresponding to the drive frequency and applies the signal to the inverter-driven pulse wave generating circuit (VCO; voltage c〇_lled GsdliatGr) 118 formed by the analog IC. . The inverter drive arterial wave generating circuit 118 generates a pulse wave based on the frequency command signal. This generated driving pulse wave is applied to the IGBTs HM and 1〇5 via the driver 119 as the gate control signal and vg2. Negative judgment unit (1) shirt streaming power _ quantity and change judgment: the type of coffee. If it is determined that the thinness is 邰113, the relay switching circuit 12 causes the contact between the contact & and the contact b to be turned on (10) non). If the relay switch circuit (3) is used to make the switch and the contact of the switching power switch i 10 from the circuit of the above 4=, then the steel of the steel will have an excessive m, so the branch is fired. ^Because the inverter is heating the aluminum pan.... Therefore, the heat is more, the inductance is also 4 sense and should be =, the number of turns of the circle is higher than the iron pin to drive the inverter. "When the high-frequency drive load determination unit (1) of (10) 60 Hz is determined to be the case, the contact switch relay circuit 1 of 1295 is used to make the contact switch b of the switching relay switch 11 () The inner coil i〇8a and the outer coil i〇8b, the first capacitor 109a and the second capacitor i〇9b are connected in series to form a resonant circuit. Therefore, the loop of the induction heating coil 1〇8 The number is 8 turns (20 turns of the inner coil 108a + 6 turns of the outer coil 1〇8b), and the capacity of the resonant capacitor 109 is approximately equal to the capacity ci of the first capacitor 1〇9a. When it is 8 turns, the capacity ci is set such that the resonance frequency becomes a value around 59 kHz. The input power control unit 112 controls the operating frequency of the inverter 〇6 to perform a certain input control to obtain use. If the pot 107 is an aluminum pot, the quota input is 2 watts (2 kW). At this time, the switching frequency of the inverter 1 〇 6 is 6 。. At this time, each part of the The timing waveform becomes as shown in Fig. 12. In Fig. 12, iq It is the inverter current (the current flowing in the resonance circuit), VQ is the 咼 frequency voltage output by the inverter ι 6 (the high frequency voltage applied in the resonance circuit), and VG1 is the gate signal applied to the IGBT 104. VG2 is a gate signal applied to the IGBTs 1 and 5. When the load determination unit 113 determines that the pot 1〇7 is an iron pot, the relay switching circuit 120 causes the switching relay switch 11 to be connected to the snack joint. Then, the inner coil 1 8a and the second capacitor 109b are connected in series to form a resonant circuit. Therefore, the number of turns of the induction heating coil 108 is 20 (the number of turns of the inner coil 1 〇 8a) Further, the capacity of the resonant capacitor 109 is equal to the capacity C2 of the second capacitor 1〇9b. When the number of turns of the induction heating coil 108 is 2 turns, the capacity 〇2 is set.
12959泳而丨fd〇c 成使共振頻率成為在22仟赫附近之值。 輸入電力控制部112控制該換流器1〇6之動作頻率而 進行種輸入一疋之控制，以使一使用者所設定的火力保 持著。鍋107若是一種鐵製鍋時，則定額輸入是3仟 (3kW)，此時的換流器106的切換頻率是25仟赫。此時， 各部的時序波形成為如圖13中所示者。 、 在上述習知的構成中，感應加熱調理器的負载是鋁制 鍋日守’雖然可藉由6〇仟赫之切換頻率來驅動該換流器 以對紹製鋼進行加熱’但由於成為6〇仟赫的高頻驅動。而 換流器106之切換損失(作為切換元件用之IGBT之切換損 失)較大而不適合。感應加熱調理⑽負載是鐵製鋼時，、^ 然換流器106之切換頻率成為25仟赫之低頻，但由於感應 線圈108之圈數成為20圈而變少，則換流器電流尖峰成^ 安(A)，換流器106的穩態(steady state)損失(igbt穩 悲才貝失）變大而不適合。 * 而且，銘製銷和鐵製鋼時由於須進行該感應加熱 108的圈數切換，則感應加熱線圈1〇8之構造會較複雜。 又，藉由感應加熱線目108的圈數切換，則銘製鋼和 鍋中的加熱位置會改變。特別是在鐵製鍋時，由於只_ 側線圈108a進行加熱，則成為一種對鍋底中心部之=部性 加熱。 又，特開2001-160484號公報中揭示一種應 理器’其可對感應加熱線圈的圈數未切換之銘^鐵^ 鋼進行加熱。然而，此種感應加熱調理器之缺 I2959®>5^pif.d〇c 鐵製鋼加熱時不能使充份之換流器電流流過，因此不能獲 得高火力。 鑑於上述之事情，本發明的目的是提供一種感應加熱 5周理裔’其可對感應加熱線圈的圈數未切換之多種不同種 進行加熱，且可力求使換流器的切換損失和穩態損 失減輕。 斤用的丰段 Λ 的感應加熱調理器具備：直流電源電路；感應12959 swims and 丨fd〇c causes the resonance frequency to be around 22 仟. The input power control unit 112 controls the frequency of operation of the inverters 〇6 to perform a type of input control so that the firepower set by a user is maintained. If the pot 107 is an iron pot, the quota input is 3 仟 (3 kW), and the switching frequency of the inverter 106 at this time is 25 kHz. At this time, the timing waveform of each part becomes as shown in FIG. In the above conventional configuration, the load of the induction heating conditioner is an aluminum pot, although the inverter can be driven by a switching frequency of 6 kHz to heat the steel. 〇仟 HF's high frequency drive. However, the switching loss of the inverter 106 (the switching loss of the IGBT used as the switching element) is large and unsuitable. When the induction heating conditioning (10) load is iron steel, the switching frequency of the inverter 106 becomes a low frequency of 25 kHz, but since the number of turns of the induction coil 108 becomes less than 20 turns, the current peak of the inverter becomes ^ Ann (A), the steady state loss of the inverter 106 (igbt stability) is not suitable. * Moreover, the construction of the induction heating coil 1〇8 is complicated by the fact that the number of turns of the induction heating 108 is switched when the pin and the iron steel are made. Further, by switching the number of turns of the induction heating wire 108, the heating position in the steel and the pot is changed. In particular, in the case of an iron pot, since only the side coil 108a is heated, it is a kind of partial heating of the center portion of the bottom of the pan. Further, Japanese Laid-Open Patent Publication No. 2001-160484 discloses a processor which can heat the steel of the induction heating coil whose number of turns is not switched. However, the lack of such an induction heating conditioner I2959®>5^pif.d〇c iron steel heating does not allow sufficient current to flow through the converter, so high firepower cannot be obtained. In view of the above, an object of the present invention is to provide an induction heating for 5 weeks, which can heat a plurality of different types of the number of turns of the induction heating coil, and can strive to reduce the switching loss and steady-state loss of the inverter. .丰的丰段 Λ Induction heating conditioner with: DC power circuit; induction
It、!！共振電容器所構成之共振電路;換流器，其將 =电源電路而來的直流電壓變換成高頻電壓以供給至 LI*路’以及控制裝置’其控制該換流器之切換頻率。 換成第1頻率及其2倍之第2頻率且予以輸出。 =流H以全橋式電路來構成且具有··第！臂’立由 f ^ 2切換元件以串财式連接 和第4切換元件以串列 及弟二 =叙中性點和第2臂之中性點之間ί接二:ί 路此共振電路之共振電宏哭以六曰-r ^ 上述之控鮮ι可娜的方式構成。 狀綠"J 冓成上可切換第1控制狀態和第2杵制 狀恶，其中第1控制狀態中第1 ：^弟，制 進行開/關(on-off)且與此同步地使第、70可父互地 地進行開/關，第2控制狀態中第件交互 1切換元件的導通期間較第2 元件中第 開關以1切換元件導通期間中使第3切換 13 I29590?pif.doc 此種切換以進行料現象，此㈣裝置對應於 依攄第=電容器的容量切換。 種類而使第i _、於負载的 :第;:。又，換流ΐ 前的Γ梅1㈣㈣），職妹*可為先 此時可使換流器電流的二 換W裔的穩態損失減低。 .可使 的效果 力邊===t調理器可達成的效果是，即使感應 加:=;;=Γ行-種與負載的種類無關的 …、、了使換Μ杰的切換損失和穩態損失減輕。 為讓本發明之±麵其他目的、舰和優點 2下了文特舉較佳實施例，並配合所附圖式，作 【實施方式】 (第1實施例） 以下，將參照圖1至圖5來說明本發明的第i •圖1係第1實施例的電路方塊圖。圖i中，全:：、古 包路1疋由整平電容ϋ 2和直流電源電路3所構成，其交 I2959fl>5^pif.d〇c 流輸入端經由交流電源線4、5和雜訊濾波器6而連接至 200伏:t單相交5電源7。全波整流電路1的正側輸出端經 由電抗器8和高速二極體9而連接至整平電容器2的一 端。全波整流電路1的貞側輸出端連接至整平電容器2的 另一端。NPN電晶體1〇,電抗器8和高速二極體9同時構 成-種作為直流電壓可變裝置用的截流器(ch〇pper)u。此 NPN電晶體ig的集極連接至電抗器8和高速二極體9之 共同接點，射極連接至全波整流電路丨㈣側輸出端。然 後，直流電源線12 , 13連接至整平電容器2的二端。 換流器14 φ單相全波橋式電路所構成。錢電源線 12’ 13之間並列連接著第！ IGBT15和第2 IGBTl6的串列 ，路（第1臂）以及第3 IGBT17和第4 IGBT18的串列電路 (第2臂）clGBT15、16、17和18是切換元件。又，IGBT15、 17和18的集極、射極之間分別連接著空轉(freewheei) 一極體 15a、16a、17a 和 18a。又，第 2 IGBT16 第 4 IGBT18 的集極和射極之間分別連接著緩衝(snubber)電容器19和 20 〇 ^第UGBT15和第2IGBT16的串列電路的中性點以及 第3 IGBT17和第4 IGBT18的串列電路的中性點之間連接 著鍋21加熱用的感應加熱線圈22和第1共振電容器23 所形成之串列電路。第2共振電容器24和切換用繼電器開 ，25所形成的串列電路是與第丨共振電容器23並列連接 著。因此，藉由此2個串列電路而構成共振電路26。此時， 感應加熱線圈22的圈數設定成60圈。第丨共振電容器23 15 I2959057pif.d〇c 的容量C23設定成鋁製鍋加熱時所用的容量，容量C24設 定成鐵製鍋加熱時所用的容量。容量C24設定成較容量 C23有更大的充份容量。 控制電路27是由高速微電腦所構成，例如，其可由 32位元之RISC微電腦或數位信號處理(DSP)微電腦所構 成，且此控制電路27具備··輸入電力控制部28，負載判 定部29，換流器電壓可變部30以及換流器驅動脈波產生 部31。輸入電壓檢出電路32檢出交流電源線4、5之間的 交流輸入電壓。已檢出的交流輸入電壓施加至輸入電力控 制部28。輸入電流檢出電路33經由變流器34而檢出交流 輸入電流。已檢出的交流輸入電流施加至輸入電力控制部 28和負載判定部29。換流器電流檢出電路35經由變流器 36而檢出該換流器14中所流過的電流。已檢出的換流器 電流施加至負載判定部29。 輸入電力控制部28參照該輸入電壓檢出電路32所檢 出的交流輸入電壓以控制該換流器電壓可變部3〇。換流器 電壓可變部30經由截波器控制電路37和驅動器％來控制 遠截々il裔11，使此截流裔11在作用上成為升壓截流器。 截流器控制電路37依據該換流器電壓設定信號來產生基-極(base)信號且經由驅動器38而施加至電晶體1〇之基極。 因此，截流器11使已設定的直流電壓VDC施加至直流電 源線12，13。又，截流裔11在升壓動作的同時亦進行功 率因數(power factor)改善動作，以使輸入電流檢出電路33 所檢出的父流輸入電流的波形被控制成可對輸入電壓檢出 I2959i0^pif.d〇c 電路32所檢出的交流輸入電壓的波形進行追縱。 負載判定部29由輸入電流檢出電路33所檢出的交流 輸入電流和換流器電流檢出電路35所檢出的換流哭電流 來判定絲2!的種類，此判定信號施加至換流器驅動°财: 產生部。而且，負載判定部29中若已判定此鍋21是紹 製鍋時’則藉由繼電器切換電路39使切換用繼電器開關關 閉。負載判定部29中若已判定此銷21是鐵製銷時，則藉 由繼電器切換電路39使切換用繼電器開關25導通。 又，輸入電力控制部28由輸入電壓檢出電路32所檢 出的交流輸入電壓和輸入電流檢出電路33所檢出的交流 輸入電流來算出-輸入電力，且對換流器14的驅動頻率 (切換頻率)進行㈣，使該輸人電力成為制者已設定的 设定輸入電力。紐輸出此—鮮已受控制的驅動信號。 = 脈波產生部31。換流器驅動 L產生。p 31依據由輸入電力控制 已 :頻:信號和負載判定物所判定的結果以產二 加至驅動态4〇。驅動器40輸出以下將詳述之 閘極k號VG1至VG4 〇 其-人’參照圖2至圖5來說明本實施例的作用。 雷政人前’切麵繼電關關25關閉。因此，共振 成串列連疋接的一狀種Λ感應加熱線圈22、第1共振電容器23 應線圈22的圈數6〇圈時該共振頻率可 成為在59仟赫附近。 τ I29590^Pi,doc 在使用者設定此調理火力之後，若操作（即，使電源投 起始按鍵(start button)，則控制電路27之負載判定部29 首先進行鍋21的種類(材質）的判定。即，若電源已投入， 則輸入電力控制部28經由換流器驅動脈波產生部31和驅 動為40以施加閘極信號VG1至VG4至IGBT15至18中， 以65仟赫之切換頻率使換流器14開始受到驅動。然後， 切換頻率慢慢地下降。又，問極錢VG1至VG4的產生 方法將容後再述。換流器14的切換頻率成為63仟赫時， 負載判定部29由交流輸入電流和換流器電流來判定此鍋 2+1的種類。即，換流器14以63仟赫之切換頻率而被驅動 雖然這是以㈣鍋作為負載時的—種辦法，但此鋼21 若是鋁製鍋時，交流輸入電流和換流器電流同時成為較各 =的規定電流還大。又，此鋼21若是鐵製鋼時，交流輸入 電流和換流器電流同時較各別的規定電流明顯地減小或幾 乎不流過電流。鑑於此種差異點，則負載判定部四判定此 鍋21是否為鋁製鍋或鐵製鍋。此判定結果施加至換流器驅 動脈波產生部31。 " 又，若判定此鍋21是鋁製鍋，則接收此判定作號，1 ，用繼電器開關25仍保持著關閉狀態。另一方^換^ 器驅動脈波產生部31參照該負載判定部29的判定結果而· 進行以下的動作。即，換流ϋ驅動脈波產生部31具有分別 與6個構成三相換流器所用之電晶體Trl至加相關聯之 埠(port)”U 相上(Trl)，，、，，u 相下(Tr2)，，、，，v 相上(Tr3)，，、，，^ 相下(Tr4)”、”W相上(Tr5)”以及” w相下(Tr6)”。由各別的 1295905¾^ 埠以輸出圖2(b)至(g)中所示的驅動脈波。如圖2(a)所示， 這些驅動脈波是由波形產生用計數器所產生的上升邊緣和 下降邊緣相專的三角波在與設定成此三角波的尖峰值的 75%和25%的門限(threshold)值TH1和TH2相比較時所得 到者。這些圖2(b)所示的驅動脈波(Trl用），圖2(c)所示的 驅動脈波(Tr2用），圖2(d)所示的驅動脈波(Tr3用）以及圖 2(e)所示的驅動脈波(Tr4用）施加至驅動器4〇。藉此可由驅 動器40輸出閘極信號V(H、VG2、VG3和VG4·且分別施 加至 IGBT15、16、17 和 18。 & 具體而言，閘極信號VG1在一周期中270度的區間處 於咼位準，其餘的9〇度區間處於低位準。閘極信號VG2 具有一種與閘極信號VG1成反相的配列。閘極信號 在閘極信號VG1由90度到18〇度期間處於高位準，其它 期間處於低位準。閘極信號VG4具有一種與閘極信號 成反相的配列。IGBT15、1ό、η和1S分別在問極信號 VG1、VG2、VG3和VG4處於高位準期間導通。 化 圖4係此鍋21是鋁製鍋時被加熱的時序波形圖。 4(a^是換流器電流IQ之波形。圖4(b)是換流器14所輪二 的高頻電壓VQ。圖4(e)至圖4(f)是上述之閘極信號V(^ VG2、VG3 和 VG4 〇 ' 此處，就換流器14的動作來說明。若第i IGBTi5 第4IGBT18導通，則在第UGBT15、感應加熱線圈^ 第1共振電容器23和第4 lGBTi8所形成的路徑 ☆ IQ(+)會流過此感應加熱線圈22，同時會對第丨共捃電= 129590%^ 器23進行充電。其次’雖然第4 IGBT18關閉且第3 IGBT17 導通’但此期間中存在著第4 IGBT18和第3 IGBT17同時 成為關閉時的空滯時間（dead time)，可防止此臂發生短 路。第4 IGBT18關閉時，其空滯時間中此緩衝(snubber) 電容器20充電之後，第1 iGBT15、感應加熱線圈22、第 1共振笔谷器23和空轉一極體17a所形成的路徑中流過一 種延遲電流。 若第3 IGBT17導通，則現在第1共振電容器23、感 應加熱線圈22、空轉二極體15a和第3 IGBT17所形成的 路徑中電流IQ㈠會流過此感應加熱線圈22。其次，雖然 第3IGBT17關閉且第4IGBT18導通，但在此期間中亦會 存在著空滯時間。第3 IGBT17關閉時，其空滯時間中此 緩衝(snubber)電容器20充電之後，第！共振電容器23、 感應加熱線圈22、空轉二極體15a，整平電容器2以及空 轉二極體18a所形成的路徑中流過一種延遲電流。 若第4IGBT18導通，則在第1 iGBT15、感應加熱線 圈22、第1共振電容器23和第4 IGBT18所形成的路徑中 電流IQ(+)會流過此感應加熱線圈22，同時會對第1共振 電容器23進行充電。其次，雖然第1 IGBT15關閉且第2 IGBT16導通，但即使在此期間中亦存在著空滯時間(dead time)。第1 IGBT15關閉時，其空滯時間中此緩衝細祕⑽ 電容器19充電之後，第4IGBT18、感應加熱線圈22、第 1共振電容器23和空轉二極體16a所形成的路徑中流過一 種延遲電流。 20 12959场？pi· 若第2 IGBT16導通，則現在第i共振電容器23、感 應加熱線圈22 '第2 IGBT16和空轉二極體他所形成的 路徑中電流IQ(-)會流過此感應加熱線圈。其次，雖然 第2 IGBT16關閉且第i IGBT15導通，但在此期間中亦會 存在著空滯時間。第2 IGBT16關閉時，其空滯時間中此 緩衝(snubber)電容器19充電之後，共振電容器23、感應 加熱線圈22、空轉二極體15a，整平電容器2以及空轉二 極體18a所形成的路徑中流過一種延遲電流。 _ 藉由以上第2控制狀態的動作重複地進行，則如圖4 所示，換流器14的輸出電壓VQ之頻率成為換流器抖的 切換頻率的2倍。換流器電流IQ亦成為2倍的頻率。本 實施例中，鋁製鍋的2仟瓦加熱時的換流器電流IQ的頻 率是與先前一樣設定在60仟赫，但此時換流器14的切換 頻率成為一半只有30仟赫。 ' 又，藉由截流器11以進行功率因數改善時，直流電壓 VDC藉由截流态11而上升至300伏(V)。在未進行功率因 φ 數改善時，直流電壓VDC是282伏(交流200V之尖峰值)。 又，設定輸入中控制用的輸入一定之控制是與先前一樣地 進行使換流器14的切換頻率(驅動頻率）可改變。例如，控 制電路27中内藏的波形產生用計數器所產生的上升邊^ 和下降邊緣相等的三角形波的尖峰值可改變。即使在此種 情況下，門限(threshold)值TH1、TH2亦設定成尖峰值的 75%、25%。 另外，若判定此鍋21是鐵製鍋，則接收此判定信號， 21 I29590^pif.doc 使該切換用繼電器開關25導 波產生部31參照該負载t另―方面，換流器驅動脈 的動作1，若切換用繼電器開關的25判，果而進行=下 26成為城應加熱線圈22 ¥通，^共振電路 共振電容器24的並列電路所 共振電容11 23、第2 容量成為C23+C24。容量C24麵容旦 =路=電， 於足夠大，則共振電容器容量幾半、2⑶相比較之下由It,!! A resonant circuit composed of a resonant capacitor; an inverter that converts a DC voltage from the power supply circuit into a high-frequency voltage to be supplied to the LI* path' and a control device that controls the switching frequency of the inverter. It is replaced with the first frequency and twice the second frequency and output. = Flow H is constructed with a full bridge circuit and has ·· The arm 'is switched from the f ^ 2 switching element to the serial type and the fourth switching element to the series and the second two = the neutral point and the second arm between the neutral points. Resonance electric macro crying is composed of six 曰-r ^ the above-mentioned control of the new 可可娜. The green state can be switched between the first control state and the second control state, wherein the first control state is the first: ^, the system is on-off and synchronized with this. The first 70 can be turned on/off alternately with each other. In the second control state, the first component of the switching element is turned on during the conduction period of the first component, and the third switch is switched to the third switch 13 I29590?pif. Doc This type of switching is used to perform the material phenomenon. This (4) device corresponds to the capacity switching of the =1 = capacitor. For the type, the i-th, the load: the first::. In addition, before the commutation Γ 1 1 1 (4) (4), the actor* can be the first to reduce the steady-state loss of the converter current. The effect force can be achieved ===t The effect that the conditioner can achieve is that even if the induction is added:=;;=Γ--the type of the load is irrelevant..., and the switching loss and stability of the replacement State loss is reduced. In order to make the preferred embodiments of the present invention, the ship, and the advantages of the present invention, and the accompanying drawings, the first embodiment will be described below with reference to FIG. 5 is a block diagram showing the first embodiment of the present invention. In Fig. i, all::, Gubao Road 1疋 is composed of leveling capacitor ϋ 2 and DC power supply circuit 3, and its I2959fl>5^pif.d〇c stream input terminal is connected via AC power lines 4, 5 and The filter 6 is connected to 200 volts: t single-phase 5 power supply 7. The positive side output terminal of the full-wave rectifying circuit 1 is connected to one end of the leveling capacitor 2 via a reactor 8 and a high speed diode 9. The x-side output of the full-wave rectifying circuit 1 is connected to the other end of the leveling capacitor 2. The NPN transistor 1 〇, the reactor 8 and the high-speed diode 9 are simultaneously formed as a shunt for the DC voltage variable device. The collector of this NPN transistor ig is connected to the common junction of the reactor 8 and the high speed diode 9, and the emitter is connected to the side output of the full wave rectifier circuit 四 (four). Then, the DC power lines 12, 13 are connected to the two ends of the leveling capacitor 2. The inverter 14 is composed of a φ single-phase full-wave bridge circuit. Money power cord 12' 13 is connected in parallel! The series of the IGBT 15 and the second IGBT 16 and the circuit (the first arm) and the serial circuits (the second arm) of the third IGBT 17 and the fourth IGBT 18 (the second arm) clGBT 15, 16, 17, and 18 are switching elements. Further, freewheels 15a, 16a, 17a, and 18a are connected between the collector and the emitter of the IGBTs 15, 17 and 18, respectively. Further, the second IGBT 16 is connected to a neutral point of the snubber capacitor 19 and the tandem circuit of the UGBT 15 and the second IGBT 16 and the third IGBT 17 and the fourth IGBT 18 between the collector and the emitter of the fourth IGBT 18, respectively. A series circuit formed by the induction heating coil 22 for heating the pan 21 and the first resonant capacitor 23 is connected between the neutral points of the tandem circuit. The second resonant capacitor 24 and the switching relay are turned on, and the serial circuit formed by 25 is connected in parallel with the second resonant capacitor 23. Therefore, the resonant circuit 26 is configured by the two serial circuits. At this time, the number of turns of the induction heating coil 22 is set to 60 turns. The capacity C23 of the second resonance capacitor 23 15 I2959057 pif.d〇c is set to the capacity used for heating the aluminum pot, and the capacity C24 is set to the capacity used for heating the iron pot. The capacity C24 is set to have a larger capacity than the capacity C23. The control circuit 27 is composed of a high-speed microcomputer. For example, it can be composed of a 32-bit RISC microcomputer or a digital signal processing (DSP) microcomputer, and the control circuit 27 includes an input power control unit 28 and a load determination unit 29, The inverter voltage variable unit 30 and the inverter drive pulse wave generating unit 31. The input voltage detecting circuit 32 detects the AC input voltage between the AC power lines 4, 5. The detected AC input voltage is applied to the input power control unit 28. The input current detecting circuit 33 detects the AC input current via the converter 34. The detected AC input current is applied to the input power control unit 28 and the load determination unit 29. The inverter current detecting circuit 35 detects the current flowing in the inverter 14 via the converter 36. The detected inverter current is applied to the load determining unit 29. The input power control unit 28 refers to the AC input voltage detected by the input voltage detecting circuit 32 to control the inverter voltage varying unit 3A. The inverter voltage variable unit 30 controls the remote finder 11 via the chopper control circuit 37 and the driver %, so that the cutoff 11 acts as a boost interceptor. The shut-off control circuit 37 generates a base-base signal based on the inverter voltage setting signal and is applied to the base of the transistor 1 via the driver 38. Therefore, the interceptor 11 applies the set DC voltage VDC to the DC power lines 12, 13. Further, the cut-off 11 performs a power factor improvement operation at the same time as the boosting operation, so that the waveform of the parent current input current detected by the input current detecting circuit 33 is controlled to detect the input voltage I2959i0. ^pif.d〇c The waveform of the AC input voltage detected by circuit 32 is tracked. The load determination unit 29 determines the type of the wire 2! by the AC input current detected by the input current detection circuit 33 and the commutation crying current detected by the inverter current detection circuit 35, and this determination signal is applied to the commutation Drives the money: the production department. When the load judging unit 29 has determined that the pan 21 is a pan, the relay switching circuit 39 turns off the switching relay switch. When the load determining unit 29 has determined that the pin 21 is an iron pin, the relay switching circuit 39 turns on the switching relay switch 25. Further, the input power control unit 28 calculates the -input power from the AC input voltage detected by the input voltage detecting circuit 32 and the AC input current detected by the input current detecting circuit 33, and drives the frequency of the inverter 14. (Switching frequency) (4), the input power is set as the set input power that the manufacturer has set. The button outputs this signal that is already controlled. = pulse wave generating unit 31. The inverter drive L is generated. p 31 is based on the result of the input power control: frequency: signal and load judgement, and is added to the drive state 4〇. The driver 40 outputs the gate k numbers VG1 to VG4 which will be described in detail below. The human body is explained with reference to Figs. 2 to 5 with reference to Figs. 2 to 5 . Lei Zhengren’s front cut-off relay closure 25 was closed. Therefore, the resonance frequency of the one-type induction heating coil 22 and the first resonance capacitor 23 which are connected in series with the resonance coil can be around 59 kHz when the number of turns of the coil 22 is 6 turns. τ I29590^Pi, doc After the user sets the conditioning firepower, if the operation is started (that is, the power button is pressed to start the button), the load determining unit 29 of the control circuit 27 first performs the type (material) of the pot 21. That is, if the power source has been input, the input power control unit 28 drives the pulse wave generating portion 31 and the drive 40 via the inverter to apply the gate signals VG1 to VG4 to the IGBTs 15 to 18 at a switching frequency of 65 kHz. The inverter 14 is driven to start. Then, the switching frequency is gradually lowered. Further, the method of generating the extremely expensive money VG1 to VG4 will be described later. When the switching frequency of the inverter 14 is 63 kHz, the load is determined. The portion 29 determines the type of the pot 2+1 from the alternating current input current and the inverter current. That is, the inverter 14 is driven at a switching frequency of 63 kHz, although this is a method in which the (four) pot is used as a load. However, if the steel 21 is an aluminum pot, the AC input current and the converter current become larger than the specified current of each =. Moreover, if the steel 21 is iron steel, the AC input current and the converter current are simultaneously Each specified current is significantly reduced or In view of such a difference, the load determining unit 4 determines whether the pan 21 is an aluminum pan or an iron pan. This determination result is applied to the inverter driving pulse wave generating unit 31. " When it is determined that the pan 21 is an aluminum pan, the determination is received, and the relay switch 25 is kept in the closed state. The other switch driver pulse generating unit 31 refers to the determination result of the load determining unit 29. And the following operation is performed. That is, the commutating enthalpy driving pulse wave generating unit 31 has a port "U" (Trl) associated with each of the six transistors Tr1 to PLUS which are used for the three-phase inverter. , , , , , u phase (Tr2),,,,,v phase (Tr3),,,,,^ phase down (Tr4)", "W phase up (Tr5)", and "w phase down (Tr6) ). The respective 12957053⁄4^ 埠 outputs the drive pulse waves shown in Figures 2(b) to (g). As shown in Figure 2(a), these drive pulse waves are generated by the waveform generation counter. The triangular wave of the rising edge and the falling edge is compared with the threshold values TH1 and TH2 set to 75% and 25% of the peak value of the triangular wave. The driving pulse wave (for Trl) shown in Fig. 2(b), the driving pulse wave (for Tr2) shown in Fig. 2(c), and the driving pulse wave (for Tr3) shown in Fig. 2(d) And the driving pulse wave (for Tr4) shown in Fig. 2(e) is applied to the driver 4A. Thereby, the gate signal V(H, VG2, VG3, and VG4· can be output from the driver 40 and applied to the IGBTs 15, 16, 17 respectively. And 18. In particular, the gate signal VG1 is in the 270 level in the interval of 270 degrees in one cycle, and the remaining 9 degrees interval is in the low level. The gate signal VG2 has an arrangement that is inverted from the gate signal VG1. The gate signal is at a high level during the gate signal VG1 from 90 degrees to 18 degrees, and at a low level during other periods. The gate signal VG4 has an arrangement that is inverted from the gate signal. The IGBTs 15, 1 ό, η, and 1S are turned on while the gate signals VG1, VG2, VG3, and VG4 are at a high level, respectively. Fig. 4 is a timing waveform chart in which the pot 21 is heated when it is an aluminum pot. 4 (a^ is the waveform of the inverter current IQ. Fig. 4(b) is the high frequency voltage VQ of the inverter 2, and Fig. 4(e) to Fig. 4(f) are the above-mentioned gate signal V. (^ VG2, VG3, and VG4 〇 ' Here, the operation of the inverter 14 will be described. When the fourth IGBT 18 of the i-th IGBTi5 is turned on, the UGBT 15 , the induction heating coil ^ the first resonant capacitor 23, and the fourth lGBTi 8 are formed. The path ☆ IQ(+) will flow through the induction heating coil 22, and will charge the second 丨 = 129590% 。 23. Secondly, although the 4th IGBT 18 is turned off and the 3rd IGBT 17 is turned on, but during this period There is a dead time when the fourth IGBT 18 and the third IGBT 17 are simultaneously turned off, and this arm can be prevented from being short-circuited. When the fourth IGBT 18 is turned off, after the snubber capacitor 20 is charged in the dead time, A delay current flows through a path formed by the first iGBT 15, the induction heating coil 22, the first resonance pen holder 23, and the idling pole body 17a. When the third IGBT 17 is turned on, the first resonance capacitor 23 and the induction heating coil 22 are now The current IQ (1) in the path formed by the idling diode 15a and the third IGBT 17 flows through the induction plus The coil 22. Next, although the third IGBT 17 is turned off and the fourth IGBT 18 is turned on, there is also a lag time during this period. When the third IGBT 17 is turned off, after the snubber capacitor 20 is charged in the lag time, the snubber capacitor 20 is charged! A delay current flows through the path formed by the resonant capacitor 23, the induction heating coil 22, the free-wheeling diode 15a, the leveling capacitor 2, and the free-wheeling diode 18a. If the fourth IGBT 18 is turned on, the first iGBT 15 and the induction heating coil 22 are provided. The current IQ(+) in the path formed by the first resonant capacitor 23 and the fourth IGBT 18 flows through the induction heating coil 22, and simultaneously charges the first resonant capacitor 23. Second, the first IGBT 15 is turned off and the second The IGBT 16 is turned on, but there is a dead time even during this period. When the first IGBT 15 is turned off, the buffer is fine in the lag time (10) after the capacitor 19 is charged, the fourth IGBT 18, the induction heating coil 22, and the A delay current flows through the path formed by the resonant capacitor 23 and the free-wheeling diode 16a. 20 12959 Fields pi· If the second IGBT 16 is turned on, now the ith resonant capacitor 23, induction plus The current IQ(-) in the path formed by the coil 22' the second IGBT 16 and the free-wheeling diode flows through the induction heating coil. Second, although the second IGBT 16 is turned off and the i-th IGBT 15 is turned on, it also exists during this period. The lag time. When the second IGBT 16 is turned off, the path formed by the resonant capacitor 23, the induction heating coil 22, the idling diode 15a, the leveling capacitor 2, and the idling diode 18a after the snubber capacitor 19 is charged in the lag time A delay current flows through it. When the operation of the second control state is repeatedly performed, as shown in Fig. 4, the frequency of the output voltage VQ of the inverter 14 becomes twice the switching frequency of the inverter. The inverter current IQ also becomes twice the frequency. In the present embodiment, the frequency of the inverter current IQ at the time of heating 2 watts of the aluminum pan is set to 60 kHz as before, but at this time, the switching frequency of the inverter 14 becomes half 30 kHz. Further, when the power cut is improved by the current interceptor 11, the direct current voltage VDC rises to 300 volts (V) by the cutoff state 11. When the power factor φ is not improved, the DC voltage VDC is 282 volts (the peak value of 200 V AC). Further, the control for setting the input for control in the input is performed in the same manner as before, so that the switching frequency (drive frequency) of the inverter 14 can be changed. For example, the peak value of the triangular wave equal to the rising edge and the falling edge generated by the waveform generating counter built in the control circuit 27 can be changed. Even in this case, the threshold values TH1 and TH2 are set to 75% and 25% of the peak value. Further, when it is determined that the pan 21 is an iron pan, the determination signal is received, 21 I29590^pif.doc, the switching relay switch 25 guided wave generating unit 31 refers to the load t, and the inverter drives the pulse. In the first action, if the switching relay switch is 25, the result is = the lower 26 becomes the city heating coil 22, and the resonant circuit of the resonant circuit resonance capacitor 24 has the resonant capacitor 11 23 and the second capacity is C23 + C24. Capacity C24 face capacity = road = electricity, if it is large enough, the resonance capacitor capacity is a few and a half, compared with 2 (3)
猶感應熱線圈22的圈數即使是6〇 圈時’錢振鮮村成為22仟赫。 疋When the number of turns of the heat coil 22 is still 6 圈, 'Qian Zhen Xian Village becomes 22 仟.疋
哭所產生部31具有分別與構成三相換流 6個電晶體Trl至加有關之谭(_”u相上 Γ 目下㈣、”V 相上CM)’’、，，V 相下(Tr4)，，、，，W L及” W相下(Tr6)，，4各別的埠以輸出圖3(b) 動？波。如圖3⑷所示，這些驅動脈波是 一/ >產生料數續產生的±升邊緣和下降邊緣相等的 二角波在與設^成此三角波的尖峰㈣观的門限The crying generating portion 31 has a tan (_"u phase (4), "V phase CM" '', and V phase (Tr4), respectively, which are related to the three-phase commutation of the six transistors Tr1. ,,,,, WL and "W phase down (Tr6),, 4 each 埠 to output the dynamic wave of Figure 3 (b). As shown in Figure 3 (4), these drive pulse waves are a / > Continued generation of the ± rising edge and the falling edge are equal to the threshold of the peak (four) view of the triangular wave
_S_)值TH1相比較時所得到者。這些圖聊斤示的 驅動脈波(Trl用），圖2(e)所示的驅動脈波(Τι>2用），圖2⑷ 所示的驅動脈波(Tr3用）以及圖2(e)所示的驅動脈波_用） 施加至驅動㈣。藉此可由驅動^ 4G輸出閘極信號則、 VG2、VG3 和 VG4 且分別施加至 IGBT15、16、17 和 18。 具體而言，閘極信號VG1在一周期中270度的區間處 於同位準，其餘的90度區間處於低位準。閘極信號VG2 具有一種與閘極信號VG1成反相的配列。閘極信號VG3 22 I2959ft^pif.d〇c 在閘極信號VGl由90度到18〇度期間處於高位準， 期間處於健準。閘極錢VG4具有—軸閘極信號卿 成反相的配列。IGBT15、16、17和18分別/ 从 極信號VG卜VG2、VG3 * VG4處於高位準期^導口通閑 圖5係銷21是鐵製銷時加熱的時序波形圖。圖 是換流器電流IQ。圖5(b)是換流器21所輸出的高 VQ。圖5⑷至圖5(f)是上述的閘極信號仰、呢2、、 和 VG4 〇_S_) The value obtained when TH1 is compared. These diagrams show the driving pulse wave (for Trl), the driving pulse wave (for ιι 2) shown in Fig. 2(e), the driving pulse wave (for Tr3) shown in Fig. 2(4), and Fig. 2(e). The drive pulse shown is applied to the drive (4). Thereby, the gate signal can be output from the driver 4G, VG2, VG3 and VG4 and applied to the IGBTs 15, 16, 17, and 18, respectively. Specifically, the gate signal VG1 is in the same level in the interval of 270 degrees in one cycle, and the remaining 90 degree interval is in the low level. The gate signal VG2 has an arrangement that is inverted from the gate signal VG1. The gate signal VG3 22 I2959ft^pif.d〇c is at a high level during the gate signal VG1 from 90 degrees to 18 degrees, during which time it is in a healthy position. The gate money VG4 has an alignment of the axis gate signal. IGBTs 15, 16, 17, and 18, respectively, are derived from the pole signal VG, VG2, VG3*, VG4, and are in a high-level period, and the guide pin is free. Fig. 5 is a timing waveform diagram of the heating when the iron pin is used. The figure is the converter current IQ. Fig. 5(b) is the high VQ outputted by the inverter 21. Figure 5 (4) to Figure 5 (f) are the above-mentioned gate signal, 呢, 2, and VG4 〇
其次，就換流器14的動作來說明。若第11(}6丁15和 第4 IGBT18導通（高頻電壓VQ是+VDC)，則在第工 IGBT15、感應加熱線圈22、第2共振電容器24、，切換用 繼電器開關25和第4 IGBT18所形成的路徑中電流IQ(+) 會流過此感應加熱線圈22。又，此處省略第丨共振電容器 23之說明。其次，第1 IGBT15和第4 IGBT18關閉時，雖 然第2 IGBT16和第3 IGBT17導通，但此期間中IGBT15 至18存在著全部成為關閉時的空滯時間，可防止各臂短 路。因此’在此種空滯時間中感應加熱線圈22、第1共振 電容器23、空轉二極體17a，整平電容器2以及空轉二極 體16a所形成的路徑中流過一種延遲電流。 若第2 IGBT16和第3 IGBT17導通（高頻電壓vq是 -VDC) ’則此時在第3 IGBT17、切換用繼電器開關25、第 2共振電容器24、感應加熱線圈22和第2 IGBT16所形成 的路徑中電流IQ㈠會流過此感應加熱線圈22。其次，第2 IGBT16和第3 IGBT17關閉時，雖然第1 IGBT15和第4 23 I2959®?Pif.doc IGBT18導通，但此_中仍存在著空 此種空滯時間中第1共振電容器23、感庳力因此，在 轉二極體15a’整平電容器2以及空轉二:體 的路徑中流過一種延遲電流。 也18a所开y成 藉由重複進行以上的第i控制狀態的 示’換流器14的輸出電壓Vq之頻率成為與換流口哭回關 切換頻率相同，換流器電流IQ亦具有相同，'率-杏 施例中’換流器14的切換頻率設定成25仔赫。、因此，& 頻電壓VQ和換流n電流IQ的鮮亦成為2s仟赫。在鐵 製銷的情況下，雖然感應加熱線圈22的圈數是⑼圈，但 共振電路26(感應加熱線圈22和第2共振電容器％所形 成的串列電路）中由於可施加—種具有名製銷加°熱時的2 倍振幅(VDCx2)的高頻電壓Vq，則感應加熱線圈22中可 流過充份的換流器電流IQ，因此可進行高火力加熱。 又，該鐵製鍋加熱時，在形成3仟瓦(kW)的高火力之 情況下，藉由截流器11可使直流電壓VDC上升至300V 乃至350V。此時的換流器電流IQ如圖5所示成為15A尖 峰值。 因此，依據本實施例，換流器14是以一種全橋式電路 所構成，其具有由第1和第2 IGBT15和16串列連接而成 的第1臂以及由第3和第4 IGBT17和18串列連接而成的 第2臂。此換流器14的第1臂的中性點和第2臂的中性點 之間連接著共振電路14。此共振電路14的共振電容器在 構成上可使容量切換。控制電路27可切換成第1控制狀態 24 I2959Q577pi,d〇c (鐵^製銷加熱)和第2控制狀態，其中第1控制狀態中第1 和第2IGBT15和16交互地導通/關閉且第3和第4IGBT17 和以亦以同步方式交互地導通/關閉；第2控制狀態中第 1和第2 IGBT15和16交互地導通/關閉，使第1 IGBT15 ‘通期間較第2IGBT16的導通期間還長，第1 IGBT15導 通期間=使第3 IGBT17之切換元件導通/關閉且同時亦同 步地使第4IGBT18導通/關閉。 依據第2實施例，鍋21是鐵製鍋時，換流器14以第 1頻率來動作，且鍋21是鋁製鍋時，換流器14以第1頻 率之2倍頻率來動作，則可達成高火力的加熱。又，換流 ，14由=只使頻率切換，則可使切換損失減低。又，換流 器14以第1頻率來動作時(第1控制狀態），由於可使輸出 振幅成為先⑽2倍’則在第2辭動作時麵加熱線圈 的圈數仍與原來的相同’換流器電流的尖锋值因此可變 小。於是’換流器14的穩態損失卿τ的穩態損失)可下 降。 圈數銘製銷時，由於感應加熱線圈的 圈數，必進㈣換，則感應加熱線圈21的構造較簡單，因 此可較廉仏地製成。又，感應加熱線圈21 未變化，義製綱喝製崎加熱 局部加熱的情況。 h化’不會發生 然後，在直流(DC)400伏以下的直 於可進行銘製銷之2仟瓦加熱和鐵製銷=夺： 能以廉價的-伏切換元件_ J力路以 25 I2959097pi,doc 為換流器U。 (第2實施例） =6至圖8是本發明的第2實施例。與 的部份以相_符號來表示，以下針對不同的ς 用r:同容之土圖:中的連接方式是以 ,、、、用之共振包今态41來替換第丨共振 。此共振電容器41的容量C41在感應加 團 。。 數是60圈時設定成使共振頻率成為^千赫、。、的圈 流過:=由：是由非磁性金屬所製成’大的換流器電流 。厂有上子現象而發生橫向偏差。此鋼上浮的浮力 2頻率的平方根成反比。此銷上浮時更可藉 來抑制該換流器電流。 貝丰 、☆此處，負載判定部29若判定此鍋21是銘製銷時 叉此鋁製鍋，該換流器驅動脈波產生部31具有分別與6 個構成—相換流裔所用之電晶體Tri至Tr6相關聯之埠 (P〇rt)’’U 相上(Trl)，，、，，u 相下(Tr2)，，、，，V 相上(IY3)，，、，，v 相下(Tr4)，，、，，W相上(Tr5)”以及，，w相下(Tr6)”。由各別的 埠以輸出圖7(b)至圖7(g)所示的驅動脈波。如圖7(a)所示， 各驅動脈波是由波形產生用計數器所產生的上升邊緣和下 降邊緣相等的三角波在與設定成此三角波的尖峰值的 62.5%和37·5%的門限(threshold)值TH1和TH2相比較時所 得到者。這些圖7(b)所示的驅動脈波(Trl用），圖7(c)所示 的驅動脈波(Tr2用），圖7(d)所示的驅動脈波(Tr3用）以及 26 I2959ft^if.d〇c 圖7⑷所不的驅動脈波(Tr4用）施加至驅動器仙。藉此可 由驅動器40輸出閘極信號VG卜VG2、彻和vg4且分 別施加至IGBT15、16、17和18。 古具體而言，閘極錢V G i在一周期中22 5度的區間處 於南位準其餘的135度區間處於低位準。閘極信號vg2 具有一種與閘極信號VG1成反相的配列。閘極信^ VG3 在閘極信號VG1由45度到18〇度期間處於高位準二其它 期間處於低位準。閘極信號VG4具有一種與問極信號彻 • &反相的配列。腿丁15、^?和18分別在閘極信號 VG1、VG2、VG3和VG4處於高位準期間導通。 圖8係此鍋21是鋁製鍋時被加熱時的時序波形圖。圖 8(a)是換流器電流IQ之波形。圖8(b)是換流器14所輸出 的高頻電壓VQ。圖8(c)至圖8(f)是上述之閘極信號VG1、 VG2、VG3 和 VG4。 。 此處’就換流器14的動作來說明。若第1 igb 丁15和 第4 IGBT18導通，則在第1 IGBT15、感應加熱線圈22、 φ 第1共振電容器41和第4 IGBT18所形成的路徑中電流 IQ(+)會流過此感應加熱線圈22，同時會對第！共振電容 器41進行充電。其次，雖然第4IGBT18關閉且第3 igBT17 導通，但此期間中存在著弟4 IGBT18和第3 IGBT17同時 成為關閉時的空滯時間(dead time)，此空滯時間之後會與 第1實施例一樣流過一種延遲電流。 若第3 IGBT17導通，則第1共振電容器41、感應加 熱線圈22、空轉二極體…和第3 IGBT17所形成的路經 27 I2959iQ5?pifd〇c 中電流iq(-)會流過此感應加熱線圈22。其次，共振電容 器41、空轉二極體i7a、第1 IGBT15和感應加熱線圈22 所形成的路徑中電流IQ(+)會流過此感應加熱線圈22。然 後，第1共振電容器41、感應加熱線圈22、空轉二極體 15a和第3 IGBT17所形成的路徑中電流IQ㈠會流過此感 應加熱線圈22而產生共振。然後，雖然第3 IGBT17關閉 且第4 IGBT18導通，但在此期間中亦會存在著空滯時間。 此空滯時間之後與第1實施例一樣會流過一種延遲電流。 若第4 IGBT18導通，則在第1 IGBT15、感應加熱線 圈22、第1共振電容器41和第4 IGBT18所形成的路徑中 電流IQ(+)會流過此感應加熱線圈22，同時會對第1共振 電容器41進行充電。其次，雖然第1 IGBT15關閉且第2 IGBT16導通，但在此期間中亦存在著空滯時間（dead time)。此空滯時間之後與第1實施例一樣會流過一種延遲 電流。 若第2 IGBT16導通，則現在第1共振電容器4卜感 應加熱線圈22、第2 IGBT16和空轉二極體18a所形成的 路徑中電流IQ㈠會流過此感應加熱線圈22。其次，共振 電容器41、第4 IGBT18、空轉二極體16a和感應加熱線 圈22所形成的路徑中電流IQ(+)會流過此感應加熱線圈 22。然後，第1共振電容器41、感應加熱線圈22、第2 IGBT16和空轉二極體18a所形成的路徑中電流IQ㈠會流 過此感應加熱線圈22而產生共振。然後，雖然第2 IGBT16 關閉且第1 IGBT15導通，但在此期間中亦會存在著空滯 28 I2959iQ3pif.doc 樣會流過一種延遲 時間。此空滯時間之後與第1實施例一 電流。 错由重複進行以上的第i控制狀態的動作， 示’換流器Μ的輸出電壓VQ之頻率成為換流器=斤 換頻率的2倍’換流㈣流IQ之頻率職為城= 倍。本實施例中，㈣朗2仟瓦加熱時的換流器' 4 換頻率雖然設定成22仟赫，但高頻電壓v ☆刀Next, the operation of the inverter 14 will be described. When the eleventh (th) and the fourth IGBT 18 are turned on (the high-frequency voltage VQ is +VDC), the IGBT 15, the induction heating coil 22, the second resonant capacitor 24, the switching relay switch 25, and the fourth IGBT 18 are provided. The current IQ(+) flows through the induction heating coil 22. The description of the second resonance capacitor 23 is omitted here. Second, when the first IGBT 15 and the fourth IGBT 18 are turned off, the second IGBT 16 and the 3 IGBT 17 is turned on, but during this period, all of IGBTs 15 to 18 have a hysteresis time when they are turned off, which prevents each arm from being short-circuited. Therefore, in this dead time, induction heating coil 22, first resonance capacitor 23, and idle two A delay current flows through the path formed by the pole body 17a, the leveling capacitor 2, and the free-wheeling diode 16a. If the second IGBT 16 and the third IGBT 17 are turned on (the high-frequency voltage vq is -VDC), then the third IGBT 17 is at this time. The current IQ (1) in the path formed by the switching relay switch 25, the second resonant capacitor 24, the induction heating coil 22, and the second IGBT 16 flows through the induction heating coil 22. Next, when the second IGBT 16 and the third IGBT 17 are turned off, 1st IGBT15 and 4 23 I 2959®? Pif.doc IGBT18 is turned on, but there is still the first resonance capacitor 23 in the lag time of this _, the susceptance force, therefore, the leveling capacitor 2 and the idling body 2 in the rotating diode 15a' A delay current flows through the path. Also, the frequency of the output voltage Vq of the inverter 14 is repeated by the above-mentioned ith control state, and the frequency is the same as the switching frequency of the switching port. The current IQ of the current is also the same, and the switching frequency of the inverter 14 is set to 25 GHz. Therefore, the frequency of the & frequency voltage VQ and the commutating n current IQ are also 2s 仟In the case of the iron pin, although the number of turns of the induction heating coil 22 is (9) turns, the resonance circuit 26 (the tandem circuit formed by the induction heating coil 22 and the second resonance capacitor %) has When the high-frequency voltage Vq of 2 times amplitude (VDCx2) is added to the heating pin, the sufficient inverter current IQ can flow through the induction heating coil 22, so that high-fire heating can be performed. When the pan is heated, in the case of forming a high firepower of 3 watts (kW), by the interceptor 11 The DC voltage VDC can be raised to 300 V or even 350 V. At this time, the inverter current IQ becomes a 15 A peak value as shown in Fig. 5. Therefore, according to the present embodiment, the inverter 14 is constituted by a full bridge circuit. It has a first arm in which the first and second IGBTs 15 and 16 are connected in series, and a second arm in which the third and fourth IGBTs 17 and 18 are connected in series. The resonance circuit 14 is connected between the neutral point of the first arm of the inverter 14 and the neutral point of the second arm. The resonant capacitor of this resonant circuit 14 is configured to switch the capacity. The control circuit 27 can be switched to the first control state 24 I2959Q577pi, d〇c (iron heating) and the second control state, wherein the first and second IGBTs 15 and 16 are alternately turned on/off and the third in the first control state. The fourth IGBT 17 and the fourth IGBT 17 are alternately turned on/off in a synchronous manner. In the second control state, the first and second IGBTs 15 and 16 are alternately turned on/off, and the first IGBT 15 'on period is longer than the second IGBT 16 during the on period. The first IGBT 15 is turned on = the switching element of the third IGBT 17 is turned on/off, and the fourth IGBT 18 is also turned on/off in synchronization. According to the second embodiment, when the pot 21 is an iron pot, the inverter 14 operates at the first frequency, and when the pot 21 is an aluminum pot, the inverter 14 operates at twice the frequency of the first frequency. High heat power can be achieved. Further, the commutation, 14 by = only switching the frequency, the switching loss can be reduced. Further, when the inverter 14 is operated at the first frequency (first control state), since the output amplitude can be made first (10) twice, the number of turns of the surface heating coil in the second speech operation is the same as the original one. The sharp value of the current of the current is therefore small. Thus, the steady state loss of the steady state loss of the inverter 14 can be lowered. When the number of turns is pinned, the number of turns of the induction heating coil must be changed (four), and the structure of the induction heating coil 21 is simple, so that it can be made cheaper. Further, the induction heating coil 21 is not changed, and the system is heated to locally heat. The h' does not occur, then, under DC (DC) 400 volts, it can be used for the production of 2 watts of heating and iron pin = singer: can be used for cheap - volt switching components _ J force road to 25 I2959097pi, doc is the inverter U. (Second Embodiment) = 6 to 8 is a second embodiment of the present invention. The part of and is represented by the phase_symbol. The following is for different ς. Use the r: the same earthy map: the connection method is to replace the third resonance with the resonance state of the current state 41. The capacity C41 of this resonance capacitor 41 is inductively added. . When the number is 60 laps, the resonance frequency is set to ^ kHz. The circle flows through: = by: is made of non-magnetic metal 'large converter current. The factory has a sub- phenomenon and a lateral deviation occurs. The square root of the buoyancy 2 frequency of this steel is inversely proportional. This pin can be used to suppress the current of the converter when it is floating. In this case, when the load judging unit 29 judges that the pot 21 is a pin, the load determining unit 29 forks the aluminum pot, and the inverter driving pulse wave generating unit 31 has a configuration for each of the six constituents. The transistor Tri to Tr6 is associated with the 埠 (P〇rt) ''U phase (Trl),,,, u phase (Tr2),,,,,V phase (IY3),,,,,v Phase (Tr4), ,,,, W phase (Tr5)", and, w phase down (Tr6)". The drive pulse waves shown in Figs. 7(b) to 7(g) are outputted by the respective turns. As shown in Fig. 7(a), each of the driving pulse waves is a threshold of 62.5% and 37.5% of the peak value set by the rising edge and the falling edge generated by the waveform generating counter. The value obtained when comparing the threshold values TH1 and TH2. The drive pulse wave (for Trl) shown in Fig. 7(b), the drive pulse wave (for Tr2) shown in Fig. 7(c), the drive pulse wave (for Tr3) shown in Fig. 7(d), and 26 I2959ft^if.d〇c The driving pulse wave (for Tr4) that is not shown in Fig. 7(4) is applied to the driver. Thereby, the gate signals VGb, VG2, and vg4 can be outputted from the driver 40 and applied to the IGBTs 15, 16, 17, and 18, respectively. In ancient times, the gate money V G i is at a low level in the interval of 22 5 degrees in one cycle and the remaining 135 degrees in the south. The gate signal vg2 has an arrangement that is inverted from the gate signal VG1. The gate signal VG3 is at a low level during the other period of the gate signal VG1 from 45 degrees to 18 degrees. The gate signal VG4 has an arrangement of <& inversion with the question mark signal. The legs 15, 15, and 18 are turned on during the high level of the gate signals VG1, VG2, VG3, and VG4, respectively. Fig. 8 is a timing waveform chart when the pan 21 is heated in an aluminum pan. Figure 8(a) shows the waveform of the inverter current IQ. Fig. 8(b) is a high frequency voltage VQ output from the inverter 14. 8(c) to 8(f) are the above-described gate signals VG1, VG2, VG3, and VG4. . Here, the operation of the inverter 14 will be described. When the first igb 15 and the fourth IGBT 18 are turned on, the current IQ(+) flows through the induction heating coil in the path formed by the first IGBT 15, the induction heating coil 22, the φ first resonant capacitor 41, and the fourth IGBT 18. 22, at the same time will be the first! The resonant capacitor 41 is charged. Next, although the fourth IGBT 18 is turned off and the third igBT 17 is turned on, during this period, there is a dead time when the IGBT 18 and the third IGBT 17 are simultaneously turned off, and this lag time is the same as in the first embodiment. A delay current flows through. When the third IGBT 17 is turned on, the path formed by the first resonant capacitor 41, the induction heating coil 22, the idling diode, and the third IGBT 17 through the current Iq(-) in the 27 I2959iQ5?pifd〇c flows through the induction heating. Coil 22. Next, a current IQ(+) in the path formed by the resonant capacitor 41, the idling diode i7a, the first IGBT 15, and the induction heating coil 22 flows through the induction heating coil 22. Then, the current IQ (1) in the path formed by the first resonance capacitor 41, the induction heating coil 22, the idling diode 15a, and the third IGBT 17 flows through the induction heating coil 22 to resonate. Then, although the third IGBT 17 is turned off and the fourth IGBT 18 is turned on, there is also a lag time in this period. After this lag time, a delay current flows as in the first embodiment. When the fourth IGBT 18 is turned on, the current IQ(+) flows through the induction heating coil 22 in the path formed by the first IGBT 15, the induction heating coil 22, the first resonant capacitor 41, and the fourth IGBT 18. The resonance capacitor 41 is charged. Next, although the first IGBT 15 is turned off and the second IGBT 16 is turned on, there is also a dead time during this period. After this lag time, a delay current flows as in the first embodiment. When the second IGBT 16 is turned on, the current IQ (1) in the path formed by the first resonant capacitor 4, the second IGBT 16, and the idling diode 18a flows through the induction heating coil 22. Next, a current IQ(+) in the path formed by the resonant capacitor 41, the fourth IGBT 18, the free-wheeling diode 16a, and the induction heating coil 22 flows through the induction heating coil 22. Then, the current IQ (1) in the path formed by the first resonance capacitor 41, the induction heating coil 22, the second IGBT 16, and the idling diode 18a flows through the induction heating coil 22 to resonate. Then, although the second IGBT 16 is turned off and the first IGBT 15 is turned on, there is also a lag in the period. A delay time will flow through the I2959iQ3pif.doc. This lag time is followed by a current of the first embodiment. The error is repeated by performing the above-described operation in the i-th control state, and the frequency of the output voltage VQ of the 'inverter 成为 is twice that of the inverter=the frequency of the switching.' The frequency of the commutation (four) stream IQ is the city = double. In the present embodiment, the inverter 4 when the (4) lang 2 watts is heated is set to 22 kHz, but the high frequency voltage v ☆ knife
率44仟赫，換流器電流IQ則成為4倍頻率88仔為赫。頻 因此，依據第2實施例，換流器14 一方面以。 之切換頻率來驅動’且另一方面大約9(H千赫之高頻電2 為換流器電流IQ而可流過該感應加熱線圈，這樣二， 製鍋的上浮現象受到抑制。 ’使在呂 (第3實施例） 圖9和圖10係本發明的第3實施例。與上述第1电> 例相同的部份以相同的符號來表示，以下針對不同的 來說明。 圖9中，與圖1不同之處是圖9中更設有一種零越過 (zero cross)檢出電路42 ,控制電路27中設有換流器相位 差檢出部43。零越過(zero cross)檢出電路42檢出換流器 電流檢出電路35中已檢出的換流器電流的零越過點。此零 越過檢出信號V1 〇施加至換流器相位差檢出部4 3。此換流 器相位差檢出部43對該零越過檢出信號vl〇和換流器驅 動脈波產生部31而來的閘極信號VG1進行比較，以檢出 其相位差且輸出一種栢位差檢出脈波DIF。此種相位差檢 29 129590^^0° 出脈波DIF施加至換流器驅動脈波產生部31。 圖1〇係換流器相位差檢出的時序波形圖。換流器14 需具有感應性，以使緩衝電容器19，20的短路模式(shunt mode)不會發生。圖10⑻顯示此換流器電流IQ。圖1〇(b)DB 顯示該零越過檢出信號V10，其顯示第i IGBT15中所流 過的電流的零越過點。圖10⑷顯示一種施加至第丨IGBTl5 中的閘極信號VG卜換流ϋ相位差檢出部43檢出零越過 檢出信號νιο的上升時點和閘極信號VG1的上升時點的 相位差，即，高頻電壓VQ和換流器電流IQ的相位差，且 輸出-，相位差檢出脈波DIF。相位差檢出脈波撕施加 至換流器驅動脈波產生部3卜換流器驅動脈波產生部Η 在使緩，電容器19,2G的短路模灯會發生的情況下控制 _流器14的切換頻率，以使相位差檢出脈波dif的脈 波寬度不會在所設定的時間以下。 因此，依據第3實施例，鋼21是鐵製或銘製時，由於 ^可使換流H 14維持著感應性，則換流器“可安全地 中對的第1控制狀態At a rate of 44 kHz, the inverter current IQ becomes 4 times the frequency of 88 Hz. Frequency Therefore, according to the second embodiment, the inverter 14 is used on the one hand. The switching frequency is used to drive 'and on the other hand about 9 (H Hz high frequency power 2 is the inverter current IQ and can flow through the induction heating coil, so that the floating phenomenon of the pot is suppressed. (Embodiment 3) Fig. 9 and Fig. 10 show a third embodiment of the present invention, and the same portions as those in the above-described first embodiment are denoted by the same reference numerals and will be described below with reference to Fig. 9. The difference from FIG. 1 is that a zero cross detection circuit 42 is further provided in FIG. 9, and the inverter phase difference detecting portion 43 is provided in the control circuit 27. Zero cross detection is performed. The circuit 42 detects the zero crossing point of the inverter current detected in the inverter current detecting circuit 35. This zero is applied to the inverter phase difference detecting portion 43 over the detected signal V1. The phase difference detecting unit 43 compares the detection signal v1 零 with the gate signal VG1 from the inverter driving pulse wave generating unit 31 to detect the phase difference and output a berth difference detection. Pulse wave DIF. This phase difference detection 29 129590 ^ ^ 0 ° pulse wave DIF is applied to the inverter driving pulse wave generating portion 31. The timing waveform of the inverter phase difference detection. The inverter 14 needs to be inductive so that the shunt mode of the snubber capacitors 19, 20 does not occur. Figure 10 (8) shows the converter current IQ. 1〇(b)DB shows that the zero crosses the detection signal V10, which shows that the zero of the current flowing in the i-th IGBT 15 crosses the point. Fig. 10(4) shows a gate signal VG applied to the second IGBT 155. The phase difference detecting unit 43 detects a phase difference between the rising point of the detected signal νιο and the rising point of the gate signal VG1, that is, the phase difference between the high-frequency voltage VQ and the inverter current IQ, and the output-phase The pulse wave DIF is detected by the difference. The phase difference detection pulse wave is applied to the inverter driving pulse wave generating unit 3, and the inverter driving pulse wave generating unit Η is made slow, and the short-circuiting lamps of the capacitors 19 and 2G are generated. In this case, the switching frequency of the streamer 14 is controlled so that the pulse width of the phase difference detected pulse wave dif is not less than the set time. Therefore, according to the third embodiment, the steel 21 is made of iron or inscribed. Since the commutation H 14 can maintain the inductivity, the inverter can be safely paired First control state
Si: 2 第狀態中對銘製銷進行加執， 但亦可在弟丨控制狀態中對導磁 … ί銷進行加熱。又，亦可在第2㈣狀態中對鋼 电阻率亦低的鋼製銷進行加熱。此時，換流 =、 控制狀態中可加熱的鐵製鍋和不 鍋:的第1 定義成鐵類畜哉4名、古-1 β 1、尚、”心孝冉為鐵類銷或 負載。換“ 14的第2控制狀態t可加熱的紹 30 I2959©?if.d〇c 製鍋和銅製鍋總稱為鋁類鍋或定義成鋁類負載。 =本㈣已哺佳實關揭露如上，、然其並非用以 限^本發明，任何熟習此技藝者，在不獅本發明 内，當可作些許之更動與潤飾，因此本 乾圍當視後附之申請專利範圍所界定者 保遂 【圖式簡單說明】 ' / 圖1係第1實施例的電路方塊圖。 生圖 圖2係第i實施例中銘製鋼加熱時咖丁驅動波形產 生圖 圖3係第1實施例中鐵製錢加熱時咖T驅動波形產 圖4係第1實施例中鋁製鍋加熱 圖5係第1實施例中鐵製銷加熱時的日::形圖。 圖6係第2實施例的電路方塊圖。、、，形圖。 生圖。 圖8係第2實施例中銘製鋼加熱時 圖9係第3實施例的電路方塊圖。、波形圖 圖1 〇係第3實施例中相位差檢出時序 圖11係先前例子之電路方塊圖。 乂 圖12係先前例子中鋁製鍋加熱時 圖7係第2實施例中__熱時咖驅動波形^ 圖 圖 3 直流電源電路 31 12959*0^ρίί<ΐ00 11 截流器 14 換流器 15 至 18 第1至第4 IGBT(第1至第4開關ϋ 15a 至 18a 空轉二極體 21 锅 19、20 緩衝電容器 22 感應加熱線圈 23 第1共振電容器 24 第2共振電容器 25 切換用繼電器開關 26 共振電路 27 控制電路(控制裝置，高速微電腦) 28 輸入電力控制部 29 負載判定部（負載判定裝置） 30 換流器電壓可變部 31 換流器驅動脈波產生部 32 輸入電壓檢出電路 33 輸入電流檢出電路 35 換流器電流檢出電路 37 截流器控制電路 38 驅動器 39 繼電器切換電路 40 驅動器 41 第1共振電容器 32 12959ft7pif doc 42 零越過檢出電路 43 換流器相位檢出部(相位檢出裝置)Si: 2 In the first state, the inscription pin is added, but the magnetic pin can also be heated in the control state of the sister. Further, in the second (fourth) state, the steel pin having a low steel resistivity may be heated. At this time, the commutation =, the iron pot that can be heated in the control state, and the non-pot: the first definition is 4 iron objects, the ancient-1 β 1, Shang, "Xin Xiaoyu is an iron pin or load. Change the "second control state of 14" t can be heated 30 I2959©?if.d〇c The pot and the copper pot are collectively referred to as aluminum-type pots or as aluminum-type loads. = This (4) has been exposed to the above, but it is not intended to limit the invention. Anyone who is familiar with this skill can make some changes and refinements in the invention of the lion. The following is a description of the circuit diagram of the first embodiment. Fig. 2 is a diagram showing the generation of the kiwi drive waveform when the steel is heated in the i-th embodiment. Fig. 3 is a diagram showing the heating of the iron T-drive waveform in the first embodiment in the first embodiment. FIG. Fig. 5 is a view showing a day when the iron pin is heated in the first embodiment; Figure 6 is a block diagram of the circuit of the second embodiment. ,,, and shape. Health map. Fig. 8 is a block diagram showing the circuit of the third embodiment in the case where the steel is heated in the second embodiment. Waveform diagram Fig. 1 Phase difference detection timing in the third embodiment Fig. 11 is a circuit block diagram of the previous example. Fig. 12 is a heating example of the aluminum pot in the previous example. Fig. 7 is a second embodiment of the present invention. __热日咖驱动 waveform ^ Fig. 3 DC power supply circuit 31 12959*0^ρίί<ΐ00 11 interceptor 14 inverter 15 18 to 1st to 4th IGBT (1st to 4th switch ϋ 15a to 18a idling diode 21 pot 19, 20 snubber capacitor 22 induction heating coil 23 first resonance capacitor 24 second resonance capacitor 25 switching relay switch 26 Resonance circuit 27 control circuit (control device, high-speed microcomputer) 28 input power control unit 29 load determination unit (load determination device) 30 inverter voltage variable unit 31 inverter drive pulse wave generation unit 32 input voltage detection circuit 33 Input current detection circuit 35 Inverter current detection circuit 37 Shutdown control circuit 38 Driver 39 Relay switching circuit 40 Driver 41 First resonance capacitor 32 12959ft7pif doc 42 Zero crossing detection circuit 43 Inverter phase detection unit (phase Detection device)
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