201233032 六、發明說明 【發明所屬之技術領域】 本發明關於變化開關頻率而控制輸出的電源裝置。 【先前技術】 -近年來隨著地球環保意識之高漲，要求電源裝置之高 效率化，而使用能抑制開關損失之共振型轉換器。另外， 作爲電源裝置之高機能化之對策，係取代先前使用之類比 控制方式，而改爲電源裝置之數位控制方式。 共振型轉換器，係藉由開關電路產生矩形波狀電壓， 將該矩形波狀電壓之High(高位準）時間率維持於50%之同 時，變化頻率而調整輸出電力。爲實現共振型轉換器之輸 出電力之微小變化，需要提高矩形波狀電壓之頻率調變分 解能力，需要使矩形波狀電壓之週期微小變化。爲使矩形 波狀電壓之週期微小變化，通常係提升時脈信號之頻率。 專利文獻1揭示不提升時脈信號之頻率而縮小矩形波狀電 壓之週期變化的方法。 [習知技術文獻] [專利文獻] •專利文獻1 :特開2004-1 94484號公報 【發明內容】 (發明所欲解決之課題） . 但是，爲實現共振型轉換器之輸出電力之微小變化， -5- 201233032 而提高時脈信號之頻率的方法，會導致消費電力增加或成 本上升。 專利文獻1揭示的方法，需要頻率倍增器，而導致成 本上升之問題。 本發明目的在於解決該問題，提供可以控制輸出電力 之微小變化之同時，可以減低成本或消費電力的電源裝 置。 (用以解決課題的手段） 爲解決上述問題.，達成本發明之目的而構成如下。 亦即，特徵爲具備：開關電路，於直流端子間被連接 著直流電源，於交流端子間輸出矩形波狀電壓；整流電 路，用於對輸入交流端子間之電流實施整流，而輸出至由 直流負荷及平滑電容器並聯連接而成的直流端子間；變壓 器，具有連接於上述開關電路之交流端子間的1次繞線， 及連接於上述整流電路之交流端子間的2次繞線，對上述 1次繞線及2次繞線實施磁性耦合；共振電容器及共振電 感器，其被串聯連接於上述1次繞線及/或上述2次繞 線；及控制手段，係以使上述矩形波狀電壓之頻率變化的 方式，針對上述開關電路具有之開關元件進行控制；將上 述直流電源之電力供給至上述直流負荷的電源裝置；上述 控制手段，係使上述矩形波狀電壓之Hi gh時間率維持於 基本之特定値之同時，每隔上述控制手段所具備之時脈信 號之1時脈週期，而變化上述矩形波狀電壓之週期。 201233032 【實施方式】 參照圖面詳細說明本發明實施形態。 (第1實施形態） 圖1表示本發明第1實施形態之電源裝置1之構成電 路圖。該電源裝置1係設置於交流電源5與直流負荷6之 間，由交流電源5對直流負荷6供給電力。 電源裝置1，係具備：AC-DC轉換器2，其輸入交流 電源5之交流電力，而輸出直流之連結電壓νίΙΝΚ ; DC-DC轉換器3，其以該連結電壓VLINK作爲電源，實施絕緣 之同時對直流負荷6供給直流電力；及控制手段4，其控 制彼等AC-DC轉換器2與DC-DC轉換器3。 AC-DC轉換器2，係藉由橋式連接之整流二極體D11 〜D14，實施交流電源5之交流電壓之全波整流。全波整 流後之電壓，係輸入至由平滑電感器L1，升壓開關元件 Q10、升壓二極體D10、連結電容器C1構成之升壓截波 (chopper)電路7。連結電容器C1之兩端間之電壓成爲 AC-DC轉換器2之輸出之連結電壓VL1NK。 控制手段4,係對升壓截波電路7之N型MOSFET (Metal-Oxide-Semiconductor-Field-Effect T r an s i s t o r)構成 之升壓開關元件Ql〇實施控制。具備平滑電感器LI及升 壓開關元件Q 1 0之主要目的在於，藉由控制手段4控制 升壓開關元件Q10而使來自交流電源5之輸入電流成爲 201233032 和交流電源5之交流電壓槪略相似之正弦波狀。藉由該控 制來改善交流電源5之交流電壓與輸入電流間之功率因子 (Power Factor)。該控制之詳細說明如後述說明之圖2。 又，控制手段4亦進行如後述說明之D C - D C轉換器 3之控制。 DC-DC $專換器3，係具備：由橋式連接之開關元件 Q1〜Q4構成之開關電路8;共振電容器Crl與共振電感 器Lrl串聯連接的繞線N1;和繞線N2產生磁性耦合之變 壓器T1;橋式連接之二極體D21〜D24;及平滑電容器 C2 〇 基於變壓器T1之洩漏電感量或配線電感量，亦有省 略共振電感器Lrl之情況。 又，將開關元件Q1與開關元件Q2串聯連接而成者 標記爲第1開關腳部，將開關元件Q3與開關元件Q4串 聯連接而成者標記爲第2開關腳部。 構成開關電路8之全橋式連接之開關元件Q1〜Q4係 藉由控制手段4進行開/關控制，而於節點Ndl-Nd2間產 生矩形波狀電壓。將該矩形波狀電壓施加於共振電容器 Cr 1、共振電感器Lr 1、繞線N 1之串聯連接體而使共振電 流流入繞線N 1。在和繞線N 1產生磁性耦合之繞線N2所 感應之電流，係被橋式連接之二極體D21〜D24實施整 流，再經由平滑電容器C2實施平滑，而將直流電力供給 至直流負荷6。該控制及動作之詳細如後述圖3之說明。 二極體D2 1〜D24標記爲全橋式，其之具體構成如 201233032 下。二極體D21之陽極與二極體D22之陰極被連接，二 極體D2 1與二極體D22被串聯連接而構成，形成第1二 極體腳部。另外，二極體D23之陽極與二極體D24之陰 極被連接，二極體D23與二極體D24被串聯連接而構 成’形成第2二極體腳部。 第1二極體腳部與第2二極體腳部被並聯連接，第1 二極體腳部與第2二極體腳部之兩端子間成爲直流端子 間。另外，二極體D21與二極體D22之串聯連接點，和 二極體D23與二極體D24之串聯連接點之間係成爲交流 端子間。第1二極體腳部與第2二極體腳部之兩端子間的 直流端子間，如上述說明，係被連接於平滑電容器C2以 及直流負荷6。 二極體D21與二極體D22之串聯連接點，和二極體 D23與二極體D24之串聯連接點之間的交流端子間，係被 連接於繞線N2。 於開關元件Q 1〜Q 4，分別並聯、並以逆向（逆並聯） 連接二極體D1〜D4。其中，使用MOSFET作爲開關元件 Q1〜Q4時，可以利用MOSFET之寄生二極體作爲逆並聯 連接之二極體D1〜D4。 電源裝置1具備：檢測輸入電壓的電壓感測器21; 檢測連結電壓VLINK的電壓感測器22 ;及檢測輸出電壓的 電壓感測器23。另外，電源裝置1具備：檢測輸入電流 的電流感測器24 ;及檢測輸出電流的電流感測器25。 上述電壓感測器21〜23及電流感測器24〜25係連接 -9- 201233032 於控制手段4，控制手段4係參照電壓感測器2 1〜2 3及 電流感測器24〜25檢測出之各電壓、電流之資訊進行控 制》 如上述說明，藉由控制手段4對升壓開關元件Q 1 0 及開關元件Q 1〜Q4進行數位控制。因此，於圖1表示， 由控制手段4使控制信號線連接於升壓開關元件Q 1 0及 開關元件Q 1〜Q4之個別之閘極端子的模樣。但是此乃關 於控制關連者，實際之控制信號線係被供給轉換爲控制必 要之電壓的控制信號電壓。 例如控制手段4大略於1 2 V之電源動作，連結電容 器C1之兩端被施加大略38 OV之直流電壓。因此，欲設 定開關元件Ql、Q3成爲ON(導通）時，開關元件Ql、Q3 之閘極（Ql)、（Q3)需要施加較380V大12V之電壓。由控 制手段4無法直接供給如此高之電壓，因此介由轉換電路 (未圖示）轉換爲高的信號電壓之後，供給至開關元件 Ql、Q3 之閘極（Ql)、（Q3)。 (AC-DC轉換器之電路動作） 參照圖2說明AC-DC轉換器2之電路動作。於此說 明交流電源5之電壓以交流正負反轉時之單側之丨極性之 情況。交流電源5之電壓反轉，另一側之逆極性之動作僅 極性相反而可以類推，因此省略圖示。 另外，於圖2 ’圖2(a)、（b)表示開關元件Qi〇設爲 〇Nfc狀態（模態a) ’以及開關元件Qi〇設爲〇FF狀態（模態 -10- 201233032 b)之電路動作。於圖2(a)、（b)箭頭虛線表示電流之流向 及路徑。 (模態a) 於如圖2(a)所示模態a ’開關元件Q1 〇設爲on狀 態。交流電源5之電壓介由二極體D11及二極體D14施 加於平滑電感器L1，交流電源5之能量被儲存於平滑電 感器L 1。 (模態b) 於如圖2(b)所示模態b ’開關元件Q10設爲OFF 時，成爲模態b之狀態。於模態b ’儲存於平滑電感器L1 的能量會介由二極體D11及二極體D14及升壓二極體 D10放出至連結電容器C1。 以下重複模態a及模態b。 交流電源5爲商用電源頻率之5 0Hz〜60Hz，開關元件 Q10則於大略20KHz〜ΙΟΟΚΗζ實施開/關。因此，於圖 2，當交流電源5之極性保持圖示狀態未變化之期間，開 關元件Q 1 0係以數百次〜數千次重複進行開/關。 又如上述說明，交流電源5之極性反轉之情況並未被 圖示，極性反轉之情況係介由二極體D 1 2及二極體D 1 3 和上述同樣進行。 (DC-DC轉換器之電路動作） -11 - 201233032 以下參照圖3A〜3D說明DC-DC轉換器3 作。圖3 A〜3 D分別表示「模態A」〜「模態】 動作。 (模態A) 圖3 A表示模態A之狀態。於模態A，開關: Q4爲ON之狀態。於節點Ndl〜Nd2間，節點 加正向之連結電容器C1之電壓VLINK。共振電 與共振電感器Lrl產生之共振電流係由連結電容 向繞線N1。繞線N2感應之電流，係介由二極 D24流入平滑電容器C2及輸出之直流負荷6。| 箭頭之虛線表示電流之流向及路徑。 雖標記爲共振電容器Crl與共振電感器Lrl 振電流，嚴格講應爲包含變壓器T 1之洩漏電感 電感量的共振電路之電流。以下簡單標記爲「共 Crl與共振電感器Lrl之共振」。 另外，一次側具備繞線N1及二側側具備繞 變壓器T1，係再二次側感應出和一次側之交流 線比（N2/N1)大略成比例之交流電壓。 (模態B) 圖3 B表示模態B之狀態。藉由流入之電流 存於共振電容器Crl，共振電容器Crl與共振電 產生之共振電流流入結束後，成爲模態B »於模 之電路動 」之電路 元件Q 1、 Ndl被施 容器Crl 器C1流 體 D21、 令圖3 A， 產生之共 量或激發 振電容器 線N2的 電壓之繞 使電荷儲 感器Lrl 態B，變 -12- 201233032 壓器Τ1之激發電流流入繞線N1。繞線N2之電壓，係較 輸出之平滑電容器C2之電壓低，二極體D21、d24之存 在，因此電流未流入繞線N2。 (模態C) 圖3C表示模態C之狀態。設定開關元件Ql、q4成 爲OFF時，成爲模態C之狀態。於模態A，共振電流流 入終了前設定開關元件Ql、Q4成爲OFF時，可以省略模 態B。於模態C，流入開關元件Q1、Q4的共振電感器 Lrl之電流，係流入二極體D2、D3，流入建結電容器 C1。此時，於節點Nd 1〜Nd2間，使節點Nd2成爲正向而 產生連結電容器C1之電壓VLINK。 之後，移行至模態D之前，設定開關元件Q2、q3成 爲ON。 (模態D) 圖3 D表示模態D之狀態。模態D之狀態乃繞線N i 之電流反轉者。於模態D，設定開關元件Q2、Q3成爲 ON ’因此，於模態C之最終階段，共振電感器Lr 1之能 量全部吐出時’電流、亦即能量由連結電容器C1流向共 振電感器Lrl。 於模態D，於節點Ndl〜Nd2間，使節點Nd2成爲正 向而被施加連結電容器C1之電壓VLinic。共振電容器Crl 與共振電感器Lrl產生之共振電流會由連結電容器ci流 -13- 201233032 向繞線N 1。但是’箭頭之虛線表示之電流之流向及路徑 係和模態A相反。 又，繞線N2感應之電流，係介由二極體D22、D23 流入平滑電容器C2及輸出之直流負荷6。 模態D爲模態A之對稱動作。因此，於上述模態 A，以開關元件Ql、Q4作爲開關元件Q2、Q3，或者以二 極體D21、D24作爲二極體D22、D23，將電流之流向逆 向思考即可。 之後，進行模態B、C之對稱動作，之後，再度回至 模態A。又，對稱動作之圖示及說明則省略。 於上述電源裝置1，係以DC-DC轉換器3作爲共振 型轉換器，變化開關元件Q 1〜Q4之開關頻率，而變化於 節點Ndl〜Nd2間產生之矩形波狀電壓之頻率，據此而控 制輸出。 (矩形波狀電壓之產生方法） 依據圖4(a)〜（c)說明節點Ndl〜Nd2間之矩形波狀電 壓之產生方法。又，圖4(a)〜（c)所不項目爲’（1)控制手 段4之基本時脈信號，（2)開關元件Ql、Q4之個別之閘極 波形之閘極：Ql、Q4，（3)開關元件Q2、Q3之個別之閘 極波形之閘極：Q2 ' Q3，（4)節點Ndl〜Nd2間之矩形波 狀電壓。橫軸爲經過之時間。 <<(a)10時脈週期>> 圖4(a)表示時脈信號、開關元件Q1〜Q4之閘極信 -14- 201233032 號、節點Nd 1〜Nd2間之矩形波狀電壓。針對開關元件 Q1〜Q4，將ON時間設爲4時脈週期，將OFF時間設爲6 時脈週期，在Q1及Q4之ON狀態，與Q2及Q3之ON 狀態之間，設有Q 1〜Q4全爲OFF狀態之1時脈週期之怠 惰時間（dead time)。如此則，矩形波狀電壓之High(高電 位、正、1)時間與Low(低電位、負、0)時間均爲5時脈 週期，High時間率爲50%，週期成爲10時脈週期。 <<(b)9時脈週期>> 圖4(b)表示爲較圖4(a)僅稍微縮小輸出電力，而將矩 形波狀電壓之頻率調高1階時之波形。開關元件Q 1、Q4 之ON時間設爲3時脈週期，開關元件Q2、Q3之OFF時 間設爲5時脈週期，分別縮短。如此則，矩形波狀電壓之 High時間爲4時脈週期，Low時間爲5時脈週期，High 時間率爲44%，週期成爲9時脈週期。 <<(c)8時脈週期> > 圖4(c)表示相較於圖4(b)，將矩形波狀電壓之頻率更 調高1階時之波形。針對開關元件Q1〜Q4，將ON時間 設爲3時脈週期，OFF時間設爲5時脈週期，矩形波狀電 壓之High時間及Low時間同時成爲4時脈週期，High時 間率爲50%，週期成爲8時脈週期。 如上述說明，本實施形態中，由圖4(a)至（b)，將矩 形波狀電壓之High時間縮短1時脈週期，由圖4(b)至 (c)，將矩形波狀電壓之Low時間縮短1時脈週期。如此 則，由圖4 (a)至（c)欲漸漸縮小輸出電力時，可將矩形波 -15- 201233032 狀電壓之週期漸次縮短1時脈週期。 又’此例中，「基本上特定値」爲「基本上50%，但 其僅爲一例，在能達成發明效果之範圍內可適當變更該 値。 共振電容器Crl、共振電感器Lrl及變壓器T1之激 發電感量引起之共振頻率，係較開關元件Q1〜Q4之開關 頻率低。基本上，該共振頻率與開關頻率接近則DC-DC 轉換器3之輸出電力變大。 於如圖4(a)〜（c)所示情況下，隨著矩形波狀電壓之 時脈週期由10時脈週期變爲9時脈週期、再變爲8時脈 週期，頻率則進行逆數漸漸變高。因此，開關頻率遠離共 振頻率，則輸出電力降低。亦即，輸出電力漸漸縮小。 如上述說明，·習知技術，欲使矩形波狀電壓之頻率僅 變化1級時，係每隔1時脈週期同時變化矩形波狀電壓之 High時間與Low時間，因此，矩形波狀電壓之週期以每 隔2時脈週期呈現變化。 相對於此，本發明中，欲使矩形波狀電壓之頻率僅變 化1級時，係每隔1時脈週期交互變化矩形波狀電壓之 High時間與Low時間，因此，可以每隔1時脈週期變化 矩形波狀電壓之週期。如此則，矩形波狀電壓之頻率之調 變分解能力可以提升2倍，可以進行輸出電壓或輸出電流 之更微細控制。 矩形波狀電壓之High時間率爲50% ’亦即矩形波狀 電壓之High時間與Low時間之間隔較好是相等。開關元 -16- 201233032 件Q1、Q4與開關元件Q2、Q3之動作，理想上爲保持對 稱動作而需要使圖4之閘極Ql、Q4與閘極Q2、Q3之波 形成爲對稱。因此，矩形波狀電壓之High時間率較好是 5 0%。 於圖4(b)，矩形波狀電壓之High時間率爲稍微不同 於50%之値，但是該程度以下之差異於實用上大都無問201233032 VI. Description of the Invention [Technical Field] The present invention relates to a power supply device that controls an output by varying a switching frequency. [Prior Art] - In recent years, as the global environmental awareness has increased, power supply devices have been required to be more efficient, and a resonance type converter capable of suppressing switching loss has been used. In addition, as a countermeasure for the high performance of the power supply device, it is replaced by the digital control method of the power supply device instead of the analog control method previously used. In the resonance type converter, a rectangular wave voltage is generated by a switching circuit, and the high (high level) time rate of the rectangular wave voltage is maintained at 50%, and the output power is adjusted by changing the frequency. In order to achieve a small change in the output power of the resonant type converter, it is necessary to increase the frequency modulation and decomposition capability of the rectangular wave voltage, and it is necessary to slightly change the period of the rectangular wave voltage. In order to make a small change in the period of the rectangular wave voltage, the frequency of the clock signal is usually raised. Patent Document 1 discloses a method of reducing the periodic variation of a rectangular wave voltage without increasing the frequency of the clock signal. [PRIOR ART DOCUMENT] [Patent Document 1] JP-A-2004-1 94484 SUMMARY OF THE INVENTION (Problems to be Solved by the Invention) However, in order to realize a small change in the output power of the resonance type converter , -5- 201233032 The method of increasing the frequency of the clock signal leads to an increase in consumption power or an increase in cost. The method disclosed in Patent Document 1 requires a frequency multiplier, which causes a problem of an increase in cost. SUMMARY OF THE INVENTION An object of the present invention is to solve the problem and to provide a power supply device which can control a small change in output power while reducing power consumption or power consumption. (Means for Solving the Problem) In order to solve the above problems, the object of the present invention is achieved as follows. That is, it is characterized in that it has a switching circuit in which a DC power supply is connected between the DC terminals, and a rectangular wave voltage is output between the AC terminals; and a rectifier circuit for rectifying the current between the input AC terminals and outputting to the DC a DC terminal between the load and the smoothing capacitor connected in parallel; the transformer has a primary winding connected between the AC terminals of the switching circuit, and a secondary winding connected between the AC terminals of the rectifier circuit, for the above 1 Magnetic coupling is performed on the secondary winding and the secondary winding; a resonant capacitor and a resonant inductor connected in series to the primary winding and/or the secondary winding; and a control means for making the rectangular wave voltage The frequency change method is controlled by the switching element included in the switching circuit; the power of the DC power source is supplied to the power supply device of the DC load; and the control means maintains the Hi gh time rate of the rectangular wave voltage at Basically, at the same time, the moment is changed every one clock cycle of the clock signal provided by the above control means Wavy cycle voltage. 201233032 [Embodiment] Embodiments of the present invention will be described in detail with reference to the drawings. (First Embodiment) Fig. 1 is a circuit diagram showing a configuration of a power supply device 1 according to a first embodiment of the present invention. The power supply device 1 is provided between the AC power source 5 and the DC load 6, and the AC power source 5 supplies power to the DC load 6. The power supply device 1 includes an AC-DC converter 2 that inputs AC power of the AC power source 5 and outputs a DC connection voltage νίΙΝΚ. The DC-DC converter 3 uses the connection voltage VLINK as a power source to perform insulation. At the same time, DC power is supplied to the DC load 6; and the control means 4 controls the AC-DC converter 2 and the DC-DC converter 3. The AC-DC converter 2 performs full-wave rectification of the AC voltage of the AC power source 5 by the bridge-connected rectifying diodes D11 to D14. The voltage after the full-wave rectification is input to a boost chopper circuit 7 composed of a smoothing inductor L1, a boosting switching element Q10, a step-up diode D10, and a connection capacitor C1. The voltage between the both ends of the connection capacitor C1 becomes the connection voltage VL1NK of the output of the AC-DC converter 2. The control means 4 controls the boosting switching element Q1〇 formed by the N-type MOSFET (Metal-Oxide-Semiconductor-Field-Effect T r an s i s t o r) of the boosting chopper circuit 7. The main purpose of providing the smoothing inductor LI and the boost switching element Q 1 0 is to control the boosting switching element Q10 by the control means 4 so that the input current from the alternating current power source 5 becomes 201233032 and the alternating current voltage of the alternating current power source 5 is slightly similar. Sinusoidal. By this control, the power factor between the AC voltage of the AC power source 5 and the input current is improved. The details of this control will be described in Fig. 2 which will be described later. Further, the control means 4 also controls the D C - D C converter 3 as will be described later. The DC-DC $ multiplexer 3 has a switching circuit 8 composed of bridge-connected switching elements Q1 to Q4, a winding N1 in which the resonant capacitor Crl is connected in series with the resonant inductor L1, and a magnetic coupling with the winding N2. The transformer T1; the bridge-connected diodes D21 to D24; and the smoothing capacitor C2 are based on the leakage inductance of the transformer T1 or the wiring inductance, and the resonance inductor Lrl is omitted. Further, the switching element Q1 and the switching element Q2 are connected in series to be referred to as a first switching leg portion, and the switching element Q3 and the switching element Q4 are connected in series to be labeled as a second switching leg portion. The switching elements Q1 to Q4 constituting the full bridge connection of the switching circuit 8 are subjected to on/off control by the control means 4, and a rectangular wave voltage is generated between the nodes Nd1 - Nd2. The rectangular wave voltage is applied to the series connection of the resonance capacitor Cr 1 , the resonance inductor Lr 1 , and the winding N 1 to cause the resonance current to flow into the winding N 1 . The current induced by the winding N2 magnetically coupled to the winding N1 is rectified by the bridge-connected diodes D21 to D24, and smoothed by the smoothing capacitor C2 to supply DC power to the DC load 6 . The details of this control and operation will be described later with reference to FIG. 3. The diodes D2 1 to D24 are marked as a full bridge type, and the specific configuration thereof is as follows 201233032. The anode of the diode D21 is connected to the cathode of the diode D22, and the diode D2 1 and the diode D22 are connected in series to form a first diode leg. Further, the anode of the diode D23 and the cathode of the diode D24 are connected, and the diode D23 and the diode D24 are connected in series to form a second diode leg. The first diode body and the second diode leg are connected in parallel, and the first terminal of the second diode and the second terminal of the second diode are between the DC terminals. Further, the series connection point of the diode D21 and the diode D22 and the series connection point of the diode D23 and the diode D24 are between the alternating current terminals. The DC terminals between the first diode body and the two terminals of the second diode leg are connected to the smoothing capacitor C2 and the DC load 6 as described above. The series connection point of the diode D21 and the diode D22 and the alternating current terminal between the diode D23 and the diode D24 are connected to the winding N2. The switching elements Q1 to Q4 are connected in parallel and connected in parallel (anti-parallel) to the diodes D1 to D4. When a MOSFET is used as the switching elements Q1 to Q4, the parasitic diode of the MOSFET can be used as the diodes D1 to D4 connected in antiparallel. The power supply device 1 includes a voltage sensor 21 that detects an input voltage, a voltage sensor 22 that detects a connection voltage VLINK, and a voltage sensor 23 that detects an output voltage. Further, the power supply device 1 includes a current sensor 24 that detects an input current, and a current sensor 25 that detects an output current. The voltage sensors 21 to 23 and the current sensors 24 to 25 are connected to the control means 4, and the control means 4 are detected by the reference voltage sensors 2 1 to 2 3 and the current sensors 24 to 25. Controlling the information of each voltage and current is performed. As described above, the boosting switching element Q 1 0 and the switching elements Q 1 to Q4 are digitally controlled by the control means 4. Therefore, as shown in Fig. 1, the control means 4 connects the control signal line to the respective gate terminals of the boosting switching element Q1 0 and the switching elements Q1 to Q4. However, this is for controlling the connected person, and the actual control signal line is supplied with a control signal voltage that is converted to control the necessary voltage. For example, the control means 4 operates at a power supply of approximately 1 2 V, and a DC voltage of approximately 38 OV is applied across the junction capacitor C1. Therefore, when the switching elements Q1 and Q3 are to be turned ON, the gates (Q1) and (Q3) of the switching elements Q1 and Q3 need to apply a voltage of 12 V larger than 380V. Since the control means 4 cannot directly supply such a high voltage, it is converted into a high signal voltage via a conversion circuit (not shown), and then supplied to the gates (Q1) and (Q3) of the switching elements Q1 and Q3. (Circuit Operation of AC-DC Converter) The circuit operation of the AC-DC converter 2 will be described with reference to Fig. 2 . Here, the case where the voltage of the AC power source 5 is reversed on the one side when the AC is positive and negative is reversed. The voltage of the AC power source 5 is reversed, and the reverse polarity operation on the other side can be analogized only by the opposite polarity, and thus the illustration is omitted. 2(a) and (b) show that the switching element Qi 〇 is set to the 〇Nfc state (modal a) ' and the switching element Qi 〇 is set to the 〇FF state (modal -10- 201233032 b) The circuit action. The arrows in Fig. 2(a) and (b) indicate the flow direction and path of the current. (Mode a) The modal a ' switching element Q1 〇 is set to the on state as shown in Fig. 2(a). The voltage of the AC power source 5 is applied to the smoothing inductor L1 via the diode D11 and the diode D14, and the energy of the AC power source 5 is stored in the smoothing sensor L1. (Modal b) When the modal b' switching element Q10 is turned OFF as shown in Fig. 2(b), it is in the state of modal b. The energy stored in the smoothing inductor L1 in the mode b' is discharged to the junction capacitor C1 via the diode D11 and the diode D14 and the step-up diode D10. The modal a and the modal b are repeated below. The AC power source 5 is 50 Hz to 60 Hz of the commercial power source frequency, and the switching element Q10 is turned on/off at approximately 20 kHz to ΙΟΟΚΗζ. Therefore, in Fig. 2, the switching element Q 1 0 is repeatedly turned on/off in hundreds to thousands of times while the polarity of the AC power source 5 remains unchanged. As described above, the case where the polarity of the AC power source 5 is reversed is not shown, and the polarity inversion is performed by the diode D 1 2 and the diode D 1 3 in the same manner as described above. (Circuit Operation of DC-DC Converter) -11 - 201233032 The DC-DC converter 3 will be described below with reference to Figs. 3A to 3D. Fig. 3 A to 3 D respectively show the "modal A" to "modal" operations. (Mode A) Fig. 3 A shows the state of modal A. In modal A, the switch: Q4 is ON. Between Ndl and Nd2, the node is connected to the voltage VLINK of the capacitor C1 in the forward direction. The resonant current generated by the resonant power and the resonant inductor Lrl is connected to the winding N1 by the connecting capacitor. The current induced by the winding N2 is determined by the diode D24. The DC load 6 flows into the smoothing capacitor C2 and the output. | The dotted line of the arrow indicates the direction and path of the current. Although the resonant current of the resonant capacitor Cr1 and the resonant inductor Lrl is marked, it should be strictly included that the leakage inductance of the transformer T 1 is included. The current of the resonant circuit is simply labeled as "resonance of the common Crl and the resonant inductor Lrl". Further, the primary side is provided with a winding N1 and the two sides are provided with a winding transformer T1, and the secondary side induces an AC voltage which is slightly proportional to the primary side AC line ratio (N2/N1). (Modal B) Figure 3B shows the state of Modal B. When the current flowing in is stored in the resonant capacitor Cr1, the resonant capacitor Crl and the resonant current generated by the resonant current flow in, and the circuit elements Q1, Nd which are modal B»modes are applied to the Cr1 C1 fluid. D21, let FIG. 3 A, the voltage of the generated common or excited vibration capacitor line N2 is wound to make the charge storage device L1 state B, and the excitation current of -12-201233032 voltage device 流入1 flows into the winding N1. The voltage of the winding N2 is lower than the voltage of the output smoothing capacitor C2, and the diodes D21 and d24 are present, so that the current does not flow into the winding N2. (Mode C) Figure 3C shows the state of modal C. When the switching elements Q1 and q4 are set to OFF, they are in the state of the modal C. In the mode A, the mode B can be omitted when the switching elements Q1 and Q4 are turned OFF before the resonance current flows. In the modal C, the current flowing into the resonant inductor Lrl of the switching elements Q1, Q4 flows into the diodes D2, D3, and flows into the junction capacitor C1. At this time, between the nodes Nd 1 to Nd2, the node Nd2 is made positive, and the voltage VLINK connecting the capacitor C1 is generated. Thereafter, before switching to the modal D, the switching elements Q2 and q3 are set to be ON. (Modal D) Figure 3 D shows the state of modal D. The state of modal D is the current reversal of winding N i . In the mode D, the switching elements Q2 and Q3 are set to ON. Therefore, when the energy of the resonant inductor Lr1 is completely discharged in the final stage of the mode C, the current, that is, the energy flows from the connection capacitor C1 to the resonant inductor Lrl. In the modal D, the voltage VLinic of the connection capacitor C1 is applied between the nodes Nd1 to Nd2 with the node Nd2 in the forward direction. The resonant current generated by the resonant capacitor Crl and the resonant inductor Lrl is transmitted from the junction capacitor ci -13 - 201233032 to the winding N 1 . However, the flow of the current indicated by the dotted line of the arrow is opposite to the modality A. Further, the current induced by the winding N2 flows into the smoothing capacitor C2 and the output DC load 6 via the diodes D22 and D23. Modal D is the symmetrical action of modal A. Therefore, in the above mode A, the switching elements Q1 and Q4 are used as the switching elements Q2 and Q3, or the diodes D21 and D24 are used as the diodes D22 and D23, and the flow of current can be reversely considered. After that, the symmetry of the modes B and C is performed, and then, the mode A is returned again. Moreover, the illustration and description of the symmetrical operation are omitted. In the power supply device 1, the DC-DC converter 3 is used as a resonance type converter, and the switching frequency of the switching elements Q1 to Q4 is changed, and the frequency of the rectangular wave voltage generated between the nodes Nd1 to Nd2 is changed. And control the output. (Method of Generating Rectangular Wave Voltage) A method of generating a rectangular wave voltage between the nodes Nd1 to Nd2 will be described with reference to Figs. 4(a) to 4(c). 4(a) to (c), the items are: (1) the basic clock signal of the control means 4, and (2) the gates of the individual gate waveforms of the switching elements Q1, Q4: Ql, Q4, (3) The gate of the individual gate waveforms of the switching elements Q2 and Q3: Q2 'Q3, (4) The rectangular wave voltage between the nodes Nd1 to Nd2. The horizontal axis is the elapsed time. <<(a) 10 clock cycle>> Fig. 4(a) shows a rectangular wave shape between the clock signal, the gate of the switching elements Q1 to Q4, and the number of the gates N1 to Nd2. Voltage. For the switching elements Q1 to Q4, the ON time is set to 4 clock cycles, and the OFF time is set to 6 clock cycles. Q1 is set between the ON state of Q1 and Q4 and the ON state of Q2 and Q3. Q4 is all the dead time of the 1 clock cycle of the OFF state. Thus, the High (high potential, positive, 1) time and the Low (low potential, negative, 0) time of the rectangular wave voltage are both 5 clock cycles, the High time rate is 50%, and the period becomes 10 clock cycles. <<> (b) 9-cycle period>> Fig. 4(b) shows a waveform in which the output power is slightly reduced compared to Fig. 4(a), and the frequency of the rectangular waveform voltage is increased by one step. The ON time of the switching elements Q 1 and Q4 is set to 3 clock cycles, and the OFF times of the switching elements Q2 and Q3 are set to 5 clock cycles, which are respectively shortened. Thus, the high time of the rectangular wave voltage is 4 clock cycles, the Low time is 5 clock cycles, the High time rate is 44%, and the period is 9 clock cycles. <<(c) 8 clock cycle>> Fig. 4(c) shows a waveform when the frequency of the rectangular wave voltage is further increased by one step as compared with Fig. 4(b). For the switching elements Q1 to Q4, the ON time is set to 3 clock cycles, the OFF time is set to 5 clock cycles, and the High time and the Low time of the rectangular wave voltage are simultaneously 4 clock cycles, and the High time rate is 50%. The period becomes 8 clock cycles. As described above, in the present embodiment, the high time of the rectangular wave voltage is shortened by one clock period from FIGS. 4(a) to 4(b), and the rectangular wave voltage is obtained from FIGS. 4(b) to (c). The Low time is shortened by 1 clock cycle. In this way, when the output power is gradually reduced by FIG. 4(a) to (c), the period of the rectangular wave -15-201233032 voltage can be gradually shortened by one clock period. Further, in this example, "substantially specific 値" is "substantially 50%, but it is only an example, and the 値 can be appropriately changed within a range in which the effect of the invention can be achieved. Resonance capacitor Cr1, resonance inductor Lrl, and transformer T1 The resonance frequency caused by the excitation inductance is lower than the switching frequency of the switching elements Q1 to Q4. Basically, the resonance frequency is close to the switching frequency, and the output power of the DC-DC converter 3 becomes larger. As shown in Fig. 4 (a) In the case of ~(c), as the clock period of the rectangular wave voltage changes from 10 clock cycles to 9 clock cycles and then to 8 clock cycles, the frequency is gradually increased. When the switching frequency is far from the resonance frequency, the output power is reduced. That is, the output power is gradually reduced. As described above, the conventional technique is to change the frequency of the rectangular wave voltage by only one level, every other clock cycle. At the same time, the High time and the Low time of the rectangular wave voltage are changed, and therefore, the period of the rectangular wave voltage changes every 2 clock cycles. In contrast, in the present invention, the frequency of the rectangular wave voltage is changed only by one. Level The 1 clock period alternately changes the High time and the Low time of the rectangular wave voltage. Therefore, the period of the rectangular wave voltage can be changed every 1 clock cycle. Thus, the modulation of the frequency of the rectangular wave voltage can be improved. 2 times, the output voltage or the output current can be more finely controlled. The high time rate of the rectangular wave voltage is 50% 'that is, the interval between the High time and the Low time of the rectangular wave voltage is preferably equal. - 201233032 The operation of Q1, Q4 and switching elements Q2, Q3, ideally to maintain the symmetrical operation, the waveforms of the gates Q1, Q4 and the gates Q2, Q3 of Fig. 4 need to be symmetrical. Therefore, the rectangular wave voltage The high time rate is preferably 50%. In Fig. 4(b), the High time rate of the rectangular wave voltage is slightly different from 50%, but the difference below this degree is practically unquestioned.
I 題。 如上述說明，於控制手段4被連接輸出電壓之檢測用 電壓感測器23，及輸出電流之檢測用電流感測器25。控 制手段4，以使檢測出之輸出電壓成爲和目標値一致的方 式進行開關頻率之調整，則可以對輸出進行定電壓控制， 以使檢測出之輸出電流成爲和目標値一致的方式進行開關 頻率之調整，則可以對輸出進行定電壓控制。 適當選擇該定電壓控制與定電流控制，則可以對輸出 進行定電流定電壓控制。 如上述說明，本發明第1實施形態之電源裝置1，係 每隔〗時脈週期交互變化矩形波狀電壓之High時間及 Low時間，可以更微細調整開關頻率，因此可以進行精密 的定電壓控制或定電流控制。 於圖1之第1實施形態說明使用AC-DC轉換器2作 爲由交流電源獲得直流電力之手段，但是已經具備直流電 源（DC電源）而可以獲得直流電力時，AC-DC轉換器2並 非本發明之必須之要素。 -17- 201233032 (第2實施形態） 以下表示第2實施形態。 圖5表示採用本發明電源裝置1的電氣自動車110之 電源系統槪要之構成圖。電源裝置1係連接於，連接於交 流電源109的即插式充電連接器108，及二次電池105。 於二次電池 105連接著DC-DC轉換器100，DC-DC 轉換器100則對連接著電裝機器101的補機電池106供給 電力。 另外，於二次電池105連接著DC-DC轉換器102， DC-DC轉換器102則對動力用馬達104之驅動用的變頻 器（inverter)103供給電力。 另外，於二次電池105連接著急速充電連接器107, 急速充電連接器107則連接於急速充電器等外部直流電 源，而對二次電池1 05進行充電。 電源裝置1對直流電力之輸出進行定電流定電壓控制 之同時，使用連接於即插式充電連接器108之交流電源 109之電力對二次電池105進行充電。於第2實施形態， 藉由使用電源裝置1，可以進行電流或電壓之微細控制之 同時，對二次電池105進行充電。 電源裝置1亦適用複合型自動車（hybrid car)或電氣 自動車以外的電動車輛等。 (其他實施形態） 上述說明之共振型DC-DC轉換器3，係設定開關元 -18- 201233032 件成爲4元件之全橋式連接電路，設定二極體成爲4元件 之橋式連接電路，而予以組合者。但是，將2元件之半橋 式電路或中心抽頭電路予以組合之電路方式亦可獲得同樣 效果。 開關元件Q10(包含Q1〜Q4)係說明MOSFET，但亦可 爲 IGBT(Insulated Gate Bipolar Transistor)。 開關元件Q1〜Q4非由MOSFET構成時，於開關元件 Q1〜Q4分g!j並歹IJ而且逆向（逆並聯）添加二極體D1〜D4。 開關元件Q1 ~ Q4由MOSFET構成時，可以不具備逆並聯 —•極體D1〜D4。 於圖1之由二極體D2 1〜D24構成之整流電路，可以 取代二極體D23、二極體D24，分別替換爲第1分壓電容 器及第2分壓電容器。被取代者亦可爲二極體D21及二 極體D22。 可以取代圖1之共振電容器Crl，將開關元件Q3及 開關元件Q4替換爲共振電容器。此時，於開關元件q3 及Q4’與被替換之共振電容器、共振電感器Lrl之間會 產生共振。又’被取代者亦可爲開關元件Q1及開關元件 Q2。 於圖1 ’係於變壓器T1之一次側電路具備共振電容 器Cr 1及共振電感器Lr丨，但亦可於變壓器τ丨之二次側 電路具備。亦可於變壓器T1之一次側及二次側之雙方電 路具備。 於圖1 ’僅爲變壓器T1之一次側繞線及電路之構 -19- 201233032 成，和二次側相當之構成則可爲磁性耦合者。例如使用本 發明之電源裝置於IH(感應加熱，Induction Heating)系統 時，可以進行被加熱導體之發熱量之微細控制。 於上述說明中，於圖4(b)說明矩形波狀電壓之High 時間率稍微不同於50%之値，該程度以下之差異不會有問 題。因此，High時間設爲5時脈週期、Low時間設爲4 時脈週期，High時間率設爲56%，週期設爲9時脈週期 亦可獲得和上述同樣之效果。 於圖4(a)〜（c)，怠惰時間（開關元件Q1〜Q4全爲 OFF)係以數位式由時脈信號產生，但亦可由以數位式由時 脈信號產生之信號，以類比式產生怠惰時間而可以獲得閘 極信號。 此情況下，閘極信號之上升或下降不同步於時脈信 號’但矩形波狀電壓之High時間與Low時間成爲時脈週 期之整數倍，此點和於圖4(a)〜（c)同樣，可以適用於本 發明特徵之每隔1時脈週期交互變化上述High時間及 Low時間之方法。 另外’爲提升外觀上之頻率調變分解能力，亦可並用 以不同週期交互重複矩形波狀電壓之週期而產生之 Dithering(抖動）。例如交互重複圖4(a)所示週期1〇時脈 週期’及圖4(b)所示週期9時脈週期，則產生外觀上之週 期爲9.5時脈週期之矩形波狀電壓，可獲得圖4(a)與（b)之 中間之輸出。 -20- 201233032 (和比較電路例之對比） 藉由Dithering交互重複圖4(a)之1〇時脈週期，及圖 4(c)之8時脈週期，可以產生外觀上之週期爲9時脈週期 之矩形波狀電壓。 但是，此情況下，於輸出會被重疊9時脈週期之2倍 的18時脈週期之脈動。通常之Dithering之輸出會被重疊 分數調波。 另外，上述本實施形態之圖4(b)之週期9時脈週期之 矩形波狀電壓時，可獲得未重疊分數調波之輸出。 (本實施形態，本發明之補足） 以上，本實施形態之電源裝置，係於DC-DC轉換器 3藉由控制手段4將矩形波狀電壓之Hi gh時間率大致維 持於50%之同時，提升頻率之調變分解能力，而提供可以 對輸出電力進行微細控制的電源裝置。 無須提高時脈頻率，或無須使用倍增器（multiplier )，因此可以提供低成本，能實現低消費電力的電源裝 置。 另外，本實施形態之電源裝置，可以廣泛使用於藉由 矩形波狀電壓流入共振電流，以數位式變化頻率，控制輸 出的系統。例如於上述IH(感應加熱）系統使用本實施形態 之電源裝置’可以進行被加熱導體之發熱量之微細控制。 (發明效果） -21 - 201233032 依據本發明提供之電源裝置，其可以控制輸出電力之 微小變化，而且可以減低成本或消費電力。 【圖式簡單說明】 圖1表示本發明第1實施形態之電源裝置之構成電路 圖。 圖2表示本發明第1實施形態之AC-DC轉換器2之 電路動作說明圖。 圖3A表示本發明第1實施形態之DC-DC轉換器3之 電路動作說明之模態A之狀態圖。 圖3B表示本發明第1實施形態之DC-DC轉換器3之 電路動作說明之模態B之狀態圖。 圖3C表示本發明第1實施形態之DC-DC轉換器3之 電路動作說明之模態C之狀態圖。 圖3D表示本發明第1實施形態之DC-DC轉換器3之 電路動作說明之模態D之狀態圖。 圖4表示本發明第1實施形態之DC-DC轉換器3之 動作波形之矩形波狀電壓之波形圖。 圖5表示採用本發明電源裝置的第2實施形態之電氣 自動車之電源系統槪要之構成圖。 【主要元件符號說明】 1 :電源裝置 2 : AC-DC轉換器 -22- 201233032 3 : DC-DC轉換器 4 :控制手段 5 :交流電源 6 :直流負荷 7 :升壓截波電路 8 :開關電路 2 1、2 2、2 3 :電壓感測器 24、25 :電流感測器 101 :電裝機器 100、102: DC-DC 轉換器 103 :變頻器 1 〇 4 :動力用馬達 1 0 5 :二次電池 1 〇 6 :補機電池 107 :急速充電連接器 108 :即插式充電連接器 1 〇 9 :交流電源 1 10 :電氣自動車 C 1 :連結電容器 C 2 :平滑電容器I. As described above, the control means 4 is connected to the detection voltage sensor 23 for outputting voltage and the current detecting means 25 for outputting current. The control means 4 adjusts the switching frequency so that the detected output voltage becomes the target 値, and the output voltage can be controlled to constant voltage so that the detected output current becomes the same as the target 値. The adjustment can be used to control the output voltage. When the constant voltage control and constant current control are properly selected, constant current and constant voltage control can be performed on the output. As described above, in the power supply device 1 according to the first embodiment of the present invention, the high-time and low-time of the rectangular wave voltage are alternately changed every clock cycle, and the switching frequency can be finely adjusted, so that precise voltage control can be performed. Or constant current control. In the first embodiment of FIG. 1, the AC-DC converter 2 is used as a means for obtaining DC power from an AC power source. However, when a DC power source (DC power source) is provided and DC power can be obtained, the AC-DC converter 2 is not The essential element of the invention. -17-201233032 (Second embodiment) Hereinafter, a second embodiment will be described. Fig. 5 is a view showing the configuration of a power supply system of an electric automatic vehicle 110 using the power supply unit 1 of the present invention. The power supply device 1 is connected to a plug-in charging connector 108 connected to the AC power source 109, and a secondary battery 105. The DC-DC converter 100 is connected to the secondary battery 105, and the DC-DC converter 100 supplies electric power to the backup battery 106 to which the electrical equipment 101 is connected. Further, the DC-DC converter 102 is connected to the secondary battery 105, and the DC-DC converter 102 supplies electric power to an inverter 103 for driving the power motor 104. Further, the secondary battery 105 is connected to the rapid charging connector 107, and the rapid charging connector 107 is connected to an external DC power source such as a rapid charger to charge the secondary battery 105. The power supply device 1 performs constant current constant voltage control on the output of the direct current power, and simultaneously charges the secondary battery 105 using the electric power of the alternating current power source 109 connected to the plug-in charging connector 108. In the second embodiment, by using the power supply device 1, the secondary battery 105 can be charged while performing fine control of current or voltage. The power supply device 1 is also applicable to a hybrid car or an electric vehicle other than an electric car. (Other Embodiments) The resonant DC-DC converter 3 described above is a bridge-connected circuit in which a switching element -18-201233032 is a four-element full-bridge connection circuit and a diode is set as a four-element. To be combined. However, the same effect can be obtained by combining a two-element half bridge circuit or a center tap circuit. The switching element Q10 (including Q1 to Q4) is a MOSFET, but may be an IGBT (Insulated Gate Bipolar Transistor). When the switching elements Q1 to Q4 are not composed of MOSFETs, the switching elements Q1 to Q4 are divided into g!j and 歹IJ, and the diodes D1 to D4 are added in reverse (anti-parallel). When the switching elements Q1 to Q4 are formed of MOSFETs, the anti-parallel-•polar bodies D1 to D4 may not be provided. The rectifier circuit composed of the diodes D2 1 to D24 in Fig. 1 can be substituted for the diode D23 and the diode D24, and replaced with the first voltage dividing capacitor and the second voltage dividing capacitor, respectively. The replaced person may also be a diode D21 and a diode D22. Instead of the resonant capacitor Crl of Fig. 1, the switching element Q3 and the switching element Q4 can be replaced with a resonant capacitor. At this time, resonance occurs between the switching elements q3 and Q4' and the replaced resonant capacitor and resonant inductor Lrl. Further, the replaced person may be the switching element Q1 and the switching element Q2. The primary side circuit of the transformer T1 in Fig. 1 is provided with a resonant capacitor Cr 1 and a resonant inductor Lr 丨, but may be provided in a secondary side circuit of the transformer τ 。. It can also be provided on both the primary side and the secondary side of the transformer T1. Figure 1 ' is only the primary winding of the transformer T1 and the structure of the circuit -19-201233032. The composition corresponding to the secondary side can be a magnetic coupler. For example, when the power supply device of the present invention is used in an IH (Induction Heating) system, fine control of the amount of heat generated by the heating conductor can be performed. In the above description, the high time rate of the rectangular wave voltage is slightly different from 50% in Fig. 4(b), and the difference below this degree is not problematic. Therefore, the High time is set to 5 clock cycles, the Low time is set to 4 clock cycles, the High time rate is set to 56%, and the cycle is set to 9 clock cycles, and the same effects as described above can be obtained. In FIGS. 4(a) to 4(c), the idle time (all of the switching elements Q1 to Q4 are OFF) is digitally generated by a clock signal, but may be analogously generated by a digital signal generated by a clock signal. A gate signal can be obtained by generating a slack time. In this case, the rise or fall of the gate signal is not synchronized with the clock signal 'but the High time and the Low time of the rectangular wave voltage become integer multiples of the clock period. This point is shown in Figures 4(a) to (c). Similarly, a method of alternately changing the High time and the Low time every other clock cycle of the feature of the present invention can be applied. In addition, in order to improve the frequency modulation decomposition capability of the appearance, it is also possible to use Dithering which is generated by periodically repeating the period of the rectangular wave voltage in different periods. For example, by repeating the period 1 〇 clock period ' shown in FIG. 4( a ) and the period 9 clock period shown in FIG. 4( b ), a rectangular wavy voltage having an appearance period of 9.5 clock cycles is obtained. The output in the middle of Figures 4(a) and (b). -20- 201233032 (Compared with the comparison circuit example) By repeating the 1〇 clock period of Figure 4(a) and the 8th clock period of Figure 4(c) by Dithering interaction, the appearance period is 9 Rectangular wavy voltage of the pulse period. However, in this case, the output is superimposed on the pulsation of the 18-cycle period which is twice the 9-cycle period. Usually the output of Dithering will be overlapped by fractional wave modulation. Further, in the case of the rectangular wavy voltage of the period of the period 9 of the period of Fig. 4 (b) of the present embodiment, the output of the non-overlapping fractional modulation can be obtained. (Embodiment of the present invention) In the power supply device of the present embodiment, the DC-DC converter 3 maintains the Hi Gh time rate of the rectangular wave voltage at approximately 50% by the control means 4. It improves the frequency modulation and decomposition capability, and provides a power supply device that can finely control the output power. There is no need to increase the clock frequency or use a multiplier, so it can provide a low-cost power supply that can achieve low power consumption. Further, the power supply device of the present embodiment can be widely used in a system in which a rectangular wave voltage flows into a resonance current and the frequency is changed in a digital manner to control the output. For example, in the above-described IH (induction heating) system, the power supply device of the present embodiment can be used to finely control the amount of heat generated by the heating conductor. (Effect of the Invention) - 21 - 201233032 A power supply device according to the present invention can control a small change in output power, and can reduce cost or consume power. BRIEF DESCRIPTION OF THE DRAWINGS Fig. 1 is a circuit diagram showing the configuration of a power supply device according to a first embodiment of the present invention. Fig. 2 is a view showing the circuit operation of the AC-DC converter 2 according to the first embodiment of the present invention. Fig. 3A is a view showing a state of mode A of the description of the circuit operation of the DC-DC converter 3 according to the first embodiment of the present invention. Fig. 3B is a state diagram showing a mode B of the circuit operation of the DC-DC converter 3 according to the first embodiment of the present invention. Fig. 3C is a view showing a state of mode C of the description of the circuit operation of the DC-DC converter 3 according to the first embodiment of the present invention. Fig. 3D is a view showing a state of mode D of the description of the circuit operation of the DC-DC converter 3 according to the first embodiment of the present invention. Fig. 4 is a waveform diagram showing a rectangular wave voltage of an operation waveform of the DC-DC converter 3 according to the first embodiment of the present invention. Fig. 5 is a view showing the configuration of a power supply system of an electric vehicle according to a second embodiment of the power supply device of the present invention. [Main component symbol description] 1 : Power supply unit 2 : AC-DC converter-22- 201233032 3 : DC-DC converter 4 : Control means 5 : AC power supply 6 : DC load 7 : Boost chopper circuit 8 : Switch Circuit 2 1 , 2 2 , 2 3 : Voltage sensor 24 , 25 : Current sensor 101 : Electrical equipment 100 , 102 : DC-DC converter 103 : Inverter 1 〇 4 : Power motor 1 0 5 : Secondary battery 1 〇6 : Backup battery 107 : Rapid charging connector 108 : Plug-in charging connector 1 〇 9 : AC power supply 1 10 : Electric automatic car C 1 : Connecting capacitor C 2 : Smoothing capacitor
Crl :共振電容器 D1〜D4:二極體（逆並聯二極體） D10 :升壓二極體 D11〜D14:二極體（整流二極體） -23- 201233032 D21、D22:二極體（第1二極體腳部） D23、D24:二極體（第2二極體腳部） L1 :平滑電感器 Lrl :共振電感器 Nl 、 N2 :繞線 N d 1、N d 2 :節點 Q1、Q2 :開關元件（第1開關腳部），閘極，閘極波形 Q3、Q4 :開關元件（第2開關腳部），閘極，閘極波形 Q 1 〇 :開關元件、升壓開關元件 T1 :變壓器 VLINK :連結電壓（電壓）Crl: Resonant capacitors D1 to D4: Diode (anti-parallel diode) D10: Boost diode D11 to D14: Diode (rectifier diode) -23- 201233032 D21, D22: Diode ( 1st diode foot) D23, D24: diode (2nd diode leg) L1: smoothing inductor Lrl: resonant inductor Nl, N2: winding N d 1 , N d 2 : node Q1 , Q2 : Switching element (1st switch leg), gate, gate waveform Q3, Q4 : Switching element (2nd switch leg), gate, gate waveform Q 1 〇: switching element, boost switching element T1: Transformer VLINK: connection voltage (voltage)
Vo :輸出電壓 -24-Vo : output voltage -24-