200915744 九、發明說明： 【發明所屬之技術領域】 本發明是有關於一種發射機及接收機及其調整方法， 特別是指一種降低本地振盪洩漏及同相/正交相不匹配之發 射機及接收機及其調整方法。 【先前技術】 請參閱圖1 ’圖1係繪示一種習知的直接升頻式（ direct up-conversion)發射機，其包含二數位至類比轉換器 11、12、二低通濾波器13、14、二混頻器15、16、一加總 器17、一功率放大器18及一天線19。其中，—數位基頻 信號BBIt依序進行數位至類比轉換、低通濾波及與一同相 (in-phase)本地振盪信號L〇It混頻，以產生一類比同相射 頻信號RFIt，而另一數位基頻信號BBQt則依序進行數位至 類比轉換、低通濾波及與一正交相（quadrature-phase )本 地振盈彳s號L0Qt混頻，以產生一類比正交相射頻信號尺卩仏 。此二射頻信號RFIt、RFQt將進行加總及功率放大，以發 射到外界。 所述之本地振盪信號L0It& 1^0(^的理想相位差是9〇 度’但實際上會存在一相位偏移et ’且同相路徑上的方塊（ 包括數位至類比轉換器11和低通濾波器13)及正交相路徑 上的方塊（包括數位至類比轉換器12和低通濾波器14)也 會存有增益偏移（在圖1中以一振幅偏移叫來表示），此種 現象稱為同相/正交相不匹配（I/Q mismatch )或同相/正交 相不平衡（I/Q imbalance)。此外，此二本地振盪信號L〇It 200915744 及LOQt有可能會分別從相對應之二混頻器15、16洩漏到 二射頻信號RFIt、RFQt，此種現象稱為本地振盪洩漏（ local oscil丨ation leakage)或本地振逢饋通（local oscillation feedthrough )。所述之同相/正交相不匹配及本地振盪洩漏將 會降低此發射機所發射信號的信噪比，且很可能導致資料 漏失。 請參閱圖2，此為美國專利第6970689號所揭露之一種 用於降低本地振盪洩漏的發射機。此發射機包含一混頻器 21、一功率放大器22、一信號強度測量電路23及一控制信 號產生電路24。混頻器21具有多個操作狀態，且所述之操 作狀態分別對應到不同的本地振盪洩漏程度。信號強度測 量電路23是用來測量功率放大器22的輸出信號中本地振 盪洩漏成分的強度，其包括一整流器（Rectifier )(圖未示 )及一比較為（圖未示）。控制信號產生電路24則輸出一 控制信號來改變混頻器21的操作狀態。 在校正混頻器21的期間，功率放大器22將會提高其 增益，而控制信號產生電路24會改變混頻器21的操作狀 態，並儲存操作狀態的資訊及信號強度測量電路Μ測量到 的強度’並重複上述步驟直到混頻器21的所有操作狀態都 被使用過’然後再將混頻器21設定在使本地振衫漏程度 最小的操作狀態。 或者是在权正混頻器21的坤間，说、玄κ丄 J期間功率放大器22將會 提高其増益’而控制信號產生雷改24姓 王土电路24待續改變混頻器21 的操作狀態，直到信號強度測量雷故？ 又』里电路23測量到的強度小於 200915744 一預設的臨界值’然後再固定混頻器21的操作狀態。 請參閱圖 3，圖 3 係 ISSCC 2006 / SESSION 20 / WLAN/WPAN / 20.4 “A Highly Linear Direct-Conversion Transmit Mixer Trans conductance Stage with Local Oscillation Feedthrough and I/Q imbalance Cancellation Scheme”所揭露的一種用於降低本地振蘯泡漏及同相/正交相 不匹配的發射機。此發射機包含二數位至類比轉換器301、 302、二低通濾波器303、304、二互導級305、306、二混 頻器307、308、一加總器309、一功率放大器310、一天線 311、一包跡檢測器（envelope detector ) 312及一可變增益 放大器313。二數位的基頻信號BBIt、BBQt分別被轉換成 二類比的射頻信號RFIt、RFQt，再被加總及功率放大以發 射到外界。 包跡檢測器312及可變增益放大器313依序對功率放 大器3 10的輸出信號進行包跡檢測及放大，以產生一基頻 漣波。當二基頻信號BBIt、BBQt為弦波信號且頻率為Fbb 時，基頻漣波的頻譜成分則出現在Fbb處（由於本地振盪洩 漏）及2xFBB處（由於同相/正交相不匹配），且其頻譜分析 可以顯示出本地振盪洩漏的程度及同相/正交相不匹配的程 度。 本地振盪洩漏可被分為兩種：基頻本地振盪洩漏及射 頻本地振盪洩漏。基頻本地振盪洩漏導因於二數位至類比 轉換器301、302、二低通濾波器303、304及二互導級305 、306中的元件偏移（device offset)，而射頻本地振盪洩漏 200915744 導因於寄生電容或互感的直接耦合。其中，這兩種本地振 盪洩漏需要各自被降低。 然而所述isscc論文並沒有說明如何調整二互導級 305、306及二基頻#號BBIt、BBQt的相位及振幅來降低發 射機的本地振盈戌漏和同相/正交相不匹配，也沒有提及如 何降低接收機的本地振盪洩漏和同相/正交相不匹配的問題 〇 【發明内容】 因此，本發明之目的即在提供一種降低發射機或接收 機的本地振盈泡漏的調整方法及一種降低發射機或接收機 的同相/正交相不匹配的調整方法。 於是，本發明降低發射機或接收機的本地振盪洩漏的 調整方法，包含以下步驟： 檢測本地振盪洩漏的程度； 判斷一第一調整方向是否正確，如果是，則維持該第 一調整方向，否則，使該第一調整方向反向；及 根據該第一調整方向調整一第一控制信號。 本發明降低發射機或接收機的同相/正交相不匹配的調 整方法，包含以下步驟： 檢測同相/正交相不匹配的程度； 丄判斷第一调整方向是否正確，如果是，則維持該第 凋整方向’否％，使該第一調整方向反向；及 Ί亥第一調整方向調整一第一控制信號。 而本發明之另-目的即在提供一種發射機及_種接收 200915744 機，可以降低本地振盪茂漏。 於是’本發明發射機包含： -第-混頻器’將—基頻信號與一本地振盪信號混頻 ，以產生一射頻信號； 一檢測單兀，根據該射頻信號產生反應本地振盪洩漏 程度的一檢測信號；及 一調整單元，輸出一控制信號來改變該第一混頻器的 操作狀態，且根據該檢測信號判斷該控制信號的一調整方 向是否使本地漏的程度變小，如果本地㈣茂漏的 程度變小，則維持該調整方向，否則，使該調整方向反向 ，並根據該調整方向調整該控制信號。 本發明接收機包含： 一混頻器，將一射頻信號與一本地振盪信號混頻，以 產生一基頻信號； 檢則單元根據該基頻信號產生反應本地振盪洩漏 程度的一檢測信號；及 ^調整單兀，輸出一控制信號來改變該混頻器的操作 狀悲’且根據該檢測信號判斷該控制信號的一調整方向是 使本地振盪洩漏程度變小，如果本地振盪洩漏程度變小 ^則維持该調整方肖，否則，使該調整方向反向，並根據 '^調整方向’調整該控制信號。 而本發日月> £ 为—目的即在提供一種發射機及一種接收 機’可：降低同相/正交相不匹配。 於是，本發明發射機包含： 200915744 -補償單元，對—第—基_號及n 行相位及振幅補償，以產生二輸出信號，· 土, D〜 二數位至類比轉換器，分別對該補償單元的二輸出信 號進行數位至類比轉換； 二低通濾波器，分別對該二數位 信號進行低《波； 二混頻器’將該二低通渡波器的輸出信號分別與一同 相本地振盪信號及一正交相本地 射頻信號； ㈣遽混頻，以產生二 一第一加總器，對該二射頻信號進行加總； 一檢測單元’根據該第一加嫡 门4 , 加…态的輪出信號產生反應 同相/正交相不匹配程度的一檢測信號；及 。-調：單元，輸出至少一控制信號來改變該補償單元 的操作狀悲，且根據該檢測信號判 ^ μ 。现刘辦母—控制信號的每一 调正方向是否使同相/正交相不匹配的程度變小，如果同相/ 正交相不匹配的程度變小，則維持該等調整方向，否則， 至少使該專調整方向之盆中夕 r-, 门之/、中之一反向，並根據該等調整方 向調整該等控制信號。 本發明接收機包含： '一混頻器，其中一去技 At lt3-.. 、 、—射頻^號與—同相本地振盪 ^號混頻，以產生一基頻传轳 J。唬’而其中另-者將該射頻信 就與-正父相本地録信號混頻，以產生另—基頻信號； 二低通遽波器，分別斜号_ 才。玄一混頻益的輪出信號進行低 通濾波； - 10 200915744 一類比至數位轉換器，分別對該二低通濾波器的輪出 信號進行類比至數位轉換’以產生一第一基頻信號及—第 二基頻信號； 一補彳Μ单元，對該第一基頻信號及該第二基頻信號進 行相位及振幅補償，以產生二輸出信號； 一檢測皁元，根據該補償單元的二輸出信號，產生反 應同相/正交相不匹配程度的一檢測信號；及 凋玉單元，輸出至少一控制信號來改變該補償單元 的操作狀態，且根據該檢測信號判斷每一控制信號的每— 調整方向是否使同相/正交相不匹配的程度變小，如果同相/ 正交相不匹配的程度變小，則維持該等調整方向，否則， 至少使該等調整方向之其中之一反向，並根據該調整方向 調整該等控制信號。 【實施方式】 有關本發明之前述及其他技術内容、特點與功效，在 、下配口參考圖式之二個較佳實施例的詳細說明中，將可 清楚地呈現。 凊參閱圖4 ’所示係本發明發射機之一較佳實施例，包 3 „„補償單凡4〇、二數位至類比轉換器41、42、二低通濾、 波斋43、44、二混頻器45、％、―加總_ 4丄及-調整單元49。所述之補償單元仙對二數位的基頻 =號BBIt、BBQt進行相位及振幅補償，且具有多個操作狀 〜’、中不同的操作狀態對應到不同程度的同相/正交相 不匹配。在本實施例中，補償單元4Q包括二增益級彻、 200915744 =: 增益級401將基頻信號卿乘以-可 為補償單元4°的一輸出信號。增益級將 虎卿乘以-可變因數Yt，而加總器彻將基頻信 唬BBQt與增益級4〇2的輪 耵掏出4唬加總，以當作補償單元4〇 的另一輸出信號。二數位至類比轉換器41、42分別對增益 級4〇1及加總器403的輸出信號進行數位至類比轉換，而 二低通遽波器43、44分別對二數位至類比轉換器4卜42 的輸出信號進行低通遽波。混頻器45將低通滤波器43的 輸出信號與-同相本地振盪信號咖混頻，以產生一同相 射頻信號RFIt ’而混頻器46將低通濾波器44的輸出信號 與正交相本地振璗信號L〇Qt混頻，以產生—正交相射頻 ^虎RFQt。每-混頻器45、46皆具有多個操作狀態，且不 同操作狀態對應到不同程度的本地振盪浪漏。加總器47對 二混頻器45、46的輪出信號進行加總，檢測單元48再根 據加總器47的輸出信號’產生位於基頻的—反應本地振盈 洩漏程度的檢測信號及另一反應同相/正交相不匹配程度的 檢測信號。在本實施例中，檢測單元48包括一混頻器48ι '—可變增盈放大器482、一類比至數位轉換器483及—快 速傅利葉轉換器484，以依序對加總器47的輸出信號進行 自身混頻、放大、類比至數位轉換及快速傅利葉轉換，來 產生檢測信號。當二基頻信號BBlt、BBQt是弦波信號且其 頻率是FBB時，混頻器481的輸出信號將在ρΒΒ處（由於本 地振盪洩漏）及在2xFBB處（由於同相/正交相不匹配）具 有頻譜成分’且其頻譜分析可以顯示出本地振盪洩漏的程 12 200915744 度及同相/正交相不匹配的程度。在另一實施例中，混頻器 481也可以被替換為一包跡檢測器，且在其它實施例中，可 變增益放大器482在不那麼需要時也可以被省略。調整單 元49輸出四控制信號I(n)、Q(n)、χ(η)、γ(η)來分別改變 二混頻器45、46的操作狀態及二可變因素Xt、Yt，以降低 本地振盪洩漏及同相/正交相不匹配的程度。 請參閱圖5 ’所示係本實施例用以改變每一控制信號 I(n)、Q(n)、χ(η)、γ⑻的調整方法，包含以下步驟·· 步驟50,該檢測單元48產生反應本地振盪洩漏程度的 檢測信號或反應同相/正交相不匹配程度的檢測信號。 步驟51，該調整單元49根據先前調整方向所得到之檢 測信號，判斷控制信號的調整方向是否正確，如果是，跳 到步驟53，否則，跳到步驟52。在本實施例中，對於二混 頻器45、46的控制信號I(n)、Q(n)，是根據所檢測的本地 振盪洩漏程度’如果變小，表示調整方向正確，而對於補 償單元40的控制信號x(n)、Y(n)，是根據所檢測的同相/正 交相不匹配程度，如果變小，表示調整方向正確。 步驟52，該調整單元49使調整方向反向。 步驟53，該調整單元49根據調整方向，調整控制信號 。在本實施例中，對於二混頻器45、46的控制信號Ι(η)、 Q(n)及補償單元40的控制信號Χ(η)、Υ(η)，其中每—控制 信號的調整量與控制信號的調整方向及控制信號的調整級 距（step )成正比。 除以上步驟之外，本實施例用以改變每一控制信號1(幻 13 200915744 、Q(n)、X(n)、Υ(η)的調整方法更可以包含以下步驟： 步驟54，該調整單元49判斷是否滿足一結束條件，如 果是，則固定控制信號並結束調整，否則，跳到步驟5〇以 重覆步驟50〜53。在一實施例中，是否滿足結束條件可以是 根據執行步驟50〜53的次數來決定，也就是說，當重覆步 驟50〜53達到一預設次數時，即滿足結束條件。然而在另 一實施例中’是否滿足結束條件也可以是根據本地振盪浪 漏程度及同相/正交相不匹配程度來決定，意即當本地振盈 浪漏程度及同相/正交相不匹配程度分別小於所對應之預設 程度時’即滿足結束條件。 在本實施例中，可進行步驟50〜53來分別調整控制信 號Ι(η)、控制信號Q(n)、控制信號χ(η)及控制信號Υ(η)， 最後再進行步驟54以判斷是否繼續調整。 請參閱圖6，所述係本發明接收機之一較佳實施例，包 含二混頻器61、62、二低通濾波器63、64、二類比至數位 轉換器65、66、一補償單元67、一檢測單元68及一調整 單元69。 混頻器61將一類比的射頻信號與一同相本地振盪信號 L〇Ir混頻，以產生一基頻信號，而混頻器62將射頻信號與 一正交相本地振盈信號L〇Qr》、昆頻，以產生另一基頻信號。 每—混頻器6卜62具有多個操作狀態，且不同操作狀態對 應到不同程度的本地振m二低通濾波器63、64分別 對二混頻H 61、62的輸出信號進行低㈣波。二類比至數 位轉換器65、66分別料-似, 方J對—低通濾波态63、64的輸出信號 14 200915744 轉換。補償單元67對二類比至數位轉換器 ’輪出信號進行相位及振幅補償，且具有多個操作 ‘…、中’不同操作狀態對應到不同程度的同相/正交相 不匹配。在本實施例中，補償單元67包括二增益級671、 力〜、器673。增益級671將類比至數位轉換 的輸㈣號乘以-可變因數&，增益級672將類比至數位 轉換器66的輸出信號乘以—可變因數 '，而加總器673將 ’曰'血、.及671、672的輸出信號加總，以輸出一基頻信號 BBIr。=償單元67將類比至數位轉換器66的輸出信號二 輸出，#作另—基頻信號BBQr。檢測單it 68根據二基頻信 號BBIr、BBQr，產生一反應本地振堡沒漏程度的檢測信號 及另一反應同相/正交相不匹配程度的檢測信號。在一實施 例中★測單元68可包括—快速傅利葉轉換器68卜快速 傅利葉轉換器可將二基頻信號BBIr、BBQr視為一複數信號 BBIr+jxBBQr，來進行快速傅利葉轉換，以產生檢測信號。 有理心之射頻k號是沒有本地振盡沒漏及同相/正交相 不匹配的情形時’例如：當此射頻信號是調整後的發射機 所產生，且二基頻信號BBIt、BBQt是弦波信號且頻率是 fbb時，則二基頻信號BBIr、BBQr具有在Dc處（由於本 地振盪洩漏）及在·Fbb處（由於同相/正交相不匹配）的頻 譜成分’且其頻譜分析可以顯示出本地振盪洩漏的程度及 同相7正交相不匹配的程度。調整單元69可輸出四控制信號 I(n)、Q⑻、X(n)、Y(n)來分別改變二混頻器61、62的操作 狀態及二可變因素Xr、Yr，以降低本地振盪洩漏及同相/正 15 200915744 父相不匹配的程度。調整早元69的動作與發射機的調整單 元49的動作類似，此處將不再多加說明。值得注意的是， 在發射機及接收機的實施例中，調整單元49、69可分別接 收位於基頻用以反應本地振盪洩漏程度及同相/正交相不匹 配程度的檢測信號’且是以數位方式來實現，因此將具有 容易實現的優點。 惟以上所述者，僅為本發明之較佳實施例而已，當不 能以此限定本發明實施之範圍，即大凡依本發明申請專利 範圍及發明說明内容所作之簡單的等效變化與修飾，皆仍 屬本發明專利涵蓋之範圍内。 【圖式簡單說明】 圖1是一習知的發射機之方塊圖； 圖2是另一習知的發射機之方塊圖； 圖3是又一習知的發射機之方塊圖； 圖4是本發明發射機的較佳實施例之方塊圖； 圖5是本發明調整方法的較佳實施例之流程圖；及 16是本發明接收機的較佳實施例之方塊圖。 16 200915744 【主要元件符號說明】 40·· 補償單元 器 401 ....... 增益級 49·· 調整單元 402 增益級 5 0〜54 — 步驟 403 加總器 61、 62 ·· 混頻器 41、 42 ·· 數位至類比轉換 63 ' 64 · · 低通濾波器 器 65、 66 ·· 類比至數位轉換 43、 44 · 低通濾波器 器 45、 46 ·· 混頻器 67·· 補償單元 47·· 加總器 671 增益級 48·· 檢測單元 672 增益級 481 混頻器 673 加總器 482 可變增益放大器 68.· 檢測單元 483 類比至數位轉換 681 快速傅利葉轉換 器 器 484 快速傅利葉轉換 69·· 調整單元 17200915744 IX. Description of the Invention: [Technical Field] The present invention relates to a transmitter and a receiver and an adjustment method thereof, and more particularly to a transmitter and receiver for reducing local oscillation leakage and in-phase/quadrature phase mismatch Machine and its adjustment method. [Prior Art] Please refer to FIG. 1. FIG. 1 illustrates a conventional direct up-conversion transmitter including two-bit to analog converters 11, 12 and two low-pass filters 13. 14. Two mixers 15, 16, a summaster 17, a power amplifier 18, and an antenna 19. Wherein, the digital baseband signal BBIt is sequentially digital-to-analog converted, low-pass filtered, and mixed with an in-phase local oscillating signal L〇It to generate an analog in-phase RF signal RFIt, and another digit The baseband signal BBQt is sequentially digital-to-analog-converted, low-pass filtered, and mixed with a quadrature-phase local 彳sL0Qt to produce an analog quadrature-phase RF signal size. The two RF signals RFIt and RFQt will be summed and amplified to be transmitted to the outside world. The local oscillation signal L0It& 1^0 (the ideal phase difference of ^ is 9 ' degrees but there will actually be a phase offset et ' and the blocks on the in-phase path (including the digital to analog converter 11 and the low pass) The filter 13) and the blocks on the quadrature phase path (including the digital to analog converter 12 and the low pass filter 14) also have a gain offset (indicated by an amplitude offset in Figure 1). This phenomenon is called I/Q mismatch or I/Q imbalance. In addition, the two local oscillator signals L〇It 200915744 and LOQt may be separately The corresponding two mixers 15, 16 leak to the two RF signals RFIt, RFQt, and this phenomenon is called local oscillation leakage or local oscillation feedthrough. / Orthogonal phase mismatch and local oscillator leakage will reduce the signal-to-noise ratio of the signal transmitted by this transmitter, and may result in data loss. Please refer to Figure 2, which is one of the methods disclosed in U.S. Patent No. 6,970,689. Local oscillator leaking transmitter The transmitter comprises a mixer 21, a power amplifier 22, a signal strength measuring circuit 23 and a control signal generating circuit 24. The mixer 21 has a plurality of operating states, and the operating states respectively correspond to different operating states. The local oscillation leakage level. The signal strength measuring circuit 23 is for measuring the intensity of the local oscillation leakage component in the output signal of the power amplifier 22, and includes a rectifier (not shown) and a comparison (not shown). The control signal generating circuit 24 then outputs a control signal to change the operational state of the mixer 21. During the correction of the mixer 21, the power amplifier 22 will increase its gain, and the control signal generating circuit 24 will change the mixer 21. The operating state, and store the information of the operating state and the signal strength measuring circuit Μ the measured strength 'and repeat the above steps until all operating states of the mixer 21 have been used' and then set the mixer 21 to local The operating state with the least degree of vibration leakage. Or in the middle of the right mixer 21, the power amplifier 22 will increase during the period.増益' and the control signal produces a lightning reform 24 surname Wang Tu circuit 24 to continue to change the operating state of the mixer 21 until the signal strength measurement is thunder? The power measured by the circuit 23 is less than 200915744 a preset threshold value ' Then, the operating state of the mixer 21 is fixed. Please refer to FIG. 3, which is ISSCC 2006 / SESSION 20 / WLAN/WPAN / 20.4 "A Highly Linear Direct-Conversion Transmit Mixer Trans conductance Stage with Local Oscillation Feedthrough and I/Q The imbalance Cancellation Scheme discloses a transmitter for reducing local vibrating bubble leakage and in-phase/quadrature phase mismatch. The transmitter includes two digits to analog converters 301, 302, two low pass filters 303, 304, two mutual transconductance stages 305, 306, two mixers 307, 308, a totalizer 309, a power amplifier 310, An antenna 311, an envelope detector 312 and a variable gain amplifier 313. The two-digit baseband signals BBIt and BBQt are respectively converted into two analog RF signals RFIt, RFQt, which are summed and amplified to be transmitted to the outside world. The envelope detector 312 and the variable gain amplifier 313 sequentially detect and amplify the output signal of the power amplifier 3 10 to generate a fundamental frequency chop. When the two fundamental frequency signals BBIt and BBQt are sinusoidal signals and the frequency is Fbb, the spectral components of the fundamental frequency chopping appear at Fbb (due to local oscillation leakage) and 2xFBB (due to in-phase/quadrature phase mismatch). And its spectrum analysis can show the degree of local oscillation leakage and the degree of in-phase/quadrature phase mismatch. Local oscillation leakage can be divided into two types: fundamental frequency local oscillation leakage and RF local oscillation leakage. The fundamental frequency local oscillation leakage is caused by the component offset of the binary to analog converters 301, 302, the two low pass filters 303, 304 and the two transconductance stages 305, 306, and the RF local oscillation leakage 200915744 Direct coupling due to parasitic capacitance or mutual inductance. Among them, the two local oscillation leakages need to be reduced each. However, the isscc paper does not explain how to adjust the phase and amplitude of the two mutual transconductance stages 305, 306 and the two fundamental frequency #BBIt, BBQt to reduce the local excitation leakage and the in-phase/quadrature phase mismatch of the transmitter. There is no mention of how to reduce the local oscillation leakage of the receiver and the problem of in-phase/quadrature phase mismatch. SUMMARY OF THE INVENTION Accordingly, it is an object of the present invention to provide an adjustment for reducing the local vibration bubble of a transmitter or receiver. The method and an adjustment method for reducing the in-phase/quadrature phase mismatch of the transmitter or receiver. Therefore, the method for adjusting the local oscillation leakage of the transmitter or the receiver includes the following steps: detecting the degree of local oscillation leakage; determining whether a first adjustment direction is correct, and if yes, maintaining the first adjustment direction, otherwise And reversing the first adjustment direction; and adjusting a first control signal according to the first adjustment direction. The invention reduces the in-phase/quadrature phase mismatch adjustment method of the transmitter or the receiver, and includes the following steps: detecting the degree of in-phase/quadrature phase mismatch; determining whether the first adjustment direction is correct, and if so, maintaining the The first direction is 'n%, the first adjustment direction is reversed; and the first adjustment direction is adjusted by the first adjustment direction. Another object of the present invention is to provide a transmitter and a receiver for the 200915744, which can reduce local oscillation leakage. Thus, the transmitter of the present invention comprises: - a first mixer - mixing a baseband signal with a local oscillating signal to generate a radio frequency signal; and a detecting unit for generating a degree of local oscillation leakage according to the radio frequency signal a detection signal; and an adjustment unit, outputting a control signal to change an operation state of the first mixer, and determining, according to the detection signal, whether an adjustment direction of the control signal makes a local leakage degree smaller, if local (4) If the degree of leakage becomes small, the adjustment direction is maintained. Otherwise, the adjustment direction is reversed, and the control signal is adjusted according to the adjustment direction. The receiver of the present invention comprises: a mixer that mixes a radio frequency signal with a local oscillating signal to generate a baseband signal; and the detecting unit generates a detection signal that reflects the degree of local oscillation leakage according to the baseband signal; ^Adjusting the single 兀, outputting a control signal to change the operational sorrow of the mixer' and determining an adjustment direction of the control signal according to the detection signal is to make the local oscillation leakage become smaller, if the local oscillation leakage becomes smaller ^ Then, the adjustment mode is maintained. Otherwise, the adjustment direction is reversed, and the control signal is adjusted according to the '^ adjustment direction'. And the present day &month; £ is for the purpose of providing a transmitter and a receiver' to: reduce the in-phase/quadrature phase mismatch. Thus, the transmitter of the present invention comprises: 200915744 - compensation unit, phase-and-amplitude compensation for -first base_number and n-line to generate two output signals, earth, D~two digits to analog converter, respectively for the compensation The two output signals of the unit are digital-to-analog conversion; the second low-pass filter respectively performs low-waves on the two-digit signals; the two mixers respectively output the output signals of the two low-pass ferrites with an in-phase local oscillation signal And a quadrature phase local RF signal; (4) 遽 mixing to generate a second first adder, summing the two radio signals; a detecting unit 'according to the first twisting gate 4, adding The turn-off signal produces a detection signal that reflects the degree of in-phase/quadrature phase mismatch; - a modulation unit that outputs at least one control signal to change the operational sorrow of the compensation unit, and judges μ according to the detection signal. Now, if the positive direction of the control signal is such that the in-phase/orthogonal phase mismatch becomes smaller, if the degree of in-phase/orthogonal phase mismatch becomes smaller, the adjustment direction is maintained. Otherwise, at least The one of the gates of the special adjustment direction is reversed, and one of the gates is reversed, and the control signals are adjusted according to the adjustment directions. The receiver of the present invention comprises: 'a mixer, wherein one of the techniques At lt3-.., - the RF number and the in-phase local oscillation number are mixed to generate a fundamental frequency.唬' and the other one mixes the RF signal with the --father-local recording signal to generate another-baseband signal; the second low-pass chopper, slash _. The round-off signal of Xuanyi Mixing is low-pass filtered; - 10 200915744 A class-to-digital converter that performs analog-to-digital conversion on the round-out signal of the two low-pass filters respectively to generate a first fundamental frequency signal And a second fundamental frequency signal; a complementary unit, phase and amplitude compensation of the first fundamental frequency signal and the second fundamental frequency signal to generate two output signals; a detecting soap element, according to the compensation unit a second output signal, generating a detection signal that reflects the degree of mismatch between the in-phase/quadrature phase; and the fading unit, outputting at least one control signal to change an operation state of the compensation unit, and determining each control signal according to the detection signal – whether the adjustment direction makes the degree of mismatch between the in-phase/orthogonal phases become smaller. If the degree of mismatch between the in-phase/orthogonal phases becomes smaller, the adjustment direction is maintained. Otherwise, at least one of the adjustment directions is reversed. And adjust the control signals according to the adjustment direction. [Embodiment] The foregoing and other technical contents, features, and advantages of the present invention will be apparent from the detailed description of the preferred embodiments of the present invention. Referring to Figure 4', there is shown a preferred embodiment of the transmitter of the present invention. The package 3 „„compensation unit 4〇, the second digit to analog converter 41, 42 , the second low pass filter, the wave fast 43 , 44 , The second mixer 45, %, "total_4" and - adjustment unit 49. The compensation unit performs phase and amplitude compensation on the fundamental frequency of the two digits = BBIt, BBQt, and has a plurality of operational states ~', and the different operational states correspond to different degrees of in-phase/quadrature phase mismatch. In the present embodiment, the compensation unit 4Q includes two gain stages, 200915744 =: the gain stage 401 multiplies the fundamental frequency signal by - and can be an output signal of the compensation unit 4°. The gain stage multiplies Tiger Qing by the -variable factor Yt, and the adder adds the baseband signal BBQt and the gain of the gain stage 4〇2 to 4唬 to be the other of the compensation unit 4〇 output signal. The two-bit to analog converters 41 and 42 respectively perform digital-to-analog conversion on the output signals of the gain stage 4〇1 and the adder 403, and the two low-pass choppers 43 and 44 respectively pair the two-digit to analog converters. The output signal of 42 is low pass chopped. The mixer 45 mixes the output signal of the low pass filter 43 with the in-phase local oscillating signal to generate an in-phase RF signal RFIt' and the mixer 46 localizes the output signal of the low pass filter 44 to the quadrature phase. The vibrating signal L〇Qt is mixed to generate a quadrature phase radio frequency RFQt. Each of the mixers 45, 46 has a plurality of operational states, and the different operational states correspond to different degrees of local oscillation leakage. The adder 47 sums the rounding signals of the two mixers 45, 46, and the detecting unit 48 generates a detection signal of the local vibration leakage level at the fundamental frequency according to the output signal of the adder 47 and another A detection signal that reflects the degree of mismatch between the in-phase/orthogonal phases. In the present embodiment, the detecting unit 48 includes a mixer 48'', a variable gain amplifier 482, an analog-to-digital converter 483, and a fast Fourier transformer 484 for sequentially outputting the output signal of the adder 47. Perform self-mixing, amplification, analog to digital conversion, and fast Fourier transform to generate the detection signal. When the two fundamental frequency signals BBlt, BBQt are sine wave signals and their frequency is FBB, the output signal of the mixer 481 will be at ρΒΒ (due to local oscillation leakage) and at 2xFBB (due to in-phase/quadrature phase mismatch) It has a spectral component' and its spectral analysis can show the degree of local oscillation leakage and the degree of in-phase/quadrature phase mismatch. In another embodiment, the mixer 481 can also be replaced with an envelope detector, and in other embodiments, the variable gain amplifier 482 can be omitted when not needed. The adjusting unit 49 outputs four control signals I(n), Q(n), χ(η), γ(η) to change the operating states of the two mixers 45, 46 and the two variable factors Xt, Yt, respectively, to reduce Local oscillation leakage and the degree of in-phase/quadrature phase mismatch. Referring to FIG. 5', the method for adjusting each control signal I(n), Q(n), χ(η), γ(8) is included in the embodiment, and includes the following steps: Step 50, the detecting unit 48 A detection signal that reflects the degree of local oscillation leakage or a detection signal that reflects the degree of in-phase/orthogonal phase mismatch. In step 51, the adjusting unit 49 determines whether the adjustment direction of the control signal is correct according to the detection signal obtained by the previous adjustment direction. If yes, the process goes to step 53, otherwise, the process goes to step 52. In the present embodiment, the control signals I(n), Q(n) for the two mixers 45, 46 are based on the detected degree of local oscillation leakage 'if it is smaller, indicating that the adjustment direction is correct, and for the compensation unit. The control signals x(n) and Y(n) of 40 are based on the detected in-phase/quadrature phase mismatch, and if they become smaller, the adjustment direction is correct. In step 52, the adjusting unit 49 reverses the adjustment direction. In step 53, the adjusting unit 49 adjusts the control signal according to the adjustment direction. In the present embodiment, the control signals Ι(η), Q(n) for the two mixers 45, 46 and the control signals Χ(η), Υ(η) of the compensation unit 40, wherein each control signal is adjusted The amount is proportional to the adjustment direction of the control signal and the adjustment step of the control signal. In addition to the above steps, the adjustment method for changing each control signal 1 (Fantasy 13 200915744, Q(n), X(n), Υ(η) in this embodiment may further include the following steps: Step 54, the adjustment The unit 49 determines whether an end condition is satisfied, and if so, fixes the control signal and ends the adjustment; otherwise, it jumps to step 5 to repeat steps 50 to 53. In an embodiment, whether the end condition is satisfied may be according to the execution step. The number of times 50 to 53 is determined, that is, when the repeating steps 50 to 53 reach a preset number of times, the end condition is satisfied. However, in another embodiment, whether the end condition is satisfied or not may be based on the local oscillation wave. The degree of leakage and the degree of in-phase/orthogonal phase mismatch are determined, that is, when the local vibration leakage degree and the in-phase/orthogonal phase mismatch degree are respectively less than the corresponding preset degree, the end condition is satisfied. In the example, steps 50 to 53 may be performed to separately adjust the control signal η(η), the control signal Q(n), the control signal χ(η), and the control signal Υ(η), and finally step 54 is performed to determine whether to continue adjusting. Please refer to Figure 6 The preferred embodiment of the receiver of the present invention comprises two mixers 61, 62, two low pass filters 63, 64, two analog to digital converters 65, 66, a compensation unit 67, and a detection The unit 68 and an adjusting unit 69. The mixer 61 mixes an analog RF signal with an in-phase local oscillation signal L〇Ir to generate a baseband signal, and the mixer 62 combines the RF signal with a quadrature phase. The local vibration signal L〇Qr′′ and the Kun frequency are used to generate another fundamental frequency signal. Each mixer 6 has 62 operating states, and different operating states correspond to different degrees of local oscillator m low pass filtering. The outputs 63 and 64 respectively perform low (four) waves on the output signals of the two mixing frequencies H 61 and 62. The two analog-to-digital converters 65 and 66 respectively generate a similar signal, and the output signals of the low-pass filtered states 63 and 64 are 14 200915744 Conversion. The compensation unit 67 performs phase and amplitude compensation on the two analog-to-digital converter 'round-off signal, and has multiple operations '..., medium' different operational states corresponding to different degrees of in-phase/quadrature phase mismatch. In this embodiment, the compensation unit 67 includes two gain stages 67. 1. Force ~, 673. Gain stage 671 multiplies the analog-to-digital conversion of the input (four) by the -variable factor & the gain stage 672 multiplies the output signal analogous to the digital converter 66 by the -variable factor', The adder 673 sums the output signals of the '曰' blood, and the 671 and 672 to output a fundamental frequency signal BBIr. The compensation unit 67 outputs the analog output to the output signal of the digital converter 66. - a baseband signal BBQr. The detection unit it 68 generates a detection signal reflecting the degree of no leakage of the local Zhenbao and another detection signal reflecting the degree of mismatch of the in-phase/orthogonal phase according to the two fundamental frequency signals BBIr, BBQr. In an embodiment, the measuring unit 68 may include a fast Fourier converter 68, and the fast Fourier converter may treat the two fundamental frequency signals BBIr and BBQ as a complex signal BBIr+jxBBQr for fast Fourier transform to generate a detection signal. . The rational RF k number is when there is no local vibration and no leakage and in-phase/orthogonal phase mismatch. For example: when the RF signal is generated by the adjusted transmitter, and the two fundamental signals BBIt and BBQ are strings When the wave signal and the frequency is fbb, the two fundamental frequency signals BBIr and BBQr have spectral components at Dc (due to local oscillation leakage) and at ·Fbb (due to in-phase/quadrature phase mismatch) and their spectrum analysis can Shows the degree of local oscillation leakage and the degree of mismatch in the in-phase 7 orthogonal phase. The adjusting unit 69 can output four control signals I(n), Q(8), X(n), Y(n) to change the operating states of the two mixers 61, 62 and the two variable factors Xr, Yr, respectively, to reduce local oscillation. Leakage and in-phase/positive 15 200915744 The degree of parental mismatch. The action of adjusting early element 69 is similar to the action of adjustment unit 49 of the transmitter and will not be described again here. It should be noted that in the transmitter and receiver embodiments, the adjusting units 49, 69 can respectively receive the detection signals at the fundamental frequency to reflect the degree of local oscillation leakage and the degree of in-phase/quadrature phase mismatch, and are digital. The way to achieve this will therefore have the advantage of being easy to implement. The above is only the preferred embodiment of the present invention, and the scope of the invention is not limited thereto, that is, the simple equivalent changes and modifications made by the scope of the invention and the description of the invention are All remain within the scope of the invention patent. BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a block diagram of a conventional transmitter; FIG. 2 is a block diagram of another conventional transmitter; FIG. 3 is a block diagram of still another conventional transmitter; Figure 4 is a block diagram of a preferred embodiment of the adjustment method of the present invention; and 16 is a block diagram of a preferred embodiment of the receiver of the present invention. 16 200915744 [Description of main component symbols] 40·· Compensation unit 401 ....... Gain stage 49·· Adjustment unit 402 Gain stage 5 0~54 — Step 403 Adder 61, 62 ·· Mixer 41, 42 ·· Digital to analog conversion 63 ' 64 · · Low-pass filter 65, 66 ·· Analog to digital conversion 43, 44 · Low-pass filter 45, 46 ·· Mixer 67·· Compensation unit 47·· Adder 671 Gain stage 48·· Detection unit 672 Gain stage 481 Mixer 673 Adder 482 Variable gain amplifier 68.· Detection unit 483 Analog to digital conversion 681 Fast Fourier converter 484 Fast Fourier transform 69·· Adjustment unit 17