SE447527B - HIGH FREQUENCY FERROR RESONANCE ENERGY FOR A DEVICING AND HIGH VOLTAGE CIRCUIT IN A TV RECEIVER - Google Patents

Description

u \ _44? 527 2 Since the transformer is operated with the low frequency 60 Hz, a relatively large and heavy transformer must be used. Furthermore, the high voltage is then supplied independently by means of a relatively large sweep return transformer which is so constructed that it can allow relatively large amounts of energy.

In other regulator circuits for television receivers, which circuits are known per se, the peg voltage is regulated by arranging a sweep return transformer which is itself operated under ferrous resonance conditions. Sweep return pulses are coupled to the primary winding of the sweep return transformer, and the ultra-high voltage winding is then tuned to the desired frequency. Since the B + supply is taken from a separate energy source, such as the AC mains, an individual control circuit must be provided if the B + voltage is also to be regulated. If the E + voltage is unregulated, other circuits may be necessary with the help of which a constant raster width can be maintained.

In many typical high voltage circuits operated by means of sweep return transformers, the high voltage gives rise to a peak voltage which is considerably less than the required ultra-potential in order to reduce the number of rpm turns of the high voltage winding. A high voltage multiplier then raises the voltage to the required level. Since many voltage multipliers are designed in such a way that both polarities of the AC voltage must be used, it is desirable that the multiplier be driven by an AC voltage whose positive and negative polarity parts are substantially equal to each other. If one polarity is much smaller than the other, the capacitors and diodes that are active when this polarity applies will contribute with very little voltage.

This is the case when the sweep return pulse is used as the supply source for a voltage multiplier circuit. A six-fold multiplier, in which six diodes and six capacitors are used, is required in order to obtain a three-fold multiplication when a sweep return pulse with a low operating ratio is fed to a voltage multiplier. According to a preferred embodiment of the invention, a ferro-resonance energy source for a deflection and high voltage circuit in a television receiver comprises an alternating voltage source. A ferroresonance transformer has a magnetic core, a first winding which is connected to the AC voltage source, and a high voltage winding which is wound around a core portion of the magnetic core and which is connected to a high voltage socket for generating a high voltage. A second winding is wound around the core portion of the magnetic core and is connected to a scanning supply voltage socket to give rise to a scanning supply voltage. Means impart at least one winding of the ferro-resonance transformer sufficient capacitance to generate circulating currents to saturate the core portions during the high voltage winding and the second winding during each period of the alternating voltage. In this way, a regulated high voltage and a regulated scanning supply voltage are obtained. A deflection switch is connected to a deflection winding to generate sweep and sweep regression intervals during each deflection period. A sweep voltage source is connected to the deflection winding to give rise to a scanning current in the deflection winding. The first means connects the regulated scan supply voltage to the sweep voltage source. An ultra-outlet adds an ultra-acceleration potential. High-voltage means are connected to the high-voltage socket and to the ultra-socket to generate the ultra-acceleration potential from the regulated high-voltage.

The invention will be described in detail in the following with reference to the accompanying drawings, in which Fig. 1 shows an electrical circuit diagram of a high frequency ferro-resonance energy source for a deflection and high voltage circuit in accordance with the invention, Fig. 2 shows the design of a high-frequency circuit. frequency ferrose resonance transformer core with associated winding used in the circuit of Fig. 1, Fig. 3 shows a cross section of the transformer according to Fig. 2 taken along line 3-3, Fig. 4 shows another core arrangement of the transformer according to Fig. 2, and Fig. 5 shows waves associated with the circuit shown in Fig. 1.

According to Fig. 1, a mains AC voltage of, for example, 447 527 4 120 volts AC voltage, 60 hertz, is connected to the terminals 21 and 22 and thence to the input terminals 23 and 24 of a full-wave bridge rectifier 25. A current limiting resistor 30 is connected between the terminals 21 and 23. A direct voltage of, for example, +150 volts is obtained at a socket 26 and is filtered by means of a capacitor 27 which is connected between the socket 26 and a socket 28 which constitutes the common earth current return socket and which is not isolated from the AC mains. A resistor 29 is connected to the socket 26 for the supply voltage% + 150 volts. A small DC voltage of, for example, +20 volts is obtained at the cathode (terminal 32) of a zener diode 33 which is connected to the resistor 29.

A square wave oscillator 35 for square wave energy comprises a sine wave oscillator 36, a receiving coupled square forming stage 37 and a power output stage 38. The sine wave oscillator 36 is self-oscillating and comprises a transistor 39 with a collector electrode connected to an LC resonant circuit 40 and a winding 4la of a switching transformer 41. Collector voltage for the transistor 39 is obtained from the voltage source of +20 volts which is connected to a terminal 48 of the winding 4la via a resistor 42. A bypass capacitor 43 is connected to the terminal 48. The resistor 42 reduces the voltage of 20 volts to a voltage of 17 volts, and capacitor 43 helps to remove the ripple from the rectified input voltage with the frequency 16 Hz. The AC voltage feedback which has the task of keeping the oscillator 38 self-oscillating is obtained from a capacitor 44 which is connected between the tank circuit 45 and the base electrode of the transistor 39. DC bias for the base electrode is obtained from the voltage divider resistors 46 and 47 which are connected to the resistor 42.

Oscillator 36 generates a high frequency sine voltage in winding 41a. The resonant frequency of the tank circuit 45 may, for example, be so selected that it is close to the horizontal deflection frequency 1 / TH, which is about 15.75 kHz. The horizontal sweep return pulses 67, which are obtained from a horizontal sweep return transformer 68, are connected to a synchronizing input terminal 447 527 69 of the oscillator 56. The sweep return pulses 67 are switched to current from the terminal 69 to the emitter 71 of the resistor 71 and a capacitor 71 emitter. and 72. The sweep return pulses 67 synchronize the frequency of the oscillator 56 to the horizontal deflection frequency by turning off the transistor 59 within the horizontal sweep return range.

The high frequency sinusoidal voltage of the winding 41a of the oscillator 56 is connected by means of a winding 41b of the transformer 41 to the base electrodes of the receiving-connected transistors H9 and 50 through the respective resistors 51 and 52. A central terminal of the winding blb is grounded. The square forming step 57 converts the sine voltage formed by the oscillator 56 into a square wave voltage of the same frequency. Said square wave voltage is more suitable than a sine wave when it comes to driving the power output stage 58.

The high frequency square wave voltage generated by the square forming stage 57 is coupled from a winding 55a of a switching transformer 55 to the base electrodes of the received-coupled power output transistors 54 and 55 through a winding 55b of the transformer 55 and through the respective counter. _. ,. ---, | w ----.-_-, .. _... (v _. ._ via .- \, ........._.... ~ a ... íø, _. are connected in parallel, are connected between a central terminal of the winding 55b and the common connection point of the emitters of the transistors 54 and 55. The resistor 58 and the capacitor 59 provide a negative bias voltage at the base electrodes of the power output transistors.

A diode 60 is connected across the 5% collector-emitter electrodes of the transistor, the cathode of the diode 60 being connected to the collector of the transistor 54. Similarly, a diode 61 is connected across the collector-emitter electrodes of the transistor 55, the cathode of the diode 61 being connected to the collector of the transistor 55.

The diodes 60 and 61 have the task of limiting the peak voltage of undesirable voltage peaks which could damage the transistors.

The power output stage 58 emits a high frequency square AC voltage wave 64 at its output sockets 62 and 65, i.e. the collector electrodes of transistors 54 and 55, respectively. 447 527 6 The voltage 64 serves as a source of unregulated energy or as an excitation voltage for a high frequency ferrous resonance transformer 65. An input winding or primary winding 65a is connected across the output terminals 62 and 65 of the power output stage 38. The supply voltage for the power output stage is unregulated. volt DC at terminal 26, which is connected to a center terminal 66 on primary winding 65a.

The high frequency ferroresonant transformer 65 includes a primary winding 65a, a low voltage secondary winding 65b, a high voltage secondary winding 65c and a magnetic core 165. As shown in Fig. 2, the magnetic core 165 consists of two core portions 165a and 165b. The core portion l65a is designed as a C-shaped member, while the core portion l65b is designed as a relatively thin rectangular disk of magnetic material with a relatively large ratio between surface area and volume.

As shown in Fig. 2, the primary winding 65a is wound around the center section of the C-shaped core portion 115a. The low voltage secondary winding 65b is wound around the disk-shaped core portion 165b. The high voltage winding 65c is wound concentrically around the low voltage winding 65b. Each of the secondary windings 65b and 'Q from v; v. : a; e - ;: ':'? .. f. The other cylindrical coil shapes 265b and 2650. Other suitable winding arrangements may alternatively be used, whereby for example the high voltage winding 65c may be layer wound directly over the low winding winding 65b. As shown in Fig. 4, the core 165 may alternatively be constituted by a rectangular core formed by two C-shaped cores 765 and 865, the ends of which abut each other. at the core legs, the legs 765a and 865a of the two C-shaped cores having reduced cross-sectional area. The low voltage winding 65b and the high voltage winding 65c, not shown in Fig. 4, are then wound concentrically around the legs or legs 765a and 865a, as shown in Fig. 2, the input or primary winding 65a being wound around the opposite leg. in the manner shown in Fig. 2.

As shown in Fig. 1, a conductor in the low voltage secondary winding 65b is connected to a terminal 101, and a 447 527 7. another conductor is connected to a ground current return reference socket 102 which is conductively isolated from the mains AC voltage. This outlet can be at ground potential. The low voltage secondary winding 65b is connected to a scan supply voltage terminal 501 through a half-wave rectifier 401. The high frequency alternating voltage generated by the low voltage winding 65b across the terminals 101 and 102 is half-wave rectified by the rectifier 401 and a voltage is applied by a rectifier B1. for example + 120 volts DC is obtained at the scan supply voltage socket 501.

Second sockets of the direct voltage secondary winding 65b are taken out of said winding and are each connected to a rectifier ÄO5-405 to produce low direct voltages of +50 volts, +72 volts and + 20 volts at the respective sockets 303-505.

The filter capacitors 503-505 are connected to each of the cathodes of the diodes 1403-405.

A horizontal deflection circuit 75 includes a conventional horizontal oscillator and drive circuit 74, a deflection sweep switch 76 including a attenuation diode 77 and a horizontal output transistor 78, a horizontal sweep output capacitor 70 and a series capacitor 80 and a terminal 80. over the sweep capacitor 81 serves as the source of sweep voltage for the horizontal deflection winding 80. During each horizontal sweep interval, the sweep switch 76 is current conducting, switching the sweep voltage Vf over the horizontal deflection winding 80 and thereby giving rise to the required current.

In order to obtain the sweep voltage Vt, the sweep capacitor 81 is connected to the B + scan supply voltage terminal 301 through the primary winding 68a of the sweep return transformer 68. The average DC voltage value of the sweep voltage Vt thus becomes substantially equal to the E-bias voltage supply voltage.

When during the horizontal sweep regression interval the sweep switch 76 is disabled, the horizontal deflection winding 80 and the horizontal sweep regression capacitor 79 resonate for a half oscillation period. The horizontal sweep return pulses generated in the primary winding 68a of the sweep return transformer 68 are transformer coupled to the secondary windings 68b and 68c of the sweep return transformer. The terminals 82b 83 and 83 of the secondary winding 68b connect sweep return pulses to the circuits in question, such as the quenching and horizontal sync circuits. The secondary winding 680 serves as the source of sweep return pulses 67 used to synchronize the oscillator 36 of the high frequency square wave effect oscillator 35.

In many conventional energy supply circuits for tele + vision receivers, the ultra-acceleration potential is obtained from rectified sweep regression pulses. In the circuit of Fig. 1, however, it is the high frequency AC voltage generated across the high voltage secondary winding 65c that gives rise to the ultra voltage.

This high AC voltage generated across terminals 106 and 107 * is rectified and multiplied by a voltage multiplier fl circuit 84 formed of three diodes 85-87 and three capacitors 88-90. The cathode of the diode 87 is connected to an ultra-socket U, at which socket an ultra-acceleration potential of, for example, +27 kV DC voltage is obtained. An intermediate high DC voltage generated at the cathode 85 of the diode can serve as the focusing voltage for the focusing electrode of a cathode ray tube in a television receiver.

Since the ferroresonance transformer 65 provides both a high voltage in the high voltage secondary winding 65c and a low voltage in the secondary winding 65b, both the ultor acceleration potential and the E + scan supply voltage will be regulated without the need to resort to relatively complicated electronic control circuitry. To regulate the voltages across the secondary windings 65b and 65c, a resonant capacitor 91 can be connected to the low voltage secondary winding 65b at the terminal 101 in the manner shown in Fig. 1 or to some other winding wound around the thin, disc-shaped core portion 165b.

The capacitance value of the capacitor 91 is selected so that the capacitor 91 and the low voltage secondary winding. 65b comes in resonance close to the frequency of the excitation source, ie. close to the high frequency AC voltage 64 frequency of 15.75 kHz. Under certain conditions, the capacitor 91 may be omitted to reduce the cost, as will be described below. If there is sufficient winding capacitance in the high-voltage secondary winding, no additional capacitor is needed. The circulating resonant current flowing in the winding 65b and the capacitor 91 help to magnetically saturate the core below both the low voltage winding 65b and the high voltage winding 65c during each half period of the oscillation of the circulating current. By measuring the core in this way, the induced voltage in both secondary windings 65b and 65c will be regulated.

As shown in Fig. 5a, the voltage V65a, which occurs across the primary winding 65a of the high frequency ferrous resonance transformer 65, is a symmetrical square wave voltage with the frequency, 75 kHz and with the period TH = 65.5 microseconds. The high frequency voltage across the secondary winding 65c is also a relatively symmetrical square wave voltage V65c, as shown in Fig. 5b. The square wave generated by the low voltage winding 65b across the receptacles 101 and 102 is shown in Fig. 5c, the portion having a flat top appearing when the rectifier 401 is conducting.

The input current i65a to the primary winding 65a (from the socket 26) is shown in Fig. 5d and constitutes a relatively constant current, except near the switching moments of the transistors SÄ and 55, near the leading and trailing edges of the wave V65a in Fig. 5a.

The resonant current or the circulating current ic, which flows in the capacitor 91 and the low voltage secondary winding 65b between the terminals 101 and 102 and which is illustrated in Fig. Se, helps to magnetically saturate the core portion 115b under both the high voltage secondary winding 65c and the low voltage secondary winding 65c. Saturation occurs near the top portions 94 and 95 of the circulating current wave.

To enable saturation of the core portion l65b while the core portion l65a below the primary winding 65a remains unsaturated, the cross-sectional area of the disk l65b in Fig. 2 is made smaller than the cross-sectional area of the C-shaped core 165a. As can be seen from the sectioned view taken along line 5-5 of Fig. 2 in Fig. 3, the cross-sectional area a = (t) - (H), i.e. the product of the thickness t and the width w of the disc l65b, relatively much 447 527 less than the cross-sectional area A = I - f, i.e. the product of sides w and f of the C-shaped core l65a. For the values given below, the ratio a / A is equal to about 0.19.

It may be desirable to limit the temperature increase in the saturation core portion 165b, which is below the low voltage cross-winding 65b and the high-voltage cross-winding 65c, after the circuit of Fig. 1 has been magnetized. The saturation flow density Bsat of the core material decreases with increasing temperature. Since in a ferroresonance transformer the voltages generated across the secondary windings 65b and 65c are a function of Bsat, it may be desirable to limit the temperature rise by providing a secondary winding and core structure with increased cooling capacity. This temperature rise can occur as a result of increased core losses in high frequency operation and as a result of relatively large IQR losses on the basis of such factors as the relatively large non-circulating high frequency saturation current.

As shown in Figs. 2 and 3, the core portion l65b consists of a thin disk having the thickness t, the width w and the length 1.

With typical values given below, the ratio of surface area to volume of the disc l65b will be comparatively large, for example 40: 1, thereby enabling improved cooling of the thin disc.

Furthermore, the inner diameter D of the cylindrical coil mold 265b is also larger than the thickness t of the disk 165b, whereby the windings 65b and 65c can be wound loosely around the disk core portion 165b with a relatively large air space between the windings and the core. A core configuration of this type is described in U.S. Pat. No. 2,262,245.

If the increase in core temperature is of relatively small significance, a core with a more conventional shape can be used for the core portion l65b, such as a core with a square or circular cross-section, whereby the windings 65b and 65c can be more tightly wound around the core.

As shown in Fig. Sb, the high voltage V65c over the high voltage winding 65c is a relatively symmetrical 447 527. 11 square wave, the size of the positive portion 92 being approximately equal to the size of the negative portion 93. With a voltage swing between peak to peak of e.g. 18 kv of V65c only three rectifiers and two or three capacitors are needed in the high voltage multiplier circuit 84 according to Fig. 1 in order to e.g. shall receive an ultra voltage of +27 kV. Diodes 85 and 87 align the positive portions of the V65C, while diode 86 aligns the negative portion. The ultimate voltage is thus approximately twice as large as the positive magnitude of the V65c added to the negative magnitude. The capacitor 90 can be omitted if the current-conducting coatings on the picture tube provide a sufficient filtration capacity. Conventional high voltage sources that rectify positive sweep return pulses equal to the size of the positive portion 92 in Fig. 5b may need five or six diodes and associated capacitors in the multiplier circuit to obtain the same ultra voltage.

In the described high-frequency ferroresonant transformer system which constitutes an embodiment of the invention, a regulated B + scanning supply voltage and thus a regulated sweep voltage Vt is provided, and further a regulated high voltage is obtained. With an arrangement of this type, the design criteria for the horizontal bending circuit 73 can be significantly simplified. For example, if the sweep return transformer 68 no longer needs to transfer load energy to the electrode, the size of the sweep return transformer can be significantly reduced, since a relatively small load current flows in the sweep return transformer. If only small direct current flows through the horizontal output transistor 48, its requirements with respect to size, current and voltage as well as the ability to dissipate heat will be reduced to a high degree. If the circuit is redesigned in a suitable manner, a low impedance deflection winding 80 can be used, which winding in turn would require a much lower peak voltage of 200 volts as an example, instead of the relatively large horizontal sweep return pulse of 1000 volts now normally used .

With the high-frequency ferro-resonance transformer 65, which supplies both the B + scan supply voltage and the ultra-high voltage, relatively good image width stability can be obtained without the need to resort to electronic control circuits or individual series resistors. At the same time as the radiation load of the ultra-outlet U increases, the ultra-acceleration potential will decrease. The increased direct current potential of the load current flowing in the high voltage winding 65c demagnetizes the core portion l65b slightly and displaces the operating point of said core portion away from greater saturation slightly towards the knee of the transformer bra hysteresis loop, thereby reducing the high voltage slightly. However, since the low voltage winding 65b and the high voltage winding 65c both operate with a common saturation core portion, the increased load current flowing in the winding 65c will also reduce the B + scanning voltage, thereby obtaining a significant raster width control.

Explained in another way, it can be said here that the existing leakage inductances in the transformer 65 give an increased voltage drop with increased video beam load, whereby both the high voltage and the B + voltage are reduced. The leakage inductance between the high and low voltage secondary windings 65b resp. 65c and the degree and position of the core saturation are adjusted to achieve control of the grid width.

A relatively large number of winding turns is required to generate the comparatively high voltage across the high voltage secondary winding 65c. By appropriate selection of such factors as winding configuration, layer spacing and current conductor dimensions, the leakage capacitance or the distributed capacitance between the windings can be made large enough for the high voltage winding 65c to resonate and thereby saturate the core portion l65b, thereby adjusting of the voltages 1 both the high and low voltage secondary windings are obtained. This distributed resonant capacitance is illustrated in Fig. 1 by means of a capacitor 665 which is connected across the high voltage winding 65c, although in reality the total capacitance is distributed along the winding turns. Since the distributed capacitance 665 and the high voltage winding 65c provide the circulating current for saturating the core portion 165b, the capacitor 91 447 527 13 is no longer needed. This eliminates the risk of aging or faults that occur in individual components and which can lead to an increase in the high voltage. Furthermore, the use of a ferroresonance transformer to provide the high voltage provides a built-in high voltage protection capability, since changes in a winding inductance or in the capacitance value of a resonant capacitor usually result in a loss of ferrous resonance operation and a reduction in the winding voltage.

Since the ferroresonance transformer 65 gives rise to both the high voltage and the B + scan supply voltage, a relatively large increase in the core temperature after the initial secretion magnetization can be allowed without this having a significant effect on the raster width. In view of the fact that the high-voltage secondary winding 65c and the low-voltage winding 65b are wound around a common core portion l65b, a decrease in Bsat with an increase in the core temperature will mean that both the ultra-voltage and the B + voltage are reduced, whereby a significant degree of raster width control.

As examples of typical values of a high frequency ferroresonant transformer 65 of the type illustrated in Figs. 1-5 using the distributed capacitance 665 associated with the high voltage winding 65c, the following may be mentioned: The core 165: the cross-sectional area of the C-shaped core portion l65a = 2.52 rounds, the length of the outer leg = 5.1 cm, the length of the middle section = 7.1 cm; the thickness of the thin disc-shaped core portion l65b t = 2.79 mm, the width w = 15.5 mm, the length = 7.1 cm, the cross-sectional area = 45.2 mmz. The core material is a ferrite with Bsat = about 4000 gauss at 25 ° C, for example Ferroxcube5E2A from Ferroxcube Corporation, Saugerties, New York, USA, or RCA 540 from RCA Corporation, Indianapolis, Indiana, USA.

Primary winding 65a: 30/40 nylon-insulated lacquer copper-colored wire layer-wound with four layers, provided with center socket, bifilar-wound, 200 winding turns in total, no insulating layers between the winding layers, winding length = 5.94 cm.

Low voltage winding 65b: cylindrical coil shape 265b with inner diameter D = 18.2 mm, outer diameter 21.6 mm, and length 42.55 mm. The winding 65b is made of 25/38 nylon-insulated 447 527 e 14 rows of lacquer copper wire, bifilar, layer-wound, with 190 windings »a total of four layers comprising about 48 turns in each layer.

A fifth shield comprising four winding turns gives a filament voltage of about 6.3 volts and 900 milliamps for the cathode ray tube. Winding length = 42.55 mm.

High-voltage winding 65c: cylindrical coil shape 265c with a diameter of 29.21 mm, a ring thickness of 1.52 mm and a length of 26.67 mm. Lindníngen 65c: lacquer copper wire 0.100? mm, part-wound with 52 layers with 147 winding turns in the first 51 layers and 43 winding turns in the last layer, the layers being separated by means of a mylar insulator with a thickness between 0.05 mm and 0.10 mm. Total number of winding turns = 4600, winding length = 19 mm.

Claims (16)

447,527 Patent claims A ferro-resonance energy source for a deflection and high voltage circuit in a television receiver comprising an AC voltage source, characterized by a ferro-resonance transformer (65) comprising a magnetic core (165), a first winding (65a) connected to said alternating voltage source (37, 38), a high voltage winding (65c) wound around a core portion (165) of said magnetic core (165) and connected to a high voltage terminal (106) for generating a high voltage (V65c), a second winding (65b) wound around a core portion (l65b) of said magnetic core (165) and coupled to a scanning supply voltage socket (501) for generating a scanning supply voltage, and means (91 or 665) for providing a sufficient capacitance of at least one winding (65b or 65c) of said ferroresonance transformer (65) for generating circulating currents to saturate the core portions during said high voltage winding and said second winding in each period of said alternating voltage to provide a regulated high voltage and a regulated scanning supply voltage, further by a deflection winding (80), a deflection switch (78) connected to said deflection winding for generating sweep and sweep return intervals in each deflection deflection period (80). coupled source (81) for sweep voltage for generating scan current in said deflection winding, first means (68a) for coupling said regulated scan supply voltage (at 501) to said sweep voltage source, an ultro terminal (U) for providing an ultra acceleration potential, and high voltage means (84) coupled to said high voltage terminal (106) and to said ultra-terminal (U) to generate said ultra-acceleration potential from said regulated high voltage. 2. An energy source according to claim 1, characterized in that said high voltage winding (65c) and said second winding (65b) together use a common saturation portion (l65b) of said core. 3. An energy source according to claim 2, characterized in that said high voltage winding (65c) is wound concentrically with said second winding (65b). An energy source according to any one of claims 1 to 3, characterized in that said means for providing a sufficient capacitance comprises the distributed capacitance (665) of said high voltage winding (65c). 5. An energy source according to any one of claims 1 - 3, characterized in that said means for providing a sufficient capacitance comprises a capacitor (91) connected to said second winding (65b). 6. An energy source according to any one of claims 1-5, characterized in that a sweep return transformer (68) is connected to said deflection winding (80) to produce sweep return pulses (67). A single-pin socket according to claim 6, wherein said first means comprises a first winding (68a) of said sweep return transformer (68). 8. An energy source according to any one of claims 1 to 7, characterized in that said high voltage winding (65c) is magnetically coupled to said second winding (65b) to a sufficient degree that a relatively constant raster width is maintained when the beam current load at the ultra outlet (U) varies. Energy source according to claim 8, characterized in that increased radiation exposure demagnetizes said core (165) in such a way that a reduction is obtained in said ultra-acceleration potential (27 kv) and scanning supply voltage (E +) to such an extent that a relatively constant raster width maintained. An energy source according to any one of claims 1 to 9, characterized in that said high voltage (V65c) comprises a relatively symmetrical, high alternating voltage. 11. ll. Energy source according to claim 1, characterized in that said high voltage means comprises multiplying means (84) for adding a first multiple of the magnitude of a first polarity of said high alternating voltage (vósc) to a second multiple of the magnitude of a second polarity of said high AC voltage in order to obtain said ultra-acceleration potential 447 527 17 (27 kV). 12. An energy source according to claim 11, characterized in that said multiplying means (8Ä) adds twice the magnitude of a first polarity of said high AC voltage. 13. (V65b) to the magnitude of a second polarity of said high AC voltage. An energy source according to any one of claims 1 to 12, characterized in that said AC voltage source comprises a square wall generator (37). leeward. Energy source according to any one of claims 1 - 15, characterized in that the frequency of said alternating voltage 14. (V65a) is equal to the horizontal deflection frequency. Energy source according to any one of claims 1 to 14, characterized in that said core portions (l65b), around which said high voltage winding (65c) and said second winding (65b) are wound, comprise a disc (165b) of magnetic material with a comparatively large ratio of surface area to volume to effect cooling of said disk. An energy source according to claim 15, characterized in that said core comprises a substantially rectangular core (l65a), that said disc (l65b) forms a leg of said core, that said first winding (65a) is wound around a non-rectangular core. saturating leg of said core and that said high voltage winding and second winding (65b) are loosely wound around said disk (l65b).