SE409148B - FILTER DEVICE INCLUDED IN THE RECEIVER OF A PULSE DOPPER RADAR - Google Patents

Description

71091194 '.2. the receiver mixed signal will thus contain a number of desired and undesired frequency components.

It is previously known to arrange filters (so-called Doppler filters) in single-line registers of pulse Doppler radar, the task of which is to suppress as much as possible the frequency components which originate from undesired targets, ie mainly those from land, sea and precipitation which are low-frequency. The Doppler filter can consist of a digital filter, which filters out the female components, the frequency of which is less than a certain value corresponding to a certain target speed. Such a Doppler filter has, within the frequency band determined by the period time T of the radar transmitter, a certain characteristic, which is now illustrated in the attached figure 1 with dashed lines. It is then desirable that for no frequencies, for example less than 1'8T the filter u shows smaller ~ English »while for higher frequency values ¶ PP ln I ic _ the filter_ exhibits bandpass character whereby any movable mills, whose radial velocity is greater than the cutter speeds However, the usefulness of the Doppler filter is limited by the size of its barrier band. For example, if the upper cut-off frequency of the filter cut-off band fmaxzl / BT and if the period time T downward of the radar pulses is limited by the desired range Rmax, T = 2Rmax / c, where c = the propagation speed, and the highest graffiti velocity within the filter cut-off band becomes vmax = Äc / lßllnnax radar wavelength. For example, for Ä: 'ldm (S-band) 1 and Rmax = 10.10'm, rmaxz becomes 2nMs, which means that only nominal gradients can be suppressed by the filter, while remaining graffiti with higher frequency components is affected.

In the event that the radar operates in low PRF mode, ie the period time T is adjusted so that all radar echoes of interest are reflected and received before the next radar pulse is transmitted, this means that the target's Doppler frequency fd may be greater than the pulse repetition frequency 1 / T. This in turn means, as can be seen from Figure 1, that the target echo can also be suppressed by the Doppler filter for certain so-called blind speeds, more specifically for the speeds which give a Doppler frequency which constitutes a minimum of the frequency 1 / T. It is known to avoid suppression of such target cones by introducing staggering, i.e. the period time T is caused to vary from one pulse to a next of the transmitted radar pulses.

Another known method of eliminating the unwanted graffiti spectrum is to perform rate compensation pre-filtering in the Doppler filter. This means that the graffiti speed is estimated, for example by phase measurement, during the successive sweep. By, for example, controlling the receiver oscillator's local oscillator, the graffiti spectrum can be shifted so that its dominant component assumes the value zero and is thus located within the filter barrier band. However, the method presupposes that the graffiti spectrum has a dominant component that is correctly estimated. 7709119-7 I __ _ 3.

The present invention is based on this previously known method, but has the further advantage that, since the estimation of the speed of the moving graffiti is carried out after filtering the ground graffiti, this graffiti will not affect the estimation. The object of the present invention is to provide a Doppler filter device included in the receiver of a pen Doppler radar by means of which device filtering of both low frequency ground and sea graffiti and filtering of remaining graffiti with higher Doppler frequency can be performed using digital filters of i and known kind.

The invention, the features of which appear from the appended claims, will be described in more detail with reference to the accompanying drawings.

Figure 1 shows a frequency diagram, which illustrates both the frequency spectrum of a received radar signal and a certain selected filter characteristic.

Figure 2 shows for explanatory purposes a block diagram of certain units included in a radar receiver, which precede the Doppler filter device according to the invention.

Figure 5 shows in the form of a block diagram the principle of the Doppler filter device according to the invention. Figure 1 shows in more detail the appearance of a digital filter known per se included in the device according to Figure 3.

Figure 5 shows a block diagram of an embodiment of the Doppler filter device according to the invention.

The frequency diagram according to Figure 1 shows a graffiti spectrum, as well as the spectrum of an incoming target echo at a certain distance from the radar. The filter characteristics of a digital filter included in the Doppler filter device are indicated by dashed lines and have barrier bands for low frequencies, for example for frequencies <1 / 8T for frequencies between T / BT and 1 / T and in between a passband.

The filter characteristic is then periodic with a period 1 / T. The spectrum of the desired measure is denoted s and the moving scale how a dominant spectrum sm whose average frequency is denoted fm. Doppler filter trebe, the structure of which will be explained in more detail in connection with Figures Ä and 5, has the task of suppressing both the ground graffiti spectrum sg and the spectral sm of the dominant moving graffiti originating mainly from precipitation (rain and snow).

In order to better elucidate the signal processing and the structure of the filter device according to the invention, it will first be described in more detail in connection with Figure 2 that the filter axxordixillgexx is further described. A signal Mt) = = cos [21 '[(fo + i'd) t + q): | : from the radar receiver's SlI-våixlurnn Frequency-vxl fd lltgiíz- Idoppler frequency for the desired target. The signal A (t) is applied to two channels, I and Q, which each contain a mixer B1 and H2, respectively, and a digital converter AD1 and AD2, respectively. To each mixer B1, B2 is supplied a reference signal cos 2-TEfot respelrtive its Zlïfot from a reference oscillator OSC 'in the receiver. Thereby the output signals cos (2'11'fd + k [>) and sin (2 "¶fd + kP) are obtained, which are each applied to analog-digga'talomvaxxdlarex1 AD1, AD2. In these transducers, the signals at the sampling time points tu are sampled by clock pulses from a clock circuit CL so that the output signals XI = cos (2TIïfdtn + \ P) _ XQ = Sí11 (2 "f <1l711 + q7) in the respective channels I and Q are obtained.

The signals XI fl- n) and Xq fl zn) he are both represented by the signal xun) = cje fl ffarn, xp = o 'Sonxplingsticllnxznl: terna can fall in so that regular sannplírng ntförzes, ie tn = nT (n = 1, 2, 3, .... ) or that the time between succc-ssíva szunplixxgs pulses varies within a certain time interval NT,: nen -salnma szunplingsxnxöxlstei 'reappears after the sampling time point in = NT, so-called stuggering: In the latter case, sampling takes place at the times V NT + tk, where 'V = 0.1 ... and k = 0, 1, 2, .., N-1. The principle of the Doppler filter device according to the invention is apparent from the block filter according to Figure 3. The filter device contains a first digital filter DI-'l of a type known per se; ] ämpligen_ett-irnnšversnlfílier, which dímensiunernis so that it elixniln-rzlr- mark- oeh sea graffiti, ie graffiti: down low: speed, jäsnföi- -; filter charactercristilcelz elllištl figure 1. l-Iftersolll filterkaral: The leristics of a digi »i spoken filter is periodic with a period equal to the inverse value of the sampling frequency, the blocking band returns at frequencies corresponding to . In the case that the sampling freedom varies (st-uggerizng), the characteristic of the filter DI- “J is irregular and a certain position on its barrier band can not be determined except for slightly low frequencies corresponding to the dark and sea graffiti. Thus, in this latter case, the filter DI- “l laan is generally not dimensioned so that lilotter-llxecl mveket- low llast (lnark and lake) and graffiti with higher speed can be filtered out at the same time. Filirei's input signal is b-eteclfnad x1 (tn) and its output signal yo fl n).

To the output of the filter DFI is connected lml-oclíet lllx ', - whose xxpplrvggnzxd: lširnmrc shall be described in connection with figure §. The block HK performs an estimate of the graffiti remaining after the filtering in the first filter DI-'i and performs a velocity coefficient of average frequency fm of the donating graffiti filter. 7709119-7 5.

This compensation means that the total frequency component in the input signal y1 (tn) is moved in frequency so that the remaining moving ball root falls below a certain cut-off frequency, for example below the value 1 / ST in the diagram according to Figure 1. The subsequent digital filter DF2 connected to block HK is dimensioned according to the same principle as the first filter Idö, which is dimensioned so that its barrier band coincides with the graffiti whose frequencies have a low value (land and sea graffiti). As a result, the dimensioning problem for the second filter DF2, as staggering is used, has been transformed into the relatively simple dimensioning problem that applies to the first filter DF1. The filter DF2 thus eliminates the remaining moving graffiti (precipitation) and the only condition is that the remaining graffiti has a predominant spectrum whose average frequency fm can be estimated in the block HK.

Each filter DFI, DF2 consists of a digital filter of a per se known appearance shown in Figure k. The filter according to Figure 4 contains a number of delay circuits, for example three circuits DL1 - DL3, each with the delay T = the period time of the radar pulses. The output of each delay circuit is connected to a multiplier MUO - NU3 with the coefficients L0 (n), L1 (n), L2 (n) and L3 (n) for the filter DF1 and with the coefficients K0 (n), K1 (n), K2 (n), K3 (n) for the filter DF2, where the index (n) indicates that the coefficient values may vary for different sampling times. All multiplier outputs are connected to an adder circuit ADD. In the following, only the case of staggering is considered as the time interval between two suessive sampling pulses varies in accordance with what has been mentioned above. The case of regular sampling is a special one when tn = nl.

The speed compensation ee of the output signal yi (tn) from the filter DF1 proposed according to the inventive idea will first be described in terms of signal, after which a suitable embodiment of the block HK and the subsequent filter DP2 (figure Z) will be described in more detail in connection with figure 5.

During staggering, the time varies between suocessive sampling pulses, but variations are periodic with the period NT, which means that the filter characteristics of the filters DF1, DF2 are repeated after the time now NT, 2NT ..... If inputs to the filter DF1 ej2Trfd (vNT + tn), the output of the filter DFI becomes l: is x (tn) = r '1-. 1 Lí (n) eJ21tfD (Väl + tn_. Y1 (tn) = 23 1 = 0: g Lime -jzrrfnçflr-tnd) e je fi ffbbrx + u) T i = 0 Cn (ID) e i filtret DF1. j2TffD (VN + n) T, where rl = number of delay circuits II 17o9119-7). 6_ Thus applies in this. case that mnplitxxd and fm: in uitsglxznlenz yl fl zn), represented by the factor (In fl d) are time-dependent.

The velocity coexistence11 means that 5-1 (lividcfrzxs with the signal Cn (i'nl) efjgïrfxnxßyïlldlyr where fm is the result of a measurement of the non-conductivity frequency fd after filter DFI. F ) (vr: + n) n 'x “%) =% The output from filter DF2 is given by K (n) Cn-1 (few) e -JIQ-nzud _ finn)' 1 'T <> r fi 3: 2 tu = Z 1 1 = 0 Cn-l (få) _-; 2rr (fd - fm) (vx + n) a '. e where r2 = the number of delay curves in the filter DF2.

From the l fl rbrjfclcen for yIÜ-n), XQU-n) and y fifl u) frunxgår ntí 'a) about dopplc-všrekvuxlsr-xl fdï-O, the signal c-lilllixlxrraxs in the proposed fillmc-L DFI, ty (2n (|' d) z! hihi) can be made = 0. 1 = 0 b) if (luppl.crfrelivcuxsen fd =, 4 O and correct- uppslaarizl-uíng: w this frolcx-c-lrs of the dominant k] ol.lorspeld21'aí- ulfiírls in lsloclwt HK ie fm x fd, becomes: ínsiggnxanlølx to the filter lu-'e xgüu) = 05mm "'ÜÜÜ"' * “W signal and which can be fi filtered in: the filter Dl-'Q on the sanuna sill-t annu signal ejgmltn awayfiltreraaes 1 film Dm for m wo. , villu-'L represenlerax 'cn- lägf1'ck \' <- nt For the calculation of the filter coefficients 1.161) and K (n) the following requirements are set for _ l the output signals yl fl u) and y, ¿(tn): y1 (tn) * '10 when the insignia consists of markkl'oí.ter, ie fD "r0. In particular it is required that yl (tu) = Oíör fn = MK, K = 1 ,. ... rr AfK is selected within marlclzlottcrspclctmms: Irekvensområde. This requirement can be met by selecting cn (lAfK) = o K = 1, ...... r1 í n = 1, ...... N avs filter 1 eliminates the ground line. lalvafiolleu en (Ark) = o leads to the following equation system for calculating filter coefficients.

L. n) n) -j2TEÅIK (nT - tn_) lr 1 (i 21 :. G = 0 1 1 = 0 »1709119-7 .'7- If ÅfK is chosen symmetrically liring frelcvensen 0 leads the above connection to real time-dependent kocfi ' ie.ie11tv. \ 1 'Li n. y2 (tn)' Å-'O when ínsignalerx consists of a moving globe with the frequency fd. lšlottrelzs mcdelfrelnrens iir measured to finn; fd. It is especially required that yèftn) = O for: FD - fm = SfK, K = 1, ... r2. GfK is chosen so that fm + SfK ends up within the frequency range of the moving graffiti. This leads to equabiooxassysiexnel: r - n 2 I, (n) CM (fm + SfK) e -Jevr. gfK. 1. 1 e o 1 <= 1, ... 1-2 1,: o Cn-l fm) (n) for determination of the filter coefficient Kl.

Figure 5 shows a hlocksclxenxn over a ull'öríngs.l'nrn1 of the block. IIK according to Figure 3 to ästudliolunus. luxstighetskmnpensering 'Lillsnnunauxls med de laädo filters lll-l, lll-ll.

From the expression for x2 (tn) (rnligb above it appears! Koxnpexxseringaeli signalxniissigt nl ~ is passed through a divide 'output signal yl fl n) from the filter DFI nzed the factor Cn (f | n) ej2 fm (vwnylu where fm represents * an estimated value: Lv average rela -nsexl of the donlineraxxzle spectrum sd of the moving lilol filter From the filter lll-'1 a signal Re {y1 ('tn)} is obtained over the l-l channel and a signal I111 {y'1 (t11)} over (i-kaarzxlell Connected to each channel I and Q is a phase meter FK for networking the phase difference Atpxnellai). Two successive and filtered samples are to be performed in a known manner by first measuring both sample epoxlexes of the sample value in the II and Q channels. a value of fasx-'inlccln LP! relative to a certain reference is obtained.

Then, in the same way, the phase angle * P2 is read for: read sample value and the difference Åq) = \ p2 -Lpl is formed. For each stagger sequence tn, the sample values are obtained, which give a sequence of phase differences llxpn nxellan two consecutive and filtered samples yl fl n). This sequence ALpn is supplied to an accumulator S for by the nxottaglm. fnsslcillxmads- values. Å fl pn Imdei 'time period NP, corresponded to a complete staggering sequence. The accumulator is in itself lišíml and can consist of, for example, a feedback solar unit.

With M1 and M2, two memory units are denoted, for example, the PROll-nu prog nnen (progrannnabl e- read-only-nlemory fl The memory M1 consists of a matrix, in which the values of the coefficients Cn (fun) are recorded for different values of the phase difference Ål? And for the different values For each value pair fn, Acpfíu 'is thus a certain value of the coefficients C11 (fn) since fm is calculated when the value of / llP = 2 “- fm where' ll is liåincl. llínnese11heter1 M2 consists of a matrix ti form of ell 'Pllflll in which sine and eos; inu. ~' = ~ viirdenn for different vinlslnr * P åix- listed, vlllm angles are obtained from uekulnulzxiion H. lli | 1nesex1lnrl = se11 M1 how (only a nlgiivugg iivor which lširdel. 1 / (ï1x (1'n |) | upplrâimler alu-n two entrances ilver which .1'.11g'z "| 111: s \ 'ii1-del- Åtp respectively klockpnlsrerna cl upptrâicler, de latter vid sznupel'cíclpunlrtcrna in innan hver 'interval 9N'l'. llínnescnlxetcn bl !! hur. 77091 19 "? .8. an I- and a Q-utoäno over which values - sin res ective cos naxt raids where s s P Il Q is the accumulated phase. A multiplier number HU is connected to the two outputs of the unit M2 and to the output of the unit M1 for multiplying the factor 1 / ICn (fm) [by the sine and cosine values of the accumulated phase Q, respectively.

Thus, at the I and Q outputs of the multiplier HU, the two components -Si fl * v m v are obtained. .. _ ... _ och, v: lku m 'erforderliga for at bilda det lwmlælexu lCnliml {[Cu ñnl 1 e -j2¶ïmCVNT + fn).

C11 (1'1n) value According to the signal description above, the factor Cn (fd) - e j21rfd (vNT + t “) must be multiplied by the factor --3--. e-Jgïtfm (0Ä1 + tn) Cn (fm) for speed coupling.

The complex multiplier MM connected to the output of the filter DF1 and the multiplier MU performs this complex multiplication in a manner known per se, since the I and Q components of the complex factors are available as signal values over each channel. The corresponding signal component of 'Cnüa) e j21t (f <1 - fm) (v. \' R + tn) is thus obtained over the I and Q outputs from the multiplier MK. x (tn) =. 2 cnrfnd The filter DF2 consists of a digital transverse filter Dïh of such construction as shown in figure Ä. In order to achieve good graffiti suppression within the entire speed range, it is generally necessary to select different filter coefficients Kí n) for different uPPhätta frequencies fm. The filter coefficients Ki n are then determined from the relationship above. Therefore, a memory MF is connected to the multipliers included in the filter DPh, for example in the form of a FROM in which the coefficients Kl n for each value of Amp and for each time in are fixedly written in matrix form. The memory MP is therefore connected with its bed controllers partly to the output of the phase measuring circuit FK over which the value of ÅKP appears, partly to the clock pulse generator (not shown), which generates the clock pulses c1 in step with the staggered sequence tn (within each input VN "). coefficient telma Kl n depending on'Ŷ) and tn is output to the multiplier now included in the filter DFÅ and over the filters' outputs I and Q the quadrature components of the desired filtered signal y2 (tn) are obtained.

The filters DFI, DF2 are, as mentioned above, constructed as it appears from figure h. However, this figure does not end! the structure of a knnnI, e.g. the I-knnnlen and the multipliers MUO - NU2 multiply the signal components of X1 (tn), x2 (tn) in this kann] by the corresponding components of the coefficients hi n, K] H. Corresponding filter circuits are present in the second channel Q and at the complex multiplication in the multipliers MUO - MU2 the values in the I and Q channels are mixed. The filters DF1, DF2 have

Claims (3)

1. 77091 19-? 9 for a given order (determined by the number of delay circuits DL1 - DL3) a given passband, the width of which can be extended in a known manner by selecting higher order filters. CLAIMS 1 Filter device included in the receiver of a pulse Doppler radar for reducing unwanted graffiti signals within a certain lower and a certain higher speed range of a received target echo, which is a response to radar pulses transmitted from the radar with irregular pulse repetition frequency ("staggering"). first and a second digital filter, of which the cut-off frequency of the first filter between passband and latch is selected so that the graffiti signals in the lower velocity range fall within the latch band of the first filter but so that the desired measurement co-signal falls within its passband, k ä nne - drawn by a circuit device (HK) connected to the output of the first filter for estimating a dominant frequency component (fm) of said graffiti signals within the higher speed range and for performing a frequency shift of those from the first filter (DF1) filtered the graffiti signals so that the average frequency of graffiti the signals within the higher speed range assume a value lower than the value before said movement, the output of the circuit arrangement being connected to the input of the second digital filter (DF2), the latch band substantially coinciding with the latch band at low frequencies of the first filter (DFI) for suppressing the aforementioned movement within the higher speed range of the graffiti signals. Filter device according to claim 1, characterized in that said circuit device (HK) contains a phase measuring circuit (FK) for determining, within each sequence of transmitted radar pulses, a sequence of phase differences (ßón) between two successive and filtered filters in the first filter, a first memory unit (M1) for forming the inverted value of the coefficients (Cn) equal to the sample values of the output signal (yl (tn)) obtained from the first filter corresponding to a certain phase difference, a multiplier circuit (M) connected between the first and second filters (DFI, DF2) and to said memory unit (M1) for multiplying the output signal by said inverted value and a second memory unit (MF) connected to said phase measuring circuit and to the second digital filter for each measured phase difference and for each sampling time (tn) within a stagger sequence store associated coefficient values (K1 (“)) for the second filter. 7709119-7 '10. Filter device according to claim 1-2, characterized in that said multiplier circuit (MK) consists of a complex multiplier with two input pairs, each of which corresponds to the I and Q channels of the radar receiver, the first input pair being connected to the output of the first filter (DFI), that an accumulator (S) is connected to the output of the phase measuring circuit (FK) to form an average value ($) of said sequence of phase differences (Adn), a third memory unit (M2) being arranged to form the sine and cosine values of the mean value, that an additional multiplier (MU) is connected to said first memory unit '(M1) and to said third memory unit (M2) to multiply said sine and cosine values by said inverted value, each multiplying values over the corresponding I-ocšPQ channel, the second input pair of the complex multiplier is applied. PROMISED PUBLICATIONS: US 4,035,799 (343-7.7)