CH632850A5 - PULSE DOPPLER RADAR FILTER ARRANGEMENT. - Google Patents

Description

The object of the invention is to obtain a filter arrangement in the Emp-Q. The signals X] (tn) and XQ (tn) can each create a pulse Doppler radar, with which one can be represented by the signal X (tn) = e> 2 nfdtn, filtering out the low ground and sea clutter 0 = 0.

frequencies and a filtering out of the other interference spots 40. The sampling torques can be selected such that a regular sampling known at a higher Doppler frequency is carried out by means of digital filters, i.e. tn = nT (n = 1, interpretation can take place. 2,3, ...), or such that the time between successive

This object is changed by a filter arrangement, as defined in the sampling pulses within a certain time-characterizing part of patent claim 1, if NT, but the same sampling pattern is solved after. 45 sampling moment tn = NT reappears, which is the

The invention is thus based on the so-called “staggering” mentioned above. In the last-mentioned known method, however, shows the additional advantage that the case applies that the sampling at the times yNt + tk is because the calculation of the moving interference spots is carried out according to that in which y = 0.1, ... and k = 0,1,2, ..., N -1.

The ground clutter is filtered out, these clutter The principle of the Doppler filter arrangement according to the invention

do not affect the calculation. 50 voltage is shown in FIG. 3. The filter arrangement contains a

Further features and expediencies of the invention first digital filter DFi of a type known per se, expediently result from the description of exemplary embodiments, a transversal filter which is dimensioned such that it is based on the figures. From the figures show: - the echoes of soil and sea clutter eliminated, ie

1 shows a frequency diagram in which, on the one hand, the Fre interference spots at low speed or the filter frequency spectrum of a received radar signal and other characteristics according to FIG. 1. Since the filter characteristic of a digital side has a certain selected filter or sieve characteristic. filters are shown periodically with a period equal to the inverted stik; Value of the sampling frequency, it reappears periodically,

Fig. 2 is a block diagram for explanatory purposes when a regular scan is applied. In the case of certain units which contain a changing scanning frequency (staggering) in a radar receiver, the characteristic is, these units of the Doppler filter 60 according to the invention being irregular, and no specific filter arrangement can precede them; Location of its stop band except for very low frequencies,

3 shows the principle of a Doppler filter according to the invention which corresponds to the soil and sea clutter, indicated in the form of a block diagram; will. In the latter case, the DFi filter cannot

Fig. 4 details of a digital filter of known design, my dimensioned so that clutter with very low-that is included in the arrangement of FIG. 3, and 65 ger speed (ground and sea) and clutter with

5 shows a block diagram of an embodiment of the higher speed which can be simultaneously eliminated according to the Doppler filter arrangement. nen. The input signal of the filter is denoted by xi (tn) and its off in the frequency diagram shown in FIG. 1 is an interference signal denoted by yi (tn).

632 850

The block HK is connected to the output of the filter DFi, the structure of which is explained in more detail with reference to FIG. 5. The block HK carries out a calculation of the interference signals which remain in the first filter DFi after the filtering or screening process and carries out a speed compensation of the main frequency fm of the dominating interference spectrum. This compensation includes that all frequency components of the incoming signal yi (tn) are changed in frequency so that the remaining moving interference spots fall below a certain frequency limit, for example below the value 1/8 T in the diagram according to FIG. 1. The subsequent digital filter DF2, which is connected downstream of the block HK, is dimensioned according to the same principle as the first filter DFi, which is dimensioned in such a way that its stop band coincides with the interference spots, the frequencies of which have a low value (ground and sea interference spots). As a result, the dimensioning problem for the second filter DF2 is converted into the relatively simple dimensioning problem that applies to the first filter DFi by using the “staggering”. The filter DF2 thus eliminates the remaining moving clutter (precipitation), and the only assumption is that the remaining clutter signals have a dominant spectrum, the center frequency fm of which can be calculated in block H K.

Each filter DFi, DF2 consists of a digital filter known per se, the design of which is shown in FIG. 4. The filter of Figure 4 contains a number of delay circuits, at 10

for example three circuits DL1-DL3, each with a delay T equal to the period of the radar pulses. The output signal of each delay circuit is applied to a multiplier MU0-MU3, with the coefficients Lo (n), Li (n) and Lî (n) for the filter DFi and with the coefficients Ko (n), Ki (n), K2 (n), K3 (n) for the filter DF2, the index (n) indicating that the value of the coefficients can change for the different sampling moments tn. The output signals of all multipliers are fed to an adder circuit ADD. In the following, only the staggered version is considered, where the time interval between successive scanning pulses changes in accordance with the above information. The case with regular sampling is a special case in which t "= n.T.

1 s The speed compensation of the output signal yi (tn) from the filter DFi proposed according to the invention is first described with regard to the signals and then a suitable embodiment of the block HK and the subsequent filter DF2 (FIG. 3) is explained in more detail with reference to FIG. 20 explained.

In the staggering, the time changes between successive sampling pulses, but the change is periodic with the period NT, which means that the characteristics of the filters DFi, DF2 are repeated after the times 25 NT, 2NT, ... If the input signal of the filter DFi has the value x (tn) = e-i2 nfd (vNT + w), the output signal of the filter DFi is:

yl (t) = r Lt <n> e j 2Ttf d (-UNT + =

n i = 0

= Z Li (n) e-j2nfD (nT-tn-i) ej2nfD <uN + n> T i = 0

= Cn (fD) ej21IfD ^ N + n> T '

where n is the number of delay circuits in the filter 40 that the signal yi (t ") is divided by the signal C" (fm) ei2 DFi. nfm (vN + n) T, where fm is the result of a measurement of the mean

In this case, the amplitude and the phase of the sequence of the dominant interference spectrum behind the output signal yi (t "), represented by the factor Cn (fd), is filter DFi. So the following applies:

are time-dependent. The speed compensation includes

x, (t) = 2. n

-n

(fd)

ej2JI (fd - fm) (VN + n) T

The output signal of the filter DF2 is given by r2

y2 (tJ = 1 Ki • '

An) cn-l (fd} e-j2n (fd ~ fm). I

T

i = 0 Cn-1 (f m ')

j2ü (fd - fm) (vN + n) T,

where r2 is the number of delay circuits in filter DF2 that:

is. a) if the Doppler frequency fd »0, the signal in the first filter

The expressions for y i (tn), X2 (tn) and y2 (tn) show that DFi can be eliminated because

Cn (fd)

E

i = 0

Li ^ ^ - 0 can be made

632850

b) if the Doppler frequency fd # 0 and a correct calculation of this frequency of the dominant clutter spectrum was carried out in block HK, i.e. fm = »fd, the input signal of the filter DF2 has the value x2 (tn) = e> 2 n (fd ~ fm) (vN H

n) T

yi (tn) »0 if the incoming signal consists of interference signals, i.e. fD »0. Specifically, it is required that yi (tn) = 0, because fD = AfK, K = 1, ... n. AfK is chosen within the frequency range of the soil interference spectrum. This 5 requirement can be met by making the following choices:

which represents a low frequency signal that can be eliminated in filter DF2 in the same way as signal eJ'2 nfdtn that was eliminated in filter DFi for fd 0.

To calculate the filter coefficients Lj (n) and K; (n), the following requirements are placed on the output signals yi (tn) and y2 (tn):

Cn (AfK) = 0 K = i, ... ri, n = i, ... N

i.e. Filter DFi eliminates the floor clutter. The equation C „(AfK) = 0 leads to the following system of equations for the calculation of the filter coefficients Lj (n):

rl l

i = 0

L. 1

(n) e-j2ïïAfK

(nT

tn-i) _

= 0

If AfK is chosen symmetrically around the frequency 0, the interfering spots telfrequency was measured to fm ~ fd. In particular, the relationships given above lead to real and special demands are made that y2 (tn) = 0 for fD - fm = SfK within time-dependent coefficients L; (n). the frequency range of the moving noise spots. These

The signal y2 (tn) »0 if the input signal results from condition signals the following system of equations consists of moving noise spots with the frequency fd. The 25

T2 / \ C. (f + <5f

^ T,. (n) n-l "m K)

z ki • c — m—

l-O n-i 'm e ~ j211. ôfK

T = 0 K = i,. . . r 2

for the determination of the filter coefficients K | (n).

FIG. 5 shows a block diagram of an embodiment of the block HK shown in FIG. 3 for obtaining speed compensation, together with the two filters DFi, DF2. According to the above statements, the expression for X2 (tn) shows that the compensation with respect to the signals is carried out by dividing the output signal yi (tn) from the filter DFi by the factor Cn (fm) e'2 nfm (vN + n) T where fm represents a calculated value of the center frequency of the dominant spectrum sd of the moving clutter. A signal Re {yi (tn) 1 on the 1-channel and a signal Im yi (tn) on the Q-channel are obtained from the filter DFi. A phase measuring circuit FK is connected to each channel I, Q in order to measure the phase difference A® between two successive filtered samples. This is carried out in a known manner by first measuring the two components of the sample value in the I and Q channels, a value of the phase angle ®i being obtained relative to a specific reference value. Then the phase angle ®2 for the next sample value is measured in the same way, and the difference A® = ®i - ®2 is formed. Sampling values are obtained for each staggering sequence t ", which result in a sequence of phase differences A®n between two successive sifted or filtered sampling values yi (tn). This sequence A® “is fed to an accumulator S for the received phase difference values A®n during the time period NT, which corresponds to a complete staggering sequence. The accumulator is known per se and can, for example, consist of a feedback summation device.

Mi and M2 are two storage units, for example PROMs (programmable read-only memories). The memory Mi consists of a matrix in which the values of the coefficients Cn (fm) for different values of the phase difference 40 and for different values of tn are written in the staggered sequence. A specific value of the coefficients Cn (fm) is thus obtained for each pair of values tn, A®, since fm is calculated from the value A®211T • fm, in which T is known.

45 The storage unit M2 consists of a matrix in the form of a PROM in which the sine and cosine values are listed for different angles 4>, these angles being obtained from the accumulator S. The storage unit Mi has only one output at which the value 1/1 Cn (fm) | appears, however, two inputs at which the input value A® and the clock pulses ci appear, the latter at the sampling times tn in each interval vNT.

The memory unit M2 has an I and Q output at which the values - Sinus ® and Kosinus ® appear, 55 ® being the accumulated phase. A multiplier MU is connected to the two outputs of unit M2 and to the output of unit Mi by a factor of 1/1 Cn (fm) | with the sine or multiply by the cosine values of the accumulated phase. Consequently, two components appear at the I or Q output of the multiplier MU 60

- Sine (j)! Cn (fm) I

respectively.

Cosine <{) j Cn (fm) I

which are required to form the complex value 1 / Cn (fm) e ~ '2 nfm <vNT + tn) 2U.

632 850 6

According to the description above applies to the multiplier MU connected, this complex mult'iplica

Signal handling that the factor Cn (fd) • eJ2 nfd (vNT + tn> to act out because the I and Q components of the complex facto-

Speed compensation with the factor 1 / Cn (fm) • e ~ -i2 ren are available as signal values on the respective channel. At I-nfm (vNT + tn) must be multiplied. The complex multiplication and Q output of the multiplier MK thus become the entrer MK, the one with the output of the filter DFi and with the multilingual signal components of x (tn) = C] 1 (£ d). ei2ïï (fd - fm) (vNT + tn)

received, namely the above.

The filter DF2 contains a digital transversal filter DF4 of the design shown and shown at the outputs I and Q of the filter shown in FIG. 4. The quadrature components of the desired filtered pressure within the entire speed range signals y2 (tn) around a good interference spot.

to achieve, it is generally necessary to form different filter coefficients. The filters DFi, DF2 are, as mentioned according to FIG. 4, efficient K, (n) for different measured frequencies fm. However, this figure shows only the interpretation for one choose. The filter coefficients Kj (n) are determined from the relationship given above, for example the I-channel, and the multiplier. With the multipliers contained in the filter DF4 20 MU0-MU2 multiply the signal components of xi (t "), a memory unit MF is connected and X2 (tn) in this channel with the corresponding components, for example in the form of the PROM and the coefficients of the coefficients L; (n) and K; (n). The corresponding filters Ki (n) for each value of AO and each time t "in circuits are contained in the other channel Q, and in the

Matrix form. The memory unit MF complex multiplication in the multipliers MU0-MU2 is for this with its two control inputs on the one hand with the 25, the values in the channels I and Q are mutually output of the phase measuring circuit FK, at which the value of is mixed. The filters DFi, DF2 appear for a given Ord-AO, and on the other hand connected to the clock pulse generator (determined by the number of delay circuits (not shown) which clock pulses ci to the right gen DL1-DL3) a given pass band, whose width time with which the stagger sequence tn (within each Inter- can be expanded in a known manner by using filters with valls vNT). The values of the coefficients Ki (n) chosen from high order 30.

AO and tn depend on the multipliers in the filter DF4

G

1 sheet of drawings

Claims (4)

632850 2 PATENT CLAIMS echo signals or echo signals within one 1. Filter arrangement in the receiver of a pulse Doppler-determined lower and higher speed range, ie to reduce unwanted interference echo signals which corresponds to the speed range of a received len within a certain lower and a higher target object echo, which corresponds to the response signal of the speed range which corresponds to the speed range Forms radar pulses, which corresponds to a received target echo from the radar with an irregular range, which pulse repetition frequency (“staggering”) is emitted — which forms the response signal of radar pulses from the radar, whereby the arrangement emits a first and a second digital film with an irregular pulse repetition frequency. ter contains and the frequency limit between the pass-det, the arrangement of a first and a second and the stop band of the first filter is selected so that the digital filter contains and the frequency limit between the io interference Spot echo signals within the lower speed pass band and the stop band of the first filter is selected to fall within the stop band of the first filter, but that the target echo signals within the lower speed fall within its pass band, the range of the desired target echo signal into the stop band of the first filter Filters fall, such “clutter echo signals” can be caused, for example, by the desired target echo signal in its passage through the ground, the sea or rain. band falls, characterized by a circuit arrangement 15 It is one of the tasks of a radar receiver to suppress radar (HK), which is connected to the output of the first filter (DFi) to suppress echoes or so-called “interference spots” with means for frequency conversion the interference echo signals transmitted by the first filter caused by reflections on irrelevant target objects (DFi) become higher, for example on the ground, at sea or through Speed range such that the center frequency of the precipitation (rain or snow), and only the desired water jam echo signal takes on a value that is lower than the moving target, for example an aircraft, before the value before the transfer, the output of which is detected . For this purpose the speed difference Circuit arrangement is connected to the input of the second digital filter of the undesired target objects with respect to the one or more (DF2), the blocking band mainly used desired target objects. If a coherent coincides with the stop band of the first filter (DFi) in the case of a pulse type Doppler radar of a known type, a pulsed high-low frequency is transmitted to suppress the clutter echo frequency signal with a specific carrier frequency. which after reflection are on a moving target with a range of motion. returns at a certain changed frequency fo ± fj, 2. Filter arrangement according to claim 1, characterized in that the change fd depends on the Doppler shift, net that the circuit arrangement is a phase measurement circuit, i.e. the radial speed of the moving target object rela- (FK) for determining a sequence of phase differences 30 tiv to the radar station. The incoming echo signal is in (Acl) n) between successive samples of the first of the receiver mixed with the carrier frequency fo, the filters (DFi) being obtained for each sequence of radar pulse Doppler frequency fd emitted. If the emitted sen, a first storage unit (Mi) to form the inverted signal (the carrier frequency fo) were not pulsed, then a pure value of coefficients (Cn), which receive the samples of the sinusoidal signal, the frequency of which is the Doppler frequency fd output signals (yi (tn)) of the first filter at a certain 35. Since the transmitted signal corresponds to a pulse frequency fp = th phase difference, a multiplier 1 / T, where T is the period time, is pulsed, the receiver (MK) inputs between the first and second filters (DFi, DF2) pulsed signal, which is sinus modulated, which is laid and connected to the memory unit (Mi), has a frequency to the modulation size that is equal to the multiplication of the output signal by the inverted value, Doppler frequency fj. Furthermore, the received signal contains and a second memory unit (MF) which contains the pha frequency components which originate from undesired target objects and are connected to the second digital filter, which leads to the fact that the received signal is not, for storing coefficient values (Ki (n)), which is modulated to be purely sinusoidal. The received and in the Emp-second filter (DF2) belong, for each measured catch mixed signal, consequently contains a number of phase differences and for each sampling moment (tn) within desired and undesired frequency components, a staggering sequence of those with irregular pulse 45 Es is already known, filters (so-called Doppler filters) in repetitive radar pulses. to provide a receiver for pulse Doppler radar devices, 3. Filter arrangement according to claim 2, characterized gekennzeich- wherein the task of these filters is, in a poss net, that the multiplier circuit (MK) consists of a complex, high extent, the frequency components to under-multiplier with two input pairs, of which press that originate from the undesired target objects, each pair of the I- or Q-channel of the radar receiver mainly contains the low frequency components, which speaks of the fact that the first input pair is caused by the exit of the ground, the sea and precipitation -The first filter (DFi) is connected to an accumulator (S). The Doppler filter can consist of a digital filter, the output of the phase measurement circuit (FK) to form one that eliminates the components whose frequencies are lower mean (<P) of this sequence of phase differences (A <Dn) than a certain value that of a particular target It is connected that a third storage unit is provided which corresponds to 55 speed. Such a Doppler filter shows the formation of the sine and cosine values of this mean value within the period time T of the radar transmitter and that a further multiplier (MU) with the first specific frequency band has a specific characteristic, Memory unit (Mi) is connected, for multiplying this the sine and cosine values shown in dashed lines in FIG. 1 of the attached drawing with the inverted value, which is shown. It is desirable for the filter for low multiplied values to have a band-stop characteristic for the second input pair of the complete frequencies, which is less than 1/8 T for the multiplier on the assigned channels I or Q, the filter for high fre - be fed. has a band filter characteristic, which leads to The result is that any moving target objects whose radial speed is greater than that of the "Störflek-65 ken" can be detected. The use of a Doppler filter is, however, limited by the size of the pass band. The invention relates to a filter arrangement in the receiver. If, for example, the upper frequency limit of a pulse Doppler radar to suppress unwanted filter stop band fmax == »1/8 Tact and the period time T 3,632,850 the radar pulses are limited downwards by the desired range of the spot frequency spectrum together with the spectrum of an Rmax, then T = 2 Rmax / c, where c is the propagation arriving target echoes at a certain distance velocity; the highest clutter speed shown by the radar station. The filter characteristic of a digital filter contained in the filter stop band is vmax = Xcl 16 Rmax, where X = radar wave of the Doppler filter arrangement. For example, if X = 1 dm (the S-band) and Rmax 5 are shown in broken lines and show a stop band for = 10 • 104 m on the one hand, vmax »2 m / s, which implies that low frequencies, for example for frequencies < 1/8 T only the ground clutter can be suppressed by the filter and on the other hand for frequencies between 7/8 T and 1 / T and can, while the other clutter with higher frequencies in between a pass band. The filter characteristic is unaffected by quenz components. then periodically with a period 1 / T. The spectrum of the If the radar is operated in the lower PRF mode, the desired target echo is denoted by s, i.e. the period T is adjusted in such a way that all the echo of interest of the moving interference spot has a dominant speculating radar echo before the transmission of the next radar spectrum sd, the center frequency fm of which is designated fd. The impulses are reflected and received, so it becomes Doppler filter, the construction or design of which implies that the Doppler frequency fd of the target object is larger than that shown in FIGS. 4 and 5 is explained in more detail, which can be as the pulse repetition frequency 1 / T. As can be seen from 15 tasks, on the one hand the ground clutter spectrum sg and Fig. 1, this means that the target - on the other hand - to suppress the spectrum sm of the dominating moving ject echo from the Doppler filter for so-called blind-speed clutter, which mainly suppresses baseness can be more precisely for such speed beats (rain or snow). doppler frequencies result in multiples for a better understanding of signal handling and frequency 1 / T. It is already known to prevent the suppressing of the filter arrangement according to the invention from effecting such target object echoes by introducing a so-called “staggering” of the units initially described on the basis of FIG. 2, that is to say the periods that precede the filter arrangement . At input A, the time T changes from a transmitted radar pulse to a signal A (t) = cos 12 n (fo + fd) t + Ol from the duplicate following. xer of the radar receiver. The frequency fd is the Doppler frequency Another known method for eliminating the sequence for the desired target body. The signal A (t) unwanted interference spectrum consists of one of the two channels I and Q, each of which is a mixer To perform speed compensation. The Bi or B2 is included, estimated together with an analog / interference speed, for example by digital converter ADi or AD2. The mixer Bi or B2 is subjected to phase measurement during successive sweeps - a reference signal cosine 2 nfot from a reference oscillator. For example, by controlling the local oscillator 30 OSC in the receiver. Then the off-of the receiver, the interference spectrum can be shifted so shifted signals cosine (2 nfd + <I>) and sine (2 FIfd + O) are obtained, that its dominant component has the value 0 which is then assigned to the analog / digital converter ADi or AD2 is accepted and consequently within the suppression band of the leads. The signals to the filters are located in these converters. In this method, however, it is assumed that sampling torques tn are sampled from a clock pulse by means of clock pulses — that the interference spectrum is a dominant component 35 device CL, so that the output signals X | = Has a cosine that can be easily calculated. (2 nfdtn + O) in channel I or Xq = sine (2 IIfdtn + O) in channel