RU2805384C1 - Method for beam control in active phased array antenna - Google Patents

Abstract

FIELD: radio engineering.

SUBSTANCE: invention can be used in measuring technology, radar and communications. The essence of the claimed solution lies in the fact that in the method of forming an APAA beam, which consists in the fact that in each m-th channel of the APAA, a linearly frequency modulated (chirp) signal is generated according to the formula

in this case, once for each channel, the control processor and processing of radio signals of the transceiver modules calculates the values of: the estimated carrier frequency of the emitted chirp signal calculated carrier frequency of the received chirp signal and calculated initial phase of the chirp signal of the m-th channel, while in the transmitting and receiving modes the calculated carrier frequencies of the m-th channel are determined by the expressions accordingly. In this case, the calculated initial phase of the chirp signal of the m-th channel in the transmission mode is determined by the expression where ∆f is the frequency deviation of the chirp signal, τ is the duration of the chirp signal, – time shift of the m-th channel signal, d – grating pitch, – carrier frequency of the channel taken as the reference one, for which m=0.

EFFECT: increase in the speed of frequency tuning and change the position of the APAA beam, reduced energy consumption and simplification of the APAA design while reducing the requirements for the computing power of the control system.

1 cl, 3 dwg

Description

Field of technology

The invention relates to radio engineering and can be used in measuring technology, radar and communications.

Prior Art

The development of radar systems and the improvement of their technical and economic characteristics follows the path of using high-precision digital beamforming circuits, which make it possible to control the phase distribution in the grating aperture in real time. A known broadband antenna with phase scanning for emitting a linearly frequency-modulated (chirp) pulsed radio frequency (RF) signal with the required scanning angles [1], containing many subarrays, within which the delay time error at maximum scanning angles is negligible, one of the subarrays is a reference, a set of controlled phase shifters, the outputs of which are connected to the inputs of the subarrays, and which have control inputs, an RF signal distribution circuit for supplying a chirp pulse to the phase shifters, a phase shifter control device that ensures the setting of a zero initial phase and then a linear increase in the phase during the duration of the pulse. The disadvantage of this method is the difficulty of ensuring a low error of the phase shifters [2].

Digital computational synthesizers (DCS) can be used for digital signal generation in channels and digital control of beam parameters in an active phased array antenna (APAA). An antenna array with digital beamforming is known [3] (US 5943010). The antenna array contains multiple DACs with digital-to-analog converters (DACs) that produce a transmitting signal, which are controlled by a preprocessor that sets phase, frequency and time delay values for each DAC. Each DDS is clocked from a common clock generator, which establishes phase relationships between all DDSs. The formation of signals generated by each DDS and the establishment of the required phase relationships between these signals are controlled by a common receive/transmit processor.

The disadvantage of this patent is the lack of algorithms for calculating the required delay times and phase corrections for each channel. The second disadvantage is the small frequency range of the generated signals, limited above by the maximum frequency of the DAC output signal.

A fully digital APAA with a large instantaneous bandwidth [4] (CN113253210A) is known, which is a single APAA that includes many subarrays, each of which contains multiple channels, each channel contains a transceiver component, a digital phase shifter, a frequency conversion component and a frequency shifter phase shifter , connected in series, in which a frequency shifter is used to calculate compensation parameters for the initial phases and frequencies of the required transmitted and received signals, performing signal correction and providing constant amplitude and phase mode for all signal propagation paths, and a method for controlling a fully digital APAA with a large instantaneous width band, which consists of calibrating the APAA in an anechoic chamber in receive and transmit modes, calculating compensation parameters for initial phases and frequencies, compensating for non-identity of devices and channels, performing these operations for various frequencies and scanning angles, and recording the obtained parameters in a memory device.

The disadvantage of this control method is the complexity of its application for modulated signals, since this method does not take into account the connection of the initial phase and frequency compensation parameters to be calculated, as well as the compensation parameters for the non-identity of devices included in the APAA channels, with the signal modulation parameters.

The second disadvantage of this method is the need to experimentally determine the parameters for compensation of initial phases and frequencies using specialized anechoic chambers in relation to the required emitted signals, which increases the cost and time of designing an APAA.

There is a known method for achieving ultra-wideband beam control with a real-time delay for active electronically scanned arrays [5] (US 9479232), in which the oscillator consists of a DDS, which uses a time delay component generating an intermediate frequency signal with a first time delay, which causes a delay of the first RF signal, in which the time delay is in accordance with the beam steering algorithm. In the next APAA channel, the DDS generates an intermediate frequency signal with a second time delay and an additional phase shift calculated based on the value of the first time delay, and so on along the APAA channels.

The disadvantage of this method is that the delay time resolution is limited by the value inverse to the DDS sampling frequency. Moreover, in DDS with parallel operation of digital blocks, the delay time resolution, that is, the minimum time for changing the frequency of the output oscillation of the DDS, increases in proportion to the number of parallel operating digital code converters [6]. Since the permissible delay time error depends on the field of practical application of APAA [5], the use of method [5] turns out to be limited. Moreover, the need to generate a time delay and phase shift when generating signals for each DDS increases the number of control signals for the DDS, which complicates the APAA circuit.

A wideband active phased array antenna system and a method for producing one or more beams using an array of antenna elements [7] (US 7345629) are known, which uses multiple signal or beam synthesizer assemblies to generate a plurality of elementary transmitted signals, each of which contains a phase shift and amplitude settings, using the frequency, frequency, amplitude and phase parameters responsible for controlling the signal without processing with a time delay. In this method, by controlling the phase shift of the output signal, the DDS plays the role of a phase shifter and time delay device by establishing the dependence of the frequency phase in the form

φ _c ( t )= f ( t )× t _d ,

where φ _c ( t ) is the instantaneous phase correction value, f ( t ) is the instantaneous frequency of the signal, t _d is the required time delay. The disadvantages of this method are: the need to adjust the phase φ _c ( t ) for each instant depending on the required time dependence of the frequency f ( t ); a small frequency range of the generated signals, limited above by the maximum frequency of the signal of the synthesized digital digital signal.

An APAA system with beam control using a virtual time delay is known [8] (US2002/0175859), containing many radiating elements, in which, to control the beam, a set of signals with a preset frequency shift is generated and these signals are subsequently supplied to the corresponding antenna elements. For this purpose, mixers are used, one of the inputs of which is supplied with a chirp signal from the main chirp signal shaper, and the others are supplied with signals from sources that provide a virtual time delay of both received and emitted signals; signal shapers that provide a time delay are used DDS and ADC that generate signals in the form of oscillations with frequencies that differ from each other by an amount that is a multiple of ΔF, where the ΔF value is proportional to the required signal delay.

The disadvantage of this method of controlling the APAA beam is the appearance in the signal spectrum of regular components generated by the DDS, with a frequency depending on ΔF and the DDS clock frequency. Thus, when scanning the APAA beam, the spectrum of the emitted signal will contain undesirable components that worsen the characteristics of the APAA.

The closest to the claimed method is a method for digitally forming the radiation pattern (RP) of an active phased array antenna when emitting and receiving a linearly frequency-modulated signal [9].

Figure 1 shows a block diagram that implements the prototype method.

The first disadvantage of this method is the increased requirements for the performance of the pattern formation processor, in which the input chirp signal is multiplied by complex coefficients that make it possible to compensate for changes in the phase of the chirp signal depending on the number of the digital PPM. And since the compensating coefficients are a function of time, the pattern formation processor must, when digitally forming a pattern, calculate the compensating coefficients for each s-th sample during the duration of the chirp signal formation, which significantly increases the number of computational operations.

The second disadvantage of this method is the need to digitally generate two quadrature components of the chirp signal, which also increases the performance requirements of the pattern generation processor, and the use of quadrature modulators in each transmitting channel of the APAA complicates and increases the cost of the APAA design.

The technical problem of the claimed invention is to increase the speed of frequency tuning and changing the position of the APAA beam while reducing the requirements for the computing power of the control system.

The technical result consists in solving the specified technical problem.

The specified technical result is achieved in the method of beam formation in an active phased antenna array (APAA), which consists in the fact that in each m -th channel of the APAA, a linearly frequency modulated (chirp) signal is generated according to the formula

,

Where - calculated carrier frequency of the emitted chirp signal, – chirp signal deviation, – duration of the chirp signal, – calculated initial phase of the chirp signal of the m-th channel,

in this case, once for each channel, the control processor and processing of radio signals of the transceiver modules calculates the values of: the estimated carrier frequency of the emitted chirp signal , the calculated carrier frequency of the received chirp signal , and the calculated initial phase The chirp signal of the mth channel, while in the transmitting and receiving mode the calculated carrier frequencies of the mth channel are determined by the expressions

, respectively,

in this case, the calculated initial phase of the chirp signal of the m -th channel in the transmission mode is determined by the expression

,
Where – frequency deviation of the chirp signal, – duration of the chirp signal, – time shift of the m-th channel signal, – grating pitch, – carrier frequency of the channel taken as the reference one, for which m =0.

The claimed invention is illustrated in graphic materials.

In fig. Figure 1 shows a block diagram that implements the prototype method, where 1 is a digital chirp signal synthesizer, 2 is a pattern generation processor, 3 is a block of digital transceiver modules (TRM). Digital PPM block 3 includes M digital PPMs 3.1, 3.2, ..., 3.M. Each digital PPM includes quadrature modulators 3.1.1, 3.2.1-3.M.1, digital-to-analog converters (DACs) 3.1.2, 3.2.2-3.M.2, up frequency converters 3.1.3, 3.2.3- 3.M.3, amplifiers 3.1.4, 3.2.4-3.M.4, transmit-receive switches 3.1.5, 3.2.5-3.M.5, antenna elements 3.1.6, 3.2.6-3 .M.6, quadrature demodulators 3.1.7, 3.2.7-3.M.7, analog-to-digital converters (ADC) 3.1.8, 3.2.8-3.M.8, down frequency converters 3.1.9, 3.2 .9-3.M.9, low noise amplifiers 3.1.10, 3.2.10-3.M.10 and protection devices 3.1.11, 3.2.11-3.M.11. Moreover, the 1st, 2nd, ..., M-th output of the DN formation processor 2 is connected, respectively, to the inputs of quadrature modulators 3.1.1, 3.2.1-3.M.1 digital PPM 3.1, 3.2, ..., 3.M, and the outputs of quadrature demodulators 3.1.7, 3.2.7-3.M.7 are connected, respectively, to the 1st, 2nd, ...., Mth inputs of the DP formation processor 2. When AFAR is emitted, the digital chirp signal synthesizer 1 generates complex samples u(s) of the chirp signal. which arrive at the input of the DP formation processor 2. In processor 2, for given phasing directions θph and sample numbers s, the input chirp signal u(s) is multiplied by complex coefficients that make it possible to compensate for changes in the chirp signal phase depending on the number of digital PPM m, the selected direction phasing θf, as well as its frequency deviation ∆f. From the 1st, 2nd, ..., M-th outputs of the DP generation processor, 2 M chirp signals are supplied to the inputs of quadrature modulators 3.1.1, 3.2.1-3.M.1 digital PPM 3.1, 3.2, ..., 3. M accordingly. Quadrature modulators 3.1.1, 3.2.1-3.M.1 carry out the transfer of the chirp signal spectrum to the region of intermediate frequencies. Next, the signals are converted into analog form using the corresponding DACs 3.1.2, 3.2.2-3.M.2, and the chirp signal spectrum is transferred to the carrier frequency region using up frequency converters 3.1.3, 3.2.3-3.M. 3 and the formation in each of the digital PPM signals, then their amplification in amplifiers 3.1.4, 3.2.4-3.M.4 and through transmit-receive switches 3.1.5, 3.2.5-3.M.5 radiation into space using appropriate antenna elements 3.1.6, 3.2.6-3.M.6. Thus, the total transmission pattern is formed.

The signals received by antenna elements 3.1.6, 3.2.6-3.M.6 are sent through transmit-receive switches 3.1.5, 3.2.5-3.M.5 to the inputs of protection devices 3.1.11, 3.2.11-3. M.11, which perform the role of protecting the sensitive receiving path of digital PPM 3.1, 3.2-3.M from leakage of probing signals during their emission and the effects of powerful interference during reception. From the output of protection devices 3.1.11, 3.2.11-3.M.11, the signals are fed to the inputs of low-noise amplifiers 3.1.10, 3.2.10-3.M.10, which raise the signal amplitude to the required level for further digitization. Next, the spectrum of chirp signals is transferred to the region of intermediate frequencies using downward frequency converters 3.1.9, 3.2.9-3.M.9 and their conversion into digital form using the corresponding ADCs 3.1.8, 3.2.8-3.M. 8. From the outputs of the latter, signals are fed to the inputs of quadrature demodulators 3.1.8, 3.2.8-3.M.8. Quadrature demodulators form a complex envelope of the received chirp signals. From the outputs of quadrature demodulators 3.1.7, 3.2.7-3.M.7, the complex envelopes of M received chirp signals are supplied to the 1st, 2nd, ..., M inputs of the DP generation processor 2, respectively. In processor 2, for a given phasing direction θph and discrete sample number s, the m-th received chirp signal is multiplied by complex coefficients that make it possible to compensate for changes in the phase of the received chirp signal depending on the digital PPM number m, the selected phasing direction θph, as well as its deviation frequency ∆f and summation of signals from the outputs of digital PPM for each s-th sample. In this case, the resulting pattern (for transmission and reception) is formed.

In fig. Figure 2 shows a device that implements the claimed method of controlling the APAA beam and contains a digital PPM block, including amplifiers 3.1.4, 3.2.4-3.M.4, antenna elements 3.1.6, 3.2.6-3.M.6, ADC 3.1.8, 3.2.8-3.M.8, down frequency converters 3.1.9, 3.2.9-3.M.9, low noise amplifiers 3.1.10, 3.2.10-3.M.10, additionally introduced frequency converters 3.1.12, 3.2.12-3.M.12, output device 3.1.13, 3.2.13-3.M.13, TsVS 3.1.14, 3.2.14-3.3.14, reference oscillator 4, processor processing of radar information and control of the PPM 5 and bidirectional information transmission lines 6.1, 6.2-6.M, wherein the output 1 of the reference oscillator 4 is connected to the input 1 of the processor for processing radar information and controlling the PPM 5, the output 2 of the reference oscillator 4 is connected to the inputs 1 of the TsVS 3.1 .14, 3.2.14-3.3.14, inputs 1 ADC 3.1.8, 3.2.8-3.M.8, inputs 1 frequency converters 3.1.12, 3.2.12-3.M.12 inputs 1 frequency converters down 3.1.9, 3.2.9-3.M.9, outputs 2.1, 2.2-2.M of the processor for processing radar information and control PPM 5 through bidirectional information transmission lines 6.1, 6.2-6.M are connected to inputs 2 of the digital computer 3.1.14 , 3.2.14-3.3.14, inputs of 2 ADCs 3.1.8, 3.2.8-3.M.8, inputs of 3 frequency converters 3.1.12, 3.2.12-3.M.12 and inputs of 3 frequency converters down 3.1 .9, 3.2.9-3.M.9 digital PPM 3.1, 3.2-3.M respectively, outputs of 3 DCVs 3.1.14, 3.2.14-3.3.14 are connected to inputs of 2 frequency converters 3.1.12, 3.2.12 -3.M.12, outputs of 4 frequency converters 3.1.12, 3.2.12-3.M.12 are connected to inputs of 1 amplifiers 3.1.4, 3.2.4-3.M.4, outputs of 2 amplifiers 3.1.4, 3.2.4-3.M.4 are connected to inputs 1 of output devices 3.1.13, 3.2.13-3.M.13, inputs 3 of output devices 3.1.13, 3.2.13-3.M.13 are connected to inputs 1 active elements 3.1.6, 3.2.6-3.M.6, outputs of 2 output devices 3.1.13, 3.2.13-3.M.13 are connected to the inputs of 1 low-noise amplifiers 3.1.10, 3.2.10-3.M .10, the outputs of 2 low-noise amplifiers 3.1.10, 3.2.10-3.M.10 are connected to the inputs of 4 downward frequency converters 3.1.9, 3.2.9-3.M.9, the outputs of 2 downward frequency converters 3.1.9, 3.2.9-3.M.9 are connected to inputs 3 of ADC 3.1.8, 3.2.8-3.M.8.

In fig. Figure 3 shows a graph of voltage dependences for a five-channel grille.

The essence of the invention is as follows.

In the general case, the expression for the chirp signal at the output of the m-th channel of the APAA in the radiation mode can be presented as [9]:

, (1)

Where – frequency deviation of the chirp signal, – duration of the chirp signal, – time shift of the m-th channel signal, – grating pitch, – carrier frequency of the channel taken as the reference one, for which m =0. Grouping the components of the instantaneous phase of expression (1) for degrees of time t equal to 0, 1 and 2, we obtain the expression

,

or

, (2)

Where – calculated carrier frequency of the mth channel,

– calculated initial phase of the chirp signal of the m -th channel.

Thus, the formation of the APAA beam is carried out by generating a chirp signal in each channel using the calculated carrier frequency and the initial phase of the chirp signal of the mth channel. In this case, in the direction determined by the angle rotation of the beam pattern, the signals emitted by all antenna elements at any time have the same instantaneous frequency and phase.

When operating in the receiving mode, taking into account the change in the sign of the delay time for the reflected chirp signal, the voltage at the output of the mth receiving channel can be described by the expression

,

or , (3)

Where – calculated carrier frequency of the mth channel in receiving mode,

– previously calculated initial phase of the chirp signal of the mth channel.

Then, when calculating the resulting pattern, the output voltage of the mth channel should be calculated for the carrier frequency and initial phase taken with a minus sign.

The device works as follows.

The PPM 5 control and radio signal processing processor generates control commands and transmits them through bidirectional information transmission lines 6.1, 6.2-6.M to the DCS 3.1.14, 3.2.14-3.3.14. In accordance with this DDS codes, chirp signals are generated at intermediate frequency (IF) with the required IF values and initial phase . Next, the chirp signal is transferred to the frequency of the emitted signal AFAR frequency converter 3.1.12, 3.2.12-3.M.12, is amplified by amplifier 3.1.4, 3.2.4-3.M.4 and passing through the output device 3.1.13, 3.2.13-3.M.13 radiated by antenna element 3.1.6, 3.2.6-3.M.6.

In this scheme, frequency converters can transfer the spectrum of the generated chirp signal both up and down in frequency, depending on the values of the channel carrier frequency and IF .

Specific values And are selected from the condition of obtaining the required level of suppression of combinational components. The coherence of the signals at the outputs of the DDS and the frequency converters of all PPMs is ensured by a single reference oscillator 4 for all channels.

Output devices 3.1.13, 3.2.13-3.M.13 provide isolation and protection of amplifiers 3.1.4, 3.2.4-3.M.4 and low-noise amplifiers 3.1.10, 3.2.10-3.M.10 can contain decoupling, switching, and filtering limiting elements, depending on the required parameters of the probing signal and the modes of changing the position of the pattern beam.

The received signals from the antenna elements through output devices 3.1.13, 3.2.13-3.M.13 are supplied to low-noise amplifiers 3.1.4, 3.2.4-3.M.4, then to downward frequency converters 3.1.9, 3.2. 9-3.M.9 and further converted into digital form in ADC 3.1.8, 3.2.8-3.M.8.

Next, the signals are converted from digital form into quadrature components in quadrature demodulators and enter the PPM 5 radio signal control and processing processor, where preliminary processing of radio signals is carried out - the formation of the necessary receiving radiation patterns (one or several simultaneously) by controlling the digital delays and amplitude multipliers received in the channels signals, accumulation, convolution, decimation and formation of data packets for the system of primary processing of radar information.

The coherence of conversion and processing of the received APAA signal is also ensured by a common reference oscillator 4 for all channels.

Expression (2) is most simply implemented when using digital computational synthesizers (DCS) as a chirp signal generator with the presence in the structure of not only a phase code accumulator, but also a frequency code accumulator. To form a chirp signal with specified characteristics using such a digital digital signal, the values of the initial frequency and phase, the frequency increment step and the duration of the chirp signal formation are specified [6]. For a linear array, the time shift must be measured relative to the center of the grating. The DACs included in the DAC convert the chirp signal generated in accordance with expression (2) from digital to analog form directly at the intermediate frequency.

Thus, there is a significant simplification of the APAA pattern formation scheme, since a separate digital chirp synthesizer and quadrature modulators in each PPM are excluded from it, and the PPM radio signal control and processing processor calculates the values of the calculated carrier frequency of the emitted chirp only once for each position of the pattern beam signal , the calculated carrier frequency of the received chirp signal , and the calculated initial phase Chirp signal of the m -th channel, and instead of multiplication operations, addition (subtraction) operations are performed, performed once during the period of chirp pulse formation, which significantly reduces the requirements for the computing power of the pattern formation processor.

At the same time, the requirements for the speed of information transfer from the pattern formation processor to the digital PPM are reduced, which increases the noise immunity of the APAA. Moreover, the elements necessary for generating a pattern for transmission can be excluded from the circuit of the pattern formation processor, since for a finite number of positions of the pattern beam the values , And can be calculated once and stored in a memory device and, if necessary, sent to the appropriate MRP. By freeing up the computing resources of the pattern formation processor, the time required to apply new values of modulation parameters and change the position of the APAA beam is reduced.

In fig. Figure 3 shows an example of voltage dependencies for a five-channel array, calculated using formulas (2). It can be seen that in the time domain, where more than one voltage exists simultaneously , these voltages change synchronously, so that the instantaneous phases of all voltages , match up.

Calculation example And .

Central array channels, beam deflection 1°.

;

;

;

.

The outer and central channels of the array (m=100), beam deflection 45°.

;

;

;

;

Literature

1. US Patent No. 4263600 dated 04/21/1981. Wide Band, Phase Scanned Antenna.

2. Active phased array antennas / Ed. DI. Voskresensky and A.I. Kanashchenkova. – M.: Radio engineering, 2004. - 488 p.

3. US Patent No. 5943010 dated 08/24/99. Direct Digital Synthesizer Driven Phased Array Antenna.

4. China Patent No. CN113253210A dated 06/29/2021. Full-digital frequency-shift phase-shift large instantaneous broadband phased array and method.

5. US Patent No. 9479232 dated October 25, 2016. Method of Achieving Ultra-Wideband True-Time-Delay Beam Steering for Active Electrically Scanned Arrays.

6. Kochemasov V., Skok D., Cherkashin A. Digital computational synthesizers - modern solutions. Part 2 // Electronics, science, technology, business. 2014. No. 4. P. 154-158.

7.US Patent No. 7345629 dated March 18, 2008. Wideband Active Phased Array Antenna System.

8. US Patent No. 2002/0175859 dated November 28, 2002. Phased Array Antenna System with Virtual Time Delay Beam Steering.

9. RF Patent No. 2516683 dated October 17, 2012. A method for digitally forming the radiation pattern of an active phased array antenna when emitting and receiving a linearly frequency-modulated signal.

Claims (10)

A method for forming a beam in an active phased antenna array (APAA), which consists in the fact that in each m-th element of the APAA, a linearly frequency modulated (chirp) signal is generated according to the formula in this case, once for each channel, the control processor and processing of radio signals of the transceiver modules calculates the values: calculated carrier frequency of the emitted chirp signal calculated carrier frequency of the received chirp signal and calculated initial phase Chirp signal of the m-th channel, in this case, in the transmission and reception mode, the calculated carrier frequencies of the m-th channel are determined by the expressions respectively, in this case, the calculated initial phase of the chirp signal of the m-th channel in the transmission mode is determined by the expression where ∆f is the frequency deviation of the chirp signal, τ is the duration of the chirp signal, – time shift of the m-th channel signal, d – grating pitch, – carrier frequency of the channel taken as the reference one, for which m=0.