RU2722408C1 - Digital receiving module of active phased antenna array - Google Patents

Abstract

FIELD: radio engineering; instrumentation.SUBSTANCE: invention relates to radio engineering instrument making and can be used in designing active phased antenna arrays (APAA) with digital generation and electronic control of beam pattern in a wide sector during broadband probing of targets. Device comprises a radiator, a low-noise amplifier, an analogue-to-digital converter, a group of M digital band-pass filters dividing the broadband spectrum into narrow-band spectra, a group of M dividers into two directions, two groups of synchronous phase detectors (SPD), consisting of M detectors each, a generator of digital complex coefficients, two groups of digital complex multipliers by M multipliers each, two digital adders, which outputs are digital receiver module outputs, at that outputs of dividers into two directions are connected to the first SPD inputs, the second inputs of which are connected to outputs of digital heterodyne APAA. Outputs of SPD are connected to second inputs of digital complex multipliers, first inputs of which are connected to outputs of generator of digital complex coefficients, input of which is connected to output of beam control system APAA. As a result, based on broader functional capabilities of the radar to provide high resolution in range provides reduced cost of production of receiving module and costs for operation of APAA, high accuracy of formation of direction-finding characteristics of APAA.EFFECT: broader functional capabilities of the radar set providing high resolution in range.1 cl, 1 dwg

Description

The invention relates to antenna technology, namely to active phased antenna arrays (AFAR) with digital formation and radiation pattern control and can be used in radar stations (radars) with a wide-angle electronic survey of the space, using broadband linear-frequency-modulated as probing pulses (LFM) signals.

Known technical solutions aimed at creating receiving modules (or receiving channels of receiving and transmitting modules). However, most of them involve either the use of narrow-band signals [1-5] or broadband signals with a relatively narrow sector of electronic scanning of the radiation pattern [6]. At the same time, solving the problems facing modern radars requires the use of signals with a wide range (up to tens of percent of the carrier frequency). Such tasks include providing radar noise immunity and increasing its resolution in range, which allows for the recognition of detected objects by their range portraits or to determine the numerical composition of a group target [7].

With a wide spectrum of the emitted LFM signal and a wide sector of electronic scanning, an additional phase incursion caused by the frequency deviation of the LFM signal is superimposed on the linear phase shift at the aperture of the linear AFAR, which leads to distortion of the radiation pattern of the AFAR in the receiving mode. Therefore, in a number of technical solutions for broadband sounding, targets limit the sector of electronic scanning of the radiation pattern. As an example, in [8], the characteristics of radars such as GBR (Ground Based Radar), SBX (Sea-Based X-band radar), AN / TPY-2, using signals with a spectral width of the order of 1 GHz (which allows providing radar resolution 0.15 m range), the electronic scanning sector of which is only 10 °, while in narrow-band mode the sector width is 120 °.

The analysis performed in [9] showed that the additional phase incursion is determined by the phase factor

, (1)

, (1)

- девиация частоты ЛЧМ-сигнала,

- длительность зондирующего импульса,

- номер излучателя линейной АФАР,

- шаг решетки,

- угол отклонения луча АФАР от нормали к ее раскрыву,

- текущее время (

).Where

- frequency deviation of the chirp signal,

- duration of the probe pulse,

- number of the emitter linear AFAR,

- lattice pitch,

- the angle of deviation of the beam AFAR from the normal to its opening,

- current time (

)

In [9], when an LFM signal is emitted in each nth element for the selected phasing direction θ _{f, it is} possible to compensate for changes in the phase of the signal by deviating the frequency Δƒ of the LFM signal by multiplying (1) by the complex conjugate coefficient

(2)

(2)

Since both the phase factor (1) and the coefficient (2) complex conjugate with it are functions of time, they should be formed synchronously. In transmission mode, synchronizing functions (1) and (2) is not a technical difficulty. This is the advantage of the method [9]. However, in [9], it was also proposed in the reception mode to compensate for the phase change of the signal from the output of the nth antenna element for the selected phasing direction θ _f by multiplying by the complex coefficient (2), while for the received signal the current time t is in the range of values

t _s ≤ t <t _{s +} τ _u, (3)

where t _s - the delay time of the signal reflected from the target.

However, since the distance to the target is unknown, the delay time t _s is also unknown. Even if the distance to the target is measured, it is measured with an error. An elementary analysis shows that when the error in measuring the delay time Δt _s <<τ _{u, the} introduced additional phase error significantly distorts the directivity pattern of the AFAR.

Thus, the method proposed in [9] for compensating phase incursions that occur when receiving an LFM signal with a frequency deviation Δƒ and when phasing the beam in the direction θ _f is not technically feasible. This is the disadvantage of the method [9].

с помощью М фильтров делится на М узкополосных участковThe considered disadvantage is eliminated in the technical solution [10], which contains a serially connected module emitter, a low noise amplifier (LNA), a mixer, an intermediate frequency amplifier (IFA), a filter of spectral components, as well as M filters for dividing the broadband spectrum of the received signal into M narrow-band sections of the spectrum, two groups of synchronous phase detectors (SFD) by M SFD in each group, two groups of analog-to-digital converters (ADC) by M ADC in each group, a constant 90 ° phase shifter, two groups of controlled phase shifters by M controlled phase shifters in each group, and two digital adder. In this case, the broadband spectrum of the received signal

using M filters is divided into M narrow-band sections

,

,

- участок ширины спектра широкополосного сигнала, для которого выполняется условие узкополосностиWhere

- portion of the spectrum bandwidth of the broadband signal for which the narrowband condition is satisfied

where c is the speed of light, L is the maximum aperture of the AFAR.

) с помощью двух групп управляемых фазовращателей. При этом вносимый фазовый сдвиг для i-го узкополосного участка спектра определяется соотношениемThe AFAR radiation pattern control is carried out for each narrow-band i-th spectrum (

) using two groups of controlled phase shifters. In this case, the introduced phase shift for the ith narrow-band portion of the spectrum is determined by the relation

,

,

where ƒ _i is the center frequency of the i-th narrow-band spectrum, n is the number of the emitter of the linear antenna array, d is the pitch of the array, c is the speed of light, θ _f is the direction of phasing of the beam.

Thus, the control of the AFAR radiation pattern with a broadband signal is reduced to control with a narrowband signal, which eliminates the appearance of additional phase errors at the aperture of the antenna caused by the frequency deviation of the probe LFM signal. This is the advantage of the technical solution [10], and its disadvantages include: the complexity of the design of the analog-to-digital module, low efficiency (EFFICIENCY) and low accuracy of phase control of the AFAR radiation pattern, due to the discreteness of the phase shifters used in the technical solution.

These shortcomings were eliminated in the technical solution [11], in which the simplification of the design of the analog-to-digital receiving module and the increase in its efficiency were achieved by eliminating two blocks of controlled phase shifters from its structure, and the improvement of the accuracy of the AFAR beam control was achieved by introducing two groups of digital complex multipliers by M multipliers in each group and shaper of digital complex weighting factors. This technical solution is the closest to the claimed and selected as a prototype. It consists of an emitter, a low-noise amplifier, a mixer, an intermediate-frequency amplifier, a filter of spectral components, a group of M dividers in two directions, a constant 90 ° phase shifter, two groups of synchronous phase detectors, consisting of M detectors each, two groups of analog-to-digital converters (ADC), consisting of M ADCs each, two groups of digital complex multipliers, consisting of M multipliers each, two digital adders, the outputs of which are the outputs of the receiving module. The second input of the mixer is connected to the output of the first local oscillator AFAR. The input of the constant phase shifter is connected to the output of the second AFAR local oscillator. The control inputs of the ADC are connected to the output of the AFAR clock pulse generator. The input of the shaper of digital complex coefficients is connected to the output of the AFAR beam control system. Selected as a prototype analog-to-digital receiving module has the following disadvantages.

Analogue filters are used as dividers of a wide range of the received signal into M narrow-band spectral regions, the parameters of which change with changing operating conditions (temperature, pressure, etc.), which leads to an uncontrolled error of the output signal, i.e. to distortions of the direction-finding characteristic of the AFAR [12, p. 46]. In addition, the presence of a large number of (M) analog filters in each receiving module leads to an increase in the cost of their production.

, что необходимо для выделения сигнала от одной цели в составе групповой, ширина спектра зондирующего сигнала должна бытьThe prototype includes a frequency converter, an intermediate frequency amplifier, a filter of spectral components, i.e. processing of the received signal is carried out at an intermediate frequency, which limits the radar's functionality to provide the required resolution in range required to recognize single targets from a long-range portrait, as well as to determine the numerical composition of a group target. So, to ensure the resolution of the radar range

, which is necessary to isolate the signal from one target as part of a group, the spectrum width of the probing signal should be

.

.

In this case, the intermediate frequency should be at least an order of magnitude higher than the signal spectrum width, i.e.

In turn, the frequency of the carrier oscillation of the probe signal should be at least an order of magnitude higher than the intermediate, i.e.

, то придется использовать миллиметровый диапазон волн.which corresponds to a wavelength of λ _about = 2 cm, unsuitable for long-range radar due to the large attenuation of the waves of the centimeter range in the atmosphere. If range resolution is required

then you have to use the millimeter wave range.

, то ширина спектра зондирующего сигналаThe composition of the receiving module of the prototype includes two ADCs for each narrowband channel. For example, if the resolution of the radar range

, then the width of the spectrum of the probe signal

.

.

, что соответствует длине волны λ_о=0,06 м. Если при этом разрешающая способность по азимуту то линейный размер апертуры антенны в горизонтальной плоскости

Исходя из условия узкополосности спектра

, примем

=4.10⁶ Гц. Тогда количество узкополосных участков спектраThen the carrier frequency of the probe signal

, which corresponds to a wavelength of λ _{about =} 0.06 m. If the resolution in azimuth is the linear size of the antenna aperture in the horizontal plane

Based on the conditions of narrow-band spectrum

accept

= 4.10 ⁶ Hz. Then the number of narrow-band sections of the spectrum

therefore, the required number of ADCs will be 2M = 250. The largest contribution to the cost of the receiving module is made by the ADC (for example, the wholesale cost of a two-channel ADC12DL3200 ADC is 2599.95 US dollars per 100 pcs. [13]), and the number of receiving modules in the AFAR can be several thousand [14, p. 119], which significantly increases the cost of its production. In addition, the characteristics of high-speed ADCs with a power consumption of at least 3 W are given in [15]. Thus, with M ADCs as part of one receiving module and with a large number of such modules as part of AFAR (up to several thousand), the cost of its operation increases significantly.

) соединен с входом i-го делителя на два направления, первый выход которого соединен с первым входом i-го синхронного фазового детектора (СФД) первой группы, а второй выход соединен с первым входом i-го СФД второй группы, две группы цифровых комплексных умножителей по М умножителей в каждой группе, к первому входу i-го умножителя каждой группы подключен i-й выход формирователя цифровых комплексных коэффициентов, вход которого соединен с выходом системы управления лучом (СУЛ) АФАР, выходы цифровых комплексных умножителей первой группы подключены к входам первого цифрового сумматора, выход которого является первым выходом приемного модуля, а выходы цифровых комплексных умножителей второй группы подключены к входам второго цифрового сумматора, выход которого является вторым выходом приемного модуля, введен аналого-цифровой преобразователь (АЦП), первый вход которого подключен к выходу малошумящего усилителя, а выход подключен к входам всех фильтров деления широкополосного спектра на М узкополосных участков спектра, которые реализованы на цифровых полосовых фильтрах с конечной импульсной характеристикой (КИХ-фильтрах), причем вторые входы СФД первой группы и вторые входы СФД второй группы подключены к соответствующим (синусным и косинусным) ортогональным выходам цифрового гетеродина АФАР, общего для всех приемных модулей, вторые входы АЦП и цифровых полосовых фильтров деления широкополосного спектра на узкополосные участки спектра подключены к выходу генератора тактовых импульсов АФАР, общего для всех приемных модулей, выходы всех СФД подключены к вторым входам соответствующих цифровых комплексных умножителей.The objectives of the invention are to reduce the cost of production of the receiving module and the cost of operating the AFAR, increasing the accuracy of forming the direction-finding characteristics of the AFAR, expanding the radar's functionality to provide high resolution in range. The solution of these problems is achieved by the fact that in a digital receiving module containing a serially connected module emitter and a low-noise amplifier (LNA), as well as M filters for dividing the broadband spectrum into M narrow-band sections of the spectrum, the output of the ith filter for dividing the broadband spectrum (

) is connected to the input of the i-th divider in two directions, the first output of which is connected to the first input of the i-th synchronous phase detector (SFD) of the first group, and the second output is connected to the first input of the i-th synchronous phase detector (second group), two groups of digital complex multipliers according to M multipliers in each group, the i-th output of the digital complex coefficient generator, the input of which is connected to the output of the AFAR beam control system (SUL), is connected to the first input of the i-th multiplier of each group, the outputs of the digital complex multipliers of the first group are connected to the inputs of the first digital an adder whose output is the first output of the receiving module, and the outputs of the digital complex multipliers of the second group are connected to the inputs of the second digital adder, the output of which is the second output of the receiving module, an analog-to-digital converter (ADC) is introduced, the first input of which is connected to the output of the low-noise amplifier, and the output is connected to the inputs of all filters dividing the broadband spectrum into M narrow band sections of the spectrum that are implemented on digital bandpass filters with a finite impulse response (FIR filters), the second inputs of the first group of the SFD and the second inputs of the second group of the connected to the corresponding (sine and cosine) orthogonal outputs of the AFAR digital local oscillator, common to all receiving modules, the second inputs of the ADC and digital bandpass filters dividing the broadband spectrum into narrowband sections of the spectrum are connected to the output of the AFAR clock pulse generator, common to all receiving modules, the outputs of all SFDs are connected to the second inputs of the corresponding digital complex multipliers.

Thus, the differences of the proposed digital receiving module AFAR from the prototype are as follows:

Instead of a 2M ADC, the proposed receiver module contains only one, which makes the design of the receiver module simpler and cheaper, in addition, the power consumed by analog-to-digital converters is significantly reduced. This difference from the prototype allows to reduce the cost of production of the receiving module and the cost of operating the AFAR.

An analog-to-digital converter is connected between the output of the low-noise amplifier and the inputs of the filter dividers of the broadband spectrum into narrow-band sections, i.e. Digital processing of the received signals is carried out without conversion to an intermediate frequency.

Until a certain time, AFAR developers had to build receiving modules (or receiving channels of receiving and transmitting modules) according to a superheterodyne circuit, since ADCs and digital-to-analog converters (DACs) with a dynamic range acceptable for radar systems could operate at frequencies of 100-200 MHz, which led to the necessity of installing frequency converters in the transmitting and receiving module, since sampling of signals at the carrier frequency was impossible [16, p. 9]. However, the current level of ADC technology makes it possible to carry out analog-to-digital conversion of signals at a speed of up to 40 gigabytes per second with 8-10-bit resolution, while the range of input analog signals is up to 13 GHz [14, p. 120]. The expediency of such a solution was indicated in a number of works [14, 17], for example, in [14, p. 183] it is noted that with high demands on the dynamic range and noise figure of the receiving path, it is advisable to implement it without frequency conversion. However, in the framework of this invention, the rejection of the frequency conversion is dictated primarily by the fact that in this case the restrictions on ensuring high resolution radar in range are removed, which is necessary for recognizing the type of a single target from a long-range portrait and (or) determining the numerical composition of a group target. There is no such approach to rejecting the frequency conversion of the received signal in the well-known technical literature. Thus, this difference from the prototype eliminates the limitations associated with the frequency conversion of the signal on the implementation of the required range resolution.

Digital bandwidth FIR filters are used as dividers of the broadband spectrum into narrow-band sections, which, in contrast to the analog filters used in the prototype, provide high accuracy of AFAR direction-finding characteristic formation and the possibility of their implementation in the form of specialized programmable logic integrated circuits [18, 19]. The analysis performed in [19] showed that the modern elemental base allows digital processing of signals in real time at frequencies up to 1.5 GHz, and in the coming years up to 20 GHz.

These distinguishing features are significant, as they provide a positive effect consisting in lowering the cost of manufacturing the receiving module and lowering the cost of operating the AFAR in comparison with the prototype due to a significant reduction in the number of expensive and consuming high power ADCs, in increasing the accuracy of the formation of direction-finding characteristics of the AFAR based on the use of digital bandwidth FIR filters, as well as in expanding the radar's functionality to provide high resolution in range by eliminating the frequency conversion of the received signal.

The invention is illustrated in figure 1, which shows a block diagram of a digital receiving module AFAR. The digital receiving module includes: 1 - emitter, 2 - low-noise amplifier, 3 - ADC, 4 - a group of M digital bandwidth FIR filters for dividing a wide spectrum into narrow-band sections of the spectrum, 5 - a group of M dividers in two directions, 6 and 7 - two groups of synchronous phase detectors (SFD), consisting of M detectors each, 8 - shaper of digital complex coefficients, 9 and 10 - two groups of digital complex multipliers, consisting of M multipliers each, and 11 and 12 - two digital adders, outputs which are the outputs of the digital receiving module. The control inputs of the ADC 3 and digital filters 4 are connected to the output of the AFAR 15 clock pulse generator. The input of the digital complex coefficients 8 is connected to the output 16 of the AFAR beam control system.

) делителя 5 сигнал поступает на первый вход i-го синхронного фазового детектора 6 первой группы, а со второго выхода i-го делителя 5 сигнал поступает на первый вход i-го СФД 7 второй группы. Цифровые коды синусоидального опорного напряжения с выхода 13 цифрового квадратурного гетеродина АФАР поступают на вторые входы СФД 6 первой группы, а цифровые коды косинусоидального опорного напряжения с выхода 14 цифрового гетеродина АФАР поступают на вторые входы СФД 7 второй группы.Digital receiving module operates as follows. Received by the emitter 1, the signal reflected from the target is fed to the input of a low-noise amplifier 2, which raises the signal level to a value sufficient for its analog-to-digital conversion of the ADC 3. The digitized signal from the output of the ADC 3 goes to the inputs M of the digital bandpass FIR filters 4. From the outputs digital bandpass FIR filters 4 narrowband signals are fed to the inputs of M dividers in two directions 5. From the first output of the i-th (

) of the divider 5, the signal goes to the first input of the i-th synchronous phase detector 6 of the first group, and from the second output of the i-th divider 5, the signal goes to the first input of the i-th SFD 7 of the second group. The digital codes of the sinusoidal reference voltage from the output 13 of the digital quadrature local oscillator AFAR are supplied to the second inputs of the AFD 6 of the first group, and the digital codes of the cosine reference voltage from the output 14 of the digital quadrature local oscillator AFAR are supplied to the second inputs of the AFD 7 of the second group.

и квадратурную

, где k - порядковый номер цифрового отсчета (

), T - период дискретизации напряжения сигнала, определяемый в соответствии с теоремой Котельникова и задаваемый генератором тактовых импульсов АФАР 15. Синфазная составляющая сигнала на выходе i-го СФД 6 первой группы может быть представлена в видеSynchronous phase detectors 6 and 7 divide the narrow-band digital signals arriving at their inputs into two quadrature components - in-phase

and quadrature

where k is the serial number of the digital reference (

), T is the sampling period of the signal voltage, determined in accordance with the Kotelnikov theorem and set by the AFAR 15 clock pulse generator. The in-phase component of the signal at the output of the i-th SFD 6 of the first group can be represented as

, (4)

, (4)

and the quadrature component

, (5)

, (5)

- центральная частота i-го узкополосного сигнала,Where

is the center frequency of the i-th narrowband signal,

,

- цифровые коды амплитуд, а

,

- цифровые коды фаз напряжений (4) и (5),

частота колебаний гетеродина АФАР, равная частоте зондирующего сигнала

,

- digital codes of amplitudes, and

,

- digital codes of voltage phases (4) and (5),

oscillator frequency of the AFAR local oscillator equal to the frequency of the probe signal

The complex envelopes of stresses (4) and (5) have the form

, (6)

, (6)

, (7)

, (7)

Digital codes (6) from the output of the i-th SFD of the first group 6 go to the second input of the i-th digital complex multiplier of the first group 9, and digital codes (7) from the output of the i-th SFD of the second group 7 go to the second input of the i-th digital complex multiplier 10 of the second group.

и

, определяющих ожидаемое направление прихода отраженного от цели сигнала. Комплексные цифровые весовые коэффициенты

для каждой центральной частоты узкополосного сигнала вычисляются в соответствии с соотношениемAt the input of block 8 of the shaper of digital complex coefficients from the output 16 of the AFAR beam control system, the values of the direction cosines of the angles

and

determining the expected direction of arrival of the signal reflected from the target. Integrated Digital Weights

for each center frequency of the narrowband signal are calculated in accordance with the ratio

, (8)

, (8)

, (9)Where

, (9)

и

- шаг решетки вдоль осей OX и OY соответственно.c is the speed of light, m and n are the numbers of rows and columns at the intersection of which are located emitters AFAR,

and

is the lattice pitch along the OX and OY axes, respectively.

As a result of multiplying the complex envelopes of stresses (6) and (7) by the complex weighting factors (8), the stresses (6) and (7) receive an additional phase shift by an amount determined by relation (9). The signal from the output of the i-th digital complex multiplier of the first group 9 goes to the i-th input of the first digital adder 11, and the signal from the output of the i-th digital complex multiplier of the second group 10 goes to the i-th input of the second adder 12. Adders 11 and 12 summarize the digital codes, respectively, in-phase and quadrature components of narrow-band sections of the broadband spectrum. As a result, from the first and second outputs of the digital receiving module, digital codes of the amplitude and phase of the signal received by this module are received in the primary radar information processing system.

Thus, the technical implementation of the proposed digital receiving module will reduce the cost of its production and the cost of operating the AFAR, increase the accuracy of the formation of its direction-finding characteristics of the AFAR and ensure the required radar resolution range.

Sources of information used in the preparation of the application:

1. US patent No. 59430/10, H01Q 3/24. 08/24/1999. Direct Digital Synthesizer Driven Phased Array Antenna.

2. US patent No. 6441783, H01Q 3/22, 08/27/2002. Circuit Module for a Phased Array.

3. RF patent No. 2454763, H01Q 21/00, June 27, 2012. Transceiver module of the active phased microwave antenna array.

4. RF patent No. 2206155, H01Q 3/34, 06/10/2003. Transceiver module of an active phased antenna array.

5. RF patent No. 2362268, Н04 В 1/38, 02/10/2009. Transceiver module of an active phased array antenna.

6. RF patent No. 2571884, H01Q 21 \ 00, 12/27/2015. Transceiver module of an active phased array antenna.

7. RF patent No. 2513041, G01S 13/52, 04/20/2014. Device for identification of air objects by the structure of a long-range portrait.

8. Abramenkov V.V., Vasilchenko O.V., Muravsky A.P. Processing of extended ultra-wideband signals in radars with electronic scanning of the bottom of the beam \ Electromagnetic waves and electronic systems, 2013, vol. 8, No. 3, p. 7-14.

9. RF patent No. 2516683, H01Q 21/00, 05.20.2014. A method for digitally generating a radiation pattern of an active phased array antenna when emitting and receiving a linear frequency-modulated signal.

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11. RF patent No. 2692417, H03M 1/12; H01Q 21/00, 06/24/2019 .. Analog-digital module of the active phased antenna array.

12. Goldenberg L.M. et al. Digital signal processing: Handbook. - M .: Radio and communications, 1985 .-- 312 p.

13. The official website of the Texas Instruments Inc campaign: ADC12DL3200 ADC cost. - URL: http://www.ti.com/product/ADC12DL3200.

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15. ADC12DL3200 6.4-GSPS Single-Channel or 3.2-GSPS Dual-Channel, 12-Bit Analog-to-Digital Converter (ADC) With LVDS Interface: Texas Instruments ADC12DL3200. SLVSDR3A -May 2018-Revised September 2018.

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Claims (1)

A digital receiving module of an active phased antenna array containing a serially connected module emitter and a low-noise amplifier (LNA), M filters for dividing the broadband spectrum into M narrow-band sections of the spectrum, the output of the ith filter for dividing the broadband spectrum (

) is connected to the input of the i-th divider in two directions, the first output of which is connected to the first input of the i-th synchronous phase detector (SFD) of the first group, and the second output is connected to the first input of the i-th synchronous phase detector (second group), two groups of digital complex multipliers by M multipliers in each group, the i-th output of the digital complex coefficients shaper is connected to the first input of the i-th multiplier of each group, the input of which is connected to the output of the AFAR beam control system, the outputs of the digital complex multipliers of the first group are connected to the inputs of the first digital adder, output which is the first output of the receiving module, and the outputs of the digital complex multipliers of the second group are connected to the inputs of the second digital adder, the output of which is the second output of the receiving module, characterized in that an analog-to-digital converter (ADC) is inserted into it, the first input of which is connected to the output LNA, and the output is connected to the first inputs of all filters for dividing broadband spectrum in the M narrow-band spectral regions that are implemented on digital bandpass FIR filters, the second inputs of the first group of AFDs are connected to the first orthogonal (sine) output of the AFAR digital local oscillator, the second inputs of the second group of AFDs are connected to the second orthogonal (cosine) output of the AFAR digital local oscillator, common to all receiving modules, and the second inputs of the ADC and digital FIR filters are connected to the output of the AFAR clock pulse generator, common to all receiving modules, the outputs of the SFD are connected to the second inputs of all digital complex multipliers.

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