RU2563872C2 - Diversity radar with third-party illumination of gsm cellular communication networks - Google Patents

Abstract

FIELD: physics, navigation.

SUBSTANCE: invention relates to diversity radar. The main and reference channels of diversity radar include amplifiers with a logarithmic gain characteristic at an intermediate frequency, and each range channel of an M-channel correlator includes a rough selection filter with band analysis of the entire possible range of Doppler frequencies and with rejection zones at frequencies of periodic components of the spectrum of signals reflected from local objects, as well as subtractors for unsuppressed residues of penetrating interference on side lobes of the beam pattern of the forward signal radar antenna. To generate a copy of the unsuppressed residue of penetrating interference, acting on range channels, similar reference signal generators include forward signal multipliers and a copy thereof delayed by the expected delay time. The multiplication result is transmitted to the inverting input of the subtractor.

EFFECT: longer range and constancy of characteristics of detecting targets with semi-active diversity radar with third-party illumination, generated by a GSM base station, owing to suppression of periodic components of the spectrum of interference signals, passive interference generated by the frame and the slot structure of the signal reflected from local objects, as well as owing to use of devices for stabilising the detection threshold and logarithmic gain.

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Description

The invention relates to diversity radar and can be used to detect, measure coordinates and recognize low-altitude targets in the backlight field of GSM base stations. The technical result of the invention is to improve the tactical and technical characteristics, namely to increase the accuracy of measuring coordinates and protect the receiving device from the influence of a direct penetrating signal of a base station (BS) and reflections from local objects, as well as the ability to recognize location information and transmit it to collection and processing points information.

A known method and method of passive coherent location of civil aviation aircraft (US patent No. 7012552 B2, 03/14/2006, B64F 1/36; G01S 13/00; G01S 13/46; G01S 13/50; G01S 13/91; G01S 19 / 48; G01S 5/12; G08B 21/00; G08G 5/00; G08G 5/04; G01S 13/93).

Known mobile system for detecting an object and a method of using signals transmitted by a mobile telephone exchange (US patent No. 6930638 B2, 1.08.2001, G01S 3/00; G0S 13/46; G0S 13/58; G0S 13/92; H04Q 7/34; G0S 3/46), comprising a receiver having first and second antennas, and processing means, the first (reference) antenna being configured to receive a direct signal from a mobile telephone base station; and the second (target) antenna is configured to receive a base station signal reflected from the object. The processing means compares the signal received from the base station of the mobile telephone with the signal reflected from the object, and determines the speed and position of the object. A system consists of a plurality of base stations for mobile telephony that transmit a signal. The disadvantage of the system is the lack of compensation for the direct signal of the base station that penetrates along the side lobes of the target antenna, which leads to a decrease in the detection range and accuracy of coordinate measurement.

Closest to the invention is a semi-active radar station using for illumination of radar surveillance radiation from cellular base stations (Eurasian patent No. 008335, June 9, 2005; G01S 13/50), containing one main connected to a directional antenna, and two reference connected to directed to the BS of the channel, and the outputs of the receiving channels are connected to the corresponding inputs of the spatial auto-compensator, after which, for spectral rejection of the directional pattern penetrating along the side lobes and radar antenna direct input of range M channels, each of which comprises a multiplier, and N combined at the input narrowband Doppler filters. The formation of the reference signal is carried out in a special driver of the reference signal connected to the output of the reference channel. Additional spectral notch provides an increase in the compensation efficiency of the penetrating signal and an increase in the range.

The semi-active radar station under consideration is not without a number of disadvantages.

и

Экспериментально полученные гармоники спектра автосвертки сигнала GSM, приведенные на фиг.1а и б, перекрывает ожидаемый диапазон доплеровских частот до 1700 Гц и становятся мешающей помехой при обнаружении маловысотных целей (МВЦ), летящих со скоростями до 900 км/ч. При этом, как показывают расчеты [2], основной лепесток спектра автосвертки сигнала GSM на 20-30 дБ превышает амплитуда первых периодических гармоник на частотах кратных 216,6 Гц и 1732,8 Гц. Известно, что мешающими помехами в разнесенных РЛС с ненаправленным подсветом являются прямые сигналы передатчика, проникающие по боковым лепесткам диаграммы направленности антенны (ДНА) целевого канала, и отражения от местных предметов, находящиеся вместе с целью в основном луче ДНА и одном элементе разрешения по дальности.Firstly, the GSM signal model presented in the description with a Gaussian frequency shift keying with a minimum frequency shift GMSK modulation does not take into account the frame periodic structure of the signal. So for the transmission of information using time and code division multiplexing (TDMA) uses 5 types of periodic time intervals [1]. At the same time, a TDMA frame with a duration of τ _s = 4615 μs, consisting of 8 slots with a duration of τ _k = 577 μs each, is the basis of the hierarchical separation of the frame structure. The periodic structure of the time intervals of the GSM standard signal leads to the fact that the auto-convolution spectrum of the GSM signal is linear and has, along with the central harmonic, a series of periodically following harmonics at multiple frequencies

and

The experimentally obtained harmonics of the spectrum of the GSM signal auto-convolution shown in figa and b overlap the expected range of Doppler frequencies up to 1700 Hz and become a nuisance when detecting low-altitude targets (MVC) flying at speeds up to 900 km / h. Moreover, as the calculations show [2], the main lobe of the auto-convolution spectrum of the GSM signal is 20-30 dB higher than the amplitude of the first periodic harmonics at frequencies multiple of 216.6 Hz and 1732.8 Hz. It is known that interfering interference in separated radars with non-directional illumination are direct transmitter signals penetrating the side lobes of the antenna pattern (BOTTOM) of the target channel, and reflections from local objects located together with the target in the main beam of the BOTTOM and one range resolution element.

In this case, the spectral harmonics of convolution of interfering and reference signals at the center frequency and frequencies that are multiples of 216.6 Hz and 1732.8 Hz will have the most powerful interference effect on the detection process. However, in the description of the prototype, as a method and means of additional compensation for penetrating noise, a filter with an amplitude-frequency characteristic (AFC) of a notch is described only at the central convolution frequency. Therefore, without compensation for the periodic harmonics of the auto-convolution spectrum, the additional coefficient of suppression of the penetrating signal by 30-40 dB declared in the prototype is hardly achievable.

Secondly, the given model of the GSM signal is based on the assumption that the envelope of the signal spectrum is constant. In fact, the envelope of the spectrum of the GSM signal is not limited to OMSK modulation and is not stationary (Fig. 2, a). In addition, the structure of the GSM signal is strictly determined by the temporal characteristics of the envelope of the signal emitted by the packets on the channel interval of the TDMA frame. The temporal envelope mask for the graph interval signal (voice and data) of the full TDMA frame is shown in FIG. 2, b. [one]. Therefore, when the content of the graph changes, the envelope of the signal spectrum also changes. This leads to the fact that the correlation coefficient of the signals will depend on the graph interval and its intensity. In addition, a change in the spectral parameters of the GSM signals with a slight discrepancy between the identity of the main and compensation paths leads to decorelation of the signals. So, if the discrepancy between the electrical parameters of the paths is currently technically extremely achievable at 1%, the maximum achievable coefficient of suppression of stationary noise interference does not exceed 30 dB. Therefore, the suppression coefficient of the interfering non-stationary GSM signal will not exceed 25 dB. In addition, changing the envelope of the signal will require stabilizing the probability of a false alarm. Therefore, along with spatial and frequency selection, to ensure the required detection performance, it is necessary to use stabilization threshold detection devices.

Thirdly, providing a constant value of the probability of correct detection under conditions when the target's bistatic effective scattering surface (EPR) varies in the range of tens of dB [6], requires the use of a device for expanding the dynamic range of the receiver. The absence of such a device in the prototype will not allow to achieve the stated reduction in the radar criticality to placement relative to the BS.

The aim of the invention is:

1. An increase in the detection range of bistatic radar targets with third-party illumination created by a GSM base station by suppressing the periodic components of passive interference generated by the frame and slot structures of the signal reflected from local objects.

2. Ensuring the constancy of detection indicators through the use of devices to stabilize the detection threshold and logarithmic amplification.

This goal is achieved by the fact that in a semi-active radar containing a series-connected survey antenna with a control device, a superheterodyne receiver of the target channel, a series-connected antenna and a superheterodyne receiver of the reference channel, a local local oscillator, to the input of which the second inputs of the superheterodyne receivers are connected, and the antenna of the target channel an overview of the space, and the antenna of the reference channel is fixed and oriented to the backlight BS, an auto-compensator, the main input of which is connected n to the output of the superheterodyne receiver of the target channel, and the compensation input is connected to the output of the superheterodyne receiver of the reference channel, M combined at the input of identical range channels, each of which contains a channel multiplier, which is the input of the range channel, and a set of N combined at the input of narrow-band filters (UPF) the outputs of which form N outputs of the corresponding long-range channel, a terminal device (DU), to the N inputs of which the same-name outputs of the long-range channels are connected, M-channel form a reference signal trimmer containing an M-tap delay line, the input of which is the input of the M-channel former, as well as a reference local oscillator, additionally introduced in each channel of the M-channel former are a reference multiplier, intermediate frequency amplifiers of the target channel and the reference channel with a logarithmic gain characteristic (INCL) at the frequency of amplification, and the output of the autocompensator is connected to the input of the INCL of the target channel, and the first inputs of all channel residents, in each range channel, a coarse selection channel filter (FGS) and a subtracting device, the output of which is connected in parallel to the input of the UPF set, the input of the coarse channel filter is connected to the output of the channel multiplier, the GPRS modem is connected to the output of the terminal device , in each channel of the M-channel reference signal driver, a mixer, a band-pass filter, as well as a coarse reference filter and a controlled amplifier, as well as select the filter and the amplitude detector, the output of which is connected to the second input of the controlled amplifier, the output of the bandpass filter of the ith channel of the reference signal shaper is connected to the first input of the reference multiplier, as well as to the second input of the channel multiplier of the same range channel, the output of the controlled amplifier of the ith the channel of the driver of the reference signal is connected to the second input of the subtracting device of the same range channel, the output of the subtracting device of the i-th range channel is connected to the input to the selector filter of the channel of the same name as the reference signal driver, the output of the reference channel amplifier is parallel connected to the input of the M-tap delay line and to the second input of the reference multipliers of all channels of the M-channel driver of the reference signal, the output of the reference local oscillator is connected in parallel to the heterodyne inputs of the mixers, and the signal inputs mixers of each channel of the driver of the reference signal are connected to the corresponding tap of the M-tap delay line.

The given set of features is absent in the studied patent and scientific and technical literature on this issue, therefore, the proposed technical solutions meet the criterion of "novelty."

The invention is illustrated in FIG. 1-7.

Figure 1 - experimental spectra of auto-reconstructions of the GSM standard signal at various frequency scales.

Figure 2 - real spectrum a) and a temporary mask b) of the GSM standard signal.

Figure 3 - radio channels of the passage of signals.

Figure 4 is a block diagram of the inventive radar.

5 is a typical amplitude characteristic and the dependence of the gain of the amplification amplifier from the input voltage.

6 is an amplitude-frequency characteristic of FGS.

7 is a block diagram of a terminal device.

The equipment of the diversity radar station in Fig. 4 consists of a series-connected survey antenna 1 with a control device and a superheterodyne receiver 2 of the target channel, serially connected antennas 3 and a superheterodyne receiver 4 of a reference channel, a local local oscillator 5, to the input of which second inputs of the superheterodyne receivers 2 and 4, and the antenna 1 of the target channel surveys the space, and the antenna 3 of the reference channel is fixed and oriented to the backlight BS, auto-compensator 6, the main the input of which is connected to the output of the superheterodyne receiver of the target channel 2, and the compensation input is connected to the output of the superheterodyne receiver of the reference channel 4, amplifiers of the intermediate frequency of the target channel 7 and the reference channel 8 with a logarithmic gain characteristic (AML) at the amplification frequency, and to the input of the AML of the target channel 7, the output of the autocompensator 6 is connected, and the output of the superheterodyne receiver of the reference channel 4, M-connected at the input and connected in parallel to the output, is connected to the input of the AMCH of the reference channel 8 UPCL of the target channel 7 identical range channels 9, each of which contains a series-connected channel multiplier 10, which is the input of the range channel, coarse channel filter (FGS) 11, subtractor 12, a set of N combined at the input of narrow-band filters (UPF) 13, outputs which form N outputs of the corresponding range channel, the terminal device (OS) 14, to the N inputs of which are connected the same outputs of the range channels, a GPRS 15 modem connected to the output of the terminal device, the M-channel the first driver of the reference signals 16, containing the M-tap delay line 17, the input of which is the input of the M-channel driver, as well as the reference local oscillator 18 and M identical channels 25, and each of the channels of the M-channel driver 25 contains a series-connected mixer 19, a strip a filter 20, a reference multiplier 21, a coarse reference filter 22 and a controlled amplifier 23, as well as a selection filter 24 and an amplitude detector 26 connected to the second input of the controlled amplifier 23, the output of a band pass track 20 of the i-th channel of the reference signal driver is connected to the second input of the channel multiplier 10 of the same range channel, the output of the controlled amplifier 23 of the i-th channel of the reference signal generator is connected to the second input of the subtractor 12 of the same range channel, the output of the subtractor 12 of the i-th range channel is connected to the input of the filtering filter 24 of the channel of the same name of the reference signal shaper, the output of the AMCH of the reference channel 8 is connected in parallel to the input of the M-tap delay line 17 and to oromu input of multipliers 21 supporting all the channels M-channel reference signal generator 16, the output of the reference oscillator 18 is connected to the parallel inputs of heterodyne mixers 19, and signal inputs of the mixers 19, each channel of the reference signal generator connected to the appropriate allotment of M-tap delay line.

и боковых лепестков

. Диаграмма направленности антенны опорного канала является остронаправленной и характеризуется коэффициентом усиления С_ОК. При этом в основном луче ДНА 32 целевого канала в одном элементе разрешения по дальности оказываются эхо-сигналы, отраженные от цели m(t) 33 и отраженные от местных предметов s(t) 34. Кроме того, через боковые лепестки ДНА антенны целевого канала 1 на входе супергетеродинного приемника целевого канала 2 попадает прямой проникающий сигнал передатчика БС n(t) 35, представляющий собой мощную проникающую помеху.The inventive diversity radar station with third-party illumination of cellular networks of the GSM standard works as follows (figure 3). An arbitrary j-base station 28 of the GSM standard cellular network generates an omnidirectional or slightly directional (depending on the type of BS antenna) radiation field 29. In the radiation field of the BS there may be airborne location objects - radar targets 30 and closely located passive interfering reflectors 31. Object detection location is carried out by the passive module of the radar station 27, containing the target and reference channels, which form the radiation patterns (BOTTOM) of the target 32 and the reference channel 37. The antenna pattern of the target channel 32 provides the required resolution in angular coordinates and is characterized by the gain of the main beam

and side lobes

. The radiation pattern of the antenna of the reference channel is pointed and characterized by a gain of _OK . In this case, in the main beam of the DND 32 of the target channel in one range resolution element, echoes are reflected from the target m (t) 33 and reflected from local objects s (t) 34. In addition, through the side lobes of the bottom of the antenna of the target channel 1 at the input of the superheterodyne receiver of the target channel 2, a direct penetrating signal of the BS transmitter n (t) 35 is received, which is a powerful penetrating interference.

Thus, an additive signal coupling acts on the receiver of the target channel 2

Through the main lobe of the bottom of the reference channel 37 at the input of the superheterodyne receiver of the reference channel 4 gets a direct BS signal n _o (t) 36.

where n ₀ (t) = N (t) exp (j [ω _o t + Σ _i Σ _j Ψ (t-iτ _k -jτ _с )]);

N (t) = U _n (t) exp [jφ _n (t)] is the complex envelope of the reference signal;

ω ₀ = 2πf _o t - carrier frequency of the base station;

Σ _i Σ _j Ψ (t-iτ _k -jτ _s ) is the phase modulation law of the GSM standard taking into account periodic, personnel and slot structures [7];

φ _n (t) is the initial phase;

;

;

m (t) = M (t) exp (j [ω _o t + Σ _i Σ _j Ψ (t-iτ _k -jτ _c -τ _r ) -2πF _d C]);

τ _r is the delay time of the echo signal relative to the reference signal;

F _d - Doppler frequency.

M (t) = U _m (t) exp [jφ _m (t)] is the complex envelope of fluctuations of the echo signal;

S (t) = S (t) exp (j [ω _o t + Σ _i Σ _j Ψ (t-iτ _k -jτ _c -τ _c )]);

S (t) = U _s (t) exp [jφ _s (t)] is the complex amplitude of the envelope of the fluctuation of the signal reflected from the local object;

τ _s is the delay time of the signal reflected from the local object relative to the reference signal. Under the condition τ _s = τ _{r, the} signal reflected from the local object becomes a passive interference.

. Процесс компенсации происходит линейно без изменения закона фазовой модуляции. Реально достижимая автокомпенсатором эффективность подавленияIn superheterodyne receivers 2 and 4, the signals are amplified and converted to an intermediate frequency ω _pc . In the auto-compensator 6, spatial compensation is carried out - suppression of the penetrating noise n (t) in the main channel by a single subtraction of the reference signal n _o (t) received by the reference channel superheterodyne receiver 4 from it. In the autocompensator, the signal n _o (t) is used as a compensation signal, correlated in amplitude and phase, so as to ensure the minimum power of the suppressed signal n (t) at the output of the autocompensator 6. This operation is equivalent to the formation of a dip in the side lobes of the bottom of the target channel 32 in the direction of the emitted BS [8] with level

. The compensation process occurs linearly without changing the law of phase modulation. Autocompensator Realizable Suppression Efficiency

unsteady penetrating interference does not exceed 25 dB when the ratio of the power of the echo signal to the power of the drooping interference 70-90 dB [4]. Since the compensation process in the autocompensator is linear, the unsuppressed remnants of the penetrating signal n _AK (t) are described by phase modulation and the complex amplitude envelope as well as the direct signal n _о (t). The amplitude of the complex envelope is determined by the formula

The output signal of the autocompensator 6 contains a signal reflected from the target m (t), a partially suppressed penetrating signal n _AK (t) and a signal reflected from the local object s (t) is fed to the input of the amplification target channel 7. Similarly, from the output of the superheterodyne reference receiver channel 4 ₀ received signal n (t) is input to the reference channel UPCHL 8. The amplifiers 7 and 8 have a logarithmic gain characteristic for the intermediate frequency and dynamic range of 80 dB gain to [9], and a bandwidth of 200 kHz, which corresponds to the spectral width of the signal to the BS. A typical amplitude characteristic and the dependence of the gain on the input voltage are shown in Fig.5. UPCL allows you to compress the amplitude of the fluctuation of the interfering oscillations to the level of the amplitudes of the internal noise, and thereby prevent overloading of the receiving device without losing sensitivity when exposed to intense interference. Features of amplification with amplification at an intermediate frequency are:

1) the linearity of amplification of weak signals before the onset of the logarithmic section, which allows for high sensitivity of the receiving device;

2) inertia-free gain;

3) the possibility of attenuation of strong signals up to the intrinsic noise of the receiver;

4) the possibility of coherent processing after amplification.

The listed features of the AMCH due to the spectrum convolution features of the GSM signal can significantly reduce the requirements for the amount of compensation of unsuppressed residues of penetrating and passive interference. At the output of the IFA of the target channel 7, the weak signal reflected from the target m (t) is amplified, and the remnants of the unsuppressed n _AK (t) and interference reflected from the local object s (t) are attenuated. In this case, the amplitudes of the echo and interfering signals are aligned, and the additive mixture of signals can be represented as:

f _ccl (t) = log [s (t)] + log [m (t)] + log [n (t)].

From the output of the UCHL of the reference channel 8, the direct BS signal is fed to the input of the M-channel driver of the reference signal 16, namely, to the input of the M-tap delay line and the second inputs of the reference multipliers 21 of each of the M channels of the driver of the reference signals 16. At the outputs of the elements of the M-tap delay line 17, the BS direct signal n _o (t) is sequentially delayed by a time Δτ _s equal to the correlation interval

where Δf _c = 200 kHz is the width of the BS spectrum, and is fed to the signal input of the mixer 19 of the channel of the same name of the reference signal driver 16. The voltage of the reference local oscillator 18 with a frequency ω _r is applied to the heterodyne input of the mixers 19. At the same time, at the input of the band-pass filter of the 20th channel of the reference signal conditioner, the reference signal lg [n _oc (t)] acts

where n _oс (t) = N _oс (t) exp (j [ω _c t + Σ _i Σ _j Ψ (t-iτ _c -jτ _c -Δτ _Зк )]);

N _oc (t) = U _oc (t) exp [jφ _n (t)] is the complex envelope of the reference signal;

ω_c= ω_be-ω_g= ω_from= 2πf_fromt is the second intermediate frequency selected from conditions f_c= (5-10) F_dmax;

F _dmax - the maximum value of the Doppler frequency.

The bandpass filter 20 has a passband Δf _pf = f _c ± 2F _dmax . The voltage from the output of the bandpass filter 20 is supplied to the second input of the channel multiplier 10 of the k-th range channel and to the first input of the reference multiplier 21. An additive signal mixture is formed in the channel multiplier 10:

An additive signal mixture is formed in the reference multiplier 21

The additive mixture of signals 5 contains:

lg [n _AK (t)] lg [n _oc (t)] - convolution of the logarithms of the reference and unsuppressed remnants of the penetrating signals. This convolution is not demodulated on the observation interval, since in the general case the delay time of the signals is different, and will be an interfering background that requires additional compensation;

lg [m (t)] lg [n _oc (t)] is the convolution logarithm of the reference and echo signals, as a result of which, under the condition τ _r = _{Δτ zk} , demodulation and compression over the spectrum of the echo signal, as well as the formation of the maximum convolution value proportional to the effective target surface (EPR), at a frequency fc ± Fд. This convolution contains information about the speed and range of detected objects. In this case, the periodic components of the convolution at frequencies fc ± Fd ± 216.6 · n ± 1732.8 · n, Hz, where n is an integer, will be 20-30 dB less (Fig. 1) [2] convolutions at a frequency fc ± Fd.

lg [s (t)] lg [n _oc (t)] is the logarithm of the convolution of the reference signal and passive interference, as a result of which, under the condition τ _s = _{Δτ sz} , demodulation and compression over the spectrum of the passive interference signal at a frequency fc occurs. In this case, the harmonic amplitude at frequencies fc ± 216.6 · n ± 1732.8 · n. less than the convolution amplitude at the frequency fc by 20-30 dB. The ratio of the powers of convolutions of the third and second terms of equation 5 is proportional to the ratio

The ratio (7) can reach values of 40 or more dB.

In the second equation, the convolution logarithm of the reference and direct BS signal is formulated:

log [n _o (t)] log [n _oc (t)] - the convolution logarithm differs from the third term in equation (5) by the value of the complex amplitude of the signals N _AK (t) and N (t). These differences up to PCL are determined by the relation

The complex amplitudes of the signals N _AK (t) and N (t) are quite large and fall under the logarithmic section of the amplitude characteristic of the PCA. When passing through the PCA, the complex amplitudes of the envelopes N _AK (t) and N (t) are limited to the logarithmic section by the amplitude characteristic and slightly differ in amplitude. In this case, relation (8) takes the form:

Given the logarithmic limitation of passive interference and (7), we can assume that

The convolution amplitudes log [n _AK (t)] log [n _oc (t)] and log [n _o (t)] log [n _oc (t)] will slightly differ in value and have the same laws of change. From equation (10) it follows that, provided τ _s = Δτ _h, the convolution amplitudes log [n _AK (t)] log [n _oc (t)] and log [s (t)] log [n _oc (t)] will be are the same. The complex amplitude of the echo signal at the output of the IFA with an intensive change in the amplitude of the target at the input (due to a change in range, EPR, or viewing angle) is limited by the logarithmic characteristic and changes insignificantly. This ensures a constant amplitude and probability of correct detection. From the outputs of channel 10 and reference 21 multipliers, the results of convolutions f ₁₀ (t) and f ₂₁ (t) are fed to the inputs of channel 11 and reference 22 FGS, respectively.

Channel filter coarse selection has the frequency response shown in Fig.6, a. The frequency response of the channel FGS has a center frequency, a passband Δf ₁₁ = ± 2F _dmax , and also notch zones at frequencies fp = fc ± 216ξ, where ξ∈ [0 ... ξ _max ], ξ _max = F _dmax / F _dmax / 216.6 - the number of zones of rejection; F _dmax = 2V _max / λ is the maximum value of the Doppler frequency, λ is the wavelength of the GSM signal, V _max is the maximum value of the speed of a low-altitude target. So at the maximum speed of a low-altitude target 500 km / h, wavelength λ = 0.32 m, the number of notch zones in one frequency direction is ξ = 4.

The most powerful harmonics of the convolution of the reference signal and passive interference lg [s (t)] lg [n _oc (t)] at frequencies that are multiples of 216.6 Hz (FIG. 6 a) fall into the rejection zones of the channel FGS.

Passive interference suppression coefficient determined by the formula

Where

S _p (f) is the normalized energy spectrum of convolution of the logarithms of the reference signal and passive noise lg [s (t)] lg [n _oc (t)];

- квадрат АЧХ зоны режекции канального ФГС 11 может достигать значений 30-40 дБ. При таком значении подавления помеховую составляющую lg[s(t)]lg[n_oc(t)] можно исключить из процесса обработки.

- the square of the frequency response of the rejection zone of the channel FGS 11 can reach values of 30-40 dB. With this suppression value, the interference component log [s (t)] log [n _oc (t)] can be excluded from the processing.

Within the ± 2F _{dmax band of the} channel FGS 11, the results of the convolution of the reference and unsuppressed remnants of the penetrating noise lg [n _AK (t)] lg [n _oc (t)], as well as the results of the convolution of the reference and echo signals lg [m (t )] lg [n _oc (t)]. In the particular case when the Doppler addition of the echo frequency is a multiple of 216.6 Hz, the central harmonics of the mutual convolution of the reference and echo signals lg [m (t)] lg [n _oc (t)] will fall into the notch zone. This disadvantage can be eliminated by setting the spatial reference beam antenna 37 to another base station 28i (Figure 3), in which the illumination field is objective and the restructuring of the local oscillator 5 in such a way that the value of the intermediate frequency ω _IF would remain unchanged. The dependence of the Doppler frequency in a bistatic radar on the geometry and target orientation relative to the base line is known [6]. In this case, the central convolution of the echo reconciliation will exit the notch zone.

The reference FGS 22 in each of the M-channels of the driver of the reference signal do not have zones of rejection and cover the frequency range of ± 2F _dmax ( _Fig.6 , b).

In the general case, the convolution of the unsuppressed remnants of the penetrating noise lg [n _AK (t)] lg [n _oc (t)] appears along with the useful convolution outside the zone of rejection of the channel FGS 11. This convolution provides an unsteady echo detection background to be excluded from further processing. For this, it is necessary to ensure the formation of an adaptive threshold detection level (AAP) of detection. The principle of implementation of the APA is to subtract the interference convolution lg [n _AK (t)] lg [n _oc (t)] in the subtractor 12 from the output voltage of the channel FGS. The first input of the subtracting device 12 receives voltage

The second input of the subtractor 12 receives the voltage from the output of the controlled amplifier 23

where K _p - regulation coefficient of the controlled amplifier 23.

Since, under the condition τ _r = _{Δτ cc,} the convolution structures log [n _AK (t)] log [n _oc (t)] and log [n _o (t)] log [n _oc (t)] are the same, when these voltages are applied to the corresponding inputs at the input of the subtracting device 12 we get the difference:

Provided

in the second term lg [n _AK (t) / n _oc (t) ^Кр ] → 0 and in equation 14 only the term due to the convolution of the echo signal should remain:

f ₁₂ (t) = log [m (t)] log [n _oc (t)].

The implementation of condition 15 is carried out under the condition that the regulatory coefficient changes according to the law

Since the logarithms of the amplitude of the signals n _AK (t) and n _oc (t) are approximately equal, then K _p → 1. In addition, due to the influence of the logarithmic amplitude characteristics, fluctuations will be insignificant. At the same time, it is impossible to effectively stabilize the probability of false alarm by compensating for non-stationary unsuppressed residuals of penetrating interference lg [n _AK (t)] in subtractor 12 without monitoring the result of compensation. To do this, from the output of the subtractor, voltage 14 through the selection filter 24 and the amplitude detector 26 is fed to the second input of the controlled amplifier 23. The selection filter 24 has a passband Δf ₂₄ = F _dmax -2F _dmax , which is outside the expected range of Doppler frequencies, which excludes the possibility of regulation the amplifier 23 on the convolution of the echo signals lg [m (t)] lg [n _oc (t)]. The detector 26 selects the envelope of the diversity voltage 14, which is fed to the second input of the adjustable amplifier 23. With an increase in the differential voltage, the regulation coefficient K _p decreases, reducing the magnitude of the differential voltage. The expected real coefficient of suppression of unsuppressed remnants of non-stationary penetrating noise n _AK (t) will be 20-25 dB. Taking into account the suppression in the auto-compensator, the total suppression coefficient of penetrating interference will be 50-60 dB. In the absence of AECL, this value would not be sufficient to provide the required indicators of detection quality. However, after passing the convolution voltage of the IFA and compressing the amplitude of the penetrating noise, the signal-to-noise ratio at the output of the narrow-band filters 13 provides the required indicators of detection quality for the following reasons:

firstly, at the output of the PCA, the convolution amplitude lg [n _AK (t)] lg [n _oc (t)] of the interfering signal slightly exceeds the convolution lg [m (t)] lg [n _oc (t)] of the echo signal, therefore, additional compensation of unsuppressed residues by 20-25 dB significantly improves the detection conditions;

secondly, as shown by calculations [2] and real measurements (Fig. 1), with coherent filtering in a set of narrow-band filters of Doppler filters 13, the central peak of the convolution of the echo signals exceeds its own side peaks by 20–25 dB;

thirdly, due to the fact that the unsuppressed convolution residues lg [n _oc (t)] lg [n _AK (t) n _oc (t) ^Kr ] are distributed in the signal band Δf _c = 200 kHz, and the useful convolution component lg [ m (t)] log [n _oc (t)] of the echo signal is coherently allocated in the Doppler filter band with the band Δf _f << Δf _s , then the selection of the useful signal against the background of unsuppressed residues will be carried out with the efficiency of the correlation accumulation coefficient K _n = Δf _s / Δf _f . With a 20 Hz narrowband filter bandwidth, the gain in signal-to-noise ratio can be 10 dB.

Moreover, as shown by calculations [2] and real measurements (Fig. 1), the level of the central peak of the convolution spectrum of the GSM signal exceeds its own side peaks that are multiples of the frame repetition rate of 216.6 Hz by 20-25 dB.

From where, due to additional compensation in the subtractor and coherent accumulation, the signal-to-noise ratio at the output of the narrow-band filter can be 30-35 dB, which is enough to detect targets with specified quality indicators.

Thus, the use of a control amplifier with a logarithmic amplitude characteristic ensures a constant amplitude of the echo signal and, therefore, the probability of correct detection, and the use of a subtractor in each range channel ensures a constant probability of false alarm.

Coherent filtering convolution of the echo signal is carried out in a set of N combined at the input of narrow-band filters 13 at the Doppler frequency. The bandwidth of each filter Δf _f determines the accumulation time T _ns = 1 / Δf _f and the speed resolution ΔV = λ / 2Δf _f . The passband of each filter from a set N of narrow-band filters 13 combined at the input is determined by the width of the central peak of the auto-spectrum and does not exceed 20-25 Hz. The number of narrow-band filters is selected from the condition

Signal analysis in the entire Doppler range of targets from F _dmin to F _dmax .

The number of range channels M is determined by the ratio

Where

- разрешающая способность РЛС по дальности;

- resolution radar range;

D _max -D _min - the measured range.

The terminal equipment of Fig. 7 contains M-switches 39 outputs of filters 38 of Doppler frequency connected to the outputs of the corresponding N combined at the input of narrow-band filters (UPF) 13 of the corresponding range channels, and the outputs of these switches are connected to the corresponding M inputs of the switch by a delay time of 40 τ _g , the output of which through the detector 44 is connected to the signal inputs of the indicators “direction-speed” 50 and “direction-range" 47 with amplitude indication. In this case, the control input of the indicator 50 is combined with the control inputs of the switches 39 and connected to the output of the review unit 41 by Doppler frequency, and the control input of the indicator 47 is combined with the control input of the switch 40 and connected to the output of the review unit 42 by the delay time, and the input of block 42 is combined with the control inputs of the indicator 50 and the M-switches 39 and is connected to the output of the unit 41 of the review at the Doppler frequency. Block 41 generates a cyclically changing digital code with values from 0 to N, the value of this code, entering the control inputs of the switches 39, determines the input numbers from the Doppler filter 38 connected to the output of each of the switches 39, that is, the value of this code corresponds to the Doppler frequency of the connected to the output of the signal switch and, accordingly, the radial speed displayed on the indicator 50. The zero value of the code ensures the transfer to a new state of the block 42, forming a digital code with a value from 0 to M, which, entering the control input of the switch 40, determines the number of the switch 40, the output signal from which is connected to the detector 44. Since the switches 39 ₁ ... 39 _m connected to the corresponding long-range channels, this code value determines the number of the delay time interval displayed on the indicator 47, since the code value sets the sweep position in the time coordinate of the indicator 47, as well as the sweep position according to the coordinate coordinate of the indicator 50 is determined by the code value from the output of block 42. The output signal of the detector 44 is supplied to the signal inputs of the indicators 50 and 47, providing an amplitude indication of the signals on the indicator screens and deciding whether there is a target in the direction of the orientation of the radar antenna diagram 1, its radial speed and delay time, characterizing the range to the target. A sweep in the direction is provided by a digital code that arrives at the corresponding inputs of the indicators via the corresponding links from the output of the antenna driver of the angular position code 43 of the antenna, the input of which is connected to the output of the antenna control device 1. The inputs of the sweeps according to the radial speed and delay time of the indicators 50 and 47 are respectively connected to the outputs of blocks 41 and 42, and the inputs of the brightness indication to the output of the detector 44. In this case, the formation of a two-dimensional scan on the indicators 50 and 47 with a brightness indication of the target position, which provides a decision on the presence of the target, its speed and range in the conditions of a changing direction of the antenna 1. The threshold circuit 45 is connected to the output of the detector 44, the output of which is connected to the control inputs of the gating unit 46 with an analog-to-digital converter (ADC) and block 48 gating codes, the outputs of these blocks are connected to the corresponding inputs of the Central computing complex (CVC) 49 radar. In this case, the signal input of block 46 is connected to the output of the detector 44, and the digital inputs of block 46 are connected to the outputs of blocks 41, 42, 43, respectively. If the output signal of the detector exceeds the threshold level, the threshold circuit 45 generates a signal for the presence of the target, which, upon entering the control input of block 46, provides the allocation of the values of the currently existing Doppler frequency codes, delay time, and direction to the target, which are received at the first input of CVC 49 and via the data bus from the output of the CVC to the GPRS 15 data transfer modem. The CVC performs the necessary calculations to form weighted coordinate estimates, accumulate decisions on exceeding the threshold and form the integral decisions on the availability of targets, trajectory calculations, antenna control during a program review, as well as the issuance of radar information. All of the listed nodes forming the terminal device 14 are known, are used in radar, technically feasible and meet the criterion of "industrial applicability". The authors do not claim to be new technical solutions of the terminal device.

Information about the study of the claimed subject matter of the invention in patent and scientific literature.

Diversity radar station with third-party illumination of GSM standard cellular networks

Patent Documentation Countries I searched IPC core index Type of patent information sources used Search depth Identified analogues USA B64F 1/36; G01S 13/00; G01S 13/46; G01S 13/50; G01S 13/91; G01S 19/48; G01S 5/12; G08B 21/00 IPS FIPS, VET Foundation for the Universal Scientific Library named after A.M. Gorky in Tver. http://books.google.com/patents/ 1993-2013 Pat. USA No. 7012552 B2, No. 6930638 B2 European Union G01S 13/00; G01S 13/46; G01S 13/50; http://worldwide.es pacenet.com/ 1993-2013 EP No. 2065729 A2 Eurasian Patent Organization G01S 3/00; G0S 13/46; G0S 13/58; G0S 13/92 http://www.eapo.org/en/patents/reestr 1993-2013 Evraz. patent No. 008335 (prototype)

Scientific and technical literature.

1. Popov V.I. The basics of cellular communications standard GSM. - M .: Ecotrades, 205-296.

2. M. Chemiakov, R. Saini, M. Antoniov, R. Zuo, and J. Edwards, SS-BSAR with transmitter of opportunity-Practical Aspects. Electrical and Computer Engineering University of Birmingham, 2006.

3. Proskurin V.I. Assessment of the requirements for the linearity of the receiving path of an active-passive radar. // Radio engineering (Magazine in the journal). 2011, No 1. S.80-83.

4. Okhrimenko EA, Parkhomenko NG, Semashko P.G. Methods for suppressing a direct signal in radars illuminated by broadcast transmitters // Electromagnetic waves and electronic systems No. 5, v.16, 2011. P.81-82.

5. Lebedev E.P., Chelpanov A.S. Kuz N.Ya. The influence of the non-identity of the reception channels on the quality of interference compensation by the correlation auto-compensator // Transactions of Academy No. 72. Kharkov: ARTA, 1966. P.57-65.

6. Averyanov V.Ya. Diversity radar stations and systems. - Mn .: Science and technology, 1978.

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Claims (1)

A GSM radar station with third-party illumination of cellular networks containing a series-connected survey antenna with a control device, a target channel superheterodyne receiver, a reference channel superheterodyne receiver, a local local oscillator, to the input of which second inputs of the superheterodyne receivers are connected, and the target channel antenna provides an overview of the space, and the antenna of the reference channel is fixed and oriented to the backlight base station, by auto A sensor whose main input is connected to the output of the target channel superheterodyne receiver, and the compensation input is connected to the output of the superheterodyne receiver of the reference channel, M combined at the input of identical range channels, each of which contains a channel multiplier, which is the input of the range channel, and a set of N combined at the input narrow-band filters (UPF), the outputs of which form N outputs of the corresponding long-range channel, a terminal device designed to decide on the availability of , its speed and range, to the N inputs of which the same outputs of the long-range channels are connected, an M-channel driver of reference signals containing an M-tap delay line, the input of which is an input of an M-channel driver, and also a reference local oscillator, characterized in that GPRS modem connected to the output of the terminal device, amplifiers of the intermediate frequency of the target channel and the reference channel with a logarithmic gain characteristic (AML) at the amplification frequency, and to the input of the AML of the target channel the output of the autocompensator is turned on, and the first inputs of all channel multipliers are connected in parallel to the output of the output channel of the target channel, the channel coarse filter (FGS) and a subtractor, the output of which is parallel connected to the input of the UPF set, are connected in parallel to each input of the channel multiplier, and the input of the channel filter of coarse selection connected to the output of the channel multiplier, each channel of the M-channel former further comprises a reference multiplier connected in series with the mixer , a bandpass filter, as well as a coarse reference filter and a controlled amplifier, as well as a selection filter and an amplitude detector, the output of which is connected to the second input of the controlled amplifier, and the output of the bandpass filter of the ith channel of the reference signal shaper is connected to the first input of the reference multiplier, and also to the second input of the channel multiplier of the same range channel, the output of the controlled amplifier of the i-th channel of the reference signal conditioner is connected to the second input of the subtractor one the nominal range channel, the output of the i-th range channel subtractor is connected to the input of the selector filter of the channel of the same name as the reference signal conditioner, the output of the reference channel AML is connected in parallel to the input of the M-tap delay line and to the second input of the reference multipliers of all channels of the M-channel reference signal shaper , the output of the reference local oscillator is connected in parallel to the heterodyne inputs of the mixers, and the signal inputs of the mixers of each channel of the driver of the reference signal are connected to the corresponding tap of the M-tap delay line.

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