RU2294052C9 - Method for correcting nonlinear distortions in electric signal amplifier - Google Patents

Abstract

FIELD: radio engineering.

SUBSTANCE: proposed method includes parallel amplification of signals in first and second amplifier channels followed by shaping output difference signal from amplified ones. Quiescent currents I₀₁ of first-channel amplifying component are set to be higher than quiescent current I₀₂ of second-channel amplifying component. In the process either first-channel input signal is attenuated by K₁ times when using bipolar transistors, or second-channel signal is attenuated by K₂ times when using tubes and field-effect transistors, or second-cannel output signal is attenuated by K₃ times in latter of mentioned cases:

where n , m are numbers of parallel-connected amplifying components (or their pairs in case of push-pull operation) of first and second channels, respectively; α = 2 for field-effect transistor, α = 3/2 for tubes.

EFFECT: reduced nonlinear distortions due to correction of third harmonic and reduction of higher harmonics.

2 cl, 8 dwg

Description

The invention relates to radio engineering, to amplifiers of electrical signals, and can be used in amplifiers of audio frequency.

A known method of reducing non-linear distortions, widely used in amplification technique, the introduction of linear negative feedback [1, p. 33].

The method increases the linearity of the amplifier. However, at the same time, the introduction of negative feedback generates interactions of nonlinearity products of various orders and signal. In addition, the known method is inherent in the manifestation of the so-called dynamic distortion.

A known method of compensating for nonlinear distortion in an amplifier of electrical signals by parallel amplification of signals on two amplifier elements with identical pass-through characteristics and subsequent subtraction of amplified signals at the amplifier load. The quiescent currents of the amplifying elements are set equal. And the input signal to them is fed in the opposite polarity (in antiphase) [4, p. 59].

The method is widely used in amplification technology and is implemented in push-pull amplifiers on a variety of amplifying elements: lamps, bipolar and field effect transistors. The method allows you to compensate for the even harmonics of the nonlinear distortion of the signal.

The disadvantage of the analogue is the inability to compensate for the odd harmonics of nonlinear distortion of the signal. However, it is known that in many applications it is the odd harmonics that are most harmful, for example, in amplification paths of AM signals, sound amplifiers, etc.

A known method of compensating for non-linear distortions in amplifiers of electrical signals by parallel amplification of signals in two amplification channels (with matching transfer (amplitude) characteristics) and subsequent formation of a differential output signal of the amplifier from the amplified channels of the signals. In this case, the input signal of one of the channels is attenuated by n-decibels, the output signal of the other channel is attenuated by 3n-decibels, and the difference output signal is formed by amplifying common-mode signals in the channels and subtracting one amplified signal from the other [3].

The method allows you to compensate for the third harmonic and reduce intermodulation distortion of the output signal.

This method is adopted as a prototype.

The invention solves the problem of expanding the arsenal of technical means aimed at increasing the linearity of amplification paths by compensating for odd harmonics.

To achieve this technical result, a method for compensating non-linear distortions in an electric signal amplifier includes parallel amplification of signals in the first and second amplification channels and subsequent generation of a differential output signal of the amplifier from the signal amplified in the channels. The method also includes the step of attenuating the input signal of one of the channels and the step of attenuating the output signal of one of the channels. In contrast to the prototype, in the method, the quiescent currents of the amplifying elements of the first channel are set to more quiescent currents of the amplifying elements of the second channel. At the same time, the input signal of the first channel is weakened - in the case of using amplifying elements with a passage characteristic approximated by an exponential function, or the input signal of the second channel is weakened - if amplifiers with a passage characteristic of an approximated power function are used, or weakening the output signal of the second channel in the last from the mentioned cases.

For the cases mentioned in the method, either the input signal of the first channel is attenuated by K ₁ time, or the input signal of the second channel is attenuated by K ₂ times, or the output signal of the second channel is attenuated by K ₃ times:

where n, m are the numbers of parallel-connected amplifying elements (or their pairs with push-pull amplification), respectively, of the first and second channels;

I ₀₁ , I ₀₂ - quiescent currents of amplifying elements of the first and second channels, respectively;

α is an exponent of the power function approximating the pass-through characteristic of the amplifying element.

(The above relations allow us to achieve the minimum level of the third harmonic in the spectrum of the output signal of the amplifier.)

The differential output signal of the amplifier is formed by amplification in the first and second channels of mutually inverted signals and the subsequent summation of the amplified signals.

(The feature allows you to most easily implement the method with transformerless push-pull cascades.)

The implementation of the method in amplifiers with different stages in parallel amplification channels is illustrated in figures 1-8.

In Fig 1. shows a diagram of a single-cycle amplifier on bipolar transistors.

Figure 2 - diagram of a single-cycle amplifier on triodes.

Figure 3 is a diagram of an amplifier with differential stages on bipolar transistors.

Figure 4 is a diagram of an amplifier with differential stages on field-effect transistors.

Figure 5 - circuit with push-pull cascades of voltage on complementary transistors.

Figure 6 - diagram of a push-pull amplifier with complementary source repeaters and bridge switching of the load.

7 is a diagram of a push-pull transformer amplifier on shielded lamps.

On Fig - diagram of the parallel connection of several amplifying elements (for example, bipolar transistors).

The pass-through characteristics of the amplifier element, approximated by a power-law function, have lamps and field-effect transistors. So the passage characteristics of the triode and pentode in the "working" areas obey the well-known "law of degree 3/2" [5]. The pass-through characteristic of the field-effect transistor has a pronounced quadratic dependence (see [1], p. 75). The pass-through characteristic of a bipolar transistor is well approximated by an exponential function (see [1], p. 34).

Thus, the pass-through characteristic of the amplifying element, conditionally reduced to the origin, can be represented as:

or

where I is the output current of the amplifying element;

U is the voltage at the input electrodes of the amplifying element;

α is an exponent;

U _T is the thermal voltage;

A and B are proportionality coefficients.

The voltage U = U ₀ + u consists of a constant component U ₀ , which serves to select the quiescent current I ₀ (operating point) and the ac signal itself (and for the harmonic signal u = V · sinωt).

Having expanded the function (1) in a series in powers of u for the input currents I ₁ , I _{2 of the} amplifying elements of the first and second channels, we will have:

where u ₁ , u ₂ - input signals, respectively, of the first and second channels;

U ₀₁ , U ₀₂ - bias voltage.

If now, in accordance with the method, put u ₁ = Ku ₂ , K≥1 and subtract one current from another, then in the load we will have:

Obviously, the odd harmonics are generated by members of the series (3) with odd powers. Therefore, setting the coefficients for the terms of the series with powers of 3, 5, 7, etc., to zero, we find the conditions for compensating for the odd harmonics:

и т.д.

etc.

или, учитывая, что

получаем

Thus, the condition for the compensation of the third harmonic (more precisely, the “cubic" nonlinearity) in accordance with the claimed method will be:

or, given that

we get

For example, for lamps (α = 3/2)

where I ₀₁ , I ₀₂ - quiescent currents of amplifying elements (lamps).

K=2.This reduces the level of other harmonics. So for

K = 2.

The level of the 5th harmonic decreases by 16 times, the second harmonic by 5 times. While the level of the first harmonic (signal) is reduced only by 25%.

In the variant of the method with bipolar transistors, after expanding the exponential function (2) in powers of u into a series, for the current difference we will have:

and for u ₂ = Ku ₁ , K≥1 we obtain the following conditions for the compensation of odd harmonics:

Thus, the condition for the compensation of the third harmonic ("cubic" nonlinearity) in this variant of the method will be:

where I ₀₁ , I ₀₂ - quiescent currents of transistors, respectively, of the first and second channels.

K=2 уровень 5-й гармоники уменьшается в 4 раза, остальных гармоник не менее чем в 3 раза.When choosing the same ratio of quiescent currents as in the above example of the method:

K = 2 the level of the 5th harmonic decreases by 4 times, of the other harmonics by at least 3 times.

In cases of using lamps or field effect transistors as amplifying elements, the compensation of odd harmonics is also achieved by attenuating the output signal of the second channel, without attenuating the input signals of the channels. In fact, for the difference between the amplified signals, we will have:

и условием компенсации третьей гармоники будет:

and the condition for compensation of the third harmonic will be:

We obtain the same conditions (4), (6), (7) for the compensation of odd harmonics when generating a differential output signal by amplifying mutually inverted signals in the first and second channels and then summing the amplified signals.

In fact, if u ₁ = -Ku ₂ , then for the sum of the currents of the first and second channels we will have:

and we get the same expression (4) for the attenuation coefficient

- in the case of lamps and field effect transistors; or (6) in the case of bipolar transistors. The considered method for generating a difference signal can be successfully implemented with push-pull cascades in channels and complementary transistors in the arms of push-pull cascades. Despite the increased level of even harmonics characteristic of this method of formation, one should not expect significant growth in the spectrum of the output signal, since they will be compensated by push-pull amplification.

Having transformed into (5) the expression for the first harmonic (provided that the third harmonic is compensated):

we find the ratio of currents that maximizes the first harmonic (or, equivalently, the transmission coefficient):

Summarizing expressions (4), (6), (7) and (8) for the case of parallel switching on of amplifying elements from n-elements (or their pairs for push-pull amplification) in the first channel and m-elements in the second amplification channel, for attenuation coefficients the following expressions take place:

And relation (8) takes the form:

Expressions (9), (10), (11) and (12) are obtained from the series expansion of the difference output signal I = nI ₁ -mI ₂ for all cases considered, similar to the way it was done above for single amplifier elements in the channels (n = m = 1).

The parallel inclusion of amplifying elements allows to obtain attenuation coefficients K ₁ , K ₂ , K ₃ equal to unity, i.e. dispense in the method without the stages of attenuation. In fact, setting K ₁ = 1, K ₂ = 1 and K ₃ = 1 in (9), (10), (11), we can obtain the required ratios for the quiescent currents of the amplifying elements in the first and second channels with parallel switching on of the elements (m ≠ n).

Figure 1 and figure 2 presents the circuit that implements the method in single-ended amplifiers, respectively, for bipolar transistors and lamps.

In Figure 1 the resistors 6 and 7, the offset is set quiescent currents of transistors 4 and 5 (I _01> I _02), and voltage divider resistors 3 and 1, 2 attenuates the input signal to the first channel K ₁ times:

Similarly, in the tube version (figure 2): resistors 14 and 15 set the quiescent currents of the lamps 16 and 17 (I ₀₁ > I ₀₂ ), and the voltage divider on the resistors 18 and 19 attenuates the input signal of the second channel with lamp 17 by a factor of ₂ :

The input signals of the in-phase channels (poles 12 and 13 in the circuits of Fig.1, 2 can be combined). The differential output signal in the amplifier of FIG. 1 is generated using a current reflector 10 (transistors 8, 9) by subtracting the output current of one channel from the output current of another channel on the load resistor 11. In the amplifier of figure 2, the differential output signal is generated in a similar way, only using current reflectors 20, 21 and 22 (see the direction of the output currents I ₁ , I ₂ channels in the circuit). In addition to participating in the formation of the difference signal, the current reflector serves as a buffer between the lamp and the load 23. As a result, the through-flow dynamic characteristic of the triode practically coincides with its through-pass static characteristic in a fairly wide frequency band (at least in the audio frequency band) and is well approximated for most lamp power function with an exponent α≈3 / 2.

In the amplifier of Fig. 3, parallel amplification channels are made on differential stages. The quiescent currents I _{01 of the} transistors 24, 25 of the first channel are set by the current source 26. The quiescent currents I _{02 of the} transistors 28, 29 of the second channel are set by the current source 27 (I ₀₁ > I ₀₂ ). Attenuator 30 (voltage divider across resistors 31, 32) attenuates the input signal of the first amplification channel. The collectors of transistors 24, 25 and 28, 29 are connected so that the differential output signal from the signals amplified in the channels is formed in loads 33 and 34 from the corresponding collector currents of the cascades.

Since the achievement of the claimed technical result from the implementation of the method in an amplifier with differential switching on of amplifying elements is far from obvious, we will show below that it takes place. And moreover, we show that the conditions for the compensation of the third harmonic exactly coincide with the conditions (6), (9).

It is known that the transfer characteristics of a differential cascade on bipolar transistors are given by the expressions [2, p. 292]:

where I _K1 , I _K2 are the collector currents of the transistors of the differential stage,

U is the input differential voltage

I ₀ is the quiescent current of transistors.

Summing the collector currents in the loads 33 and 34, we obtain the output signal of the amplifier in the form of the difference of the amplified signals in the first and second channels:

I ₃₃ = I _K24 + I _K28 or I ₃₄ = I _K25 + I _K29 .

.

.

We expand the expression for I ₃₃ in a series in powers of U:

Putting the coefficient at the third power of U ^{3 to} zero, we obtain the third harmonic compensation condition:

- coinciding with expression (6).

T.O.

In the amplifier of FIG. 4, differential stages are performed on field-effect transistors 35, 36 in the first channel and 37, 38 in the second amplification channel. An ideal field effect transistor has a quadratic pass-through characteristic (α = 2). It would seem that this guarantees the absence in the spectrum of the output signal of harmonics above the second. However, the output signals of the differential cascade and cascades with a resistor in the source circuit (for example, source repeaters) contain the third and highest harmonics. Those. The transfer characteristics of the amplification stages with field-effect transistors have third and higher order nonlinearities, which can also be compensated by the invented method.

In fact, from the well-known [6, p. 259] equation relating the input differential voltage U and the difference ΔI of the drain currents:

after simple transformations, we obtain the equation of the transfer characteristic of the differential cascade:

where I ₀ is the quiescent current (total) of the transistors,

I _DSS - saturation current (short circuit),

U _p is the cutoff voltage.

We expand expression (13) in a series in powers of U:

According to the proposed method, setting in (14) for the first amplifier amplification channel (Fig. 4) U = U ₁ , I ₀ = 2I ₀₁ ; for the second channel, U = U ₂ = U ₁ / K, I ₀ = 2I ₀₂ ; I _01> I _02, we obtain the equation of the transfer characteristic of the amplifier 4 as:

here ΔI ₁ = I _C35 -I _C36 , ΔI ₂ = I _C38 -I _C37 .

Equating the coefficient at U ^{3 to} zero, we find the conditions for the compensation of the third harmonic in the output signal:

We obtain the same expression (16) by formal substitution of α = 2 in formula (4).

T.O. attenuator attenuation coefficient 39 in the second channel

раз. Второй канал усиления содержит двухтактный каскад на комплементарных транзисторах 45, 46. Источники 49 и 50 смещения задают токи покоя I₀₁, I₀₂ каскадов (I₀₁>I₀₂) и режимы работы каскадов в классах А или АВ. Входные сигналы усилителя парафазные: U_вх1=-U_вх2. Первый и второй каналы усиливают взаимно инвертированные сигналы, а выходной сигнал усилителя формируется суммированием выходных сигналов каскада на нагрузке 51.In the amplifier of FIG. 5, the first amplifier channel contains a push-pull cascade of voltage amplification on complementary transistors 42, 44. Voltage dividers 43 on resistors 47, 48 attenuate the input signal of the first channel to

time. The second amplification channel contains a push-pull cascade on complementary transistors 45, 46. The bias sources 49 and 50 specify the quiescent currents I ₀₁ , I _{02 of the} cascades (I ₀₁ > I ₀₂ ) and the operation modes of the cascades in classes A or AB. The input signals of the amplifier are paraphase: U _in1 = -U _in2 . The first and second channels amplify mutually inverted signals, and the output signal of the amplifier is formed by summing the output signals of the cascade at the load 51.

Similarly to that considered in the amplifier of Fig. 5, the method is also implemented in amplifiers with complementary push-pull emitter (source) repeaters.

Another possible variant of the amplifier with complementary repeaters shown in Fig.6. Here, the load 52 is connected in a bridge circuit between the source followers on transistors 54, 55 and 56, 57 of the first and second channels. The output signal of the amplifier is formed as the difference between the output signals of the channels. The input signals of the amplifier are in-phase: U _in1 = U _in2 .

Attenuator Attenuation Ratio 53:

Sources 61 and 62 of bias voltages U ₀₁ and U ₀₂ specify the quiescent currents I ₀₁ and I _{02 of} transistors 54, 55 and 56, 57 (I ₀₁ > I ₀₂ ).

Finally, FIG. 7 shows a variant of a push-pull power amplification stage, the arms of which are linearized by the claimed method.

Here, lamps 63 and 64 form one arm, and lamps 65 and 66 form the other arm of the amplifier. The cathode bias resistors 67 and 68, 69 and 70 specify the quiescent currents I ₀₁ and I _{02 of the} lamps 63, 65 and 64, 66, respectively, and the voltage divider on the resistors 71 and 72 attenuates the signal supplied to the lamps 64 and 66 by a factor of ₂ , Where

In the amplifiers shown in FIGS. 1-7, instead of one amplifying element in a cascade (or a pair of elements in a push-pull cascade), parallel connection of several amplifying elements can be used. For example, as shown in FIG. 8 for bipolar transistors 73, 74, 75.

A variant of the method with attenuation of the output signal of the second amplification channel is most easily implemented in amplifiers with an output transformer (diagram not shown). Depending on the adopted method of generating a differential output signal, the transformer either subtracts or sums the channel signals at the load connected to the secondary winding of the transformer. The attenuation of the output signal of the second channel is achieved by the appropriate choice of the transformation coefficient n ₂ in the second channel with respect to the transformation coefficient n ₁ in the first channel.

To compensate for the third harmonic: n ₁ : n ₂ = w ₁ : w ₂ ≈ K ₃ : 1,

where w ₁ , w ₂ - the number of turns of the primary winding, respectively, in the first and second channels;

In a multi-stage amplifier (the diagram is not shown), the linearization of the amplification path is achieved by performing the amplification stages using a two-channel parallel amplification circuit and implementing the claimed method for compensating non-linear distortions in each stage.

Information sources

1. Shkritek P. Reference manual for sound circuitry. - M .: Mir, 1991;

2. Gusev V.G., Gusev Yu.M. Electronics.- M.: Higher School, 1991.

3. The application of Germany No. 2718172, H 03 F 1/32.

4. Voishvillo G.V. Amplifiers of low frequency on electronic tubes. - M.: Communication, 1963.

5. Vlasov V.F. Electronic and ionic devices.- M .: Communication, 1956.

6. Soklof S. Analog Integrated Circuits: Translation from English, - M.: Mir, 1988.

Claims (3)

1. A method of compensating for nonlinear distortions in an amplifier of electrical signals by parallel amplification of signals in the first and second amplification channels and subsequent formation of a differential output signal of the amplifier from the amplified signal channels, the method also includes the step of attenuating the input signal of one of the channels and the step of attenuating the output signal of one of channels, characterized in that the quiescent currents of the amplifying elements of the first channel set more quiescent currents of the amplifying elements of the second channel, at m or weaken the input signal of the first channel - in the case of using amplifying elements with a pass-through characteristic approximated by an exponential function in the channels, or weaken the input signal of the second channel - in the case of using amplifying elements with a pass-through approximating an exponential function, or weaken the output signal of the second channel - in the last of the cases mentioned. 2. The method according to claim 1, characterized in that for the cases mentioned in the method, either the input signal of the first channel is attenuated by K ₁ time, or the input signal of the second channel is attenuated by K ₂ times, or the output signal of the second channel is attenuated by K ₃ times:

where n, m are the numbers of parallel-connected amplifying elements or their pairs with push-pull amplification of the first and second channels, respectively; I ₀₁ , I ₀₂ - quiescent currents of amplifying elements of the first and second channels, respectively; α is an exponent of the power function approximating the pass-through characteristic of the amplifying element. 3. The method according to claim 1 or 2, characterized in that the differential output signal of the amplifier is formed by amplifying mutually inverted signals in the first and second channels and then adding the amplified signals.

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