RU2177167C2 - Device for synchronization of clocks - Google Patents

Abstract

FIELD: communication. SUBSTANCE: device can be employed in radio interferometry with superlong bases, in unified time and frequency service. Device has geostationary satellite-repeater, first and second ground stations. Each ground station includes time and frequency calibration instrument, generator of pseudo-noise signal, analog and digital descramblers. Generator of pseudo-noise signal of each station incorporates analog and digital scramblers. Circuit design of proposed device provides for comparison of time scales of ground stations, makes it possible to exchange analog and digital information with its protection against unauthorized access. EFFECT: increased functional reliability of device. 2 cl, 9 dwg

Description

 The invention relates to communication technology and can be used in radio interferometry with extra-long bases, in the service of a single time and frequency, as well as for the exchange of confidential discrete and analog information between ground points spaced over long distances, using a geostationary satellite repeater and protecting this information from unauthorized access.

 Clock synchronization devices are known (ed. Certificate of the USSR 591.799, 614.416, 970.300, 1.180.835, 1.244.632, 1.278.800, RF patents 2.001.423, 2.003.157, 2.040.035; BC Gubanov, A.M. Finkelshtein , P.A. Fridman, Introduction to Radio Astronomy. - M., 1983 and others).

 Of the known devices, the closest to the proposed one is "Clock synchronization device" (RF patent 2.001.423, G 04 С 11/02, 1992), which is selected as a prototype.

 The specified device provides a comparison of time scales spaced over long distances, and is based on the use of the duplex method of communication through a geostationary satellite repeater. The main advantage of the duplex communication method is that it eliminates the length of the signal path. Therefore, its accuracy mainly depends on the parameters of the onboard repeater, the type of signal used, and the technique for measuring time intervals.

 However, the known device does not provide the ability to exchange confidential discrete and analog information between ground stations.

 The aim of the invention is to expand the functionality of the synchronization device by exchanging confidential discrete and analog information between ground stations and protecting it from unauthorized access.

 This goal is achieved by the fact that in a clock synchronization device containing a geostationary satellite repeater, the first and second ground stations, each of which contains a time and frequency standard connected in series, the first local oscillator, the first mixer, the second input of which through the duplexer and the first amplifier are connected in series power is connected to the transceiver antenna, the amplifier is the first intermediate frequency, the first clipper, the second input of which is connected to the second output of the standard time and frequency, the first bl ok memory and a correlator, the output of which is the first output of the device, a second local oscillator connected in series to the first output of the time and frequency standard, a second mixer, the second input of which is connected via a switch to the first output of the pseudo-noise signal generator, a second intermediate frequency amplifier and a second power amplifier, output which is connected to the input of the duplexer, a pseudo-noise signal generator, a second clipper, the second input of which is connected in series to the third output of the time and frequency standard It is connected to the second output of the time and frequency standard, and the second memory block, the output of which is connected to the second input of the correlator, introduces the first and second amplitude limiters, a selector, a synchronous detector, a multiplier, a bandpass filter, a phase detector, and analog and digital descramblers, and between the output of the amplifier of the first intermediate frequency and the input of the first clipper include a selector and a first amplitude limiter in series between the second output of the pseudo-noise signal generator and the input of the second clipper the second amplitude limiter is turned on, a synchronous detector is connected in series to the output of the first amplitude limiter, the second input of which is connected to the output of the selector, and an analog descrambler, the output of which is the second output of the device, a multiplier is connected in series to the output of the first amplitude limiter, the second input of which is connected to the second output the first local oscillator, a bandpass filter, a phase detector, the second input of which is connected to the output of the second local oscillator, and a digital descrambler, output d which is the third output of the apparatus.

 In FIG. 1 is a structural diagram of the proposed synchronization device; figure 2 is a geometric arrangement of the satellite-relay S and two ground points A and B; figure 3 is a timing chart of the duplex method of comparing watches; figure 4 is a frequency diagram explaining the process of converting the frequency of the clock signal; figure 5 is a structural diagram of a generator of a pseudo-noise signal; figure 6 is a structural diagram of a selector; 7 is a structural diagram of an analog scrambler using the frequency inversion method; on Fig - presents the spectra of the input (a) and converted (b) signals; figure 9 shows timing diagrams explaining the principle of operation of the proposed device clock synchronization.

 The clock synchronization device contains a geostationary satellite repeater S, first A and second B ground stations, each of which contains a standard 1 time and frequency, the first 2 and second local oscillators, transceiver antenna 4, duplexer 5, first power amplifier 6, first mixer 7, first intermediate frequency amplifier 8, first clipper 9, first memory block 10, correlator 11, pseudo noise signal generator 12, switch 13, second mixer 14, second intermediate frequency amplifier 15, second power amplifier 16, second clipper 17, second a memory unit 18, a selector 19, the first 20 and second 21, amplitude limiters, synchronous detector 22, an analog descrambler 23, a multiplier 24, a bandpass filter 25, a phase detector 26 and the digital descrambler 27.

 The pseudo-noise signal generator 12 comprises a high-frequency driver 28, a discrete message source 29, a digital scrambler 30, a phase manipulator 31, an analog message source 32, an analog scrambler 33, an amplitude modulator 34 and a band amplifier 35.

 The selector 19 comprises a phase doubler 36, a first 37 and a second 38 spectrum width meters, a comparison unit 39, a threshold block 40, and a key 41.

 An analog scrambler 33 using the frequency inversion method comprises a clock 42, a divider-shaper 43, a band-pass filter 44, a balanced modulator 45, an analog switch 46, an adder 47, and a low-pass filter 48.

The device operates as follows. At point A, a pseudo-noise microwave signal a1 is generated using a generator 12 (Fig. 3) with combined phase shift keying and amplitude modulation (FMN-AM). To this end, the high-frequency voltage (Fig.9, a)
u _c (t) = U _c cos (2πf _c t + φ _c ), 0≤t≤T _c .
where U _c , f _c , φ _c , T _c - amplitude, carrier frequency, initial phase and signal duration;
from the output of the master oscillator 28 is fed to the first input of the phase manipulator 31, the second input of which is supplied with a modulating code M (t) (Fig. 9, b) from the output of the digital scrambler 30. The input of the latter is connected to a source 29 of a discrete message. At the output of the phase manipulator 31, a phase-shift (PSK) signal is generated (Fig. 9, c)
u ₁ (t) = U _c cos [2πf _c t + φ _k (t) + φ _c ], 0≤t≤T _c ,
where φ _k (t) = {0, π} is the manipulated phase component that displays the phase manipulation law in accordance with the modulating code M (t), and φ _k (t) = const for kτ _e <t <(k + 1) τ _e and can change abruptly at t = kτ _e , i.e. at the boundaries between elementary premises (k = 1, 2, .., N); τ _e N is the duration and number of chips that make up a signal of duration T _s (T _s = N).

This signal is fed to the first input of the amplitude modulator 34, to the second input of which a modulating function m (t) (Fig. 9d) is supplied from the output of the analog scrambler 33. The input of the latter is connected to the source 32 of the analog message. At the output of the amplitude modulator 34, a pseudo-noise signal is generated with combined phase shift keying and amplitude modulation (PSK-AM) (Fig. 9, e)
u ₂ (t) = U _c [1 + m (t)] cos [2πf _c t + φ _k (t) + φ _c ], 0≤t≤T _c ,
where m (t) is the modulating function that displays the law of amplitude modulation.

 Digital 30 and analog 33 scramblers implement cryptographic methods, which are the most effective methods of protecting confidential discrete and analog information.

 Cryptographic methods of information protection are special methods of encrypting, encoding and converting information, as a result of which its content becomes inaccessible without presenting a cryptogram key and reverse transformation.

With the digital method of closing the transmitted message, four main groups can be conditionally distinguished:
1) substitution - the characters of a discrete message are replaced by other characters in accordance with a predetermined rule;
2) permutation - the symbols of a discrete message are rearranged according to some rule within a given block of a transmitted discrete message;
3) analytical conversion - the encrypted message is converted according to some analytical rule;
4) combined conversion - the original discrete message is encrypted with two or more encryption methods.

With analog scrambling, the signal undergoes the following transformations:
1) frequency inversion;
2) frequency permutation;
3) temporary permutation.

As an example in FIG. 7 is a structural diagram of an analog scrambler using the frequency inversion method. The device consists of a clock generator 42 that generates a signal with a frequency of 7 kHz, a carrier splitter 3.5 kHz, an analog switch 46, a balanced modulator 45, an input bandpass filter with a passband f _min = 300-f _max = 3000 Hz, an adder 47, a balance modulator 45 and a low-pass filter 48. On Fig presents the spectra of the input (a) and converted (b) signals, where f _o = 3500 Hz.

The generated signal U ₂ (t) is supplied to the first input of the mixer 14, the second input of which is supplied with the voltage of the second local oscillator stabilized by standard 1 of frequency and time
u _g2 (t) = U _g2 cos (2πf _g2 + φ _g2 ).
In this case, the switch 13 is closed, and a similar switch at step B is open. At the output of the mixer 14, a voltage of combination frequencies is generated.

где

К₁ - коэффициент передачи смесителя;
f_пр2=f₁=f_c+f_r2 - вторая промежуточная частота;
φ_пр2 = φ_c+ φ_г2;
которое усиливается в усилителе мощности 16 и излучается через дуплексер 5 и антенну 4 в направлении ИСЗ-ретранслятор на частоте f₁. Вместе с тем этот же сигнал ограничивается по амплитуде в ограничителе амплитуды 21
u₃(t) = U_ocos[2πf_пр2t+φ_k(t)+φ_пр2], 0≤t≤T_c,
где U_о - порог ограничения;
клиппируется во втором клиппере 17 и записывается в буферный блок 18 памяти.Amplifier 15 stands out the voltage of the total frequency

Where

To ₁ - gear ratio of the mixer;
f _CR2 = f ₁ = f _c + f _r2 - the second intermediate frequency;
φ _CR2 = φ _c + φ _g2 ;
which is amplified in the power amplifier 16 and is radiated through the duplexer 5 and the antenna 4 in the direction of the satellite repeater at a frequency f ₁ . However, the same signal is limited in amplitude in the amplitude limiter 21
u ₃ (t) = U _o cos [2πf _pr2 t + φ _k (t) + φ _pr2 ], 0≤t≤T _c ,
where U _o is the limit threshold;
clipped in the second clipper 17 and recorded in the buffer block 18 of the memory.

 The operation of the clipper 17 is synchronized by the standard 1 frequency and time.

The signal received by the on-board receiver of the geostationary satellite repeater is re-emitted to points A and B at a frequency of f ₂ while maintaining phase relationships in the entire frequency band of the signal. The directivity pattern of the onboard antenna of the geostationary satellite repeater is selected so that this signal can be received at both points A and B.

где

f_пр1=f_r1-f₂ - первая промежуточная частота;
φ_gh1 = φ_u1- φ_gh2.
Это напряжение поступает на вход селектора 19, состоящего из удвоителя 36 фазы, измерителей 37 и 38 ширины спектра, блока 39 сравнения, порогового блока 40 и ключа 41. При этом на выходе удвоителя 36 фазы, в качестве которого может использоваться перемножитель, на два входа которого подается одно и то же напряжение u_пр1(t) образуется следующее напряжение
u₅(t) = U₅(t)cos(4πf_пр1t+2φ_пр1), 0≤t≤T_c,
где U₅(t)=U_пр1 ²[1+m(t)]².The signal received at point A by antenna 4 relayed by a geostationary satellite repeater (signal α ₂ )
u ₄ (t) = U ₄ [1 + m (t)] cos [2πf ₂ t + φ _k (t) + φ _pr2 ],
amplified by the first power amplifier 6 and supplied to the first input of the first mixer 7, the second input of which is supplied with the voltage of the first local oscillator 2, stabilized by the standard 1 frequency and time
u _g1 (t) = U _g1 cos (2πf _g1 + φ _g1 ).
At the output of the mixer 7, voltages of combination frequencies are generated. The amplifier 8 is allocated the voltage of the first intermediate frequency (differential frequency)

Where

f _pr1 = f _r1 -f ₂ - the first intermediate frequency;
φ _gh1 = φ _u1 - φ _gh2 .
This voltage is supplied to the input of the selector 19, which consists of a phase doubler 36, spectral width meters 37 and 38, a comparison unit 39, a threshold block 40, and a key 41. At the same time, the output of the phase doubler 36, which can be used as a multiplier, has two inputs which is supplied with the same voltage u _pr1 (t) the following voltage is formed
u ₅ (t) = U ₅ (t) cos (4πf _pr1 t + 2φ _pr1 ), 0≤t≤T _c ,
where U ₅ (t) = U _pr1 ² [1 + m (t)] ² .

Since 2φ _k (t) = {0.2π}, phase manipulation is already absent in the indicated voltage.

The spectrum width of the FMN-AM signal Δf _c is determined by the duration τ _{e of the} elementary premises Δf _c = 1 / τ _e . Whereas the width of the spectrum of its second harmonic is determined by the duration T _{c of the} signal Δf ₂ = 1 / T _c ; those. the width of the spectrum of the second harmonic Δf ₂ is N times smaller than the width of the spectrum Δf _{c of the} input signal Δf _c / Δf ₂ = N, where N is the number of chips (for the Barker code N = 7, for m - the sequence N = 1023).

 Consequently, when the phase of the spectrum is doubled, the FMN-AM signal “folds” N times. This circumstance makes it possible to detect a complex signal with combined phase shift keying and amplitude modulation and apply a new type of selection - structural selection.

The width of the spectrum Δf _{c of the} input QPSK-AM signal is measured using meter 37, and the width of the spectrum Δf _{2 of} its second harmonic is measured using meter 38. The voltages U and U ₂ are proportional to Δf _c and Δf _2, respectively, from the outputs of meters 37 and 38 the width of the spectrum goes to the two inputs of block 39 comparison. Since U>> U ₂ , a positive pulse is generated at the output of the comparison block 39, the amplitude of which exceeds the threshold voltage U _then in the threshold block 40. A positive pulse at the output of the comparison block 39 is generated when the voltages U and U ₂ the two inputs of block 39 comparison, significantly differ in magnitude from each other. With an approximate equality of voltages (U = U ₂ ), which is typical for simple signals and interference, there is no voltage at the output of comparison unit 39. The threshold voltage U _pore in the threshold block 40 is selected so that it does not exceed random interference. If the threshold level U _pores is exceeded, a constant voltage is generated in the threshold block 40, which is supplied to the control input of the key 41 and opens it. In the initial state, the key 41 is always closed. The voltage u _pr1 (t) ( _Fig.9 , f) through the public key 41 from the output of the amplifier 8 of the first intermediate frequency to the input of the amplitude limiter 20 and to the information input of the synchronous detector 22. A voltage is generated at the output of the amplitude limiter 20 (Fig. 9 , g)
u ₆ (t) = U _o cos [2πf _pr1 t-φ _k (t) + φ _pr1 ], 0≤t≤T _c ,
which is an FMN-AM signal and is supplied to the reference input of the synchronous detector as a reference voltage, to the control input of the clipper 9 and to the first input of the multiplier 24. The operation of the clipper 9 is also synchronized with the clock frequency of the frequency and time standard 1. Then, the indicated voltage is recorded by the memory unit 18.

К₂ - коэффициент передачи синхронного детектора;
пропорциональное модулирующей функции m(t).At the output of the synchronous detector 22, a low-frequency voltage is generated (Fig. 9, h)
u _н1 (t) = U _н1 [1 + m (t)],
Where

K ₂ is the transfer coefficient of the synchronous detector;
proportional to the modulating function m (t).

 This voltage is supplied to the input of the analog descrambler 23, the principle of which corresponds to the principle of analog scrambler 33, but has the opposite character. The output of the descrambler 23 (the second output of the II device) is the source information of the source 32 of the analog message.

The voltage u _r2 (t) of the local oscillator 2 is supplied to the second input of the multiplier 24, a voltage is generated at its output (Fig. 9, and)
₇ u (t) = U ₇ cos [2πf _{r2 t + φ k (t) +} φ _r2], 0≤t≤T _c,
where U ₇ = 0.5K ₃ U _o U _r1 ;
K ₃ - transfer coefficient of the multiplier;
which is an FMN signal at a stable frequency of the second local oscillator. This voltage is allocated by a band-pass filter 25 and fed to the information input of the phase detector 26, whose reference input is supplied with the voltage u _r2 (t) of the second local oscillator 3.

At the output of the phase detector 26, a low-frequency voltage is generated (Fig. 9, k)
u _n2 (t) = U _n2 cosφ _k (t),
where U _n2 = 0.5K ₄ U ₇ U _r2 ;
To ₄ - the transfer coefficient of the phase detector;
proportional to the modulating code M (t) (Fig. 9, b). This voltage is supplied to the input of the digital descrambler 27, the principle of operation of which corresponds to the principle of operation of the digital scrambler 30, but has the opposite character. The output of the descrambler 27 (the third output of the III device) is the source information of the source 29 of the discrete message.

Similar operations are carried out at point B. At the second measurement step (when transmitting a signal from point B), after some time T after the end of signal registration, the switch 13 at point A opens and closes at point B. At this point in time t ₃ ^B = t ₂ ^B + T, the generator begins to generate a new microwave signal (signal β ₃ ), which, after a corresponding conversion, is fixed at point B and emitted in the direction of the geostationary satellite repeater at a frequency f ₁ . The signal received by the onboard receiver of the satellite repeater is re-emitted back to points A and B at a frequency f ₂ while maintaining phase relationships in the entire frequency band of the signal. The relay signal is received at both points, converted to a low frequency voltage and recorded at time t ₄ ^A and t ₄ ^B, respectively (signals α ₄ , β ₄ ,). This completes the single measurement of the duplex method. Then, in the interval between the measurement acts, the pairs of signals α ₁ , α ₂ and α ₃ , α ₄ (β ₁ , β ₂ and β ₃ , β ₄ ) are subjected to correlation processing in the correlator 11.

и соответствующие им частоты интерференции F_i (i=l, 2, 3, 4), которые определяют производные этих задержек:

где

a_j, b_j (j=1, 2, 3) - время распространения сигнала между геостационарным ИСЗ-ретранслятором и пунктами А и В соответственно (фиг.2);
t_A, t_B - время задержки сигналов в излучающей аппаратуре обоих пунктов;
τ_A,τ_D - время задержки сигналов в приемно-регистрирующей аппаратуре;
ΔS - время задержки сигналов бортовым ИСЗ-ретранслятором;
Δt= t₁ ^B-t₁ ^A - искомая разность показателей часов в один и тот же физический момент времени t.Correlation processing of two pairs of registered signals at each point allows you to determine the following time delays:

and the corresponding interference frequencies F _i (i = l, 2, 3, 4), which determine the derivatives of these delays:

Where

a _j , b _j (j = 1, 2, 3) is the signal propagation time between the geostationary satellite repeater and points A and B, respectively (figure 2);
t _A , t _B - signal delay time in the radiating equipment of both points;
τ _A , τ _D - signal delay time in the receiving and recording equipment;
ΔS is the delay time of the signals by the onboard satellite repeater;
Δt = t ₁ ^B -t ₁ ^A is the desired difference of the clock indicators at the same physical time t.

из первых двух уравнений (1) получаем

где

- время задержки сигнала в атмосфере на частотах f₁ и f₂ соответственно;
ν - релятивистская поправка (эффект Саньяка);
ω - угловая скорость вращения Земли;
с - скорость света;
D - площадь четырехугольника OA'S'B', образуемого в экваториальной плоскости центром масс Земли, проекциями пунктов А', В' и геостационарного ИСЗ-ретранслятора S'.Assuming a _j and b _j linear functions with derivatives

from the first two equations (1) we obtain

Where

- the delay time of the signal in the atmosphere at frequencies f ₁ and f _2, respectively;
ν - relativistic correction (Sagnac effect);
ω is the angular velocity of the Earth;
c is the speed of light;
D is the area of the quadrangle OA'S'B ', formed in the equatorial plane by the center of mass of the Earth, the projections of points A', B 'and the geostationary satellite relay S'.

который следует в начале измерений рассчитать по приближенным эфемеридным данным, а затем уточнить по результатам текущих измерений.The correction γ for the mobility of a geostationary satellite repeater during a single measurement is most easily reduced to zero by the appropriate choice of the free parameter T

which should be calculated at the beginning of measurements by approximate ephemeris data, and then clarified by the results of current measurements.

 As for the correction δ for hardware delays, it can be found by calibration using the “zero base” method.

 The atmospheric correction ε can also be taken into account.

 The relativistic correction ν can reach 430 ns and vary within one day within a few nanoseconds. It is easy to show that for the preliminary calculation of this correction with an error of 0.1 s, the coordinates of points A and B must be known with an error of no more than 1 km, and the coordinates of the geostationary satellite repeater with an error of up to 7 km.

а ошибки измерения временной задержки τ и частоты интерференции F имеют вид:

где Δf - полоса принимаемых и регистрируемых частот псевдошумового сигнала;
Р_c, Р_ш - мощность сигнала и шума на входе приемника;
t_c - интервал когерентности сигнала при его ретрансляции.Let us estimate the errors in the measurement of time delays τ _i (i = 1,2,3,4).
The radio interferometric signal-to-noise ratio is determined by the expression

and the errors of measuring the time delay τ and interference frequency F have the form:

where Δf is the band of received and recorded frequencies of the pseudo-noise signal;
R _c , R _W - signal power and noise at the input of the receiver;
t _c is the coherence interval of the signal when it is relayed.

Then, to obtain an error, for example, σ _τ ≤ 0.1 ns, it is necessary that QΔf ≥ 5 • 10 ⁹ . For example, at Δf = 10 MHz, we obtain Q ≥ 500, which is quite achievable even with the use of small diameter terrestrial transceiver antennas. For Q = 500 and Δf ≈ 100 MHz it turns out to be sufficient and t _c = 5 • 10 ^-6 s. Such a coherence time is provided even with the instability of the local oscillator local oscillator σ _f = 2 • 10 ^-6 .
To calculate the difference of the clock readings Δt by the formula (2), it is now sufficient to exchange the received digital data between the ground points A and B, which can be done via satellite communication channels through the same geostationary satellite repeater. As shown above.

 Thus, the proposed device in comparison with the prototype provides not only the synchronization of remote time scales, but also the exchange of confidential discrete and analog information between remote ground stations and its protection against unauthorized access. Moreover, the protection of this information has three levels: energy, structural and cryptographic. The energy and structural levels are ensured by the use of complex signals with combined phase shift keying and amplitude modulation, which have high energy and structural secrecy.

 The energy secrecy of these signals is due to their high compressibility in time or spectrum with optimal processing. That allows you to reduce the instantaneous radiated power. As a result, the complex signal used at the receiving point may be masked by noise. Moreover, the energy of a complex signal is by no means small; it is simply distributed over the time-frequency domain so that at each point in this region the signal power is less than the noise power.

 The structural secrecy of complex signals with combined phase shift keying and amplitude modulation is due to the wide variety of their shapes and significant ranges of parameter changes, which makes it difficult to optimize or at least quasi-optimal processing of complex signals of an a priori unknown structure in order to increase the sensitivity of the receiver.

 Complex signals with combined phase shift keying and amplitude modulation open up new possibilities in the technique of transmitting messages on one carrier frequency and protecting them from unauthorized access. These signals allow the use of a new type of selection - structural selection. This means that there is a new opportunity to distinguish complex signals from other signals and interference operating in the same frequency band and at the same time intervals. This feature is realized by convolution of the spectrum of complex signals.

 The cryptographic level is provided by special methods of encrypting, encoding and converting confidential discrete and analog information, as a result of which its content becomes inaccessible without presenting the cryptogram key and reverse conversion.

 Therefore, the functionality of the known device clock synchronization is greatly expanded.

Claims (3)

 1. A clock synchronization device comprising a geostationary satellite repeater, first and second ground stations, each of which contains a time and frequency standard connected in series, a first local oscillator, a first mixer, the second input of which is connected to a transceiver antenna through a series-connected duplexer and a first power amplifier , an amplifier of the first intermediate frequency, a first clipper, the second input of which is connected to the second output of the time and frequency standard, a first memory unit and a correlator, the output of which The second oscillator, the second mixer, the second input of which through a switch is connected to the first output of the pseudo noise signal generator, the second intermediate frequency amplifier and the second power amplifier, the output of which is connected to the duplexer input, is connected in series with the first output of the device, connected in series to the first output of the time and frequency standard connected to the third output of the time and frequency standard, a pseudo-noise signal generator, a second clipper, the second input of which is connected to the second output of the standard time and frequency, and a second memory unit, the output of which is connected to the second input of the correlator, characterized in that the first and second amplitude limiters, a selector, a synchronous detector, a multiplier, a bandpass filter, a phase detector, analog and digital descramblers are introduced into it, and between the output of the amplifier of the first intermediate frequency and the input of the first clipper are connected in series with the selector and the first amplitude limiter, between the second output of the pseudo-noise signal generator and the input of the second clipper A second amplitude limiter, a synchronous detector is connected to the output of the first amplitude limiter, the second input of which is connected to the output of the selector, and an analog descrambler, the output of which is the second output of the device, a multiplier is connected in series to the output of the first amplitude limiter, the second input of which is connected to the second output of the first a local oscillator, a bandpass filter, a phase detector, the second input of which is connected to the output of the second local oscillator, and a digital descrambler, the output of which a third output device.  2. The clock synchronization device according to claim 1, characterized in that the pseudo-noise signal generator is made in the form of a serially connected high-frequency oscillator, a phase manipulator, the second input of which is connected through a digital scrambler to a source of a discrete message, an amplitude modulator, the second input of which is through an analog the scrambler is connected to the source of the analog message, and a strip amplifier with two outputs.  3. The clock synchronization device according to claim 1, characterized in that the selector is made in the form of series-connected first spectral width meter, a comparison unit, the second input of which is connected through the second spectral width meter to the output of the phase doubler, threshold block and key, the output of which is the output of the selector, and the second key input, the inputs of the phase doubler and the first meter of the spectral width are combined and are the input of the selector.

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