RU2350998C2 - Method of synchronising clocks - Google Patents

Abstract

FIELD: physics; communication.

SUBSTANCE: present invention pertains to communication technology and can be used in radio interferometry with very long baselines, as well as in a universal time and frequencies service. The result is achieved by that, the invention uses standard frequency and time, first and second heterodynes, pseudonoise signal generator, a switch, first and second amplifier of the first intermediate frequency, first and second power amplifiers, a duplexor, a transceiving antenna, first and second clippers, first and second buffer memory, first and second amplifiers of the second intermediate frequency, a device for measuring delays and their derivatives, +90° phase changer, third mixer, 90° phase changer, first, second, third and fourth adders, a multiplier, first and second narrow-band filters, an amplitude detector, switch, first, second and third phase inverters, and first and second band-pass filters.

EFFECT: increased noise immunity and accuracy of synchronising remote time scales through suppression of false signals (noise), received on a direct transmission channel and intermodulation channels.

6 dwg

Description

The proposed method relates to communication technology and can be used in radio interferometry with extra-long bases, as well as in the service of a single time and frequencies.

Known methods and devices for clock synchronization (ed. Certificate of the USSR No. 591799, 614416, 970300, 1180835, 1244632, 1278800; RF patents No. 20041423, 2003157, 2040035, 2177267, 2292574; BC Gubanov, A.M. Finkelshtein, P.A. . Friedman and others. Introduction to radio astronomy. - M .: 1983).

Of the known methods, the closest to the proposed one is the "Method of clock synchronization" (RF patent No. 2292574, G04C 11/02, 2005), which is selected as a prototype.

This method provides a comparison of time scales spaced over long distances, and is based on the use of the duplex method of communication through a geostationary IS3 repeater.

The main advantage of the duplex communication method is that it eliminates the length of the signal path. Therefore, its accuracy mainly depends on the parameters of the onboard transponder, the type of signal used, and the technique for measuring time intervals.

For the technical implementation of the known method, a superheterodyne receiver is used in which the same value of the second intermediate frequency ω _pr2 can be obtained by receiving signals at two frequencies ω ₂ and ω ₃ , i.e.

ω = ω _{z2 np2} -ω and ω ₂ = ω _{3 np2} -ω _r2.

Therefore, if the tuning frequency ω _{3 is} taken as the main receiving channel, then along with it there will be a mirror receiving channel, the frequency ω _{3 of} which differs from the frequency ω ₂ by ω _pr2 and is located symmetrically (mirror) with respect to the frequency of the second local oscillator ω _g2 (Fig. .four).

Conversion on the mirror channel of the reception occurs with the same conversion coefficient K _ol as on the main channel. Therefore, it most significantly affects the selectivity and noise immunity of the method.

In addition to the mirror, there are other additional (combinational, intermodulation and direct channel) receive channels.

In general terms, any Raman receive channel occurs when the following condition is met:

where ω _ki is the frequency of the i-th Raman reception channel;

m, n, i are positive integers.

The most harmful Raman reception channels are those generated by the interaction of the carrier frequency of the received signal with the harmonics of the frequency ω _{G2 of the} second small local oscillator (second third, etc.), since the sensitivity of the receiver through these channels is close to the sensitivity of the main channel. So, two combination channels with m = 1 and n = 2 correspond to frequencies:

ω _K1 = 2ω _{Г2 -ω пр2} and ω _К2 = 2ω _Г2 + ω _пр2 .

The nature of the intermodulation channels is as follows.

If two large amplitude signals at frequencies ω _I and ω _II simultaneously arrive at the input of a superheterodyne receiver, then they form a series of intermodulation frequencies on any nonlinear elements of the receiver (Fig. 5)

mω _I ± nω _II = ω _mn .

The sum (difference) of the coefficients m and n is called the order, i.e. the intermodulation frequency ω _mn is called a frequency of the order of m + n.

As can be seen from FIGS. 5 and 6, two powerful signals at frequencies ω _I and ω _II , ω _III and ω _IV hit the frequencies of intermodulation interference. With an increase in the order of the amplitude of the intermodulation noise, they quickly decrease. The more linear the receiver elements are, the smaller the amplitudes of the intermodulation noise are and the faster they decrease with increasing order. The linearity of the elements of the frequency receiver is also characterized by the magnitude of the dynamic range, i.e. the range of signal amplitudes from the minimum level equal to the level of the receiver's own noise to the maximum signal level at which non-linearity begins to appear. Since two signals are involved in the formation of intermodulation interference, the selectivity of the receiver to this interference is called “two-signal selectivity”.

If the intermodulation interference is in the receiver bandwidth, it is received as a useful signal, i.e. no filters can eliminate it.

The use of highly selective quartz filters at the second intermediate frequency, improving the selectivity on the adjacent channel, can help suppress interference from one powerful out-of-band signal, but it is powerless to help suppress intermodulation interference.

If the interference frequency is equal to the second intermediate frequency, then a direct channel is formed. The receiver elements for such interference are simple transmission links.

The known method provides the suppression of false signals (interference) received through the mirror and Raman channels.

However, the presence of false signals (interference) received through the direct channel and intermodulation channels leads to a decrease in noise immunity and accuracy of synchronization of remote time scales.

An object of the invention is to increase the noise immunity and accuracy of synchronization of remote time scales by suppressing false signals (interference) received on the direct channel and intermodulation channels.

The problem is solved in that according to the method of clock synchronization, based, in accordance with the closest analogue, on the simultaneous reception by separated ground points of noise microwave signals from an artificial Earth satellite, coherently converting them to a video frequency, digitally recording the received signals and determining the time delay of arrival of the same signal to synchronization points by the method of correlation processing of registered signals, the magnitude of which compares time scales, at in this case, at the initial time t _1, according to the clock of the first point, a microwave noise signal is generated using a code sequence, it is recorded at the same point, the generated signal is converted to frequency ω ₁ , amplified by power, emitted amplified signal in the direction of the artificial Earth satellite -retranslyator, at the same time t _1, the clock of the second paragraph with the same code sequence form the same noise microwave signal, it is recorded in the second paragraph, taking onboard equipment STCs artificial Earth relay nick signal at frequency ω _1, re-emit it at first and second points at the frequency ω ₂ while preserving phase relationships, at an arbitrary time t ₃ by the clock of the second paragraph similarly formed and retransmit noise microwave signal generated signal is converted to the frequency ω ₁ , amplify it in power, emit an amplified signal in the direction of the same satellite repeater, at the same time t _3, according to the clock of the first point, using the same code sequence form the same noise microwave signal, register it at the first point, the signal at the frequency ω _{1 is} received by the onboard equipment of the satellite repeater and re-emitted to the first and second points at the frequency ω ₂ while maintaining phase relationships, the received signal at the frequency ω _{2 is} converted in frequency using the voltage of the second local oscillator shifted in + 90 ° phase, the voltage of the second intermediate frequency is isolated, it is shifted in phase by -90 °, summed with the initial voltage of the second intermediate frequency, the resulting total voltage is multiplied with the received signal, the harmonic is isolated nical voltage at frequency ω _r2 of the second local oscillator is detected it and is used to permit further processing of the received signal differs from the closest analog by the fact that to transform the frequency emit spurious signal (noise) received by the second intermediate frequency ω _np2, inverted its phase 180 °, summarize with the original false signal (interference) and compensate for it, isolate false signals (interference) received in the frequency band Δω _П1 = ω _II -ω _I , where ω _I and ω _II are the boundary frequencies that determine the frequency band Δω _PI, location with ennuyu "left" by Δω receiver _P bandwidth hit in which two or more signals leads to the formation of intermodulation interference, invert their phase by 180 °, summed with the original false signals (noisy) and compensate them emit spurious signals (interference) taken in the frequency band Δω _P2 = ω _IV -ω _III where ω _III and ω _IV are the boundary frequencies defining the frequency band Δω _P2 located "to the right" of the passband Δω _{P of the} receiver, if two or more signals get into it, intermodulation interference, invert they are phase-aligned by 180 °, summed with the initial false signals (interference), and compensate for them.

The geometric arrangement of ground points A and B and the satellite of the relay S is shown in figure 1, where the following notation is introduced: O is the center of mass of the Earth; d is the base of the interferometer; r is the radius vector of the satellite repeater.

The timing diagram of the duplex clock comparison method is presented in figure 2, where the following notation is introduced: S, A, B - time scale of the satellite repeater and points A and B, respectively.

The structural diagram of the equipment of one of items (A) that implements the proposed method for clock synchronization is presented in Fig. 3, where the following notation is introduced: 1 - standard frequency and time, 2.1 - first local oscillator, 2.2 - second local oscillator, 3 - pseudo-noise signal generator, 4 - switch, 5 - first mixer, 6 - first intermediate frequency amplifier, 7 - first power amplifier, 8 - duplexer, 9 - transceiver antenna, 10 - first clipper, 11 - first buffer memory, 12 - second power amplifier, 13 - second mixer, 14 - per the second amplifier of the second intermediate frequency, 15 - the second clipper, 16 - the second buffer memory, 17 - the meter of delays and their derivatives, 18 - the first phase shifter + 90 °, 19 - the third mixer, 20 - the second amplifier of the second intermediate frequency, 21 - second phase shifter -90 °, 22 - adder, 23 - multiplier, 24 - narrow-band filter, 25 - amplitude detector, 26 - key, 27 - second narrow-band filter, 28 - first phase meter, 29 - second adder, 30 - first band-pass filter 31 - the second phase meter, 32 - the second bandpass filter, 34 - the third phase inverter, 35 - even fifth adder.

Clock synchronization by the proposed method is as follows.

At time t ₁ ^A, according to the clock of the first point A, a microwave noise signal is generated using a code sequence (signal α ₁ ) (figure 2):

U _s (t) = U _s cos [ω _c t + φ _k (t) + φ _s ], 0≤t≤T _c ,

where U _c , ω _c , φ _c , T _c - amplitude, carrier frequency, initial phase and signal duration;

φ _k = (t) = {0, π} is the manipulated phase component that displays the phase manipulation law in accordance with the code sequence M (t), and φ _k (t) = const for kτ _e <t <(k + 1) τ _e and can change abruptly at t = kτ _e , i.e. at the borders between elementary premises (K = 1, 2, ... N-1);

τ _e , N - the duration and number of chips that make up a signal of duration T _s (T _s = Nτ _e ),

in generator 3 using standard 1 frequency and time.

The specified signal is supplied to the input of the clipper 10, and then is registered in the buffer memory 11. Registration is synchronized by standard 1 frequency and time.

The generated signal through the switch 4 is fed to the first input of the first mixer 5, the second input of which is supplied with the voltage of the first local oscillator 2.1

u _Г1 (t) = U _Г1 cos (ω _Г1 t + φ _Г1 ).

At the output of the mixer 5, voltages of combination frequencies are generated. Amplifier 6 distinguishes the voltage of the first intermediate (total) frequency

u _CR1 (t) = U _CR1 cos [ω _CR1 t + φ _k (t) + φ _CR1 ], 0≤t≤T _s ,

;Where

;

To ₁ - gear ratio of the mixer;

ω _p1 = ω _s + ω _G1 - the first intermediate (total) frequency;

φ _CR1 = φ _s + φ _G1 ,

which, after amplification in the power amplifier 7, through the duplexer 8 and the transceiver antenna 9 is radiated in the direction of the satellite repeater at a frequency of ω ₁ = ω _pr1 .

At the same time t ₁ ^A = t ₁ ^V , the same noise microwave signal (signal β ₁ ) is formed using the same code sequence M (t) using the same code sequence M (t). Register it at the second point (signal β ₁ , which, however, is not sent for relay). Accept on-board equipment of the satellite repeater at a frequency of ω ₁ (signal α ₁ ), re-emit it to points A and B at a frequency of ω ₂ while maintaining phase relationships in the interval t _c .

Relay signal (signal α ₂ ) at a frequency of ω ₂

u ₂ (t) = U ₂ cos [ω ₂ t + φ _k (t) + φ ₂ ], 0≤t≤T _c ,

received by the transceiver antenna 9 and through the duplexer 8, the power amplifier 12 and the adders 29, 32 and 35, for which only one arm operates, is fed to the first inputs of the second 13 and third 19 mixers and multiplier 23. Voltage is applied to the second inputs of the mixers 13 and 19 second local oscillator 2.2:

u _Г2 (t) = U _Г2 (ω _Г2 t + φ _Г2 )

u _Г3 (t) = U _Г2 cos (ω _Г2 t + φ _Г2 + 90 °).

Moreover, the frequencies ω _G1 and ω _{G2 of the} first 2.1 and second 2.2 local oscillators are spaced at the second intermediate frequency

w _{G1 G2} -ω = ω _WP2.

At the outputs of the mixers 13 and 19, voltages of combination frequencies are generated. Amplifiers 14 and 20 are allocated voltage of the second intermediate (differential) frequency:

u _pr2 (t) = U _p2 cos [ω _p2 (t) -φ _K (t) + φ _p2 ],

u _pr2 (t) = U _pp2 cos [ω _pp2 (t) -φ _K (t) + φ _pp2 + 90 °], 0≤t≤T _c ,

Where

ω _p2 = ω _Г2 + ω ₂ - the second intermediate (difference) frequency;

ω _p2 = ω _Г2 + ω ₂

The voltage U _pr3 (t) from the input of the amplifier 20 of the second intermediate frequency is supplied to the input of the phase shifter 21 by -90 °, at the output of which a voltage is generated

u _pr4 (t) = U _pr2 cos [ω _pr2 t-φ _k (t) + φ _pr2 + 90 °] = U _p2 cos [ω _p2 t-φ _k (t) + φ _p2 ], 0≤t≤T _c

The _voltages u _pr2 (t) and u _pr4 (t) from the output of the amplifier 14 and the phase shifter 21 by -90 ° are fed to the two inputs of the adder 22, at the output of which the total voltage

u _Σ (t) = U _Σ cos [ω _np2 (t) -φ _k (t) + φ _np2 ] , 0≤t≤T _c ,

where U _Σ = 2U _pr2 ,

which is fed to the second input of the multiplier 23. At the output of the latter, a harmonic voltage is generated

u ₁ (t) = U ₁ cos (ω _Г2 t + φ _Г2 ), 0≤t≤T _s ,

Where

K ₂ - transfer coefficient of the multiplier,

which is allocated by a narrow-band filter 24 (tuning frequency ω _n which is chosen equal to the frequency of the second local oscillator 2.2 ω _n = ω _G2 ), is detected by the amplitude detector 25 and is fed to the control input of the key 26, opening it. In the initial state, the key 26 is always closed.

The voltage u _Σ (t) from the output of the adder 22 through the public key 26 is supplied to the input of the clipper 15, where it is clipped and recorded in the buffer memory 16. The registration is synchronized by standard 1 frequency and time.

In the second step (when transmitting the signal from point B), the switch 4 must be open and the signal α ₃ from the generator 3 through the clipper 10 enters the same storage device 11. The relay signal α _{4 is} recorded as α ₂ in the storage device 16.

At an arbitrary point in time t ₃ ^V = t ₃ ^V + Θ, the noise of the microwave signal (signal β ₃ ) is similarly generated and recorded by the clock of the second point. The generated signal is converted at a frequency of ω ₁ , amplified by power, emitted amplified signal in the direction of the same satellite repeater.

At the same time t ₃ ^B = t ₃ ^A , the same noise microwave signal (signal α ₃ ) is formed using the same code sequence according to the clock of the first point A. Register it at the first point A. Accept the on-board equipment of the satellite relay signal at a frequency of ω ₁ (signal α ₃ ), re-emit it to points A and B at a frequency of ω ₂ while maintaining phase relationships, receive a relay signal at both points, convert it to video frequency, recorded at time t ₄ ^A and t ₄ ^V, respectively (signal α ₄ , β ₄ ).

The correlation processing of two pairs of registered signals in the meter 17 determines at each point the following time delays:

τ ₁ = β ₁ ⊗ β ₂ = t ₂ ^B -t ₁ ^B = a ₁ + b ₁ + (Δ _B ^И + Δ _B ^П + ΔS) + Δt,

τ ₂ = α ₃ ⊗ α ₄ = t ₄ ^A -t ₃ ^A = a ₃ + b ₂ + (Δ _B ^AND + Δ _A ^P + ΔS) -Δt,

τ ₃ = α ₁ ⊗ α ₂ = t ₂ ^A -t ₁ ^A = a ₁ + a ₂ + (Δ _A ^AND + Δ _A ^P + ΔS),

τ ₄ = β ₃ ⊗ β ₄ = t ₄ ^B -t ₃ ^B = b ₂ + b ₃ + (Δ _B ^AND + Δ _B ^P + ΔS),

and the corresponding interference frequencies F _i (i = 1,2,3,4), which determine the derivatives of these delays:

,

,

,Where

,

a _j , b _j (j = 1, 2, 3) is the signal propagation time between the satellite and points A and B, respectively (Fig. 1);

Δ _A ^And , Δ _B ^And - signal delays in the radiating equipment of both points;

Δ _A ^P , Δ _V ^P - signal delay in the receiving and recording equipment;

ΔS - signal delay in the onboard satellite repeater;

Δt-t ₁ ^B -t ₁ ^A is the desired difference in the clock readings at the same physical moment.

,

, получаем:Putting a _j and b _j linear functions with derivatives

,

we get:

,

,

Where

,

,

,

,

,

,

,

,

Δ ' _{A, B} , Δ'' _{A, B} are the signal delays in the atmosphere at frequencies ω ₁ and ω _2, respectively;

ν - relativistic correction (Sagnac effect);

ω is the angular velocity of the Earth;

c is the speed of light;

D is the area of the quadrangle OA'S'B ', formed in the equatorial plane by the center of mass of the Earth, the projections of points A, B and the satellite S.

Corrections γ on the mobility of the satellite repeater during a single measurement is most easily reduced to zero by the corresponding choice of the free parameter Θ:

,

,

which should be calculated at the beginning of measurements by approximate ephemeris data, and then clarified by the results of current measurements.

As for the correction δ for hardware delays, it can be found by calibration using the “zero base” method.

The atmospheric correction ε is also taken into account.

At point B, the equipment works similarly, only the order of steps there is the opposite. To calculate the difference between the clock readings Δt, it is now sufficient to exchange the received digital data between the points, which can be done via ordinary telephone or telegraph communication channels.

The above operation of the device that implements the proposed method, corresponds to the reception of useful signals on the main channel at a frequency of ω ₂ (figure 4).

If a false signal (interference)

u _З (t) = U _З cos (ω _З t + φ _З ), 0≤t≤T _З ,

is taken along the mirror channel at a frequency of ω ₃ , the following voltages are allocated by amplifiers 14 and 20 of the second intermediate frequency:

u _pr5 (t) = U _pp5 cos (ω _p2 t-φ _p5 ),

u _pr6 (t) = U _pp5 cos (ω _p2 t-φ _p5 -90 °), 0≤t≤T _З ,

;Where

;

ω _p2 = ω ₃ -ω _G2 - the second intermediate (difference) frequency;

ω _pp5 = ω ₃ -ω _G2 .

The voltage u _p6 (t) from the output of the amplifier 20 of the second intermediate frequency is supplied to the input of the phase shifter 21 by -90 °, at the output of which the following voltage is generated

u _CR7 (t) = U _CR5 cos (ω _CR2 t + φ _CR5 -90 ° -90 °) = - U _CR5 cos (ω _CR2 t-φ _CR5 ), 0≤t≤T _З ,

The voltage u _p5 (t) and u _p7 (t) supplied to the two inputs of the adder 22 are compensated at its output.

Therefore, a false signal (interference) received on the mirror channel at a frequency of ω ₃ is suppressed.

For a similar reason, a false signal (interference) received on the second combination channel at a frequency of ω _{k2 is also} suppressed.

If a false signal (interference) is received on the first combinational channel at a frequency ω _k1

u _K1 (t) = U _K1 cos (ω _K1 t + φ _K1 ), 0≤t≤T _K1 ,

the amplifiers 14 and 20 of the second intermediate frequency are allocated the following voltages:

u _CR8 (t) = U _CR2 cos (ω _CR2 t + φ _CR8 ),

u _PR9 (t) = U _CR8 cos (ω _CR2 t + φ _CR8 + 90 °), 0≤t≤T _K1 ,

;Where

;

ω _CR2 = 2ω _Г2 -ω _K1 - the second intermediate (difference) frequency;

φ _pr8 = φ _Г2 -φ _K1 .

The voltage u _pr9 (t) from the output of the amplifier 20 of the second intermediate frequency is supplied to the input of the phase shifter 21 by -90 °, at the output of which a voltage is generated

u _pr10 (t) = U _pr8 cos (ω _pr2 t + φ _pr8 + 90 ° -90 °) = U _pp cos (ω _p2 t + φ _pr8 ), 0≤t≤T _K1 .

Voltages u _CR8 (t) and u _CR10 (t) are supplied to two inputs of the adder 22, at the output of which the following total voltage is formed

u _Σ1 (t) = U _Σ1 cos (ω _np2 t + φ _np8 ) , 0≤t≤T _K1 ,

where U _Σ1 = 2U _pr8 .

This voltage is supplied to the second input of the multiplier 23, at the output of which the following harmonic voltage is formed

u ₂ (t) = U ₂ cos (2ω _Г2 t + φ _Г2 ), 0≤t≤T _K1 ,

.Where

.

This voltage does not fall into the passband of the narrow-band filter 24. Therefore, a false signal (interference) received on the first combination channel at a frequency ω _K1 is suppressed.

If a false signal (interference) is received through the direct passage channel at the second intermediate frequency ω _CR2 (ω _p = ω _CR2 ), then it goes to the first input of the adder 29, is allocated by a narrow-band filter 27, is inverted by 180 ° phase in phase inverter 28 and supplied the second input of the adder 29, the output of which it is compensated.

Therefore, a false signal (interference) received through the direct channel at the second intermediate frequency ω _pr2 is suppressed by the phase-compensation method using a filter plug consisting of a narrow-band filter 27, a phase inverter 28, and an adder 29. In this case, the tuning frequency is ω _{n of the} narrow _- band filter 27 is chosen equal to the second intermediate frequency ω _CR2 (ω _n = ω _CR2 ) (figure 4).

If false signals (interference) are received via the intermodulation channel in the frequency band Δω _p1 = ω _II -ω _I ,

where ω _I and ω _II are the boundary frequencies that determine the frequency band Δω _p1 , located "to the left" of the passband Δω _{p of the} receiver, hit by two or more signals leads to the formation of intermodulation interference,

then they through the adder 29, which has only one arm, go to the first input of the adder 32, are allocated by a band-pass filter 30, are inverted in phase 180 ° in the phase inverter 31, and fed to the second input of the adder 32, at the output of which they are compensated.

Consequently, spurious signals (noise) received by intermodulation channel in the band Δω _n1 frequencies fed fazokompensatsionnym method using notch filter consisting of a bandpass filter 30, phase inverter 31 and the adder 32, and the tuning frequency ω _H1 bandpass filter 30 is selected as follows way (figure 5):

.

.

If false signals (interference) are received via the intermodulation channel in the frequency band

Δω _n2 = ω _IV -ω _III ,

where ω _III and ω _IV are the boundary frequencies that determine the frequency band Δω _p2 located "to the right" of the passband Δω _{p of the} receiver, if two or more signals get into it, it leads to the formation of intermodulation interference,

then they pass through adders 29 and 32, which have only one arm, go to the first input of adder 35, are separated by a band-pass filter 33, are inverted by 180 ° phase in phase inverter 34, and fed to the second input of adder 35, at the output of which they are compensated.

Consequently, spurious signals (interference) of the received channel in the band intermodulation Δω _n2 frequencies fazokompensatsionnym suppressed by using a notch filter consisting of a bandpass filter 33, phase inverter 34 and the adder 35. In this case the tuning frequency ω _2n bandpass filter 33 is selected as follows way (Fig.6):

.

.

The clock synchronization method allows you to:

- achieve extreme measurement accuracy (about ± 0.1 ns) using VLBI technology and relay technology, which is already widely used in practice;

- to form the microwave signals necessary for the measurement at ground points, which makes it possible to gradually increase the accuracy of measurements by optimizing the signal structure and improving the ground-based recording technique without interfering with the satellite onboard equipment;

- increase the efficiency of measurements, i.e. bring the time interval from the beginning of measurements to obtain results up to several tens of seconds (almost to the time of correlation signal processing);

- to avoid the installation on board of a satellite of highly stable time-keepers and time interval meters, to limit the on-board equipment to only a phase-stable microwave signal relay system.

The suppression of false signals (interference) received via the mirror and the second Raman channel is provided by the phase compensation method, which is implemented by the local oscillator 2.2, mixers 13 and 19, amplifiers 14 and 20 of the second intermediate frequency, phase shifters 18 and 21 at + 90 ° and -90 ° and adder 22.

The suppression of false signals (interference) received via the first combinational channel is provided by the narrow-band filtering method, which is implemented by a multiplier 23, a narrow-band filter 24, an amplitude detector 25, and a key 26.

Thus, the proposed method in comparison with the prototype provides increased noise immunity and synchronization accuracy of remote time scales. This is achieved by suppressing false signals (interference) received on the direct channel and intermodulation channels. Moreover, the suppression of false signals (interference) received through the direct channel at the second intermediate frequency ω _pr2 is provided by the phase-compensation method, which is implemented by a narrow-band filter 27, phase-inverter 28 and adder 29.

Suppression of spurious signals (noise) received by intermodulation channel in the frequency band Δω _n1 fazokompensatsionnym is provided a method which is implemented by a bandpass filter 30, the adder 31 and the phase inverter 32.

Suppression of spurious signals (interference) of the received channel in the band intermodulation Δω _n2 frequencies provided fazokompensatsionnym method which is implemented by a bandpass filter 33, phase inverter 34 and the adder 35.

Claims (1)

A clock synchronization method based on the simultaneous reception of noise-like microwave signals from an artificial Earth satellite by spaced ground points, their coherent conversion to a video frequency, digital recording of received signals and determining the time delay of the arrival of the same signal to synchronization points by correlation processing of recorded signals, in magnitude of which the time scales are compared, while at the initial time t ₁ by the hours of the first point using the code after sequences generate a noise microwave signal, register it at the same point, the generated signal is converted to frequency ω ₁ , amplify it by power, study the amplified signal in the direction of the artificial Earth satellite relay, at the same time t ₁ by the clock of the second point using the same code sequence, the same microwave noise signal is generated, it is recorded at the second point, the signal at a frequency ω _{1 is} re-emitted to the first and second signals by the onboard equipment of the artificial Earth-relay satellite points at a frequency of ω ₂ while maintaining phase relationships, at an arbitrary time t ₃ according to the hours of the second point, they similarly generate and register a noise microwave signal, the generated signal is converted to a frequency of ω ₁ , amplifies it by power, emits an amplified signal in the direction of the same artificial Earth-satellite repeater in the same time t ₃ by the clock of the first item by using the same code sequence form the same noise microwave signal, it is recorded in the first paragraph, taking onboard equipment suit sstvennogo earth satellite repeater signal at frequency ω ₁ and re-emit it at first and second points at the frequency ω ₂ while preserving phase relationships, receiving a signal at the frequency ω ₂ is converted in frequency using the voltage of the second local oscillator shifted in phase by + 90 °, isolate the voltage of the second intermediate frequency, shift it in phase by -90 °, summarize with the initial voltage of the second intermediate frequency, multiply the resulting total voltage with the received signal, emit the harmonic voltage at the frequency _n2 of the second local oscillator is detected it and is used to permit further processing of the received signal, characterized in that before converting the frequency emit spurious signals (interference) received on the bus directly passing the second intermediate frequency ω _np2, inverting its phase by 180 °, summarize with the original false signal (interference) and compensate for it, isolate the false signals (interference) received in the frequency band Δω _p1 = ω _II -ω _I , where ω _I and ω _II are the boundary frequencies that determine the frequency band Δω _p1 located " to the left of the polo s Δω _n receiver pass, entering into which leads to intermodulation of two or more signals, invert their phase by 180 °, summed with the original false signals (noisy) and compensate them emit spurious signals (interference) received in the band Δω _p2 = ω _IV- ω _III , where ω _III and ω _IV are the boundary frequencies defining the frequency band Δω _p2 located “to the right” of the frequency band Δω _{p of the} receiver, if two or more signals get into it, it causes intermodulation interference, invert them in phase by 180 °, s summarize with the original false signals (interference) and compensate for them.

Images