RU2386159C2 - Clock synchronisation system - Google Patents

Abstract

FIELD: physics.

SUBSTANCE: invention can be used in radio interferometry with very long bases, in universal time and frequency service, as well as for exchanging confidential discrete and analogue information between ground stations which are far apart using a geostationary artificial earth satellite transponder and protection of the said information from unauthorised access. The clock synchronisation system has an artificial earth satellite transponder, first and second ground stations, each of which has a time and frequency standard, first and second heterodynes, a transceiving antenna, a duplexer, a first power amplifier, a first mixer, a first intermediate frequency amplifier, a first clipper, a first memory unit, a correlator, a pseudonoise generator, a switch, a second mixer, a second intermediate frequency amplifier, a second power amplifier, a second clipper, a second memory unit, a selector, first and second amplitude limiters, first, second and third synchronous detectors, an analogue descrambler, a multiplier, band-pass filter, first, second and third phase detectors, a digital descrambler, first and second +30° phase changers, first and second -30° phase changers, first, second, third and fourth subtracting units, first and second +90° phase changers. The pseudonoise generator has a high-frequency driving oscillator, a discrete message source, a digital scrambler, a phase manipulator, an analogue message source, an analogue scrambler, an amplitude modulator and a band-pass amplifier. The selector has a phase doubler, first and second spectral width measuring devices, a comparator unit, a threshold unit and a switch. The analogue scrambler, which uses a frequency inversion technique, has a clock generator, a divider - shaper, a band-pass filter, a balanced modulator, an analogue switch, an adder and a low-pass filter.

EFFECT: increased noise-immunity and reliability of exchanging confidential information between ground stations through suppression of narrow-band interference.

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Description

The proposed system relates to communication technology and can be used in radio interferometry with ultra-long bases, in the service of a single time and frequency, as well as for the exchange of confidential discrete and analog information between ground points spaced over long distances, using a geostationary IC3 repeater and protecting this information from unauthorized access.

There are known devices for clock synchronization (autosw. USSR No. 591.799, 614.416, 970.300, 1.180.835, 1.244.632, 1.278.800; RF patents No. 2.001.423, 2.003.157, 2.040.035, 2.177.167; BC Gubanov, A. M. Finkelstein, P. A. Friedman, Introduction to Radio Astronomy (Moscow, 1983).

Of the known devices, the closest to the proposed system is the "Clock Synchronization Device" (RF patent No. 2.177.167, G04C 11/02, 2000), which is selected as a prototype.

The specified device provides a comparison of time scales spaced over a long distance, and is based on the use of the duplex method of communication between ground points through a geostationary satellite repeater.

The main advantage of the duplex communication method is that it eliminates the length of the signal path. Therefore, its accuracy mainly depends on the parameters of the onboard transponder, the type of signal used, and the technique for measuring time intervals.

The known device also allows the exchange of confidential analog and discrete information between ground stations with protection against unauthorized access and the use of two frequencies f ₁ , f ₂ and complex signals with combined phase shift keying and amplitude modulation (FMN-AM).

These signals are broadband, the width of their spectrum Δf _c is determined by the duration τ _{E of} elementary packages (Δf _c = 1 / τ _E ) and amounts to tens of megahertz.

Due to the wide frequency band used by complex PSK-AM signals, a lot of narrow-band interference gets into it. Therefore, the absence of noise suppressing blocks in the device sharply reduces the noise immunity and reliability of the exchange of confidential information between ground points located over long distances.

An object of the invention is to increase the noise immunity and reliability of the exchange of confidential information between ground points spaced over long distances by suppressing narrow-band interference.

The problem is solved in that the clock synchronization system, containing in accordance with the closest analogue a geostationary satellite repeater, first and second ground stations, each of which contains an analog descrambler, the output of which is the second output of the system, a digital descrambler, the output of which is the third output of the system connected in series with a time and frequency standard, a first local oscillator, a first mixer, the second input of which is connected through a series-connected duplexer and a first power amplifier nen with a transceiver antenna, an amplifier of the first intermediate frequency, a selector, a first amplitude limiter and a first synchronous detector, the second input of which is connected to the output of the selector, the first clipper connected in series to the output of the first amplitude limiter, the second input of which is connected to the second output of the time and frequency standard, the first memory unit and the correlator, the output of which is the first output of the system, connected in series to the first output of the time and frequency standard of the second local oscillator, the second with a mixer, the second input of which is connected via a switch to the first output of the pseudo-noise signal generator, an amplifier of the second intermediate frequency and a second power amplifier, the output of which is connected to the duplexer input, the pseudo-noise signal generator is connected in series to the third output of the time and frequency standard, the second amplitude limiter, the second clipper , the second input of which is connected to the second output of the standard time and frequency, and the second memory block, the output of which is connected to the second input of the correlator, the last A multiplier that is connected to the output of the first amplitude limiter, the second input of which is connected to the output of the first local oscillator, the bandpass filter and the first phase detector, the second input of which is connected to the second output of the second local oscillator, differs from the nearest analogue in that each ground station is equipped with two phase shifters + 30 °, two phase shifters at -30 °, two phase shifters at + 90 °, second and third synchronous detectors, second and third phase detectors and four subtraction units, and to the output The first amplitude limiter is connected in series to the first + 30 ° phase shifter, the second synchronous detector, the second input of which is connected to the selector output, the first subtraction unit, the first + 90 ° phase shifter and the second subtraction unit, the second input of which is connected to the output of the first synchronous detector, and the output is connected to the input of the analog descrambler, the first phase shifter at -30 ° and the third synchronous detector, the second input of which is connected to the output of the villages, are connected in series to the output of the first amplitude limiter ctor, and the output is connected to the second input of the first subtraction unit, the second phase shifter at +30, the second phase detector, the second input of which is connected to the output of the bandpass filter, the third subtraction block, the second phase shifter at + 90 ° and the fourth are connected to the second output of the second local oscillator a subtraction unit, the second input of which is connected to the output of the first phase detector, and the output is connected to the input of the digital descrambler, the second phase shifter at -30 ° and the third are connected in series to the second output of the second local oscillator AZOV detector, a second input coupled to an output of the bandpass filter, and an output connected to the second input of the third subtracter.

Figure 1, 10 presents a structural diagram of the proposed clock synchronization system; figure 2 is a geometric arrangement of the satellite-relay S and two ground points A and B; figure 3 is a timing chart of the duplex method of comparing watches; 4 is a frequency diagram explaining a frequency conversion process of a clock signal; figure 5 is a structural diagram of a generator of a pseudo-noise signal; figure 6 is a structural diagram of a selector; Fig.7 is a structural diagram of an analog scrambler using the frequency inversion method; on Fig - presents the spectra of the input a) and converted b) signals; figure 9 - shows a timing diagram explaining the principle of operation of the proposed clock synchronization system.

The clock synchronization system contains a geostationary IS3 repeater S, first A and second B ground stations, each of which contains a standard 1 (1.1) time and frequency reference, a first local oscillator 2 (2.1), a first mixer 7 (7.1), the second input of which through series-connected duplexer 5 (5.1) and the first power amplifier 6 (6.1) connected to the transceiver antenna 4 (4.1), the amplifier 8 (8.1) of the first intermediate frequency, selector 19 (19.1), the first amplitude limiter 20 (20.1), the first clipper 9 (9.1), the second input of which is connected to the second output the house of reference 1 (1.1) of time and frequency, the first memory block 10 (10.1) and correlator 11 (11.1), the output of which is the first I output of the system, connected in series to the first output of reference 1 (1.1) of time and frequency, the second local oscillator 3 ( 3.1), a second mixer 14 (14.1), the second input of which is connected via a switch 13 (13.1) to the first output of the pseudo-noise signal generator 12 (12.1), a second intermediate frequency amplifier 15 (15.1) and a second power amplifier 16 (16.1), the output of which connected to the input of duplexer 5 (5.1), connected in series to the third output of the standard 1 (1.1) time and frequency generator 12 (12.1) of the pseudo-noise signal, the second amplitude limiter 21 (21.1), the second clipper 17 (17.1), the second input of which is connected to the second output of the standard 1 (1.1) time and frequency, and the second block 18 (18.1) memory, the output of which is connected to the second input of the correlator 11 (11.1), connected in series to the output of the first amplitude limiter 20 (20.1), the first phase shifter 49 (49.1) + 30 °, the second synchronous detector 53 (53.1), the second input which is connected to the output of the selector 19 (19.1), the first calculation unit 57 (57.1), the first phase shifter 59 (59.1 ) by + 90 °, the second block 61 (61.1) of subtraction, the second input of which through the first synchronous detector 22 (22.1) is connected to the output of the selector 19 (19.1) and the first amplitude limiter 20 (20.1), and an analog descrambler 23 (23.1), the output of which is the second II output of the system, connected in series to the output of the first amplitude limiter 20 (20.1), the first phase shifter 51 (51.1) by -30 ° and the third synchronous detector 54 (54.1), the second input of which is connected to the output of the selector 19 (19.1), the output is connected to the second input of the first subtraction unit 57 (57.1), sequentially connecting the multiplier 24 (24.1), the second input of which is connected to the output of the first local oscillator 2 (2.1), the bandpass filter 25 (25.1), the first phase detector 26. (26.1), the second input of which is connected to the second output of the second local oscillator, 3 (3.1), the fourth block 62 (62.1) of subtraction and digital descrambler 27 (27.1), the output of which is the third III output of the system, connected in series to the second output of the second local oscillator 3 (3.1), the second phase shifter 50 (50.1) by + 30 °, a second phase detector 55 (55.1), the second input of which is connected to the output olos filter 25 (25.1), the third block 58 (58.1) subtraction and the second phase shifter 60 (60.1) + 90 °, the output of which is connected to the second input of the fourth block 62 (62.1) subtraction, connected in series to the second output of the second local oscillator 3 (3.1 ) the second phase shifter 52 (52.1) by -30 ° and the third phase detector 56 (56.1), the second input of which is connected to the output of the bandpass filter 25 (26.1), and the output is connected to the second input of the third block 58 (58.1) of subtraction.

The pseudo-noise signal generator 12 (12.1) comprises a high-frequency driver 28, a discrete message source 29, a digital scrambler 30, a phase manipulator 31, an analog message source 32, an analog scrambler 33, an amplitude modulator 34 and a band amplifier 35.

The selector 19 (19.1) comprises a phase doubler 36, a first 37 and a second 38 spectral width meters, a comparison unit 39, a threshold unit 40 and a key 41.

An analog scrambler 33 using the frequency inversion method comprises a clock 42, a divider-shaper 43, a bandpass filter 44, a balanced modulator 45, an analog switch 46, an adder 47, and a low-pass filter 48.

The system operates as follows.

At point A, using a generator 12, a pseudo-noise microwave signal α _{1 is formed} (Fig. 3) with combined phase shift keying (FMN-AM). To this end, the high-frequency voltage (Fig.9, a):

u _c (t) = U _c cos (2πf _c t + φ _c ), 0≤T≤T _c ,

where U _c , f _c , φ _c , Tp - amplitude, carrier frequency, initial phase and signal duration;

from the output of the master oscillator 28 (Fig. 5) is fed to the first input of the phase manipulator 31, the second input of which is supplied with a modulating code M (t) (Fig. 9, b) from the output of the digital scrambler 30. The input of the latter is connected to a source 29 of a discrete message .

At the output of the phase manipulator 31, a phase-shift (PSK) signal is generated (Fig. 9, c):

u ₁ (t) = U _c cos [2πf _c t + φ _k (t) + φ _c ], 0≤T≤T _c ,

where φ _k (t) = {0, π} is the manipulated component of the phase that displays the law of phase manipulation in accordance with the modulating codes M (t), and φ _k (t) = const for kτ <t <(k + 1) τ _Oe and can change abruptly at t = kττ _Oe , i.e. at the boundaries between elementary premises (k = 1, ..., N);

τ _E , N is the duration and number of chips that make up the signal with a duration of T _c (T _c = Nτ _e ).

This signal is fed to the first input of the amplitude modulator 34, to the second input of which a modulating function m (t) is supplied (Fig. 9, d) from the output of the analog scrambler 33. The input of the latter is connected to the source of the analog message.

At the output of the amplitude modulator 34, a pseudo-noise signal is generated with combined phase shift keying and amplitude modulation (FMN-AM) (Fig. 9, e):

u ₂ (t) = U _c [1 + m (t)] cos [2πf _c t + φ _r (t) + φ _c ,

where m (t) is the modulating function that reflects the law of amplitude modulation.

Digital 30 and analog 33 scramblers implement cryptographic methods, which are the most effective methods of protecting confidential discrete and analog information.

Cryptographic methods of information protection are special methods of encrypting, encoding and converting information, as a result of which the content becomes inaccessible without presenting a cryptogram key and reverse transformation.

With the digital method of closing the transmitted message, four main groups can be conditionally distinguished:

1) substitution - the characters of a discrete message are replaced by other characters in accordance with a predetermined rule;

2) permutation - the symbols of a discrete message are rearranged according to some rule within a given block of a transmitted discrete message;

3) analytical conversion - the encrypted message is converted according to some analytical rule;

4) combined conversion - the original discrete message is encrypted with two or more encryption methods.

With analog scrambling, the signal undergoes the following transformations:

1) frequency inversion;

2) frequency permutation;

3) temporary permutation.

As an example, Fig. 7 shows a block diagram of an analog scrambler using the frequency inversion method. The device consists of a clock generator 42 that generates a signal with a frequency of 7 kHz, a splitter-shaper 43 of a carrier of 3.5 kHz, an analog switch 46, a balanced modulator 45, an input band-pass filter 44, an adder 47 and a low-pass filter 48.

On Fig presents the spectra of the input (a) and converted (b) signals, where f ₀ = 3500 Hz.

The generated signal u ₂ (t) is fed to the first input of the mixer 14, the second input of which is supplied with the voltage of the second local oscillator 3, stabilized by element 1 of frequency and time:

u _g2 (t) = u _g2 cos (2πf _g2 t + φ _g2 ).

In this case, the switch 13 is closed, and the analog switch at step B is open.

At the output of the mixer 14, voltages of combination frequencies are generated. The amplifier 15 is allocated the voltage of the second intermediate (total) frequency:

u _np2 (t) = U _p2 [1 + m (t)] cos [2πf _pr2 t + φ _k (t) + φ _pr ], 0≤t≤T _c ,

;Where

;

K ₁ - gear ratio of the mixer;

f _CR2 = f _s + f _g2 - the second intermediate (total) frequency;

φ _CR2 = φ _c + φ _g2 ,

which is amplified in the power amplifier 16 and radiated through the duplexer 5 and the transceiver antenna 4 in the direction of the IS3 repeater at a frequency f ₁ = f _CR2 .

However, the same signal u ₂ (t) is limited in amplitude in the amplitude limiter 21:

u ₃ (t) = U ₀ cos [2πf _c t + φ _k (t) + φ _c ], 0≤T≤T _c ,

where U ₀ is the limit threshold;

clipped in the second clipper 17 and recorded in the buffer block 18 of the memory. The operation of the clipper 17 is synchronized by the standard 1 frequency and time.

The signal received by the on-board receiver of the geostationary satellite repeater is re-emitted to points A and B at a frequency of f ₂ while maintaining phase relationships in the entire frequency band of the signal. The directivity pattern of the onboard antenna of the geostationary IS3 repeater is selected so that this signal can be received at both points A and B.

Signal received in paragraph A by antenna 4.1 relayed by a geostationary satellite repeater

u ₄ (t) = U ₄ [1 + m (t)] cos [2πf _pr2 t + φ _k (t) + φ _pr2 ],

amplified by the first power amplifier 6.1 and fed to the first input of the mixer 7.1, the second input of which is supplied with voltage u _g2 (t) of the local oscillator 2.1.

At the output of mixer 7.1, a voltage of combination frequencies is generated. Amplifier 8.1 distinguishes the voltage of the intermediate (differential) frequency:

u _CR1 (t) = U _CR1 [1 + m (t)] cos [2πf _CR1 t + φ _k (t) + φ _CR1 ] 0≤t≤T _c ,

Where

_pp1 f = f _r2 -f _np2 - first intermediate (difference) frequency;

φ _pr1 = φ _np2 -φ _{g2 .}

This voltage is supplied to the input of the selector 19.1, consisting of a phase doubler 36, meters 37 and 38 of the width of the spectrum, block 39 comparison, threshold block 40 and key 41.

At the same time, at the output of the doubler 36 of the phase, which can be used as a multiplier, the two inputs of which are supplied with the same voltage U _p1 (t), forming the following voltage:

u ₅ (t) = U ₅ (t) cos (4πf _pr1 t + 2φ _pr1 ), 0≤t≤T _c ,

where U ₅ (t) = U ² _pr1 [1 + m (t)] ² .

Since 2φ _r (t) = {0.2π}, phase manipulation is already absent in the indicated voltage.

The spectrum width of the FMN-AM signal Δf _c is determined by the duration τ _{e of the} elementary premises Δf _c = 1 / τ _e . Whereas the width of the spectrum of its second harmonic is determined by the duration T _{s of the} signal Δf ₂ = 1 / T _s , i.e. the width of the spectrum of the second harmonic Δf ₂ is N times smaller than the width of the spectrum Δf _{c of the} input signal Δf _c / Δf _a = N, where N is the number of chips (for the Barker code N = 7, for m - the sequence N = 1023).

Therefore, when the phase of the QPSK-AM signal is doubled, its spectrum “folds” N times. This circumstance makes it possible to detect a complex signal with combined phase shift keying and amplitude modulation and apply a new type of selection - structural selection.

The width of the spectrum Δf _{c of the} input QPSK-AM signal is measured using meter 37, and the width of the spectrum Δf _{2 of} its second harmonic is measured using meter 38. Voltages U ₁ and U ₂ proportional to Δf _c and Δf _2, respectively, from the outputs of meters 37 and 38 spectral widths are fed to two inputs of the comparison unit 39. Since U ₁ >> U ₂ , a positive pulse is generated at the output of the comparison unit 39, the amplitude of which exceeds the threshold voltage in the threshold block 40. When the threshold level U _p is exceeded, a constant voltage is generated in the threshold block 40, which is supplied to the control input of the key 41 and opens it. In the initial state, the key 41 is always closed.

The voltage U _pr1 (t) ( _Fig.9 , e) through the public key 41 from the output of the amplifier 8.1 of the intermediate frequency is fed to the input of the amplitude limiter 20.1 and to the information input of the first synchronous detector 22.1. At the output of the amplitude limiter 20.1, a voltage is generated (Fig. 9, g)

u ₆ (t) = U ₀ (t) cos [(2πf _CR1 t + φ _k (t) + φ _CR1 ], 0≤t≤T _c ,

which is an FMN-AM signal and is supplied to the reference input of the synchronous detector 22.1 as a reference voltage to the control input of the clipper 9.1 and to the first input of the multiplier 24.1. The operation of clipper 9.1 is also synchronized with the clock frequency of the standard 1.1 frequency and time. Then, the indicated voltage is recorded by the memory unit 18.1.

The output of the synchronous detector 22.1 is formed of a low-frequency voltage (Fig.9, h):

u _н1 (t) = U _н1 [1 + m (t)],

Where

K ₂ is the transfer coefficient of the synchronous detector;

proportional to the modulating function m (t).

This voltage through the subtraction unit 61 is fed to the input of the analog descrambler 23.1, the principle of operation of which corresponds to the principle of analog scrambler 33, but has the opposite character. At the output of the descrambler 23.1 (second output of the II system), the source information of the analog message source is generated.

The voltage u _g2 (t) of the local oscillator 22 is supplied to the second input of the multiplier 24.1, a voltage is generated at its output (Fig. 9, and):

u ₇ (t) = U ₇ (t) cos [(2πf _g1 t + φ _k (t) + φ _g1 ],

;Where

;

K ₃ - transfer coefficient of the multiplier;

which is an FMN signal at a stable frequency f _g1 local oscillator 3.1. This voltage is allocated by a band-pass filter 25.1 and fed to the information input of the phase detector 26.1, to the reference input of which the voltage u _g1 (t) of the local oscillator 3.1 is supplied.

The output of the phase detector 26.1 is formed of a low-frequency voltage (Fig.9, k):

u _n2 (t) = U _n2 cosφ _k (t),

;Where

;

To ₄ - the transfer coefficient of the phase detector;

proportional to the modulating code M (t) (Fig.9, b). This voltage is supplied through the subtraction unit 62 to the input of the digital descrambler 27.1, the principle of operation of which corresponds to the principle of operation of the digital scrambler 30, but has the opposite character. The output of the descrambler 27.1 (third output of the III system) is the source information of the source 29 of the discrete message.

If there is narrowband interference on the air:

u _p1 (t) = U _p1 cos (2πf _p1 t + φ _p1 ),

the frequency f _p1 which is slightly different from the first intermediate frequency

f _p1 received signal:

-f f _{n1 pr1} = Δf ₁ ≤Δf _F1

where Δf _f1 is the passband of synchronous detectors 22.1, 53.1 and 54.1, then the additive mixture of the received signal u ₆ (t) and narrowband interference u _p1 (t)

u _Σ1 (t) = u ₅ (t) + u _п1 (t),

from the output of the selector 19.1 goes to the first inputs of synchronous detectors 22.1, 53.1 and 54.1.

The reference voltage u ₆ (t) from the output of the amplitude limiter 20.1 is simultaneously supplied to the second input of the synchronous detector 22.1 and to the inputs of the phase shifters 49.1 and 51.1 at + 30 ° and -30 °, at the output of which the corresponding voltage is generated:

u ₈ (t) = U ₀ cos [2πf _pr1 t + φ _k (t) + φ _pr1 + 30 °],

u ₉ (t) = U ₀ cos [2πf _pr1 t + φ _k (t) + φ _pr1 -30 °], 0≤t≤T _c ,

which are fed to the second (reference) inputs of synchronous detectors 53.1 and 54.1.

At the output of synchronous detectors 22.1, 53.1 and 54.1 in this case, the following voltages are distinguished:

u _н3 (t) = U _н1 [1 + m (t)] + U _н3 cos [2π (f _п1 -f _пр1 ) t + φ _п1 -φ _пр1 ],

u _н4 (t) = U _н1 [1 + m (t)] + U _н3 cos [2π (f _п1 -f _пр1 ) t + φ _п1 -φ _пр1 + 30 °],

u _н5 (t) = U _н1 [1 + m (t)] + U _н3 cos [2π (f _п1 -f _пр1 ) t + φ _п1 -φ _пр1 -30 °]

.Where

.

The output of the first subtraction block 57.1 produces the following differential voltage:

Δu _n1 (t) = u _n4 -u _n5 = U _n3 {cos [2π (f _n1 -f _np1 ) t + φ _n1 -φ _np1 + 30 °] -cos [2π (f _n1 -f _np1 ) t + φ _pr1 -30 °]} = U _H3 sin [2π (f _{n1 pr1} -f) t + φ _n1 -φ _pr1].

The analysis of the obtained differential voltage Δu _н1 (t) shows that it is an estimate of the noise component, which differs from the noise component in the main channel by a + 90 ° phase rotation.

The differential voltage Δu _н1 (t) from the output of the subtraction unit 57.1 is fed to the input of the phase shifter 59.1 by + 90 °, at the output of which a voltage is generated

_H2 Δu (t) = U _H3 cos [2π (f _{n1 pr1} -f) t + φ _n1 -φ _pr1]

which goes to the second input of the subtraction block 61.1. The output of the latter produces the following differential voltage:

Δu _n2 (t) = u _n3 (t) -Δu _n2 (t) = U _n1 [1 + m (t)],

in which the interference component is already absent.

This voltage is fed to the input of the amplitude detector 23.1.

If there is narrowband interference on the air

u _p2 (t) = U _p2 cos (2πf _p2 t + φ _p2 ),

the frequency f _p2 which is slightly different from the frequency f _{g1 of the} local oscillator 3.1

f _r1 -f _n2 = Δf ₂ ≤Δf _Q2,

where Δf _f2 is the passband of phase detectors 26.1, 55.1 and 56.1, then the additive mixture of the received FMN signal u ₇ (t) and narrow-band interference u _p2 (t)

u _Σ2 (t) = u ₇ (t) + u _n2 (t)

from the output of the bandpass filter 25.1 goes to the first inputs of the phase detectors 26.1, 55.1 and 56.1.

Local oscillator voltage 3.1

u _g1 (t) = U _g1 cos (2πf _g1 t + φ _g1 )

arrives at the second (reference) input of the first phase detector 26.1 and at the inputs of the phase shifters 50.1 and 52.1 at + 30 ° and -30 °, at the output of which the corresponding voltages are formed:

u ₁₀ (t) = U _g1 cos (2πf _g1 t + φ _g1 + 30 °),

u ₁₁ (t) = U _g1 cos (2πf _g1 t + φ _g1 + 30 °),

which are fed to the second (reference) inputs of the phase detectors 55.1 and 56.1.

At the output of the phase detectors 26.1, 55.1 and 56.1 in these cases, the following low-frequency voltages are distinguished:

u _н6 (t) = U _н2 cosφ _k (t) + U _н4 cos [2π (f _п1 -f _г1 ) t + φ _п2 -φ _г1 ],

u _н7 (t) = U _н2 cos [φ _k (t) + 30 °] + U _н4 cos [2π (f _n2 -f _g1 ) + φ _n2 -φ _g1 + 30 °],

u _n8 (t) = U _n2 cos [φ _k (t) -30 °] + U _n4 cos [2π (f _n2 -f _g1 ^. ) + φ _n2 -φ _g1 -30 °]

.Where

.

The output of the subtraction block 58.1 produces the following differential voltage:

Analysis of the obtained differential voltage Δu _n4 shows that it is an estimate of the noise component, which differs from the noise component in the main channel by a + 90 phase rotation.

The differential voltage Δu _n4 from the output of the subtraction unit 58.1 is fed to the input of the phase shifter 60.1 by + 90 °, at the output of which a voltage is generated:

Δu _n5 (t) = U _n4 sin [2π (f _n2 -f _g1 ) t + φ _n2 -φ _g1 + 90 °] = U _n4 cos [2π (f _n2 -f _g1 ) t + φ _n2 -φ _g1 ] ,

which goes to the second input of the subtraction block 62.1. The first input of the latter is supplied with voltage u _n6 (t) from the output of the phase detector 26.1. The following differential voltage is generated at the output of the subtraction block 62.1:

Δu _н6 (t) = u _н6 (t) -Δu _н5 (t) = U _н2 cosφ _k (t),

in which the interference component is already absent. This voltage is supplied to the input of the digital descrambler 27.1

Similarly, analog and discrete information from ground point B is transmitted through a geostationary IS3 repeater to ground point A at a frequency f ₂ .

Similar operations are carried out at point B. At the second measurement step (when transmitting a signal from point B), after some time T after the end of signal registration, switch 13 at point A opens, and at step B, switch 13.1 closes. At this point in time t ₃ ^B = t ₂ ^B + T, the generator 12.1 begins to generate a new microwave signal (signal β ₃ ), which, after appropriate conversion, is fixed at point B and is emitted in the direction of the geostationary IC3 repeater at a frequency f ₂ . The signal received by the receiver of the IS3 transponder is re-emitted back to points A and B at a frequency f ₂ while maintaining phase relationships in the entire frequency band of the signal. The relay signal is received at both points, converted to a low frequency voltage and recorded at time t ^A ₄ and t ^B _4, respectively (signal α ₄ , β ₄ ). This ends the single measurement of the duplex method. Then, in the interval between the measurement acts, pairs of signals α ₁ , α ₂ and α ₃ , α ₄ (β ₁ , β ₂ and β ₃ , β ₄ ) are subjected to correlation processing in correlator 11 (11.1).

The correlation processing of two pairs of registered signals in the interval between the measurement events in the correlators 1.19 and 2.19 determines the following time delays:

the corresponding interference frequencies Fi (i = 1, 2, 3,4), which determine the derivatives of these delays:

,

,

,Where

,

a _j , b _j (j> 1, 2, 3) - signal propagation time between the satellite and points A and B, respectively;

t _A , t _B - signal delay time in the radiating equipment of both points;

τ _A , τ _B - signal delays in the receiving and recording equipment;

ΔS is the signal delay in the on-board equipment of the IS3 repeater;

Δt = t ₁ ^B -t ₁ ^A is the desired difference in the clock reading at the same physical moment.

,Assuming a _j and b _j linear functions with derivatives

,

, из первых двух уравнений получим:

, from the first two equations we get:

Where

Δ ' _{A, B} , Δ'' _{A, B} - signal delays in the atmosphere at a frequency f ₁ and f _2, respectively;

v - relativistic correction (Sagnac effect);

ω is the angular velocity of the Earth;

c is the speed of light;

D is the area of the quadrilateral O'A'S'B ', formed in the equatorial

planes by the center of mass of the Earth, projections of points A, B and

IS3-repeater.

The correction y for the mobility of the IS3 repeater during a single measurement is most easily reduced to zero by the appropriate choice of the free parameter Θ:

which should be calculated at the beginning of measurements by approximate ephemeris data, and then clarified by the results of current measurements.

As for the correction δ, hardware delays, it can be found by calibration using the “zero base” method. The atmospheric correction ε is also taken into account.

In points A and B, the equipment works the same way, only the order of steps in them is the opposite. To calculate the difference of the clock readings Δt, it is enough to exchange the received digital data between the points, which can be done via satellite channels through the same geostationary IC3 repeater, as shown above.

The proposed system provides not only synchronization of remote time scales, but also the exchange of confidential discrete and analog information between remote ground stations and its protection against unauthorized access.

Moreover, the protection of this information has three levels: energy, structural, and cryptographic.

The energy and structural levels are ensured by the use of complex signals with combined phase shift keying and amplitude modulation, which have high energy and structural secrecy.

The energy secrecy of these signals is due to their high compressibility in time or in the spectrum with optimal processing, which reduces the instantaneous radiated power. As a result, the complex signal used at the receiving point may be masked by noise. Moreover, the energy of a complex signal is by no means small, it is simply distributed over the time-frequency domain so that at each point in this region the signal power is less than the power of noise and interference.

The structural secrecy of complex signals with combined phase shift keying and amplitude modulation is due to the wide variety of their shapes and significant ranges of parameter changes, which makes it difficult to optimize or at least quasi-optimal processing of complex signals of an a priori unknown structure in order to increase the sensitivity of the receiver.

Complex signals with combined phase shift keying and amplitude modulation open up new possibilities in the technique of transmitting messages on a single carrier frequency and protecting them from unauthorized access. These signals allow the use of a new type of selection - structural selection. This means that there is a new opportunity to distinguish complex signals from other signals and interference operating in the same frequency band and at the same time intervals. This feature is realized by convolution of the spectrum of complex signals.

The cryptographic level is provided by special methods of encrypting, encoding and converting confidential discrete and analog information, as a result of which its content becomes inaccessible without presenting the cryptogram key and reverse conversion.

Thus, the proposed clock synchronization system in comparison with the prototype provides increased noise immunity and reliability of the exchange of confidential discrete and analog information between ground points spaced over long distances, using a geostationary IS3 repeater and protecting this information from unauthorized access. This is achieved by suppressing narrow-band interference by the phase-compensation method.

Claims (1)

A clock synchronization system containing a geostationary satellite repeater, first and second ground stations, each of which contains an analog descrambler, the output of which is the second output of the system, a digital descrambler, the output of which is the third output of the system, a time and frequency reference connected in series, the first local oscillator, a first mixer, the second input of which is connected through a series-connected duplexer and a first power amplifier to a transceiver antenna, an amplifier of a first intermediate frequency, the selector, the first amplitude limiter and the first synchronous detector, the second input is connected to the output of the selector, the first clipper is connected in series to the output of the first amplitude limiter, the second input of which is connected to the second output of the time and frequency standard, the first memory unit and the correlator, the output of which is the first output systems connected in series to the first output of the time and frequency standard, a second local oscillator, a second mixer, the second input of which is connected via a switch to the first output of the generator a pseudo-noise signal torus, a second intermediate frequency amplifier and a second power amplifier, the output of which is connected to the duplexer input, a pseudo-noise signal generator, a second amplitude limiter, a second clipper, the second input of which is connected to the second output of the time standard, and connected in series to the third output of the time and frequency reference frequency, and a second memory unit, the output of which is connected to the second input of the correlator, connected to the output of the first amplitude limiter multiplier, the second input of which is connected to the output of the first local oscillator, a bandpass filter and the first phase detector, the second input of which is connected to the second output of the second local oscillator, characterized in that each ground station is equipped with two phase shifters + 30 °, two phase shifters -30 °, two phase shifters + 90 °, the second and third synchronous detectors, the second and third phase detectors and four subtraction units, and the first phase shifter + 30 °, the second sync are connected in series to the output of the first amplitude limiter a detector, the second input of which is connected to the output of the selector, the first subtraction unit, the first phase shifter + 90 ° and the second subtraction unit, the second input of which is connected to the output of the first synchronous detector, and the output is connected to the input of the analog descrambler, to the output of the first amplitude limiter in series connected the first phase shifter at -30 ° and the third synchronous detector, the second input of which is connected to the output of the selector, and the output is connected to the second input of the first subtraction unit, to the second output of the second local oscillator the second phase shifter + 30 °, the second phase detector, the second input of which is connected to the output of the bandpass filter, the third subtraction unit, the second phase shifter + 90 ° and the fourth subtraction unit, the second input of which is connected to the output of the first phase detector, and the output is connected to the input of the digital descrambler, to the second output of the second local oscillator, a second phase shifter of -30 ° and a third phase detector are connected in series, the second input of which is connected to the output of the bandpass filter, and the output is connected to the second mu input of the third block of subtraction.

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